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11863034 | MODES OF THE INVENTION Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings in detail. Purposes, specific advantages, and novel features of the invention will be made clear from embodiments and the following detailed description in connection with the accompanying drawings. In the description of the invention, when it is determined that detailed descriptions of related well-known functions unnecessarily obscure the gist of the invention, the detailed descriptions thereof will be omitted. It will be understood that, although the terms “first,” “second,” etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another. For example, a first element could be termed a second element, and a second element could similarly be termed a first element without departing from the scope of the present invention. As used herein, the term “and/or” includes any and all combinations of one or more of the associated listed items. FIG.1is a conceptual view illustrating a motor according to an embodiment. Referring toFIG.1, the motor according to the embodiment may include a rotating shaft100, a rotor200, a stator300, and a rotor location sensing apparatus400. The rotating shaft100may be coupled to the rotor200. When an electromagnetic interaction is generated between the rotor200and the stator300through the supply of current, the rotor200rotates, and the rotating shaft100rotates in conjunction with the rotation. The rotating shaft100may be connected to a steering shaft of a vehicle to transfer power to the steering shaft. The rotating shaft100may be supported by a bearing. The rotor200rotates through an electric interaction with the stator300. The rotor200may include a rotor core210and a magnet220. The rotor core210may be formed in a shape in which a plurality of plates having a circular thin steel sheet shape are stacked. A hole may be formed at a center of the rotor core210so that the rotating shaft100is coupled thereto. A protrusion configured to guide the magnet220may protrude from an outer circumferential surface of the rotor core210. The magnet220may be attached to the outer circumferential surface of the rotor core210. A plurality of magnets220may be disposed along a circumference of the rotor core210at regular intervals. The rotor200may include a can member which surrounds the magnets220, fixes the magnets220so as not to be separated from the rotor core210, and inhibits the magnets220from being exposed. Meanwhile, the rotor200may include the single-piece rotor core210having a cylindrical shape and the magnets220disposed in a single-stage on the rotor core210. Here, the single-stage refers to a structure in which the magnets220may be disposed such that there is no skew on an outer circumferential surface of the rotor200. Accordingly, the rotor core210may be formed to have the same height as a height of the magnet220on the basis of a longitudinal section of the rotor core210and a longitudinal section of the magnet220. That is, the magnet220may be implemented to cover the entire rotor core210with respect to a height direction. The stator300may be wound by a coil to induce the electric interaction with the rotor200. A detailed configuration of the stator300for winding a coil320is as follows. The stator300may include a stator core310having a plurality of teeth. The stator core310may be provided with a yoke portion having a ring shape and the teeth around which the coil is wound in a center direction from the yoke portion. The teeth may be provided at regular intervals along an outer circumferential surface of the yoke portion. Meanwhile, the stator core310may be formed by stacking a plurality of plates having a thin steel sheet shape. Further, the stator core310may be formed by coupling or connecting a plurality of divided cores to each other. The rotor location sensing apparatus400may include a magnet410and a substrate420. A housing500is formed in a cylindrical shape to provide a space therein, in which the stator300and the rotor200may be mounted. Here, although a shape or material of the housing500may be variously changed, a metal material, which may withstand high temperatures well, may be selected. An open upper portion of the housing500is covered by a cover600. FIG.2is a view illustrating a magnet. Referring toFIG.2, the magnet410may include a main magnet411, a sub-magnet412, and a sensing plate413. The magnet410is disposed above the rotor200to indicate a location of the rotor200. The sensing plate413is formed in a circular plate shape. In addition, the rotating shaft100is coupled to a center of the sensing plate413. The main magnet411is disposed at a center of the sensing plate413. In addition, the sub-magnet412is disposed on an outer side of the main magnet411and may be disposed on an edge of the sensing plate413. The main magnet411corresponds to the magnet220of the rotor200. In other words, the number of poles of the magnet220of the rotor200and the number of poles of the main magnet411are the same. For example, when the magnet220of the rotor200has six poles, the main magnet411also has six poles. A pole division region of the magnet220of the rotor200is aligned with that of the main magnet411so that a location of the main magnet411may indicate a location of the magnet220of the rotor200. The main magnet411is used to grasp an initial location of the rotor200. The sub-magnet412is used to precisely grasp a detailed location of the rotor200. For example, the sub-magnet412may have 72 poles. Sensors disposed on the substrate420sense a change in magnetic flux by the main magnet411and the sub-magnet412according to a rotation of the magnet410. The substrate420may be disposed above the magnet410. FIG.3is a view illustrating a sensing signal. Referring toFIG.3, the sensors disposed on the substrate420may sense three sensing signals T1, T2, and T3by sensing a change in an N-pole and S-pole of the main magnet411. In addition, the substrate420may further sense two sensing signals E1and E2by sensing a change in magnetic flux of the sub-magnet412. As described above, since the magnet coupled to the rotor200is directly copied to the main magnet411, the location of the rotor200may be sensed by sensing the change in the magnetic flux on the basis of the main magnet411. The sensing signals T1, T2, and T3may be used for initial driving of the motor and may feed-back U-, V-, and W-phases information, respectively. FIG.4is a view illustrating a rotor location sensing apparatus. As shown inFIG.4, a shape of the substrate420may be implemented as a ring shape corresponding to an arrangement of the main magnet411and the sub-magnet412. The substrate420may include first sensors S1and S3and second sensors S2and S4. The first sensors S1and S3and the second sensors S2and S4may be arranged on the same track having a circular shape on the basis of a center C of the magnet410. The first sensors S1and S3may include a plurality of first Hall sensors H1adjacent to each other on the circular shaped track. In addition, the second sensors S2and S4may include a plurality of second Hall sensors H2adjacent to each other on the circular shaped track. The first sensor S1and the second sensor S2, which are located relatively further inward, may be disposed along the circular shaped track disposed on the main magnet411. In other words, the first sensor S1and the second sensor S2may be disposed to correspond to the main magnet411in the radial direction of the magnet410. The first sensor S3and the second sensor S4, which are located relatively further outward, may be disposed along the circular shaped track disposed on the sub-magnet412. In other words, the first sensor S3and the second sensor S4may be disposed to correspond to the sub-magnet412in the radial direction of the magnet410. First Embodiment FIG.5is a view illustrating a first embodiment of an arrangement of a first sensor and a second sensor which are corresponding to a main magnet. Referring toFIGS.4and5, a first sensor S1and a second sensor S2disposed on an inner side of a substrate420sense a change in magnetic flux by a main magnet411. The first sensor S1may include three first Hall sensors H1. The first sensor S1may generate a continuous sensing signal having a U-phase, a V-phase, and a W-phase corresponding to a rotation of the main magnet411. The three first Hall sensors H1may be disposed to be apart from each other by a first angle R1. The second sensor S2may include three second Hall sensors H2. The second sensor S2may also additionally generate the continuous sensing signal having the U-phase, the V-phase, and the W-phase corresponding to the rotation of the main magnet411. Accordingly, the continuous sensing signal having the U-phase, the V-phase, and the W-phase may be generated even when any first Hall sensor H1of the first sensor S1fails. The three second Hall sensors H2may be disposed to be apart from each other by the first angle R1in the same manner as the first Hall sensor H1. Here, the first angle R1may be calculated by Equation 1 below, R1=R0/3 R0=360°/(Nm/2) [Equation 1] wherein R1is the first angle, R0is an electrical angle, Nm is the number of poles of the main magnet411, and a constant “3” refers to the number of a U-phase, a V-phase, and a W-phase. For example, when the magnet220of the rotor200has six poles, the number of poles of the main magnet411is six. Thus, the electrical angle R0of a corresponding motor is 120°. As a result, the first angle R1may be calculated as 40°. Here, the electrical angle indicates a physical angle (mechanical angle) of the magnet occupied by the N pole and S pole of the magnet on the basis of 360°. For example, when the magnet220of the rotor200has eight poles, the electrical angle R0of the corresponding motor is 90°. The second sensor S2may be disposed at a location shifted from a location corresponding to the first sensor S1to increase a resolution of the sensing signal. In other words, the first sensor S1and the second sensor S2may be disposed on the same track having a circular shape so as to be apart from each other by a second angle R2which is different from the first angle R1. That is, a first Hall sensor H1aand a second Hall sensor H2a, which are adjacent to each other, may be disposed along a circumference on the circular shaped track to be apart from each other by the second angle R2different from the first angle R1. Here, the second angle R2may be calculated by Equation 2 below, R2=R1±R0′/(Nm/2) [Equation 2] wherein R2is the second angle, R1is the first angle, R0′ is an electrical angle to be shifted, and Nm is the number of poles of the main magnet411. The resolution of the sensing signal due to the main magnet411may be set to 60°, and here, in a case in which a shift by an electrical angle of 30° is required to increase the resolution two times from 60° to 30°, the second angle R2may be calculated as 30° or 50° when the R1is 40°. FIG.6is a view illustrating a first embodiment of an arrangement of a first sensor and a second sensor which are corresponding to a sub-magnet. Referring toFIGS.4and6, a first sensor S3and a second sensor S4disposed on an outer side of the substrate420sense a change in magnetic flux by a sub-magnet412. The first sensor S3may include two first Hall sensors H1. The first sensor S3may generate a continuous sensing signal corresponding to a rotation of the sub-magnet412. The two first Hall sensors H1may be disposed to be apart from each other by a first angle R1. The second sensor S4may include two second Hall sensors H2. The second sensor S4may also additionally generate the continuous sensing signal corresponding to the rotation of the sub-magnet412. Accordingly, the continuous sensing signal may be generated even when any first Hall sensor H1of the first sensor S3fails. The two second Hall sensors H2may be disposed to be apart from each other by the first angle R1in the same manner as the first Hall sensor H1. Here, the first angle R1may be calculated by Equation 3 below, R1=R0×n+R3/(Ns/2)(nis an integer) R0=360°/(Ns/2) [Equation 3] wherein R1is the first angle, R0is an electrical angle, R3′ is a resolution angle, and Ns is the number of poles of the sub-magnet412. For example, when the number of poles of the sub-magnet412is 72, the electrical angle R0of a corresponding motor is 10°. When R3is 90°, the first angle R1is 10°×n+2.5°. Thus, it is very difficult to dispose the two first sensors S3physically apart from each other by the first angle R1. Accordingly, when the electrical angle R0is 10°, 10°×n+2.5° having the same phase difference may be calculated as the first angle R1. The second sensor S4may be disposed at a location shifted from a location corresponding to the first sensor S3to increase a resolution of the sensing signal. In other words, the first sensor S3and the second sensor S4may be disposed on the same track having a circular shape to be apart from each other by a second angle R2different from the first angle R1. That is, a first Hall sensor H1aand a second Hall sensor H2a, which are adjacent to each other, may be disposed along a circumference on the circular shaped track to be apart from each other by the second angle R2different from the first angle R1. Here, the second angle R2may be calculated by Equation 4 below, R2=R1±R0′/(Ns/2) [Equation 4] wherein R2is the second angle, R1is the first angle, R0′ is an electrical angle to be shifted, and Ns is the number of poles of the sub-magnet412. Accordingly, when the electrical angle R0′ to be shifted is 45° and the number of poles of the sub-magnet412is 72, the second angle R2is a value obtained by adding 1.25° to 10°×n+2.5° which is the first angle R1. As a result, as shown inFIG.6, the resolution of the sensing signal may be increased from 90° to 45° by disposing the first Hall sensor H1aand the second Hall sensor H2a, which are adjacent to each other, to be apart from each other by the value obtained by adding 1.25° to 10°×n+2.5° which is the first angle R1. Second Embodiment FIG.7is a view illustrating a second embodiment of an arrangement of a first sensor and a second sensor which are corresponding to a main magnet. Referring toFIGS.4and7, a first sensor S1and a second sensor S2disposed on an inside of a substrate420sense a change in magnetic flux by a main magnet411. The first sensor S1may include three first Hall sensors H1. The first sensor S1may generate a continuous sensing signal having a U-phase, a V-phase, and a W-phase corresponding to a rotation of the main magnet411. The three first Hall sensors H1may be disposed to be apart from each other by a third angle R3. The second sensor S2may include three second Hall sensors H2. The second sensor S2may also additionally generate the continuous sensing signal having the U-phase, the V-phase, and the W-phase corresponding to the rotation of the main magnet411. Accordingly, the continuous sensing signal having the U-phase, the V-phase, and the W-phase may be generated even when any first Hall sensor H1of the first sensor S1fails. The three second Hall sensors H2may be disposed to be apart from each other by the third angle R3in the same manner as the first Hall sensor H1. Here, the third angle R3may be calculated by Equation 5 below, R3=R0/3 R0=360°/(Nm/2) [Equation 5] wherein R3is the third angle, R0is an electrical angle, Nm is the number of poles of the main magnet, and a constant “3” refers to the number of a U-phase, a V-phase, and a W-phase. For example, when the magnet220of the rotor200has six poles, the number of poles of the main magnet411is six. Thus, the electrical angle R0of a corresponding motor is 120°. As a result, the first angle R1may be calculated as 40°. For example, when the magnet220of the rotor200has eight poles, the electrical angle R0of the corresponding motor is 90°. The second sensor S2may be disposed at a location shifted from a location corresponding to the first sensor S1to increase a resolution of the sensing signal. In other words, when a location symmetrical with respect to each first Hall sensor H1of the first sensor S1is defined as “P” inFIG.7on the basis of a reference line CL passing through a center C of a shaft, the second Hall sensors H2of the second sensor S2may be located at a location shifted from the “P” inFIG.7by a fourth angle R4in a circumferential direction. Here, the fourth angle is R4calculated by Equation 6 below, R4=R3±R0′/(Nm/2) [Equation 6] wherein R4is the fourth angle, R0′ is an electrical angle to be shifted, and Nm is the number of poles of the main magnet411. The resolution of the sensing signal due to the main magnet411may be set to 60°, and here, in a case in which a shift by an electrical angle of 30° is required to increase the resolution two times from 60° to 30°, the second angle R2may be calculated as 10°. Accordingly, in a case in which the number of poles of the main magnet411is 6, the resolution of the sensing signal may be increased from 60° to 30° when the second sensors S2are disposed by moving clockwise or counterclockwise by 10° as compared to the first sensor S1. FIG.8is a view illustrating a first sensor and a second sensor on the basis of an outer side sensor. Referring toFIGS.4and8, a plurality of sensors disposed on an outside of the substrate420may be classified into a first sensor S3and a second sensor S4. The first sensor S3and the second sensor S4sense a change in magnetic flux by a sub-magnet412. The first sensor S3may include two first Hall sensors H1. The first sensor S3may generate a continuous sensing signal corresponding to a rotation of the sub-magnet412. The two first Hall sensors H1may be disposed to be apart from each other by a third angle R3. Here, the third angle R3may be calculated by Equation 7 below, R3=R0×n+R3′/(Ns/2)(nis an integer) R0=360°/(Ns/2) [Equation 7] wherein R3is the third angle, R0is an electrical angle, R3′ is a resolution angle, and Ns is the number of poles of the sub-magnet412. For example, when the number of poles of sub-magnet412is 72, the electrical angle R0of a corresponding motor is 10°. When R3′ is 90°, the third angle R3is 10°×n+2.5°. Thus, it is very difficult to dispose the two first Hall sensors H1physically apart from each other by the third angle R3. Accordingly, when the electrical angle R0is 10°, 10°×n+2.5° having the same phase difference may be calculated as the third angle R3. In addition, when the electrical angle R0is 90° and a shift by an electrical angle of 45° is required, a fourth angle R4may be calculated as 1.25° through Equation 8 below. R4=R3±R0′/(Ns/2) [Equation 8] wherein R4is the fourth angle, R0′ is an electrical angle to be shifted, and Ns is the number of poles of the sub-magnet412. Accordingly, when the number of poles of the sub-magnet412is 72, a resolution of the sensing signal may be set to 90°, and the resolution of the sensing signal may be increased from 90° to 45° when the second sensors S4are disposed by moving clockwise or counterclockwise by 1.25° than the first sensor S3. FIGS.9A and9Bare graphs illustrating a comparison between a conventional sensing signal having a resolution of 60° (FIG.9A) and a sensing signal increased in resolution to 30° (FIG.9B) with respect to a main magnet. When the number of poles of the main magnet411is six, as shown inFIG.9A, the resolution of the sensing signal is confirmed as 60° by the first sensor S3. However, as shown inFIGS.7and9B, the resolution of the sensing signal may be increased from 60° to 30° when the second sensor S2is added and the second Hall sensors of the second sensor S2(H2inFIG.7) are disposed in such a manner that locations thereof are shifted clockwise by 10° as compared to the first Hall sensor H1of the first sensor S1. Thus, an initial driving location of the motor may be grasped more precisely. FIGS.10A and10Bare graphs illustrating a comparison between a conventional sensing signal having a resolution of 90° (FIG.10A) and a sensing signal increased in resolution to 45° (FIG.10B) with respect to a sub magnet. When the number of poles of the sub-magnet412is 72, as shown inFIG.10A, the resolution of the sensing signal is confirmed as 90° by the first sensor S3. However, as shown inFIGS.8and10B, the resolution of the sensing signal may be increased from 90° to 45° when the second sensor S4is added and second Hall sensors (H2inFIG.8) of the second sensor S4are disposed in such a manner that locations thereof are shifted clockwise by 1.25° as compared to the first hall sensor H1of the first sensor S3. Third Embodiment FIG.11is a view illustrating an extended region of a main magnet. Referring toFIGS.2and11, a main magnet411may include an extended region411aextending toward a center of a magnet410. The extended region411ais a portion corresponding to a location of a third sensor423(inFIG.13) added in parallel to a second sensor422(inFIG.13). Meanwhile, a sub-magnet412is used to precisely grasp a detailed location of a rotor200. For example, the sub-magnet412may have 72 poles. Sensors may be disposed on a substrate420. The sensors sense a change in magnetic flux according to a rotation of the magnet410. The substrate420may be disposed above the magnet410. FIG.12is a view illustrating a sensing signal. Referring toFIG.12, the sensors disposed on the substrate420may sense three sensing signals T1, T2, and T3by sensing a change in the N-pole and S-pole of the main magnet411. In addition, two sensing signals E1and E2may be further sensed by sensing a change in magnetic flux of the sub-magnet412. As described above, since the magnet coupled to the rotor200is directly copied to the main magnet411, the location of the rotor200may be sensed by sensing the change in magnetic flux on the basis of the main magnet411. The sensing signals T1, T2, and T3may be used for initial driving of a motor and may feed-back U-, V-, and W-phases information, respectively. FIG.13is a view illustrating a rotor location sensing apparatus according to the embodiment, andFIG.14is a view illustrating a first sensor, a second sensor, and a third sensor. Referring toFIGS.13and14, the substrate420may include a first sensor421, a second sensor422, and a third sensor423. The first sensor421senses a change in magnetic flux by the sub-magnet412according to a rotation of the magnet410. The second sensor422and the third sensor423sense a change in the magnetic flux by the main magnet411according to a rotation of the magnet410. The substrate420may be disposed in a form of drawing an arc corresponding to an arrangement of the main magnet411and the sub-magnet412. The first sensor421, the second sensor422, and the third sensor423may be respectively arranged on tracks O1, O2, and O3which are different from each other with respect to a center C of the magnet410. The first sensor421is disposed on an outer side of the second sensor422, and the third sensor423is disposed on an inner side of the second sensor422in a radial direction of the magnet410. The first sensor421may include a plurality of first Hall sensors421a(for example, four first Hall sensors), and the plurality of sensors may be disposed at regular intervals along an outer track O1so as to be aligned with the sub-magnet412. The second sensor422may be disposed at regular intervals along a middle track O2so that a plurality of second Hall sensors422a(for example, three second Hall sensors) are aligned with the main magnet411. The third sensor423may be disposed at regular intervals along an inner track O3so that a plurality of third Hall sensors423a(for example, three third Hall sensors) are aligned with the extended region411aof the main magnet411. FIG.15is a view illustrating the second sensor and the third sensor which are aligned and disposed in a circumferential direction of the magnet. Here, referring toFIG.15, a second Hall sensor422aof the second sensor422and a third Hall sensor423aof the third sensor423are aligned and disposed in a circumferential direction of the magnet410. The second Hall sensor422aand the third Hall sensor423aare disposed on different circular shaped tracks. In addition, since the second Hall sensor422aand the third Hall sensor423aare aligned on the basis of the circumferential direction of the magnet410, even when the third Hall sensor423ais added, it is not necessary to extend the substrate420or install a separate substrate420to connect with a cable. That is, a two-channel sensing structure may be implemented, and the limitation of an installation space may be overcome by securing a region, in which the third sensor423is mounted, inside the existing substrate420in the radial direction of the magnet410. Correspondingly, as described above, the main magnet411includes the extended region411aextending toward the center of the magnet410. The second sensor422may be electrically connected in parallel with the third sensor423. Accordingly, when an abnormality occurs in the second sensor422, the third sensor423may sense the sensing signal. As described above, the rotor location sensing apparatus and the motor according to the embodiments of the present invention has been specifically described with reference to the accompanying drawings. The above description is only an example describing a technological scope of the present invention. Various changes, modifications, and replacements may be made without departing from the spirit and scope of the present invention by those skilled in the art. Therefore, the embodiments disclosed above and in the accompanying drawings should be considered in a descriptive sense only and not for limiting the technological scope. The technological scope of the present invention is not limited by these embodiments and the accompanying drawings. The spirit and scope of the present invention should be interpreted by the appended claims and encompass all equivalents falling within the scope of the appended claims. | 26,151 |
11863035 | First of all, it is to be noted that, in the different embodiments described, equal parts are provided with equal reference numbers and/or equal component designations, where the disclosures contained in the entire description may be analogously transferred to equal parts with equal reference numbers and/or equal component designations. Moreover, the specifications of location, such as at the top, at the bottom, at the side, chosen in the description refer to the directly described and depicted figure and in case of a change of position, and these specifications of location are to be analogously transferred to the new position. FIG.1is a strongly-schematic depiction of a stator1in an oblique view. Here, the stator1comprises an essentially hollow-cylindrical laminated core2in which a plurality of receiving grooves4are arranged distributed in a circumferential direction10. Here, the receiving grooves4are continuously formed in a longitudinal direction11. InFIG.1, multiple electric conductors8are depicted in an exemplary manner before being connected to an electric winding. Analogously, it can be seen fromFIG.1in an exemplary manner that multiple electric conductors8can be bent for forming an electric coil and/or winding in a circumferential direction of the laminated core2and electric conductors8corresponding with one another can be interconnected. The receiving grooves4of the laminated core2can be open in a radial direction12in a direction of the longitudinal axis3of the stator1. Such openings can be configured as an air gap5. The areas of the laminated core2which delimit the receiving grooves4in a direction of the longitudinal axis3can be configured as a tip of a tooth6in a circumferential direction10. The opposite side of the respective receiving groove4, also referred to as yoke side, is where the groove base7is located. The precise number of receiving grooves4as well as the electric conductors8received therein is determined by the desired size and/or design of the electric machine. In principle, the receiving grooves4can have most different cross-sectional shapes, wherein corresponding rectangular cross sections of the receiving grooves4are tried and trusted for receiving electric conductors8. To insulate the individual electric conductors8relative to one another as well as to the laminated core2, it is required to flawlessly form at least one insulation layer9in a circumferential direction10as well as radial direction12in a continuously-closed manner, in particular to provide it at the lateral surface of the conductors8, wherein the electric conductors8are respectively coated with an insulation layer9, at least within the laminated core2. The essentially hollow-cylindrical laminated core2has a first and a second axial front end13a,13b.The electric conductors8in the receiving grooves4are preferably formed by metal forming rods, preferably made of copper or another material with good electrical conductivity. Here, these forming rods form a plurality of electric conductor sections La, Lb, which extend at least within the respectively-allocated receiving grooves4. These conductor sections La, Lb may, in this case, be defined by so-called I-pins or be formed by so-called hairpins, in which latter case the conductor sections La, Lb constitute the legs of these essentially U-shaped conductor segments. Therefore, the electric conductor sections La, Lb are arranged multiply in each of the receiving grooves4and the diagrammed stator winding14is constructed by predetermined electric connections between the annularly-positioned conductor sections La, Lb, which stator winding14serves to generate a circumferential magnetic field when single-phase or multiple-phase electric energy is applied to the stator1. As can be seen in an exemplary manner fromFIG.1, such a stator winding14, in the ready-to-use state, has multiple layers L of conductor sections La, Lb, which layers L are immediately adjacent in a radial direction to the longitudinal axis3of the laminated core2. The supplying of single-phase alternating current or of multi-phase alternating current (three-phase current) is done via specific connection points, not depicted in more detail, at the stator winding14, as this is widely understood. In the stator winding14in accordance with the example according toFIG.2, a total of eight layers L1 to L8 are provided. Here, the layers L1 to L8 are composed of a plurality of conductor sections La, Lb positioned in the receiving grooves4. Typically, a practical stator1has an even number of layers L, preferably from4layers, in particular between4and12layers. The diameter of the laminated core2, the number of the receiving grooves4configured, the number of the layers L, as well as the axial length of the stator1and/or laminated core2essentially depend on which performance data is required and/or which physical requirements are in place for the electric machine to be constructed. The stator winding14comprises at least two part-windings TWa, TWb. A stator1structured in accordance with the invention and/or its stator winding14can in particular have a multiple of two part-windings, in particular comprise two, four, six, eight or even ten part-windings, as this is described below. In this case, the respective part-windings can be connected in series and/or in parallel per electric phase. The winding diagram specified below enables a formation of single- or multi-phase stator windings14. In case of a multi-phase stator winding14, each so-called phase winding PW and/or each winding phase pertaining to a phase may be composed of two or more part-windings TWa to TWx, wherein x represents an even number larger than two. Here, the part-windings TWa to TWx, which constitute electrically-separate coils, can be connected in series and/or in parallel and/or be connected accordingly by a control unit, depending on the demand and/or power requirement. In the exemplary embodiment depicted inFIG.2, for the sake of better clarity only a single phase winding PW and/or a single winding phase of a stator winding14is depicted whichstator winding14is, overall, provided as a three-phase stator winding14. FIG.2shows an advantageous winding diagram for the winding phase and/or for the phase winding PW of a two-pole stator winding14in accordance with the example. This phase winding PW comprises two coil sections and/or two electric part-windings TWa, TWb. The two part-windings TWa and TWb may also be referred to as first and second winding branches and/or as first and second parallel paths. In this two-pole stator1, each of the part-windings TWa and TWb extends over the entire circumference, in particular over 360° of the annular laminated core2. In the exemplary embodiments described below, the individual part-windings extend only over a fraction of the circumference of the laminated core. The laminated core2in accordance withFIG.2, in accordance with the example, has eighteen receiving grooves4, wherein each of the receiving grooves4respectively comprises and/or receives eight conductor sections La or Lb, so that the stator winding14has a total of eight layers L, which are referred to as L1 to L8. Therefore, a stator winding14and/or a phase winding PW with a total of eight layers L is depicted. Here, the layer L1 can be understood to be the layer located closest to the air gap5—FIG.1—and/or the radially-innermost layer and the layer L8 can be understood to be the layer located closest to the groove base7—FIG.1—and/or the radially-outermost layer L8. The depicted stator winding14and/or phase winding PW has a so-called number of fractional slots q=3. This means that the number of the receiving grooves4per magnetic pole section19aor19b,per electric phase and/or phase winding PW and per part-winding TWa or TWb is exactly “three.” This is apparent inFIG.2also by means of the groups of three receiving grooves4, which are marked with cross-hatching. It can further be seen fromFIG.2that, in accordance with the example, the two depicted part-windings TWa and TWb pertain to the electric phase U—see first line inFIG.2. To construct a stator winding14with a total of three phases, the sections and/or receiving grooves4depicted as unoccupied inFIG.2are to be provided with corresponding further part-windings. In particular, the structure of the depicted phase winding PW repeats for the phase V and for the phase W in the receiving grooves4with the numbers 4 to 6 and 13 to 15 and/or 7 to 9 and 16 to 18. In particular, merely one lateral offset and/or groove offset is to be provided and/or to be taken into account, and the structure and/or the diagram of the phase winding PW for the phase U repeats thereafter for the phase windings of the phases V and W. Only for the sake of better clarity, the phase windings PW for the phases V and W are not included in the depiction in accordance withFIG.2. In particular, it must be said that merely one phase winding PW, for example for the phase U, is illustrated inFIG.2. The complete, three-phase stator winding is to be connected to a three-phase voltage source, wherein a magnetically two-pole stator1forms and/or a two-pole rotating magnetic field appears. Each of the part-windings TWa and TWb of the stator winding14and/or of the phase winding PW is formed by a plurality of serially-connected conductor sections La and Lb. These conductor sections La and Lb may form part of a conduction segment which is integrally-structured and/or configured as one piece, in particular of a so-called hairpin, as it is depicted inFIGS.4aand4b. Alternatively, the conductor sections La and Lb can also respectively be formed by so-called I-pins, as this is illustrated inFIG.5. Therefore, the conductor sections La and Lb may be defined by the legs of an essentially U-shaped conduction segment, in particular by a so-called hairpin—FIGS.4a,4b. Yet the conductor sections La and Lb may also be defined by the middle section of so-called I-pins which are electrically connected in series—FIG.5. The part-windings TWa and TWb are respectively formed by multiple serially-connected conductor sections La and Lb, wherein first and second electric connecting sections VBa, VBb between electrically serially-connected conductor sections La, Lb are allocated alternately to the first and the second axial front end13a,13bof the laminated core2—FIG.1—depending on which one is located closest. Here, the first connecting sections VBa form an electric connection between immediately-subsequent conductor sections La and Lb. In contrast to this, the second connecting sections VBb form an electric connection between a conductor section Lb and a conductor section La serially adjoining it. In accordance withFIGS.4a,4band5, therefore, the two conductor sections La and Lb are electrically connected in series by means of the connecting section VBa and thereby define a pair of forming rods16. Such a pair of forming rods16can be electrically connected in series with a serially-adjoining and/or immediately-adjacent pair of forming rods16via the connecting section VBb, so that, overall, a so-called loop winding15can be constructed, as it is illustrated in an exemplary manner inFIGS.6,7. Here, the first connecting sections VBa are depicted inFIG.2by fine solid lines and/or arrows, whereas the second connecting sections VBb are illustrated inFIG.2by relatively broad solid lines and/or arrows. The fine lines located closest may here be mentally allocated to the first axial front end13a,whereas the broad lines are mentally assigned to the second axial front end13bof the stator1(FIG.1). Here, the rectangles depicted inFIG.2using dotted hatching symbolize contact points, in particular welded points, within the second electric connecting sections VBb. The conductor sections La and Lb provided in accordance withFIG.2are thus partial sections of so-called hair-pins, as they are illustrated in an exemplary manner inFIGS.4a,4b. The zones depicted inFIG.2using diagonal hatching respectively show the beginnings of the winding of the part-windings TWa and TWb, whereas the zones with vertical hatching represent the respective ends of the winding of the part-windings TWa and TWb. These assignments of hatching also apply to the winding diagrams in accordance withFIGS.9and10. In terms of the manner of depiction used inFIG.3, which is an alternative toFIG.2, a pair of forming rods16, which is formed of the conductor sections La and Lb, is identifiable by a pair of cross and/or ring symbols. Such a pair of cross and/or ring symbols has identical numbers and a same leading letter. For example, the two cross symbols with the designation A4 define such a pair of forming rods16of conductor sections La, Lb. Further, the pairs of conductor rods in A3-A3, A2-A2 and A1-A1, for example, respectively form a pair of forming rods16having a first conductor section La in the receiving groove with the number 1 and a second conductor section Lb in the receiving groove with the number10. The electric connection between two conductor sections La, Lb of a pair of forming rods16, for example in terms of the pair of forming rods16between the identifiers A1, is referred to as connecting section VBa. In contrast to this, an electric junction and/or connecting section between a subsequent pair of forming rods16serially adjoining it and/or electrically connected in series, in accordance with the example in terms of the identifiers A2-A2, is defined and/or formed by the electric connecting section VBb. Here, the first connecting sections VBa located closest can be allocated to the first axial front end13aof the laminated core2and the second connecting sections VBb located closest can be allocated to the second axial front end13bof the laminated core2—FIG.1. As this can be seen fromFIG.2, or also fromFIG.3andFIGS.6,7, it is essential that the at least one electric part-winding TWa and/or each of the implemented part-windings TWa to TWx of a stator winding14and/or phase winding PW to be formed is formed at least by one first and one second electrically series-connected winding segment WSa, WSb. Here and also below, the index “x” respectively represents a counter variable. In this case, conductor sections La, Lb of the first winding segment WSa are electrically connected to each other by means of the first and second electric connecting sections VBa, VBb such that a helical current path17ais defined along a first radial direction18ato the longitudinal axis3of the laminated core2—see the lower section ofFIG.3. In addition, conductor sections La, Lb of the second winding segment WSb are electrically connected to each other by means of the first and second electric connecting sections VBa, VBb such that a second helical current path17bis defined. This second helical current path17bruns along an opposite, second radial direction18bto the longitudinal axis3of the laminated core2. Here, the first radial direction18acan be defined in terms of a radial diminishment relative to the longitudinal axis3, whereas the opposite second radial direction18bmay mean a radial enlargement relative to the longitudinal axis3of the laminated core2. Yet an inverse allocation of direction in terms of the current paths17aand17bis equally possible. Here, the second winding segment WSb is arranged laterally offset in terms of the circumferential direction10of the laminated core relative to the first winding segment WSa, in particular received in its entirety in receiving grooves4different from the first winding segment WSa and/or altogether different receiving grooves4. As can be gathered primarily fromFIG.2, the first and the second winding segment WSa, WSb, which, in interaction, are a component of the part-winding TWa and/or TWb, extend overall across two magnetic pole sections19aand19bimmediately subsequent in a circumferential direction10. Such a part-winding TWa and/or TWb thus has a pole covering of “two,” as this can most easily be seen fromFIG.2. In other words, each of the configured part-windings TWa to TWx extends across a pair of magnetic poles, i.e. across two pole sections19aand19bwhich are immediately subsequent in a circumferential direction10of the laminated core2. Also in stator windings14which have an even-numbered multiple of “two” as the number of poles (four-, six-, eight-pole etc.), the respectively-configured part-windings TWa to TWx always extend across a pair of magnetic poles, i.e. across two immediately-subsequent poles of the stator1to be formed, as this can also be gathered from the below exemplary embodiments in accordance withFIGS.9and10. As can further be seen fromFIGS.2and3, respectively, the first winding segment WSa and the second winding segment WSb have conductor sections La and Lb which are respectively immediately subsequent in terms of their helical current paths17a,17b,which conductor sections La, Lb are respectively arranged staggered relative to one another by a crossover with a crossover width of “one” in a radial direction to the longitudinal axis3of the laminated core2. In accordance with the example, the conductor section Lb with the identifier A1 is received in the layer L2, whereas the conductor section La with the identifier A1 is received in the layer L1. This pattern continues for any and all conductor sections La and Lb of the winding segment WSa and/or WSb. This also applies to the third winding segment WSc yet to be described below. As can further most easily be gathered from a combination ofFIG.2andFIG.3, it can be provided that, in order to create an unchorded stator winding14and/or to create unchorded part-windings TWa to TWx, the first and second electric connecting sections VBa, VBb per winding segment WSa, WSb have identical first expansion widths20a,20b(FIG.3above). These expansion widths20a,20bbetween the immediately-subsequent conductor sections La, Lb may also be referred to as number-of-grooves crossover width and/or frequently also as coil span. In accordance with the example, the expansion widths20a,20bof the first and second winding segment WSa, WSb respectively amount to ten receiving grooves. In particular, the expansion width20a,20bin accordance with the example between the electrically serially-connected conductor sections La, Lb spaced apart from one another in a circumferential direction10amounts to10, in this case. However, depending on the dimensioning and/or design of the stator1, also expansion widths20a,20bdeviating from10are possible. The part-winding TWa depicted in an exemplary manner inFIGS.2,3further comprises a third electric connecting section VBc, which builds an electric connection between the first and the second winding segment WSa, WSb. In comparison to the first and second electric connecting section VBa, VBb, this third electric connecting section VBc has a larger, second expansion width20c.In accordance with the example, this second expansion width20camounts to 11, in particular the third electric connecting section VBc expands between eleven immediately-subsequent receiving grooves. In particular, this means a relatively larger number of grooves and/or larger crossover than in the electric connecting sections VBa and VBb. As can further most easily be gathered fromFIGS.2,3, the at least one part-winding TWa and/or each of the existing part-windings TWa to TWx of a stator winding14may comprise a third winding segment WSc. This creates a number of fractional slots of q=3. Here, this third winding segment WSc is electrically series-connected to the second winding segment WSb. What is more, this third winding segment WSc, for the most part, is structurally superimposed on the first winding segment WSa. In particular, the third winding segment WSc is interleaved and/or interwoven with the first winding segment WSa, as this can be seen by means of the depictions fromFIGS.2and3, and also by means of the depiction inFIG.7. In particular, it is provided here that a third winding axis WAc of the third winding segment WSc is laterally offset in a circumferential direction10of the laminated core2by at least one and/or at least two to a maximum of six immediately-subsequent receiving grooves4relative to a first winding axis WAa of the first winding segment WSa. The third winding segment WSc, for the most part, is thus superimposed on the first winding segment WSa and/or quasi-intertwined with same in terms of its base and/or active area. A considerably smaller proportion of the surface and/or active area of the third winding segment WSc is superimposed on the second winding segment WSb and/or intertwined with same, as this can most easily be gathered from the depiction in accordance withFIG.7. A fourth electric connecting section VBd is configured to electrically connect the second winding segment WSb and the third winding segment WSc. In comparison to the first and second electric connecting sections VBa, VBb, this fourth electric connecting section VBd has a smaller, third expansion width20cin accordance with the example. In accordance with the example, this relatively smaller and/or shorter third expansion width20dis configured over a number-of-grooves crossover width of “8.” Alternatively, this fourth electric connecting section VBd may also have a relatively larger and/or longer expansion width20d.This occurs whenever the beginnings of the winding and ends of the winding of the respective part-windings TWa to TWc are selected different, in particular diametrically opposed. Here, such a shortened and/or extended, fourth electric connecting section VBd is configured exclusively if an integer number of fractional slots is larger than two and/or if three winding segments WSa, WSb and WSc serially-connected with one another are configured. In particular, it is expedient and/or characteristic that the number of the winding segments WSa, WSb, WSc, WSx is identical to the number of fractional slots q of the stator1and/or of the stator winding14. Here, the number of fractional slots is to be understood to mean the number of the receiving grooves4per pole section19aor19band per phase winding PW and/or per phase of the stator winding14. Correspondingly, the number of fractional slots amounts to q=2 if there are two winding segments WSa, WSb. In another exemplary embodiment, the number of fractional slots may amount to q=4 and four winding segments WSa, WSb, WSc, WSd are configured here. The relevant relationships may also be depicted as follows in the form of a table: Number of windingNumber of thirdNumber of fourthNumbersegments WSa toelectric connectingelectric connectingofWSx per pole pairsections (VBc) persections (VBd) perfractionaland per part-winding segmentwinding segmentslots qwindingand per pole pairand per pole pair221—33at least 11442at least 155at least 22663at least 2 It must thus also be said that the at least one part-winding TWa of the stator winding14and/or of a phase winding PW comprises a first winding segment WSa, wherein the current path17adefined by the first winding segment WSa either (i), starting from the radially-innermost layer L1 (side of the air gap) of conductor sections La, Lb, leads to the radially-outermost layer L8 (yoke side) of conductor sections La, Lb, or vice versa (ii), starting from the radially-outermost layer L8 (yoke side) of conductor sections La, Lb, leads to the radially-innermost layer L1 (side of the air gap) of conductor sections La, Lb, and is subsequently guided, without an offset in the layer L, i.e. without crossover and/or without a change in the layer L, by means of the third electric connecting section VBc, which may also be referred to as an extended and/or relatively longer connecting section, to the second winding segment WSb. The current path17bin the second winding segment WSb is thereafter (i), starting from the radially-outermost layer L8 (yoke side) of conductor sections La, Lb, guided to the radially-innermost layer L1 (side of the air gap) of conductor sections La, Lb, or vice versa (ii), starting from the radially-innermost layer L1 (side of the air gap) of conductor sections La, Lb, guided to the radially-outermost layer L8 (yoke side) of conductor sections La, Lb. In addition, it can be gathered fromFIGS.2and3, respectively, that the first winding segment WSa and the second winding segment WSb, starting from their respective beginnings of the winding21(A1) and22(A5), in a direction toward their respective ends of the winding23(A4) and24(A8), are respectively wound in the same direction, for example respectively clockwise. Alternatively, an anti-clockwise implementation running in the same direction and/or an implementation running respectively uniformly anti-clockwise would also be possible. To form a desired stator winding14and/or phase winding PW, it may be expedient that any and all winding segments WSa, WSb, WSc, WSx of a multiply-segmented part-winding TWa to TWx are electrically connected in series and, starting from their respective beginning of the winding21,22,25, in a direction toward their respective end of the winding23,24,26, are respectively wound in the same direction, for example respectively wound and/or running clockwise. As can most easily be seen fromFIG.2, the second part-winding TWb has a double-mirror-image structure and/or course relative to the first part-winding TWa. In this exemplary embodiment, the second part-winding TWb is laterally offset in a circumferential direction10by a single receiving groove4relative to the first part-winding TWa. There is thus a number-of-grooves crossover width of “one” between the part-windings TWa and TWb, which are physically arranged in parallel and interleaved with one another, as this can be seen fromFIG.2. Here, the imaginary first axis of symmetry is defined by the longitudinal axis3of the laminated core2. The imaginary second axis of symmetry runs along the circumferential direction10of the laminated core2. The first and the second part-winding TWa and TWb form a matching and/or corresponding pair of part-windings and extend across exactly one pole pair of the laminated core2, in particular across two subsequent pole sections19aand19b. A part-winding TWa, TWb, TWc and/or TWa to TWx may also be defined by the fact that the voltage induced by it, which is a vector quantity, is identical in value and phase to the induced voltages of the other part-windings within the phase winding PW. Accordingly, its matching conductors of the same receiving groove respectively take the different amplitudes of the induced voltages into account. The stator winding14and/or its part-windings TWa to TWx may be constructed by forming-rod conductors and/or conductor sections La, Lb, which are configured as so-called hairpins—FIGS.4a,FIG.4b—or as so-called I-pins—FIG.5. In a hairpin implementation in accordance withFIGS.4a,4b, respectively two conductor sections La, Lb which are immediately subsequent in the electric current paths17a,17b,17c(FIG.3) and respectively one first connecting section VBa respectively electrically connecting these conductor sections La, Lb are formed as one piece. Yet, as illustrated inFIG.5, also so-called I-pins can be used, whose middle sections respectively constitute the conductor sections La, Lb and whose first end sections, which are formed and preferably welded with one another, define the first electric connecting section VBa. The second end sections of the conductor sections La, Lb, located opposite the first end sections, define the second connecting sections VBb for the electric connection to a further I-or hairpin to be serially connected. In accordance withFIG.4a, the second connecting sections VBb may be formed such that they are aligned facing each other. Yet, in accordance withFIG.4b, the second connecting sections VBb may also be formed such that they run parallel to each other and/or point in uniform directions. The rectangles at the broad lines and/or in the middle section of the respective arrows depicted inFIG.2and inFIGS.9,10marked with dotted hatching symbolize contact points, in particular welded points, between two second electric connecting sections VBb of electrically serially-connected, essentially congruently-positioned pairs of forming rods16, as they are schematically illustrated inFIG.4a.FIG.2andFIGS.9,10thus show the use of hairpins and/or of one-piece, essentially U-shaped pairs of forming rods16. It may be expedient if each of the winding segments WSa, WSb and WSc and/or WSa to WSx extends across more than two, i.e. at least across three, immediately-adjacent receiving grooves4in terms of the circumferential direction10of the laminated core2. The first and the second expansion widths20aand20bof the first and second electric connecting sections VBa and VBb are thus larger or equal to the three-fold transverse and/or lateral distance between two immediately-adjacent receiving grooves4of the laminated core2. In the embodiment according toFIG.2, the number-of-grooves crossover widths of the connecting sections VBa and VBb and/or of the immediately-subsequent conductor sections La and Lb amount to “nine,” for example. FIG.6illustrates an exemplary embodiment of a part-winding of a stator winding14, for example a part-winding TWa. This basic structure repeats itself in a corresponding manner for the part-windings TWb to TWx. The depicted part-winding TWa is configured using forming-rod technique and comprises a plurality of electrically serially-connected hairpins. The winding diagram shown results in a number of fractional slots q=2, since the number of the immediately-subsequent receiving grooves4per magnetic pole and phase is “2.” In comparison to this, the so-called number of fractional slots q=3 in the part-winding TWa according toFIG.7. The winding segments WSa and WSb of the part-winding TWa according toFIG.6and also the winding segments WSa, WSb and WSc of the part-winding TWa according toFIG.7are respectively formed by loop windings15configured using forming-rod technique. The winding segments WSa and WSb (FIG.6) and/or the winding segments WSa, WSb and WSc (FIG.7) are respectively electrically connected in series. As can most easily be seen fromFIG.7, the third winding segment WSc is arranged interleaved relative to the first winding segment WSa, in particular, for the most part, positioned super-imposing relative to the first winding segment WSa. Its winding axes WAa and WAc are staggered relative to each other in a circumferential direction10by a number of receiving grooves4corresponding to the number of fractional slots q-1, laterally offset by two receiving grooves4in accordance with the example. According to an expedient measure as it can most easily be seen inFIGS.6and7, it may also be provided to provide the first connecting sections VBa of the individual hairpins and/or pairs of forming rods16in their middle section with an essentially S- or Z-shaped forming section28in terms of a plane27running perpendicular to the longitudinal center axis of the laminated core. This ensures that the connecting sections VBa are shaped such that a crossover by “one” in terms of immediately-subsequent conduction sections La and Lb of a hairpin are implementable in an orderly manner The winding segments WSa, WSb and/or WSc respectively configured as loop windings15inFIGS.6and7are characterized, among other things, by the fact that these winding segments WSa, WSb and/or WSc are respectively formed by a plurality of pairs of forming rods16arranged in series along the winding axis WAa, WAb and/or WAc and electrically connected in series. FIG.8shows how the part-winding TWa in accordance withFIG.7can be inserted in a laminated core2in order to thus form a partial component of a stator1. Merely for the sake of better clarity, merely half of a stator1has been illustrated here. In accordance with the example, the laminated core2in accordance withFIG.8has a total of72receiving grooves4and is provided with eight magnetic poles for implementing a stator winding with a total of three phases. FIG.9illustrates, among other things, the winding diagram of the part-winding TWa, which part-winding TWa is then depicted physically-implemented inFIGS.7and8.FIGS.7and8illustrate, among other things, a part-winding TWa, which constitutes a partial component of a stator winding to be constructed. The winding diagram shown inFIG.9is configured for provisioning four magnetic poles and designed for a laminated core2with36receiving grooves4. It should be noted merely for the sake of completeness that the winding diagram in accordance withFIG.9shows only one single phase winding PW of a stator winding14to be produced. This phase winding PW comprises four part-windings TWa, TWb, TWc, TWd, which extend over a total of 360° of the circumferential direction10of a laminated core2with a total of36receiving grooves4. These four part-windings TWa, TWb, TWc, TWd result in a four-pole stator winding14, i.e. a stator1with a number of pole pairs of “two.” The structure of the part-winding TWc is identical here to the structure of the part-winding TWa. This analogously applies to the part-windings TWb and TWd. The part-windings TWa and TWc and/or TWb and TWd, however, are arranged staggered relative to one another and/or spaced apart in terms of the circumferential direction10. The extent of this lateral offset depends on the desired number of pole pairs and number of grooves of the stator1to be produced. FIG.10shows another, and, if applicable, independent, embodiment of a stator winding14, wherein, once again, equal parts are provided with the same reference numbers and/or the same component designations as in the preceding figures. To avoid unnecessary repetition, the detailed description in the preceding figures should be noted and/or is made reference to. In particular,FIG.10shows a winding diagram for forming a chorded stator winding14, wherein here, too, only one single phase winding PW of a three-phase, four-pole stator1is depicted. The winding diagram at issue can therefore also be used to form chorded stator windings14and/or chorded part-windings TWa to TWx. To form chorded part-windings TWa, TWb, TWc, TWd, the first and second electric connecting sections VBa, VBb in each of their winding segments WS a, WSb, WSc have first expansion widths20aand20bbetween conductor sections La, Lb spaced apart from one another and electrically connected in series in a circumferential direction10which first expansion widths20aand20bare different to one another. In particular, in the exemplary embodiment shown, number-of-grooves crossover widths of15are provided in terms of the first connecting sections VBa, and number-of-grooves crossover widths of13are provided in terms of the second connecting sections VBb. Here, it is also characteristic that the number of shortened or—as elucidated below—extended second connecting sections VBb in each of the part-windings TWa, TWb, TWc, TWd, which shortened and/or extended second connecting sections VBb are configured shorter and/or longer in comparison to the first connecting sections VBa, is identical to the number of fractional slots q of the stator1. In particular, the number of shortened and/or extended second connecting sections VBb is identical to the number of the receiving grooves4per magnetic pole section19aor19b,per phase winding PW and per part-winding TWa, TWb, TWc or TWd. It must be said that in a chorded stator winding14and/or in chorded part-windings TWa to TWx, the lateral offset and/or crossover in terms of the receiving grooves4can be set optionally. InFIG.10, the cells marked with cross-hatching cross over to the “right,” but they could also cross over to the “left.” The cells colored in gray inFIG.10show stairs starting on the right. Yet, the chording of the part-windings TWa to TWx can alternatively also be done using an extended winding step. The cells marked with cross-hatching would then show stairs starting on the left. Therefore, as an alternative to the diagram according toFIG.10, also extended connecting sections VBb are possible. It should be said in summary that the winding diagram is characterized by the fact that one crossover is carried out per U-shaped pair of forming rods16and/or per hairpin and that a complete parallel branch and/or a complete part-winding TWa to TWx per phase winding PW covers at least one and/or exactly one pole pair19aand19bin a circumferential direction of the laminated core2. In an unchorded stator winding14, the U-shaped pairs of forming rods16in it have the same step width. The ends of the pairs of forming rods16radially connected in series and in a zigzag shape result in winding segments WSa to WSx which comprise, on the side of the air gap5or side of the groove base7(FIG.1) and in the relevant uppermost and lowermost layer (layer facing the stator yoke and layer facing the air gap), at least one shortened and one extended winding step with a shortened expansion width20dand/or extended expansion width20c.In this case, the uppermost or also the lower layer may contain the shortened winding step or vice versa. The winding diagram is therefore characterized by the fact that, upon implementing a chorded winding, accordingly-shortened winding steps with at least the same number as the number of fractional slots q occur. The number of fractional slots q is the number of the receiving grooves per phase, per magnetic pole of the winding and per part-winding. The winding diagram is further characterized by the fact that the U-shaped pairs of forming rods (hairpins) and/or the current paths configured with them which pertain to a parallel branch (part-winding) and are interconnected in series lead at least from an outer (uppermost or lowermost) layer to the radially opposite innermost (lowermost or uppermost) layer—expediently in the zigzag shape mentioned above—then lead to the magnetic pole adjacent in a circumferential direction with a shortened or extended winding step, without a crossover, in order to lead, from there, once again, yet in the opposite direction, from the outer layer to the opposite layer. The above-mentioned implementation with U-shaped pairs of forming rods16(hairpins) is equally feasible and/or implementable using I-shaped forming rods (I-pins). An unchorded winding is to be understood, as a rule, to mean windings and/or coils in which the coil span which refers to the circumferential direction10of the laminated core2is equal to the pole pitch. Pole pitch is understood to mean the axial distance between subsequent magnetic poles. Coil span is to be understood to mean the previously-mentioned expansion widths20aand20b.In contrast to this, the coil span is smaller (or also larger) than the pole pitch in chorded windings and/or coils. FIG.11shows another advantageous exemplary embodiment of the winding diagram. Here, equal reference numbers are used for parts already described hereinbefore, and the preceding parts of the description are analogously transferable to equal parts with equal reference numbers. The advantageous technical effects of an embodiment according toFIG.11can here be clarified in comparison to the exemplary embodiment in accordance withFIG.2. In the exemplary embodiment according toFIG.2, the voltage difference ΔU (delta U) between two immediately-abutting, i.e. directly-adjacent, conductor sections La and/or Lb within a receiving groove4can amount to up to the full phase and/or outer conductor voltage UL1-L2 (FIG.13—arrows depicted in dashed lines) if, for example, the part-windings TWa and TWb are electrically connected in parallel and form a winding phase of a delta-connected stator winding14, as this is schematically shown inFIG.13. In particular, in the embodiment according toFIG.2, the entire phase and/or outer conductor voltage UL1-L2 is applied, for example, between the immediately-adjacent conductor sections La with the designation A1 and B12 in the receiving groove4with the number 1 (for the phase U). The same applies, for example, in terms of the directly-contiguous conductor sections La with the designation B1 and A12 in the receiving groove4with the number 3 for the electric phase U, as this can also be seen by means of a combination withFIG.13(arrows depicted in dashed lines). The insulation layer9at the lateral surfaces of the conductor sections La and/or Lb must thus be designed for this relatively high difference in electric potential as it occurs in the embodiment according toFIG.2. For example, a correspondingly great thickness of the insulation layer9—FIG.1—at the lateral surfaces of the conductor sections La and/or Lb is to be provided. In contrast to this, in the advantageous embodiment according toFIG.11, the maximum voltage difference ΔU occurring between immediately-adjacent conductor sections La and/or Lb (see the upper section ofFIG.11) can be reduced and/or lowered considerably. In particular, in the embodiment according toFIG.11, which, analogously toFIG.2, relates to a two-pole stator1to be supplied using three phases and having a total of eight layers L1-L8, the maximum voltage difference AU occurring between immediately-adjacent conductor sections La or Lb can be reduced to about a third (approx. 33%) of the phase and/or outer conductor voltage UL1-L2—see, for example, the immediately-adjacent conductor sections La and/or Lb with the designation A1 and B4 in the receiving groove4with the number 1 (for the phase U) inFIG.11in combination withFIG.13. In particular, this ensures that the physical stresses on the insulation layer9can be reduced and/or the requirements for the electro-technical insulation properties of the conductor sections La, Lb be advantageously lowered. This ultimately enables the thickness of the insulation layer to be reduced and consequently the space factor of electrically-conductive material in the receiving grooves4to be increased. This advantageous reduction of the maximum difference in potential AU occurring between immediately-adjacent and/or gap-free, contiguous conductor sections La and/or Lb can be achieved by ensuring that a beginning of the winding21aof the first part-winding TWa and a beginning of the winding21bof another part-winding TWb, TWc, TWd, which is electrically connectable or connected in series or electrically connectable or connected in parallel to the first part-winding TWa, are positioned such that these beginnings of the winding21aand21b(FIGS.11,12) are arranged in receiving grooves4which belong to the same phase zone Pz. In this case, the receiving grooves4and/or all conductor sections La or Lb arranged therein which are allocated to the same phase U, V or W and belong to the same magnetic pole and/or pole section19aor19bof the stator1and are thus carrying current in the same direction in the operating mode of the stator1, count among a phase zone Pz. In the depictions inFIGS.2and9-12, the phase zones Pz for the phase U can be seen by the areas and/or receiving grooves4with cross-hatching. It may also be expedient if a beginning of the winding21aof the first part-winding TWa is positioned in the radially-innermost layer L1 and a beginning of the winding21bof the other part-winding TWb, which is to be electrically connected in series or electrically in parallel with the first part-winding TWa, is positioned in the radially-outermost layer L8, as this is illustrated inFIGS.11,12. Yet, an inverse allocation is also possible, in which the beginning of the winding21ais allocated to the radially-outermost layer L8 and the beginning of the winding21bis allocated to the radially-innermost layer L1 of the stator winding14. Therefore, the beginnings of the winding21a,21bare formed by two part-windings TWa, TWb to be electrically connected in series or electrically connected in parallel on the one hand in the radially-innermost layer L1 and on the other hand in the radially-outermost layer L8 of the stator winding14. Primarily whenever the part-windings TWa-TWx of the stator winding14are configured as unchorded windings (FIGS.2,3,9,11,12), it may be expedient if a beginning of the winding21aof the first part-winding TWa and a beginning of the winding21bof another part-winding TWb-TWx, which is electrically connectable or connected in series or electrically connectable or connected in parallel to the first part-winding TWa, are positioned and/or arranged in the same receiving groove4. As can be seen fromFIG.11in accordance with the example, the beginning of the winding21aof the part-winding TWa (Al) and the beginning of the winding21bof the part-winding TWb (B1) are selected such that they are allocated to the same receiving groove4, in accordance with the example are assigned to the groove with the number1. Analogously to the regime regarding the beginnings of the winding21a,21ba winding end26a(A12) of the first part-winding TWa and a winding end26b(B12) of another part-winding TWb-TWx connected in series or in parallel are thereafter positioned in the same, i.e. identical, receiving groove4, as this can be seen inFIGS.11and12by means of the receiving groove4with the number 12 (for the phase U). In terms of the exemplary embodiments in accordance withFIGS.11andFIG.12, in which the maximum voltage difference AU occurring between immediately-adjacent conductor sections La and/or Lb is advantageously low, it is further characteristic that, in receiving grooves4with beginnings of the winding21a,21bconfigured there (see the fields A1, B1 and/or C1, D1), or with ends of the winding26a,26bconfigured there (see the fields A12, B12 and/or C12, D12), respectively only beginnings of the winding21a,21b(A1, B1; A1, D1; B1, C1) or respectively only ends of the winding26a,26b(A12, B12; C12, D12) are configured. Therefore, in the receiving grooves4in which power connections to the electric energy supply of the stator winding14are provided, either only beginnings of the winding21a,21bor only ends of the winding26a,26bare configured, as this can be seen inFIG.11andFIG.12. In particular , a generically-identical sorting and/or grouping of the beginnings of the winding21a,21and the ends of the winding26a,26bin terms of respectively-used receiving grooves4of the laminated core2is provided. FIG.12shows a winding diagram and/or an embodiment of a stator1with four part-windings TWa-TWd and four magnetic poles, i.e. 2 pole pairs. The structural measures and/or rules described hereinbefore are analogously applicable to this embodiment. Also in this case, the respective beginnings of the winding21bof the at least one other part-winding TWb-TWx within the phase zone Pz are selected such that a marked reduction of the electric differences in potential ΔU between two adjacent conductor sections within the respective receiving grooves4occurs. Depending on the winding diagram, this reduction of the voltage load on the insulation layer9may here amount to 50%, or more than 50%, in particular around 66%, as this can be gathered fromFIG.13. According to a workable embodiment, also a mixed use of hairpins shown in an exemplary manner inFIG.4and of I-pins shown in an exemplary manner inFIG.5may be provided to construct the stator winding14. In particular, it may be provided that the first and the last conductor section La, Lb of at least one of the part-windings TWa-TWx, in particular of all part-windings TWa-TWx of the stator winding14, are respectively formed by an I-pin, in particular by I-shaped forming rods. All conductor sections La, Lb arranged in between, in contrast, are configured as electrically series-connected hairpins, in particular as U-shaped pairs of forming rods16. The winding diagram and/or the specified stator winding14presented and the stator1constructed on its basis have been described and illustrated in connection with an internal rotor and/or internal-rotor motor. Yet, this winding technology may also be applied for an internal stator and an external rotor (external-rotor motor). The exemplary embodiments show possible embodiment variants, and it should be noted in this respect that the invention is not restricted to these particular depicted embodiment variants of it, but that rather also various combinations of the individual embodiment variants with one another are possible and this possibility of variants based on the technical teaching by means of the invention at issue lies within the ability of the person skilled in the art in this technical field. The scope of protection is determined by the claims. However, the description and the drawings are to be adduced for construing the claims. Individual features or feature combinations from the different exemplary embodiments shown and described may represent independent inventive solutions. The object underlying the independent inventive solutions may be gathered from the description. Any and all specifications of value ranges in the description at issue are to be understood to comprise any and all sub-ranges of same, for example the specification 1 to 10 is to be understood to mean that any and all sub-ranges starting from the lower limit 1 and from the upper limit 10 are comprised therein, i.e. any and all sub-ranges start at a lower limit of 1 or larger and end at an upper limit of 10 or less, e.g. 1 to 1.7, or 3.2 to 8.1, or 5.5 to 10. Finally, as a matter of form, it should be noted that for ease of understanding of the structure, elements are partially not depicted to scale and/or are enlarged and/or are reduced in size. LIST OF REFERENCE NUMBERS 1stator PW phase winding 2laminated core TWa, TWb part-windings 3longitudinal axis TWc, TWd part-windings 4groove WSa first winding segment 5air gap WSb second winding segment 6tip of a tooth WSc third winding segment 7groove base L layers 8electric conductor La, Lb conductor sections 9insulation layer VBa, VBb electric connecting sections 10circumferential direction VBc, VBc electric connecting sections 11longitudinal direction WAa first winding axis 12radial direction WAb second winding axis 13a,13baxial front ends WAc third winding axis 14stator winding Pz phase zone 15loop winding 16pair of forming rods 17a,17bcurrent paths 17ccurrent path 18a,18bradial directions 19a,19bpole sections 20bfirst expansion widths 20csecond expansion width 20dthird expansion width 21;21a,21bbeginning of the winding 22beginning of the winding 23end of the winding 24end of the winding 25beginning of the winding 26;26a,26bend of the winding 27plane 28forming section | 50,638 |
11863036 | MODES FOR CARRYING OUT THE DISCLOSURE Hereinafter, an embodiment of the present disclosure will be described with reference to the drawings. First Embodiment [Structure of Stator] The structure of a stator100according to the first embodiment will be described with reference toFIGS.1to21. The stator100has an annular shape centered around a central axis C1. The stator100is an example of an “armature” in the claims. In the specification of the application, an “axial direction (central axis direction, axis direction)” means a direction (Z direction) along the central axis C1of the stator100(a rotational axis of a rotor101) as shown inFIG.1. A “circumferential direction” means a circumferential direction (A1direction, A2direction) of the stator100. A “radial direction” means a radial direction (R direction) of the stator100. An “inner radial side” means a direction (R1direction) toward the central axis C1of the stator100along the radial direction. Further, an “outer radial side” means a direction (R2direction) toward the outside of the stator100along the radial direction. The stator100configures a part of a rotary electric machine102together with the rotor101. The rotary electric machine102is configured as a motor, a generator, or a motor/generator, for example. As shown inFIG.1, the stator100is disposed on the outer radial side of the rotor101in which a permanent magnet (not shown) is provided. That is, in the first embodiment, the stator100configures a part of the inner rotor type rotary electric machine102. The radial direction is an example of a “joining direction” in the claims. As shown inFIG.2, the stator100includes a stator core10, a first insulating member20, and a coil portion30. The coil portion30includes a first coil assembly30a(non-lead side coil) and a second coil assembly30b(lead side coil). Further, the coil portion30is composed of a plurality of segment conductors40(seeFIG.4). In addition, in the first embodiment, the stator100includes a second insulating member21(seeFIG.4) that is provided separately from the first insulating member20. The stator core10is an example of an “armature core” in the claims. The second insulating member21is an example of a “joint portion insulating member” in the claims. (Structure of Stator Core) The stator core10has a cylindrical shape with the central axis C1(seeFIG.1) as the central axis. Further, the stator core10is formed, for example, by stacking a plurality of electromagnetic steel plates (for example, silicon steel plates) in the axial direction. As shown inFIG.3, the stator core10is provided with a back yoke11having an annular shape when viewed in the axial direction, and a plurality of slots12that is provided on the inner radial side of the back yoke11and that extends in the axial direction. The stator core10is provided with a plurality of teeth13on both sides of each slot12in the circumferential direction. Each slot12is a part surrounded by a wall portion11aof the back yoke11provided on the outer radial side and circumferential side surfaces13aof the two teeth13. The slot12is provided with an opening portion12athat opens to the inner radial side. The slot12opens on both sides in the axial direction. The teeth13are formed so as to protrude radially inward from the back yoke11, and a protruding portion13bconfiguring an opening portion12aof the slot12is formed on a tip end portion on the inner radial side. The opening portion12ahas an opening width W1in the circumferential direction. Here, the opening width W1corresponds to the distance between the tip end portions of the protruding portions13bof the teeth13. A width W2of a part of the slot12in which the coil portion30is disposed is larger than the opening width W1. That is, the slot12is configured as a semi-open type slot. Here, the width W2corresponds to the distance between the circumferential side surfaces13aof the teeth13disposed on both sides of the slot12in the circumferential direction. The width W2of the slot12is substantially constant in the radial direction. (Structure of Coil Portion) As shown inFIG.4, the coil portion30is configured of a flat conductor wire. For example, the coil portion30is made of copper or aluminum. As shown inFIG.2, the coil portion30is formed by the first coil assembly30aprovided on one axial side (arrow Z2direction side) and the second coil assembly30bprovided on the other axial side (arrow Z1direction side) being combined in the axial direction and joined. The first coil assembly30aand the second coil assembly30bare each formed in an annular shape centered around the same central axis C1(seeFIG.1) as the stator core10. As shown inFIG.4, in the first embodiment, the coil portion30is formed by joining in a joint portion90, a first leg portion71and a second leg portion81, described below, of the segment conductors40. The coil portion30is configured as a wave winding coil, for example. Moreover, the coil portion30is configured as a coil of eight turns. That is, the coil portion30is configured such that eight segment conductors40are disposed in parallel in the slot12in the radial direction. <Configuration of Wiring Connection of Coil Portion> As shown inFIG.5, the coil portion30is configured to generate magnetic flux by being supplied with three-phase alternating current power from a power supply unit (not shown). Specifically, the coil portions30are connected (wired) by three-phase Y-connection. That is, the coil portion30includes a U-phase coil portion30U, a V-phase coil portion30V, and a W-phase coil portion30W. The coil portion30is provided with a plurality of (for example, two) neutral points N. Specifically, the coil portion30is connected in four parallel lines (star connection). That is, the U-phase coil portion30U is provided with four neutral point connection end portions NtU and four power line connection end portions PtU. The V-phase coil portion30V is provided with four neutral point connection end portions NtV and four power line connection end portions PtV. The W-phase coil portion30W is provided with four neutral point connection end portions NtW and four power line connection end portions PtW. In the following description, when the U-phase, the V-phase, and the W-phase are not particularly distinguished for the neutral point connection end portion and the power line connection end portion, the neutral point connection end portion and the power line connection end portion are simply indicated as a “neutral point connection end portion Nt” and a “power line connection end portion Pt”. <Configuration of Coil Assembly> As shown inFIG.2, the first coil assembly30ais configured of a plurality of first segment conductors70(hereinafter, referred to as “first conductors70”) serving as the segment conductors40. It is preferable that the first coil assembly30abe configured by combining only the first conductors70. The second coil assembly30bincludes a plurality of (for example, three) power segment conductors50(hereinafter, referred to as “power conductors50”) serving as the segment conductors40, and a plurality of (for example, two) neutral-point segment conductors60(hereinafter referred to as “neutral-point conductors60”) serving as the segment conductors40, and second segment conductors80(hereinafter, referred to as “second conductors80”) that are conductors (general segment conductors40) different from the power conductors50and the neutral-point conductors60among the segment conductors40and that configure the coil portion30. That is, all of the power conductors50and the neutral point conductors60provided in the stator100are provided in the second coil assembly30b. (Configuration of Segment Conductor) As shown inFIG.6, the segment conductor40is configured as a flat conductor wire having a substantially rectangular cross section. An insulating coating40ahaving a thickness t1is provided on a conductor surface40bof the segment conductor40. The thickness t1of the insulating coating40ais set, for example, to such an extent that interphase insulating performance (insulation between the first coil end portions72and insulation between the second coil end portions82, seeFIG.2) can be ensured. Note that, inFIG.6, the size relationship such as the thickness is highlighted for the sake of explanation, and the present disclosure is not limited to this example indicated in the drawing. <Structure of First Conductor and Second Conductor> As shown inFIGS.7and8the segment conductors40include the first conductors70disposed on one axial side (Z2direction side) of the stator core10and the second conductors80that are disposed on the other axial side (Z1direction side) of the stator core10and that face the first conductors70in the axial direction. That is, the coil portion30is formed by joining the first conductors70and the second conductors80, which are divided into two in the axial direction. Here, the second conductors80are the segment conductors40other than the power conductors50and the neutral point conductors60among the segment conductors40that configure the second coil assembly30b. In the first embodiment, each first conductor70includes the first leg portion71which extends in the axial direction and which has a length L1in the axial direction. The first leg portion71extends to the other side (Z1direction side) in the axial direction. Each second conductor80includes the second leg portion81that is disposed on the Z1direction side of the first leg portion71, that extends in the axial direction, and that has a length L2that is greater than the length L1in the axial direction. The second leg portion81extends to one side (Z2direction side) in the axial direction. In the first embodiment, as shown inFIGS.7A and7B, the first conductors70are formed so as to have a U-shape (substantially U-shape) when viewed in the radial direction by connecting a pair of the first leg portions71in which the first leg portions71are disposed in the slots12different from each other. The coil pitch of the first conductors70is six. That is, the first leg portions71of the pair of first leg portions71are disposed at positions different in the circumferential direction by six slots12. That is, five slots12are provided between the slot12in which one first leg portion71of the pair of first leg portions71is disposed and the slot12in which the other first leg portion71of the pair of first leg portions71is disposed. Specifically, each first conductor70includes the pair of the first leg portions71that are disposed in different slots12and that are each linearly formed along the axial direction, and a first coil end portion72. The first leg portion71means a part disposed in the slot12from the axial position of the end face10a(seeFIG.2) in the axial direction of the stator core10, and the first coil end portion72means a part that is formed to be continuous with the first leg portion71and that is disposed on the outer axial side of the end face10aof the stator core10. The first coil end portion72has a bent shape that bends in the axial direction. Further, the first coil end portion72has a first crank part73formed in a crank shape in which the first crank part73is bent in a stepwise manner by the width of one segment conductor40in the radial direction when viewed in the axial direction. That is, the radial width of the first crank part73is twice the width of one segment conductor40. The end face10ais an example of a “one side end face” in the claims. Further, the lengths L1of the pair of first leg portions71in the axial direction are substantially equal to each other. The length L1in the axial direction means the length of the part of the first conductor70that extends linearly in the axial direction within the slot12. The length L1in the axial direction is smaller than a length L3of the stator core10in the axial direction (seeFIG.2). The length L3of the stator core10in the axial direction means an axial distance (interval) between the end face10aand the end face10bin the axial direction. Further, the end face10bis an example of the “other end face” in the claims. Similarly, as shown inFIGS.8A and8B, the second conductor80includes a pair of the second leg portions81disposed in the slot12and the second coil end portion82. The second coil end portion82also has a second crank part83. In the first embodiment, the second conductor80is formed to have a U-shape by connecting the pair of second leg portions81, which is disposed in the different slots12, to each other. The axial lengths L2of the pair of second leg portions81of the second conductor80are substantially equal to each other. Further, the axial length L2of the pair of second leg portions81of the second conductor80is larger than the axial length L1of the pair of first leg portions71of the first conductor70(L2>L1). The axial length L2means the length of the part of the second conductor80that extends linearly in the axial direction within the slot12. <Configuration of Power Conductor> As shown inFIG.9, in the power conductor50, a plurality (for example, four) of the power line connection end portions Pt of the same phase are electrically connected to each other, and a plurality of the connected power line connection end portions Pt and one power terminal member51are electrically connected. In the power conductor50, the second leg portion81joined to one of the pair of first leg portions71(seeFIG.12) and the power terminal member51are joined. The power conductor50has a function of introducing electric power into the coil portion30from the power supply unit (not shown). Specifically, the power conductor50includes an outer radial side power conductor52that is disposed on the outer radial side of the slot12(seeFIG.1) and that has the power line connection end portion Pt, and an inner radial side power conductor53that is disposed on the inner radial side and the outer axial side of the outer radial side power conductor52and that has the power line connection end portion Pt. In other words, the power conductor50is formed in a bifurcated shape. The outer radial side power conductor52and the power terminal member51are electrically connected by a lead wire54. The inner radial side power conductor53and the power terminal member51are electrically connected to each other by the lead wire54. The outer radial side power conductor52and the inner radial side power conductor53are electrically connected via the power terminal member51and the lead wire54. The lead wire54is formed of a stranded wire (conductor) and an insulating tube51ais disposed on the outer circumference, for example. The outer radial side power conductor52and the inner radial side power conductor53are each provided with the second leg portion81but are not provided with the first coil end portion72or the second coil end portion82. Further, in the outer radial side power conductor52and the inner radial side power conductor53, the lead wire54and the second leg portion81are joined via a conductor plate55. For example, the joining is performed by brazing or welding (for example, any one of resistance welding, arc welding, laser welding, or high energy beam welding). <Structure of Neutral Point Conductor> As shown inFIG.1, the neutral point conductor60includes an outer radial side neutral point conductor61and an inner radial side neutral point conductor62. As shown inFIG.5, the outer radial side neutral point conductor61and the inner radial side neutral point conductor62each include the neutral point N, and the neutral point connection end portion NtU of the U-phase coil portion30U, the neutral point connection end portion NtV of the V-phase coil portion30V, and the neutral point connection end portion NtW of the W-phase coil portion30W are electrically connected. As shown inFIG.10, each outer radial side neutral point conductor61includes two U-phase W-phase neutral point segment conductors61aand two V-phase neutral point segment conductors61b. The U-phase W-phase neutral point segment conductors61ainclude the U-phase second leg portions81connected to the first leg portions71of the first conductors70for the U-phase among the three-phase alternating current, the W-phase second leg portions81connected to the W-phase first leg portions71, and two neutral point coil end portions61cthat each connect the U-phase second leg portion81and the W-phase second leg portion81. The neutral point coil end portion61cis formed to be continuous with the U-phase second leg portion81and is formed to be continuous with the W-phase second leg portion81. The U-phase W-phase neutral point segment conductor61ais formed to have a substantially U-shape (substantially U-shape) when viewed from the inner radial side. The V-phase neutral point segment conductor61bis formed in a substantially linear shape when viewed from the inner radial side. As shown inFIG.1, the neutral point coil end portion61cis formed along the circumferential direction on the outer radial side of the second coil end portion82of the second conductor80. The neutral point coil end portion61cis formed in a substantially arc shape when viewed in the arrow Z2direction. One of the two U-phase W-phase neutral point segment conductors61ais disposed on the other outer axial side (arrow Z1direction side). As shown inFIG.10, the V-phase neutral point segment conductor61bincludes a V-phase second leg portion81connected to the V-phase first conductor70and a neutral point coil end portion61d. The neutral point coil end portion61dis formed so as to protrude from the second leg portion81in the outer axial direction (in the arrow Z1direction). The two neutral point coil end portions61dare electrically joined to each other by being joined to both of the two neutral point coil end portions61c. As shown inFIG.11, the inner radial side neutral point conductor62includes two U-phase W-phase neutral point segment conductors62aand two V-phase neutral point segment conductors62b. The U-phase W-phase neutral point segment conductors62ainclude the U-phase second leg portions81connected to the first leg portions71of the first conductors70for the U-phase among the three-phase alternating current, the W-phase second leg portions81connected to the W-phase first conductor70, and the neutral point coil end portions62cthat each connect the U-phase second leg portion81and the W-phase second leg portion81. The neutral point coil end portions62care formed to be continuous with the U-phase second leg portions81and to be continuous with the W-phase second leg portion81. As a result, the U-phase W-phase neutral point segment conductors62aare formed in a substantially U-shape when viewed from the inner radial side. The V-phase neutral point segment conductors62bare formed in a substantially linear shape when viewed from the inner radial side. As shown inFIG.12, the neutral point coil end portion62cis formed so as to protrude axially outward with respect to the second coil end portion82of the second conductor80. The neutral-point coil end portion62cis disposed close to the outer axial side of the second coil end portion82of the second conductor80, and is formed along the circumferential direction when viewed in the axial direction. One of the two U-phase W-phase neutral point segment conductors62ais disposed on outer radial side of the other U-phase W-phase neutral point segment conductor62a. The V-phase neutral point segment conductor62bincludes the V-phase second leg portion81connected to the first leg portion71of the V-phase first conductor70, and a neutral point coil end portion62d. The neutral point coil end portion62dis formed so as to protrude from the second leg portion81to the outer axial side (arrow Z1direction). The two neutral point coil end portions62dare electrically joined by being joined to both of the two neutral point coil end portions62c. (Structure of Joint Portion) As shown inFIGS.12and13, the plurality of first leg portions71is provided in one slot12such that the first leg portions71are adjacent to each other in the radial direction of the stator core10. In addition, the plurality of second leg portions81is provided in one slot12such that the second leg portions81are adjacent to each other in the radial direction of the stator core10. The joint portion90is formed by joining a first surface71aof the first leg portion71and a second surface81aof the second leg portion81, which are both described below. Further, in one slot12, the plurality of first conductors70(first leg portions71) and the plurality of second conductors80(second leg portions81) are joined. Specifically, in one slot12, a plurality of first surface disposition portions71beach provided with the first surface71a, which is described below, of the first leg portion71, and a plurality of second surface disposition portions81beach provided with the second surface81a, which is described below, of the second leg portion81are alternately arranged along the radial direction. That is, the joint portions90, which are described below, of the first leg portions71and the second leg portions81are disposed adjacent to each other in the radial direction in one slot12. Specifically, the joint portions90are configured such that the joint portions90adjacent to each other in the radial direction overlap when viewed from the radial direction. Specifically, the plurality of (all) joint portions90disposed in one slot12are configured to overlap with each other when viewed in the radial direction. That is, all the joint portions90disposed in one slot12are disposed in a state in which the joint portions90are aligned along the horizontal direction. In other words, each position of the joint portions90in the axial direction in one slot12are substantially equal to each other. As will be described below, the joint portions90are parts in which the first surfaces71aof the first leg portions71and the second surfaces81aof the second leg portions81are joined (overlapped) when viewed from the radial direction. Further, as shown inFIG.14, each of the tip end portion71cof the first leg portion71and tip end portion81cof the second leg portion81has a tapered shape. Specifically, when viewed from the circumferential direction (A direction), each of the tip end portion71cof the first leg portion71and the tip end portion81cof the second leg portion81has a tapered shape. The first surface71aprovided so as to extend in the axial direction is provided on the tip end portion71cside of each of the first leg portions71of the plurality of first conductors70. Further, the second surface81aprovided so as to extend in the axial direction is provided on the tip end portion81cside of each of the second leg portions81of the plurality of second conductors80. Specifically, each of the first surface71aand the second surface81ais provided so as to extend parallel to the axial direction. Further, the first leg portion71and the second leg portion81include the first surface disposition portion71bprovided with the first surface71aand the second surface disposition portion81bprovided with the second surface81a, respectively. The first leg portion71has a first leg portion body portion71dthat is provided to be continuous with the first surface disposition portion71bon which the first surface71ais provided. The first leg portion body portion71dis provided on the opposite side (Z2direction side) of the first surface disposition portion71bfrom the tip end portion71c. Further, the second leg portion81has a second leg portion body portion81dthat is provided to be continuous with the second surface disposition portion81bon which the second surface81ais provided. The second leg portion body portion81dis provided on the opposite side (Z1direction side) of the second surface disposition portion81bfrom the tip end portion81c. Specifically, the first surface disposition portion71bis provided to be continuous with the first leg portion body portion71dvia a first step portion71gdescribed below. Further, the second surface disposition portion81bis provided to be continuous with the second leg portion body portion81dvia a second step portion81gdescribed below. A thickness t2of the first surface disposition portion71b, which is provided with the first surface71a, in the radial direction is smaller than a thickness t3of the first leg portion body portion71din the radial direction. Specifically, the thickness t2of the first surface disposition portion71bis about half of the thickness t3of the first leg portion body portion71d. Further, a thickness t4of the second surface disposition portion81b, which is provided with the second surface81a, in the radial direction is smaller than a thickness t5of the second leg portion body portion81din the radial direction. Specifically, the thickness t4of the second surface disposition portion81bis about half of the thickness t5of the second leg portion body portion81d. The thickness t2and the thickness t4are substantially equal, and the thickness t3and the thickness t5are substantially equal. Further, the coil portion30(seeFIG.2) includes the joint portion90in which the first surface71aand the second surface81aare joined in one slot12. That is, the joint portion90is located between the end face10a(seeFIG.2) and the end face10b(seeFIG.2) of the stator core10in the axial direction. Here, in the first embodiment, as shown inFIG.14, the first surface71aand the second surface81aare joined to each other in the radial direction (R direction) at the joint portion90. Specifically, a surface part71eon the tip end portion71cside of the first surface71aand a surface part81eon the tip end portion81cside of the second surface81aare joined in the radial direction. In other words, the first surface71aand the second surface81aare joined in a state of being displaced in the axial direction. The first surface71a(surface part71e) and the second surface81a(surface part81e) are provided so as to extend parallel to the axial direction and to face each other in the radial direction. That is, each of the first surface71a(surface part71e) and the second surface81a(surface part81e) extends so as to be orthogonal to the radial direction. Further, the first surface71a(surface part71e) faces the inner radial side (R1direction side), and the second surface81a(surface part81e) faces the outer radial side. Further, in the first embodiment, a length L4(seeFIG.14) of the first surface71ain the axial direction and a length L5(seeFIG.14) of the second surface81ain the axial direction are larger than a length L6(seeFIG.14) of the joint portion90in the axial direction. The length L4of the first surface71aand the length L5of the second surface81aare substantially equal lengths. The length L6of the joint portion90means the length of the surface part71eand the surface part81ein the axial direction. Further, in the first embodiment, between the first conductor70and the second conductor80, which face each other in the axial direction, a first clearance portion74is provided between the tip end portion71cof the first leg portion71and the second leg portion81in the axial direction. Further, between the first conductor70and the second conductor80, which face each other in the axial direction, a second clearance portion84is provided between the tip end portion81cof the second leg portion81and the first leg portion71in the axial direction. Specifically, the first clearance portion74is provided between the tip end portion71cof the first leg portion71and the second leg portion body portion81dof the second leg portion81in the axial direction. Further, the second clearance portion84is provided between the tip end portion81cof the second leg portion81and the first leg portion body portion71dof the first leg portion71in the axial direction. When viewed from the circumferential direction (A direction), the first clearance portion74is surrounded by the first leg portion71and the second leg portion81that are joined to each other, and the second insulating member21adjacent on the inner radial side (R1direction side). When viewed from the circumferential direction (A direction), the second clearance portion84is surrounded by the first leg portion71and the second leg portion81that are joined to each other, and the second insulating member21adjacent on the outer radial side (R2direction side). The details of the configuration of the second insulating member21will be described below. Further, each of the first clearance portion74and the second clearance portion84is provided for each set of the first leg portion71and the second leg portion81that are joined to each other. That is, a plurality of each of the first clearance portion74and the second clearance portion84is provided side by side in the radial direction (eight in the first embodiment, seeFIG.13). Specifically, when viewed from the radial direction, the plurality of first clearance portions74overlaps with each other, and the plurality of second clearance portions84overlaps with each other. A length L7of the first clearance portion74in the axial direction is substantially equal to a length L8of the second clearance portion84in the axial direction. The length L7of the first clearance portion74means the distance in the axial direction between the tip end portion71cof the first leg portion71and the second leg portion81. Further, the length L8of the second clearance portion84means the distance in the axial direction between the tip end portion81cof the second leg portion81and the first leg portion71. Further, in the first embodiment, both the axial length L7of the first clearance portion74and the axial length L8of the second clearance portion84are larger than the thickness t2of the first surface disposition portion71bprovided with the first surface71aof the first leg portion71and the thickness t4of the second surface disposition portion81bprovided with the second surface81aof the second leg portion81, in the radial direction. The length L7of the first clearance portion74and the length L8of the second clearance portion84are each set to a length that can sufficiently absorb dimensional variations that occur in the manufacture of the first conductor70and the second conductor80, and the assembly variation that occurs when the first conductor70and the second conductor80are assembled. Further, between the first surface disposition portion71bprovided with the first surface71aof the first leg portion71and the first leg portion body portion71d, the first step portion71gincluding a corner portion inner surface71fthat faces the second clearance portion84and that has a round shape is provided. Further, between the second surface disposition portion81bprovided with the second surface81aof the second leg portion81and the second leg portion body portion81d, the second step portion81gincluding a corner portion inner surface81fthat faces the first clearance portion74and that has a round shape is provided. Specifically, the corner portion inner surface71fand the corner portion inner surface81fhave an arc shape with a curvature radius smaller than the thickness t2of the first surface disposition portion71bin the radial direction and the thickness t4of the second surface disposition portion81bin the radial direction, respectively. In this case, the first leg portion71and the second leg portion81are provided with a flat surface71hand a flat surface81hthat are provided to be continuous with the corner portion inner surface71fand the corner portion inner surface81f, respectively. Each of the flat surface71hand the flat surface81his provided so as to extend orthogonal to the axial direction. Further, as shown inFIG.13, each of the first clearance portion74and the second clearance portion84is disposed in the slot12. Specifically, the entirety of each of the first clearance portion74and the second clearance portion84is disposed in the slot12. Further, since the length L1of the pair of first leg portions71(seeFIG.7) and the length L2of the pair of second leg portions81(seeFIG.8) are different from each other, by joining the first surface71aof the first leg portion71and the second surface81aof the second leg portion81, each of the first clearance portion74and the second clearance portion84(joint portion90) is provided on the end face10aside of the axial center C2in the axial direction (seeFIG.12). As a result, each of the first clearance portion74and the second clearance portion84is provided closer to the end face10aof the stator core10than the axial center C2of the stator core10. Specifically, the edge portion of the second clearance portion84on one side (Z2direction side) in the axial direction is provided at a position substantially the same as the end face10aof the stator core10in the axial direction. Further, the edge portion of the second clearance portion84on one side (Z2direction side) in the axial direction may be provided within a range of substantially insulating creepage distance from the end face10ain the Z1direction or the Z2direction. The axial center C2is an example of a “center” of the claims. Further, the stator100includes a conductive adhesive91that adheres the first surface71aand the second surface81aat the joint portion90and that makes the first leg portion71and the second leg portion81conductive. The conductive adhesive91is a paste joining material (silver nanopaste) that contains, as conductive particles, metal particles obtained by miniaturizing silver to a nanometer level, in a solvent. Further, the conductive adhesive91is configured to be melted by heat. Further, the conductive adhesive91contains a member (resin member) that volatilizes when heated, and has a function of bringing the first surface71aand the second surface81aclose to each other when the volatilizing member is heated and the volume of the conductive adhesive91is decreased. Further, in order to join the first surface71aand the second surface81a, the first conductor70and the second conductor80are assembled in a state in which the conductive adhesive91is applied in advance to a part corresponding to at least one of the joint portion90of the first surface71aand the second surface81a(at least one of the surface part71eand the surface part81e). InFIG.14, the thickness of the conductive adhesive91is emphasized for the sake of explanation, and the present disclosure is not limited to this example indicated in the drawing. Here, in the first embodiment, the conductive adhesive91is applied to a surface part71i, which faces the second surface81awhen viewed from the radial direction, of the first surface71a, and a surface part81i, which faces the first clearance portion74when viewed from the radial direction, of the second surface81a, in addition to at least one of the surface part71eand the surface part81e, and the second clearance portion84of the first surface71awhen viewed from the radial direction. Specifically, the conductive adhesive91is applied to the entirety of each of the surface part71iand the surface part81i. That is, each of the first surface71aand the second surface81ais entirely covered with the conductive adhesive91when viewed from the radial direction. The conductive adhesive91is not applied to each of the corner portion inner surface71fand the corner portion inner surface81f. The surface part71iand the surface part81iare examples of a “part facing the second clearance portion” and a “part facing the first clearance portion” in the claims, respectively. (Configuration of First Insulating Member) As shown inFIG.4, the first insulating member20is disposed between the wall portion11aand the teeth13and the first leg portion71and the second leg portion81(segment conductor40). As shown inFIG.15, the first insulating member20has a three-layer configuration. Specifically, as shown inFIG.12, the first insulating member20includes, in the slot12: an insulating layer20athat is provided between the wall portion11aof the back yoke11and the circumferential side surface13aof the teeth13(seeFIG.4), and the first leg portion71and the second leg portion81, and that insulates the wall portion11aand the circumferential side surface13afrom the first leg portion71and the second leg portion81; and a fixing layer20cthat is provided so as to overlap with a part20bat a position (region) (P2) different from the position P1in the axial direction corresponding to the joint portion90among the insulating layer20aand that fixes the stator core10and the second leg portion81. The fixing layer20cis preferably configured as an adhesive layer containing an adhesive. In addition, the position P2includes, in the axial direction, the entire region inside the slot12of the part excluding the axial position P1, and a part near the end face10bof the stator core10(including the part outside the slot12in the axial direction) for example. And the first insulating member20is disposed so as to integrally cover the surroundings of the second leg portions81disposed in parallel in the radial direction when viewed in the arrow Z2direction. In other words, both sides in the circumferential direction and both sides in the radial direction of the second leg portions81disposed in parallel in the radial direction are covered by the first insulating member20. In this way, the first insulating member20can ensure the insulation between the joint portion90and the stator core10. The insulating layer20ais configured of a polyphenylene sulfide resin (PPS), for example. The insulating layer20amay be formed in a non-woven fabric form such as aramid paper. In addition, as shown inFIG.12, the insulating layer20ais provided across from the end face10aon one axial side of the stator core10to the end face10bon the other axial side. That is, the insulating layer20ais disposed so as to cover the wall portion11aand the circumferential side surface13ain each slot. In addition, to “cover” does not only mean to cover all parts of the wall portion11aand the circumferential side surface13a, but means a broad concept including a case in which the inner radial side part (distal end gap part) of the circumferential side surface13ais exposed, as shown inFIG.4. As shown inFIG.15, the fixing layer20cincludes a foaming agent20d(expanding agent) that foams due to heat. Specifically, the fixing layer20cis formed, for example, by mixing a plurality of capsule bodies serving as the foaming agent20dwith a thermosetting resin20e. The foaming agent20dis configured to expand the volume of the capsule body when heated to a foaming temperature T1or higher. The thickness of the fixing layer20cincreases from the thickness t6(seeFIG.16) to the thickness t7(seeFIG.17) by being heated in the manufacturing process of the stator100, for example. As a result, the fixing layer20cfills the space between the second leg portion81and the wall portion11aand the circumferential side surface13aby the foaming agent20dfoaming (expanding) when heated. Further, the thermosetting resin20eis configured to be cured by being heated to a curing temperature T2or higher which is higher than the foaming temperature T1. The thermosetting resin20eforming the fixing layer20cis, for example, an epoxy resin. The fixing layer20cis configured such that when the fixing layer20cis heated, the thermosetting resin20eis cured such that the second leg portion81and the wall portion11aand the circumferential side surface13aare bonded and fixed. As shown inFIG.12, the fixing layer20ccontaining the foaming agent20din the foamed state fills a space between at least a part of the second leg portion81, and the wall portion11aand the circumferential side surface13athat configure the slot12, at the position P2different from the position P1in the axial direction corresponding to the joint portion90. Specifically, the fixing layer20cis provided so as to overlap with the part20bof the insulating layer20aon the other axial side (Z1direction side) of the position P1in the axial direction corresponding to the joint portion90. In other words, the fixing layer20cis provided so as to overlap with the part20bof the insulating layer20aon the other axial side of the vicinity of the end face10aon the one axial side (Z2direction side). Further, the fixing layer20cis provided in the slot12so as to overlap with the part20bof the insulating layer20athat is disposed between the second leg portion81and the stator core10. For example, as shown inFIG.15, the fixing layer20cis provided so as to overlap with and sandwich the insulating layer20ain the part20bof the insulating layer20aat a position different from the axial position corresponding to the joint portion90. Further, in the first embodiment, as shown inFIG.13, the first insulating member20provided between the slot12and the coil portion30and the second insulating member21provided separately from the first insulating member20are provided. As shown inFIG.18, the joint portions90adjacent in the radial direction, among the joint portions90each in which the first surface71aof the first leg portion71of the first conductor70and the second surface81aof the second leg portion81of the second conductor80are joined between the coils adjacent in the radial direction in one slot12, are insulated by the second insulating member21that has a sheet shape and that is provided separately from the first insulating member20. The term “coils” means a linear part of the coil portion30that is disposed in the slot12after the first conductor70and the second conductor80are joined. Thus, a plurality of coils is disposed in one slot12. The second insulating member21is an example of a “joint portion insulating member” in the claims. Here, in the first embodiment, as shown inFIG.18, the second insulating member21is formed by folding one sheet-shaped insulating member such as a Nomex. The second insulating member21includes: a facing surface insulating part21athat covers facing surfaces90aof the joint portions90that are adjacent in the radial direction; and a circumferential surface insulating part21bthat is continuous from both end portions of the facing surface insulating part21ain the circumferential direction and that covers one of the circumferential surfaces90bof the joint portion90that are adjacent in the radial direction for at least the insulation distance. The facing surface90aof the joint portion90means an outer radial surface and an inner radial surface, which face each other, of the joint portions90that are radially adjacent to each other. The insulation distance means a distance (creepage distance) that is a length along the circumferential surface insulating part21bin the radial direction and that is sufficient for insulating the joint portions90, which are adjacent to each other, from each other. As shown inFIG.19, the second insulating member21includes a part21cthat covers an outer radial side of the joint portion90disposed on the outermost radial side, and a part21dthat covers the inner radial side of the joint portion90disposed on the innermost radial side. Further, in the second insulating member21, the facing surface insulating parts21athat are adjacent in the radial direction are connected to each other by the circumferential surface insulating part21bin one or the other circumferential direction. Specifically, the facing surface insulating part21aon the outer radial side among the pair of facing surface insulating parts21adisposed adjacent to each other in the radial direction, the circumferential surface insulating part21bprovided on one side in the circumferential direction, the facing surface insulating part21aon the inner radial side among the pair of facing surface insulating parts21a, and the circumferential surface insulating part21bprovided on the other side in the circumferential direction are formed to be continuous. That is, the circumferential surface90bon the A1direction side of the joint portion90and the circumferential surface90bon the A2direction side of the joint portion90are alternately covered by the circumferential surface insulating part21b. In other words, the second insulating member21is configured so as not to continuously cover the circumferential surfaces90bof the plurality of joint portions90disposed adjacent to each other in the radial direction. Thus, the second insulating member21has a meandering shape (bellows shape) when viewed from the axial direction. Further, since one second insulating member21insulates the joint portions90that are adjacent to each other in the radial direction and that are disposed in one slot12, all the joint portions90in the slot12are insulated from each other. This makes it possible to reduce the number of steps for disposing the second insulating member21as compared to the case in which the plurality of joint portions90disposed in one slot12is individually covered by the insulating member. Further, as shown inFIG.19, the second insulating member21is configured to be expandable/contractible along the radial direction. The second insulating member21is made of a flexible sheet-shaped insulating member, and is configured to not continuously cover the circumferential surfaces90bof the plurality of joint portions90disposed adjacent to each other in the radial direction. Thus, even when the first leg portion71and the second leg portion81are pressed in the radial direction or the axial direction when the first leg portion71and the second leg portion81are joined, the second insulating member21can be deformed with the movement of the first leg portion71and the second leg portion81. Here, in the first embodiment, as shown inFIG.13, the second insulating member21is provided to extend in the axial direction so as to cover both the first clearance portion74and the second clearance portion84when viewed from the radial direction. Specifically, the second insulating member21is disposed so that an edge portion on one axial side (Z2direction side) protrudes outward (toward the Z2direction side) from the end face10aof the stator core10in the central axis direction. Further, in the second insulating member21, the edge portion on the other axial side (Z1direction side) is provided on the other axial side (Z1direction side) of the edge portion on the other axial side (Z1direction side) of the first clearance portion74in the slot12. Further, as shown inFIG.13, the first insulating member20is also disposed together with the second insulating member21so as to protrude outward (toward the Z2direction side) from the end face10aof the stator core10in the axial direction. A height position h1of the part of the second insulating member21protruding outward from the end face10aof the stator core10and a height position h2of the part of the first insulating member20protruding outward from the end face10aof the stator core10are substantially equal. The protrusion amount of the first insulating member20and the second insulating member21from the end face10aof the stator core10is adjusted to a degree in which the first insulating member20and the second insulating member21are not bent by coming into contact with the first coil end portion72of the first segment conductor70. Further, as shown inFIG.20, a length L12of the second insulating member21is smaller than a length L11of the first insulating member20in the axial direction. Specifically, the length L11of the first insulating member20is larger than the length L3of the stator core10in the axial direction. The length L12of the second insulating member21is smaller than the length L3of the stator core10. The second insulating member21is provided so as to cover the joint portion90and extend from the joint portion90toward the Z1direction side and the Z2direction side. The length L12of the second insulating member21is adjusted based on the magnitude of the voltage applied to the coil portion30(based on the required creepage distance). InFIG.20, illustration of the first conductor70and the second conductor80is omitted for simplification. Further, since the length L12of the second insulating member21is smaller than the length L11of the first insulating member20, as shown inFIG.21, the first insulating member20has a part20fthat overlaps with the second insulating member21and the part20bthat does not overlap with the second insulating member21when viewed in the radial direction. Specifically, the first insulating member20overlaps with the second insulating member21in the vicinity of the end portion (end face10a) in the axial direction in the slot12. A thickness t11of the part20fof the first insulating member20that overlaps with the second insulating member21is smaller than a thickness t12of the part20bof the first insulating member20that does not overlap with the second insulating member21. A thickness t13of the second insulating member21is smaller than the thickness t11. Further, the thickness t12is obtained by adding the thickness t11to the thickness t7of two sheets (t7×2) of the fixing layer20c. Further, the second insulating member21is disposed on one axial side (Z2direction side) with respect to the fixing layer20cof the first insulating member20and between the joint portions90in the radial direction, and is configured to insulate the joint portions90from each other. Specifically, the fixing layer20cis provided so as to overlap with the part20bof the insulating layer20athat does not overlap with the second insulating member21in the radial direction. Further, the insulating layer20ais disposed in the part20fthat overlaps with the second insulating member21when viewed in the radial direction. (Conductor Insertion Step) Next, a step of inserting each segment conductor into the slot12will be described with reference toFIG.22. As shown inFIG.22, in step S1, the plurality of second conductors80is inserted in the slot12from the Z1direction side. At this time, each of the second conductors80is inserted in the slot12by each second coil end portion82being simultaneously pressed from the upper side (Z1direction side) by a plate-shaped member having a disc shape. As a result, the protrusion amount of each of the second coil end portions82of the plurality of second conductors80from the stator core10is made uniform. Next, in step S2, the second insulating member21is inserted in the slot12. Specifically, the second insulating member21is disposed on the tip end portion side of the plurality of second conductors80inserted in the slot12. Then, in step S3, the plurality of first conductors70is inserted in the slot12from the Z2direction side. At this time, the plurality of first conductors70is inserted in the slot12by the first coil end portions72being simultaneously pressed from the lower side (Z2direction side) by a plate-shaped member having a disc shape. As a result, the protrusion amount of the plurality of first coil end portions72from the stator core10is made uniform. Second Embodiment Next, a stator200and the manufacturing method of the stator200according to the second embodiment will be described with reference toFIG.1,FIG.3, andFIGS.23to31. Unlike the first embodiment described above, in the stator200according to the second embodiment, a first surface171aand a second surface181aare joined by the urging force of a spring member210. The same configurations as those in the first embodiment described above are indicated by the same reference numerals as those in the first embodiment and are shown in the drawings, and the description thereof will be omitted. [Structure of Stator] The structure of the stator200according to the second embodiment will be described with reference toFIG.1,FIG.3, andFIGS.23to30. The stator200is an example of the “armature” in the claims. As shown inFIG.1, the stator200and the rotor101form a part of the rotary electric machine202. Further, as shown inFIG.23, the stator200includes a sheet-shaped insulating member121and a coil portion130(seeFIG.1). The coil portion130includes a first coil assembly130a(non-lead side coil) (seeFIG.27) and a second coil assembly130b(lead side coil) (seeFIG.27). The coil portion130is composed of a plurality of segment conductors140(seeFIG.24). The insulating member121is an example of a “joint portion insulating member” in the claims. (Configuration of Segment Conductor) As shown inFIG.24, in the segment conductor140, a first leg portion171(second leg portion181), described below, is not covered with the insulating coating and a conductor surface140bis exposed (seeFIG.24). InFIG.24, only the first conductor170described below is shown. However, since a second conductor180is similar, illustration thereof is omitted. <Structure of First Conductor and Second Conductor> As shown inFIGS.25A,25B,26A, and26B, the plurality of segment conductors140includes a plurality of the first conductors170disposed on one axial side (Z2direction side) of the stator core10and a plurality of the second conductors180disposed on the other axial side (Z1direction side) of the stator core10. Further, the first conductor170includes the first leg portion171having a length L31in the axial direction, a first coil end portion172, and a first crank portion173. The second conductor180includes the second leg portion181having a length L32in the axial direction, a second coil end portion182, and a second crank portion183. The length L31of the first leg portion171and the length L32of the second leg portion181are substantially the same. Further, as shown inFIG.27, the stator200includes the spring member210that is provided in each of the plurality of slots12so as to be sandwiched between the coil portion130and the opening portion12a(protruding portion13b) of the slot12. That is, the spring member210is provided in a distal end clearance12bprovided on the radially inner side, inside the slot12. The spring member210is configured to press the coil portion130from the inner radial side of the coil portion130in the radial direction such that the first surface171aof the first leg portion171of the first conductor170and the second surface181aof the second leg portion181of the second conductor180are in contact with each other. A contact portion190is formed by contact between the first surface171aof the first leg portion171and the second surface181aof the second leg portion181. The contact portion190is an example of a “joint portion” in the claims. The first surface171aand the second surface181aare in contact with each other by being pressed by the spring member210without a binder being interposed between the first surface171aand the second surface181a. That is, the first surface171aand the second surface181aare not joined, and the contact state between the first surface171aand the second surface181ais maintained by the pressing force of the spring member210. Further, each of the plurality of contact portions190is disposed in the central portion of the stator core10in the axial direction, in the slot12. The spring member210is also disposed in the central portion of the stator core10in the axial direction. Specifically, the spring member210is provided so as to overlap with each of the plurality of contact portions190when viewed in the radial direction. Further, each of the first surface171aand the second surface181ais plated. That is, the plated surfaces (the first surface171aand the second surface181a) are in contact with each other. In the plating process, metals such as nickel (Ni), silver (Ag), gold (Au), and tin (Sn) are used. The plating process may be performed using a plurality of metals (for example, Ni and Ag) among the above metals. As shown inFIG.28, the first leg portion171includes a first surface disposition portion171b, a tip end portion171c, a first leg portion body portion171d, and a first step portion171e. A clearance portion171fis provided between the first step portion171eand the tip end portion181cof the second leg portion181. The clearance portion171fis an example of a “first clearance portion” in the claims. Further, the second leg portion181includes a second surface disposition portion181b, a tip end portion181c, a second leg portion body portion181d, and a second step portion181e. A clearance portion181fis provided between the second step portion181eand the tip end portion171cof the first leg portion171. The clearance portion181fis an example of a “second clearance portion” in the claims. Further, the first surface disposition portion171band the second surface disposition portion181bare provided at a central portion P3(seeFIG.27) in the axial direction of the stator core10. InFIG.28, each of the first step portion171eand the second step portion181eis shown so as not to have a corner portion inner surface having a round shape. However, similar to the first embodiment described above, each of the first step portion171eand the second step portion181emay have a corner portion inner surface having a round shape. As shown inFIG.29, between the coils adjacent to each other in the radial direction in one slot12, the sheet-shaped insulating member121is provided so as to insulate the contact portions190from each other. Here, in each contact portion190, the first leg portion171in which the conductor surface140b(seeFIG.24) is exposed and the second leg portion181in which the conductor surface140bis exposed are in contact without a bonding agent being interposed therebetween. Specifically, the insulating member121is provided between each of the plurality of (eight in the second embodiment) coils (a set of the first leg portion171and the second leg portion181that are in contact with each other) disposed in the radial direction in the slot12. Specifically, the insulating member121is formed by folding one sheet-shaped insulating member such as a Nomex. The insulating member121includes: facing surface insulating parts121athat cover facing surfaces190aof the contact portions190that are adjacent in the radial direction; and a circumferential surface insulating part121bthat is continuous from both end portions of the facing surface insulating part121ain the circumferential direction and that covers one of the circumferential surfaces190bof the contact portions190that are adjacent in the radial direction for at least the insulation distance. The facing surfaces190aof the contact portions190mean an outer radial surface and an inner radial surface of the contact portions190, which face each other in the radial direction. Further, the insulation distance is a length along the radial direction of the circumferential surface insulating part121band means a distance (creepage distance) sufficient for insulating the contact portions190adjacent to each other in the radial direction. The circumferential surfaces190bmean surfaces of the contact portions190that intersect with the circumferential direction. In other words, the circumferential surfaces190bmean surfaces extending in the radial direction and the axial direction. In addition, the insulating member121includes the contact portion insulating parts121cthat are formed such that the following are continuous: the facing surface insulating part121aon the outer radial side among a pair of the facing surface insulating parts121adisposed adjacent to each other in the radial direction; the circumferential surface insulating part121bprovided on one side in the circumferential direction; the facing surface insulating part121aon the inner radial side among the pair of the facing surface insulating parts121a; and the circumferential surface insulating part121bprovided on the other side in the circumferential direction. Further, the stator200includes a core leg portion insulating part122that is provided between the slot12and the coil portion130and that is integrally formed with the contact portion insulating part121c. That is, the core leg portion insulating part122has a sheet shape similar to the contact portion insulating part121cand is made of the same material as the contact portion insulating part121c. Further, the contact portion insulating part121cand the core leg portion insulating part122have the same thickness (not shown). Further, the contact portion insulating part121cand the core leg portion insulating part122have the same length in the axial direction. Specifically, the core leg portion insulating part122has the one side insulating part122athat is continuous with the facing surface insulating part121aon the outermost radial side and that is provided, on one side of the slot12in the circumferential direction (left side inFIG.29), between the slot12(circumferential side surface13a) and the coil portion130(circumferential surface190b). Further, the core leg portion insulating part122has the other side insulating part122bthat is continuous with the facing surface insulating part121aon the innermost radial side and that is provided, on the other side of the slot12in the circumferential direction (right side inFIG.29), between the slot12(circumferential side surface13a) and the coil portion130(circumferential surface190b). More specifically, in the one side insulating part122a(other side insulating part122b), the following parts are alternated along the radial direction: the part the is sandwiched between the circumferential side surface13aof the slot12and the circumferential surface190bof the coil portion130; and the part sandwiched between the circumferential side surface13aof the slot12and the circumferential surface insulating part121bthat covers the circumferential surface190bof the coil portion130. Further, the one side insulating part122aextends from an outer radial side end portion230aof the coil portion130in the slot12to an inner radial side end portion230b(so as to extend over the end portion230b). The other side insulating part122bextends from the inner radial side end portion230bof the coil portion130in the slot12to the outer radial side end portion230a(so as to extend over the end portion230a). That is, the coil portion130in the slot12is provided so as to be surrounded by the facing surface insulating part121aon the outermost radial side, the facing surface insulating part121aon the innermost radial side, the one side insulating part122a, and the other side insulating part122b. The core leg portion insulating part122includes an inner radial side insulating part122cthat is continuous with the one side insulating part122aand that is provided so as to cover the facing surface insulating part121aon the innermost radial side from the inner radial side. Further, the core leg portion insulating part122has an outer radial side insulating part122dthat is continuous with the other side insulating part122band that is provided so as to cover the facing surface insulating part121aon the outermost radial side from the outer radial side. Specifically, the inner radial side insulating part122cis provided so as to be sandwiched between the facing surface insulating part121aon the innermost radial side and the spring member210. That is, the coil portion130and the spring member210are insulated from each other by the facing surface insulating part121aon the innermost radial side and the inner radial side insulating part122c. The outer radial side insulating part122dis provided so as to be sandwiched between the facing surface insulating part121aon the outermost radial side and the wall portion11aof the slot12. That is, the coil portion130and the wall portion11a(stator core10) of the slot12are insulated from each other by the facing surface insulating part121aon the outermost radial side and the outer radial side insulating part122d. Further, the inner radial side insulating part122chas a length L41in the circumferential direction. Further, the outer radial side insulating part122dhas a length L42in the circumferential direction. Each of the length L41of the inner radial side insulating part122cand the length L42of the outer radial side insulating part122dis greater than half the width W2of the slot12(seeFIG.3), for example. In addition, as shown inFIG.27, each of the contact portion insulating part121cand the core leg portion insulating part122is disposed such that edge portions on both sides in the axial direction protrude outward from the end faces (10a,10b) of the stator core10in the axial direction. As a result, each of the contact portion insulating part121cand the core leg portion insulating part122is provided across the entire slot12, in the axial direction. As shown inFIG.30, the core leg portion insulating part122includes an insulating layer123aand a fixing layer123bthat includes a foaming agent123cthat foams due to heat. The foaming agent123cfoams and expands so as to fix each of the first leg portion171and the second leg portion181in at least the axial direction with respect to the stator core10. The fixing layer123bof the core leg portion insulating part122is configured to bond and fix each of the first leg portion171and the second leg portion181to the stator core10. The fixing layer123bis provided on both surfaces of the insulating layer123a. When the fixing layer123bis heated, a thermosetting resin123dis cured. Thus, it is not necessary to use a varnish or the like to fix each of the first leg portion171and the second leg portion181. InFIG.30, in order to highlight the core leg portion insulating part122, the core leg portion insulating part122is illustrated so as to have a thickness greater than the actual thickness. InFIG.30, the illustration of the stator core10and the like is omitted for simplification. Since the insulating layer123aand the fixing layer123bhave the same configurations (materials) as the insulating layer20aand the fixing layer20cof the first embodiment described above, detailed description thereof will be omitted. Further, although not shown, the contact portion insulating part121calso has the same configuration (composition) as the core leg portion insulating part122. (Stator Manufacturing Process) Next, with reference toFIG.31, a manufacturing method of the stator200will be described. As shown inFIG.31, first, in step S11, the insulating member121(contact portion insulating part121c) and the core leg portion insulating part122are integrally inserted (disposed) in the slot12. Next, in step S12, the second leg portion181(seeFIGS.26A and26B) of the second conductor180is inserted in the slot12from the other side (Z1direction side) in the axial direction. Next, in step S13, the first leg portion171(seeFIGS.25A and25B) of the first conductor170is inserted in the slot12from one side (Z2direction side) in the axial direction. At this time, the first leg portion171is disposed such that the first surface171aof the first leg portion171and the second surface181aof the second leg portion181face each other. Next, in step S14, the spring member210(seeFIG.27) is inserted in the slot12from the inner radial side through the opening portion12aof the slot12. Then, in step S15, the stator core10is heated and the fixing layer123bis heated and thus, the foaming agent123cis foamed and the fixing layer123bis expanded. In this way, the coil portion130is fixed to the slot12at least in the axial direction. The other configurations of the second embodiment are the same as those of the first embodiment described above. [Effects of First and Second Embodiments] In the first and second embodiments, the following effects can be obtained. In the first and second embodiments, as described above, between the first segment conductor (70,170) and the second segment conductor (80,180) that are joined to each other at the joint portions (90,190), the first clearance portion (74,171f) is provided between the tip end portion (71c,171c) of the first leg portion (71,171) and the second leg portion body portion (81d,181d) in the axial direction. Further, the second clearance portion (84,181f) is provided between the tip end portion (81c,181c) of the second leg portion (81,181) and the first leg portion body portion (71d,171d) in the axial direction. The term “joining” has a broad meaning including not only a state of being joined via a binder (joining material) but also a state of being only in contact without interposing a binder (joining material). As a result, even if there is a dimensional variation that occurs while manufacturing the first segment conductor (70,170) and the second segment conductor (80,180), the above variation can be absorbed by the first clearance portion (74,171f) and the second clearance portion (84,181f). As a result, the following case can be prevented: the tip end portion (71c,81c,171c,181c) of the conductor (70,80,170,180) that is manufactured to have a relatively large dimension, among the first segment conductors (70,170) and the second segment conductors (80,180), is in contact with the conductor (70,80,170,180) to be joined with and thus, movement of the other first segment conductors (70,170) and the second segment conductors (80,180) in the axial direction is stopped. As a result, even when the height of each of the coil end portions (72,82,172,182) of the first segment conductor (70,170) and the second segment conductor (80,180) are aligned, the conductors (70,80,170,180) that are manufactured to have relatively small dimensions, among the first segment conductor (70,170) and the second segment conductor (80,180), can be easily brought into contact with the conductors (70,80,170,180) to be joined with, in the joint portion (90,190). As a result, even when there is a variation in the dimensions of the first segment conductor (70,170) and the second segment conductor (80,180), it is possible to ensure a joining area of the first leg portion (71,171) and the second leg portion (81,181) while making the protrusion amount of each coil end portion (72,82,172,182) of the first segment conductor (70,170) and the second segment conductor (80,180) uniform. Since the first clearance portion (74,171f) and the second clearance portion (84,181f) are provided, the first segment conductor (70,170) and the second segment conductor (80,180) can be prevented from being contact in the axial direction. As a result, even when there is the conductor (70,80,170,180) that is manufactured to have a size larger than a design size, among the first segment conductor (70,170) and the second segment conductor (80,180), the first segment conductor (70,170) (first leg portion (71,171)) and the second segment conductor (80,180) (second leg portion (81,181)) can be prevented from being disposed on an outer side of a predetermined position in the axial direction. Further, even when there is an assembly variation when assembling the first segment conductor (70,170) and the second segment conductor (80,180), the above variation can be absorbed by the first clearance portion (74,171f) and the second clearance portion (84,181f). Further, in the first and second embodiments, as described above, each of the first surface (71a,171a) and the second surface (81a,181a) is joined to a portion of each other in the joint portion (90,190). Further, the lengths (L4, L5) of the first surface (71a,171a) and the second surface (81a,181a) in the axial direction are larger than the length (L6) of the joint portion (90,190) in the axial direction. With such a configuration, each of the first clearance portion (74,171f) and the second clearance portion (84,181f) can be easily formed. Further, in the first and second embodiments, as described above, each of the first surface (71a,171a) and the second surface (81a,181a) extends parallel to the axial direction and is provided so as to face each other in the radial direction. Further, the first surface (71a,171a) and the second surface (81a,181a) are joined to each other in the radial direction. With this configuration, since each of the first surface (71a,171a) and the second surface (81a,181a) extends parallel to the axial direction, the first surface (71a,171a) and the second surface (81a,181a) can be easily assembled without interfering with each other in the axial direction. Further, since the slots (12) in which the first leg portion (71,171) and the second leg portion (81,181) are disposed are open in the radial direction, by joining the first surface (71a,171a) and the second surface (81a,181a) to each other in the radial direction, it is possible to easily press the joint portion (90,190) with the pressing member (spring member210) via the part of each slot (12) that is open in the radial direction. Further, in the first and second embodiments, as described above, the length (L7) of the first clearance portion (74,171f) in the axial direction and the length (L8) of the second clearance portion (84,181f) in the axial direction are both larger than the thickness (t2) of the first surface disposition portion (71b,171b) provided with the first surface (71a,171a) of the first leg portion (71,171) and the thickness (t4) of the second surface disposition portion (81b,181d) provided with the second surface (81a,181a) of the second leg portion (81,181), in the direction (radial direction) in which the first surface (71a,171a) and the second surface (81a,181a) are joined. With this configuration, since both the length (L7) of the first clearance portion (74,171f) in the axial direction and the length (L8) of the second clearance portion (84,181f) in the axial direction are increased, it is possible to more surely absorb the dimensional variation and the assembly variation described above. Further, in the first and second embodiments, as described above, each of the first clearance portion (74,171f) and the second clearance portion (84,181f) is disposed in the slot (12). With this configuration, compared to the case in which each of the first clearance portion (74,171f) and the second clearance portion (84,181f) is disposed outside the slot (12), the length of the coil portion (30,130) in the axial direction can be suppressed from being increased. Further, in the first embodiment, as described above, each of the first clearance portion (74) and the second clearance portion (84) is provided closer to the one side end face (10a) of the armature core (10) in the axial direction than the axial center (C2) of the armature core (10). With this configuration, the first surface (71a) and the second surface (81a) can be joined in the vicinity of the one side end face (10a). Here, since the vicinity of the one-side end face (10a) is a part through which lubricating oil passes, the heat generated at the joint portion (90) can be cooled by the above oil described above. Further, in the first surface disposition portion (71b) provided with the first surface (71a) of the first leg portion (71) and the second surface disposition portion (81b) provided with the second surface (81a) of the second leg portion (81) are provided, the thickness (t2, t4) in the direction (radial direction) in which the first surface (71a) and the second surface (81a) is relatively small. Thus, the current density becomes relatively high and the heating value becomes relatively large. Therefore, in such a configuration, joining the first surface (71a) and the second surface (81a) in the vicinity of the one side end face (10a) is part effective in efficiently cooling the part in which the heating value is large. The meaning of the vicinity of the one-side end face (10a) includes both the position of the one-side end face (10a) itself and the vicinity of the one-side end face (10a). Further, in the first embodiment, as described above, each of the first segment conductor (70) and the second segment conductor (80) has a U shape including the pair of first leg portions (71) and the pair of second leg portions (81), respectively. The pair of first leg portions (71) and the pair of second leg portions (81) have different lengths from each other in the axial direction. Each of the first clearance portion (74) and the second clearance portion (84) are provided in the vicinity of the one side end face (10a) by the first surface (71a) of the pair of first leg portions (71) and the second surface (81a) of the pair of second leg portions (81) being joined. With this configuration, it is possible to easily provide each of the first clearance portion (74) and the second clearance portion (84) in the vicinity of the end face10a, just by making the lengths of the pair of first leg portions (71) and the pair of second leg portions (81) in the axial direction different from each other. Further, in the first embodiment, as described above, the conductive adhesive (91) is applied to the part (71i), which faces the second clearance portion (84) when viewed from the radial direction in which the first surface (71a) and the second surface (81a) are joined, of the first surface (71a), and the part (81i), which faces the first clearance portion (74) when viewed from the radial direction, of the second surface (81a), in addition to the part (71e,81e) corresponding to the joint portion (90), of at least one of the first surface (71a) and the second surface (81a). With this configuration, even when the first leg portion (71) and the second leg portion (81) are displaced in the axial direction from the predetermined positions, the first surface (71a) and the second surface (81a) can be joined by the conductive adhesive (91) that is applied to the part (71i) of the first surface (71a) facing the first clearance portion (74) and the part (81i) of the second surface (81a) facing the second clearance portion (84) when viewed from the joining direction (diameter direction) in which the first surface (71a) and the second surface (81a) are joined. As a result, even when the first leg portion (71) and the second leg portion (81) are displaced in the axial direction from the predetermined positions, the joining area of the first surface (71a) and the second surface (81a) can be ensured. Further, in the first and second embodiments, as described above, in the one slot (12), the plurality of first surface disposition portions (71b,171b) each provided with the first surface (71a,171a) of the first leg portion (71,171) and the plurality of second surface disposition portions (81b,181d) each provided with the second surface (81a,181a) of the second leg portion (81,181) are arranged alternately along the radial direction. Each of the first clearance portion (74,171f) and the second clearance portion (84,181f) is provided for each set of the first leg portion (71,171) and the second leg portion (81,181) that are joined to each other. With this configuration, in all the sets of the first leg portion (71,171) and the second leg portion (81,181) that are joined to each other in the one slot (12), with the first clearance portion (74,171f) and the second clearance portion (84,181f), the dimensional variation and the assembly variation described above can be absorbed. Further, in the first and second embodiments, as described above, the joint portion insulating member (21,121) is provided to extend in the axial direction so as to cover both the first clearance portion (74,171f) and the second clearance portion (84,181f) when viewed from the radial direction. With this configuration, it is possible to prevent the adjacent joint portions (90,190) from conducting with each other via the first clearance portion (74,171f) and the second clearance portion (84,181f). Further, in the first embodiment, as described above, the first surface disposition portion (71b) is provided so as to be continuous with the first leg portion body portion (71d) via the first step portion (71g) including the corner portion inner surface (71f) that faces the second clearance portion (84) and that has a round shape. The second surface disposition portion (81b) is provided so as to be continuous with the second leg portion body portion (81d) via the second step portion (81g) including the corner portion inner surface (81f) that faces the first clearance portion (74) and that has a round shape. With this configuration, since the first step portion (71g) and the second step portion (81g) are provided with the corner portion inner surface (71f) and the corner portion inner surface (81f) having a round shape, respectively, it is possible to suppress the concentration of stress in the first step portion (71g) and the second step portion (81g) at the time of joining and after joining of the first surface (71a) and the second surface (81a). As a result, even when the second clearance portion (84) and the first clearance portion (74) are provided so as to face each of the corner portion inner surface (71f) and the corner portion inner surface (81f), respectively, it is possible to suppress the concentration of stress in each of the first step portion (71g) and the second step portion (81g) and prevent the first leg portion (71) and the second leg portion (81) from being damaged. Further, in the first embodiment, as described above, each of the tip end portion (71c,171c) of the first leg portion (71,171) and the tip end portion (81c,181c) of the second leg portion (81,181) have a tapered shape. With this configuration, when the first leg portion (71,171) and the second leg portion (81,181) are inserted in the slot (12) in the axial direction so as that the first surface (71a,171a) and the second surface (81a,181a) are joined, the tip end portion (71c,171c) of the first leg portion (71,171) and the tip end portion (81c,181c) of the second leg portion81can be suppressed from interfering with each other. Further, in the first and second embodiments, as described above, the step of disposing the first segment conductor (70,170) and the second segment conductor (80,180) is the step of disposing the step of disposing the first segment conductor (70,170) and the second segment conductor (80,180) so that the first clearance portion (74,171f) is provided between the tip end portion (71c,171c) of the first leg portion (71,171) and the second leg portion body portion (81d,181d) in the axial direction and so that the second clearance portion (84,181f) is provided between the tip end portion (81c,181c) of the second leg portion (81,181) and the first leg portion body portion (71d,171d) in the axial direction. As a result, even if there is a dimensional variation that occurs while manufacturing the first segment conductor (70,170) and the second segment conductor (80,180), the above variation can be absorbed by the first clearance portion (74,171f) and the second clearance portion (84,181f). As a result, the following case can be prevented: the tip end portion (71c,81c,171c,181c) of the conductor that is manufactured to have a relatively large dimension, among the plurality of first segment conductors (70,170) and the second segment conductors (80,180), is in contact with the conductor (70,80,170,180) to be joined with and thus, movement of the other first segment conductors (70,170) and the second segment conductors (80,180) in the axial direction is stopped. As a result, even when the height of each of the coil end portions (72,82,172,182) of the first segment conductor (70,170) and the second segment conductor (80,180) are aligned, the conductors (70,80,170,180) that are manufactured to have relatively small dimensions, among the first segment conductor (70,170) and the second segment conductor (80,180), can be easily brought into contact with the conductors (70,80,170,180) to be joined with, in the joint portion (90,190). As a result, even when there is a variation in the dimensions of the first segment conductor (70,170) and the second segment conductor (80,180), it is possible to provide the manufacturing method of the armature (100,200) in which it is possible to ensure a joining area of the first leg portion (71,171) and the second leg portion (81,181) while making the protrusion amount of each coil end portion (72,82,172,182) of the first segment conductor (70,170) and the second segment conductor (80,180) uniform. [Modifications] It should be considered that the embodiments presently disclosed are exemplifications in all points and are not restrictive. The scope of the present disclosure is shown by the scope of the claims and not by the above description of the embodiments, and further includes the meanings equivalent to the scope of the claims and all changes (modifications) within the scope. For example, in the first and second embodiments described above, an example is shown in which each of the first surface71a(171a) and the second surface81a(181a) extends parallel to the axial direction. However, the present disclosure is not limited to this. Each of the first surface71a(171a) and the second surface81a(181a) may be tilted by a predetermined angle (for example, 5 degrees or less) with respect to the axial direction. Further, in the first and second embodiments described above, an example is shown in which the first surface71a(171a) and the second surface81a(181a) are joined to each other in the radial direction. However, the present disclosure is not limited to this. The first surface71a(171a) and the second surface81a(181a) may be joined in a direction intersecting the radial direction (for example, the circumferential direction). Further, in the first and second embodiments described above, an example is shown in which both lengths (L7, L8) of the first clearance portion74(171f) and the second clearance portion84(181f) in the axial direction are larger than the thickness (t2, t4) of the first surface disposition portion71b(171b) and the second surface disposition portion81b(181b) in the radial direction. However, the present disclosure is not limited to this. The length (L7or L8) of one of the first clearance portion74(171f) and the second clearance portion84(181f) in the axial direction may be larger than the thickness (t2, t4) of the first surface disposition portion71b(171b) and the second surface disposition portion81b(181b) in the radial direction. Further, in the first embodiment described above, an example is shown in which each of the first clearance portion74and the second clearance portion84is disposed in the slot12. However, the present disclosure is not limited to this. For example, a part of the first clearance portion74and the second clearance portion84(for example, only the second clearance portion84) may be disposed outside the slot12(seeFIG.32), or the entirety of both the first clearance portion74and the second clearance portion84and the second clearance portion84may be disposed outside the slot12(seeFIG.33). Further, in the first embodiment described above, an example is shown in which each of the first clearance portion74and the second clearance portion84is disposed in the vicinity of the end face10a(one side end face). However, the present disclosure is not limited to this. For example, each of the first clearance portion74and the second clearance portion84may be disposed in the vicinity of the end face10a(one side end face) and the end face10b(other side end face). In this case, as shown inFIG.34, a pair of first leg portions271of a first segment conductor270is configured so that the first leg portions271have different lengths from each other (seeFIG.34A), and a pair of second leg portions281of a second segment conductor280is configured so that the second leg portions281have different lengths from each other (seeFIG.34B). That is, each of the first segment conductor270and the second segment conductor280has a J-shape (substantially J-shape). Further, in the second embodiment described above, an example is shown in which the contact portion190(joint portion) is disposed in the vicinity of the central portion in the axial direction, in the slot12. However, the present disclosure is not limited to this. For example, the contact portion190(joint portion) may be disposed in the vicinity of the end face10a(one side end face) and the end face10b(other side end face). Further, in the first embodiment described above, an example is shown in which the length of the second leg portion81is longer than the length of the first leg portion71. However, the present disclosure is not limited to this. For example, the length of the second leg portion81may be shorter than the length of the first leg portion71. Further, in the second embodiment described above, the length L31of the first leg portion171and the length L32of the second leg portion181may be different from each other. Further, in the first embodiment described above, an example is shown in which the second conductor80having a long leg portion is a conductor on the lead side, and the first conductor70having a short leg portion is the conductor on the non-lead side. However, the present disclosure is not limited to this. For example, the second conductor80having a long leg portion may be the conductor on the non-lead side, and the first conductor70having a short leg portion may be the conductor on the lead side. Further, in the second embodiment described above, the second conductor180may be the conductor on the non-lead side, and the first conductor170may be the conductor on the lead side. Further, in the first embodiment described above, an example is shown in which the conductive adhesive91is applied to the entirety of each of the surface part71i(the part facing the second clearance portion) and the surface part81i(the part facing the first clearance portion). However, the present disclosure is not limited to this. For example, the conductive adhesive91may be applied only to a portion of each of the surface part71i(the part facing the second clearance portion) and the surface part81i(the part facing the first clearance portion). Further, the conductive adhesive91does not have to be applied to each of the surface part71i(the part facing the second clearance portion) and the surface part81i(the part facing the first clearance portion). In the first embodiment described above, an example is shown in which the corner portion inner surface71fand the corner portion inner surface81fhave an arc shape with a curvature radius smaller than the thickness t2of the first surface disposition portion71bin the radial direction and the thickness t4of the second surface disposition portion81bin the radial direction, respectively. However, the present disclosure is not limited to this. The corner portion inner surface71fand the corner portion inner surface81fmay have an arc shape with a curvature radius equal to or more than the thickness t2of the first surface disposition portion71bin the radial direction and the thickness t4of the second surface disposition portion81b in the radial direction, respectively. In the first embodiment above described, an example is shown in which the first insulating member20including the fixing layer20cconfigured as the adhesive layer is used. However, the present disclosure is not limited to this. For example, by using a first insulating member20including an expansive material (expansion layer) different from the adhesive layer, the wall portion11aand the circumferential side surface13aand the second leg portion81may be pressed against each other (pressing force) without being adhered, to be fixed. The same configuration may be used in the second embodiment described above. Further, in the embodiment described above, an example is shown in which the first insulating member20and the second insulating member21(joint portion insulating member) have a sheet shape. However, the present disclosure is not limited to this. It is possible to apply the present disclosure to a stator having the first insulating member20and the second insulating member21that do not have a sheet shape. Similarly, each of the insulating member121(joint portion insulating member) and the core leg portion insulating part122of the second embodiment described above may not be configured to have a sheet shape. DESCRIPTION OF REFERENCE NUMERALS 10Stator core (armature core)10aEnd face (one side end face)10bEnd face (other side end face)12Slot21Second insulating member (joint portion insulating member)30,130Coil portion40,140Segment conductor70,170,270First conductor71,171,271First leg portion71a,171aFirst surface71b,171bFirst surface disposition portion71c,171cTip end portion (tip end portion of first leg portion)71d,171dFirst leg portion body portion71fCorner portion inner surface (corner portion inner surface of first step portion)71gFirst step portion71iSurface part (part facing second clearance portion)74,171fFirst clearance portion80,180Second conductor81,181Second leg portion81a,181aSecond surface81b,181bSecond surface disposition portion81c,181cTip end portion (tip end portion of second leg portion)81d,181dSecond leg portion body portion81fCorner portion inner surface (corner portion inner surface of second step portion)81gSecond step portion81iSurface part (part facing first clearance portion)84,181fSecond clearance portion90Joint portion91Conductive adhesive100,200Stator (armature)121Insulating member (joint portion insulating member)190Contact portion (joint portion)C2Axial center (center)L1Length (length of first leg portion)L2Length (length of second leg portion)L4Length (length of first surface)L5Length (length of second surface)L6Length (length of joint portion)L7Length (length of first clearance portion)L8Length (length of second clearance portion)t2Thickness (thickness of first surface disposition portion)t3Thickness (thickness of first leg portion body portion)t4Thickness (thickness of second surface disposition portion)t5Thickness (thickness of second leg portion body portion) | 93,541 |
11863037 | DETAILED DESCRIPTION OF THE DRAWINGS FIG.1schematically illustrates, in a longitudinal section, a stator2and a rotor1of an electric motor. Further parts and details have been omitted for the sake of clarity. The stator has a schematically indicated cylindrical winding arrangement, which may consist of one or more sub-windings. The rotor1has at least one permanent magnet and a hub, which is connected to a shaft3. The magnetic poles of the rotor1or of the magnet/magnets thereof can be driven in the magnetic field of the stator2. The shaft3is usually rotatably mounted at one or more points in plain or ball bearings. The bearings may be fixedly connected for example to the stator2or to a housing (not illustrated) of the stator or the motor as a whole. InFIG.1it can be seen that the stator, which surrounds the rotor1concentrically and coaxially, has a diameter D in the radially expanded state illustrated there. FIG.2shows the elements of a motor already illustrated inFIG.1, specifically a rotor1and a winding arrangement of a stator2, wherein the rotor and stator are pulled apart from one another in the axial direction4. The stator and the rotor do not overlap one another in the axial direction in this state. The stator is radially compressed by radial compression of the winding arrangement on the whole to the diameter d, which is equal to or smaller than the outer diameter of the rotor1. Is thus clear that, due to the divisibility of the motor and the displaceability of the stator relative to the rotor, the stator is radially compressible as soon as the rotor has been removed therefrom. Irrespective of this and in addition, the rotor may also be compressible in the radial direction. In this case, the stator and rotor may also be radially compressed jointly in the assembled state, or can be displaced axially relative to one another and can both be radially compressed separately from one another. In the latter case it is useful, but not necessary, for both elements, i.e., both the stator and the rotor, to be compressible approximately to the same outer diameter. FIG.3in the upper region shows a first lead5of a winding arrangement of a motor according to the invention in the compressed state, wherein the lead extends in a spiraled manner. If a winding arrangement or sub-winding is formed from this lead extending in a spiraled manner, this winding arrangement or sub-winding can radially expand if the winding wire5is extended and later can be radially compressed again. In the lower region ofFIG.3a lead6is illustrated, which in the compressed state has a meandering form which can be extended when transitioning into an expanded state. InFIG.4aa lead5that is spiraled in the compressed state is illustrated schematically in the left-hand part and likewise in the compressed state is illustrated schematically in the form of a circular ring, which symbolizes a winding arrangement. In the right-hand part ofFIG.4a, an expanded form of the stator is illustrated in axial view, in which the winding lead(s) is/are elongated and accordingly the winding arrangement and/or the sub-windings is/are likewise expanded. The stator in the right-hand part ofFIG.4ahas the enlarged diameter D, whereas in the compressed state illustrated in the left-hand part ofFIG.4ait has the reduced diameter d. InFIG.4ba compressed lead6is illustrated in meandering form, which, as considered in the axial direction, is arranged in a circular ring form, which represents a winding arrangement of a stator. The arrangement has the compressed outer diameter d. In the right-hand part ofFIG.4bthe same stator is illustrated in the radially expanded state, wherein the winding lead(s) is/are elongated, or at least are further extended than in the compressed state. The transition between the compressed and expanded state of the stator can be implemented for example by an application of force, in that the stator is brought by means of radial external pressure into a compressed form and, when the external radial compression force is cancelled, expands resiliently again of its own accord. Conversely, the stator may also have a reduced diameter without external force application, and may be expandable by force application. As a further alternative, it may be that the winding arrangement has leads made of what are known as memory alloys, which for example in the event of temperature changes change their shape and in defined temperature ranges each have reproducible shapes. Such memory alloys may be, for example, NiTi (nickel-titanium; nitinol), NiTiCu (nickel-titanium-copper), CuZn (copper-zinc), CuZnAl (copper-zinc-aluminium), CuAlNi (copper-aluminium-nickel), FeNiAl (iron-nickel-aluminium), or FeMnSi (iron-manganese-silicon). Alloys of this type are also referred to as hyperelastic alloys. In addition to the described properties of the winding arrangement, a casting of the entire winding arrangement or individual sub-windings in a resilient material, such as a silicone elastomer or a rubber, may also be provided, which is resiliently deformable per se. There may also be no casting of the winding arrangement, or a casting in a non-resilient material, wherein the casting of individual sub-windings is performed separately and the sub-windings together with the respective casting material are movable relative to one another. Such configurations will be discussed in greater detail further below. FIG.5, in a perspective view, shows a substantially hollow-cylindrical winding arrangement, which consists of a plurality of sub-windings. Each sub-winding consists of a plurality of windings of a lead and has two electrical terminals for voltage supply and current feed. The winding arrangement as a whole may also have terminal leads or electrical terminals. Each sub-winding of the illustrated winding arrangement in the unrolled state has a rhombic basic shape. The individual sub-windings overlap one another in the circumferential direction of the winding arrangement. The individual sub-windings7,8of the winding arrangement fromFIG.5have electrical terminals9,10for supplying a current to the stator winding arrangement. InFIG.6an individual sub-winding7is illustrated, symbolized by an individual winding of a winding lead, and is designated by reference sign11. The sub-winding11has two electrical connections12,13for supplying a current. InFIG.6a hollow cylinder is illustrated schematically, over the circumference of which the part-cylindrical sub-windings are distributed in a manner overlapping one another and offset relative to one another in the circumferential direction. InFIG.7a plurality of sub-windings7,8of a winding arrangement are shown schematically in a view in the axial direction. The individual sub-windings7,8each have a radially external part7aand a radially inner part7b, wherein each radially inner part is overlaid by the following sub-winding8, more specifically by the radially external part thereof. In this way, a roof-tile-like nesting of the sub-windings is provided along the circumferential line of the stator. If the sub-windings are movable relative to one another, these can be slid further over one another in a shingle-like manner, and therefore the diameter of the overall arrangement and the circumference of the winding arrangement can be reduced. An example of a compressed state of such a compression movement is shown inFIG.8, in which in each case two windings7,8are slid one over the other, such that they overlap one another completely in the circumferential direction of the winding arrangement. This slidability of the individual sub-windings over one another is conceivable with non-cast sub-windings and also with cast sub-windings. If the individual sub-windings are cast, it is advantageous if the casting material enables easy sliding relative to one another of two bodies consisting thereof. FIG.9, in a longitudinal section, shows a motor having a radially expanded stator2and a rotor1, which has an encapsulation14in the form of a hollow cylinder, which surrounds the magnet body of the rotor and which for example also carries the bearings15,16. The shaft3of the rotor is mounted with little friction in the bearings15,16, which may be formed as plain or ball bearings. The diameter of the overall structure of the motor according toFIG.9in the expanded, assembled state ready for operation is specified by D. By contrast, the same motor having the same elements, i.e., a rotor encapsulated within an encapsulation14and a stator2having a winding arrangement, is illustrated inFIG.10in the compressed state, wherein the stator2has been displaced to such an extent in the axial direction relative to the rotor1that the rotor is located outside the stator. The stator2is then radially compressible independently of the rotor1up to the outer diameter of the rotor. FIG.11shows a design of a motor having a rotor1′ and a stator2′, wherein these are illustrated in the compressed position pulled apart axially from one another. The rotor1′ has an encapsulation, in which the magnet arrangement of the rotor, supported by two bearings, can rotate. The encapsulation of the rotor has a conical tapered portion17and a connection to a strand-shaped manipulation element18, which is fastened to the encapsulation or to a bearing and enables an axial relative movement of the rotor relative to the stator2′. At the same time, the stator2′ is connected to a second manipulation element19, for example in the form of a tube or hose, through which for example the manipulation element18can be guided. The manipulation elements18,19, which jointly form a connection element to the motor, can be jointly actuated from a remote location in order to perform a relative movement of the stator and rotor relative to one another and for example to radially expand these by means of the insertion of the encapsulation of the rotor1′ into the winding arrangement of the stator2′. FIG.12shows a particular winding arrangement consisting of four separate sub-windings20,21,22,23each cast separately in a resilient material. Each of these sub-windings is formed as part of a hollow cylinder, and the sub-windings can be assembled with their cast bodies to form an overall hollow cylinder. If a force is exerted onto the winding arrangement radially from the outside, the constellation as illustrated inFIG.13is provided, wherein the individual cast bodies and sub-windings turn radially inwardly. The individual cast bodies of the sub-windings may be connected movably to one another for example by living hinges. In the state illustrated inFIG.13, the winding arrangement in the radial direction already occupies a much smaller space than in the form illustrated inFIG.12. With a further radial compression, the individual sub-windings are further compressed radially inwardly, which is additionally made possible by a deformability of the cast body. With full compression, the form illustrated inFIG.14is provided. This can be automatically expandable back into the form illustrated inFIG.12with cancellation of the radially inwardly acting compression forces, wherein the restoring forces can be applied for example by the resiliently deformed cast bodies, but also by the winding leads themselves, or by both, jointly. If the individual sub-windings are not cast, a corresponding deformation of the winding leads can also take place reversibly within each sub-winding.FIG.15, in an illustration in the axial direction, shows two magnets24,25arranged at right angles on one another, which can be driven in the magnetic field of the winding arrangement. The magnets24,25are fixedly connected to the shaft3of the rotor. FIG.16illustrates the division of a magnet24along the surface26, whereby two segments24a,24bof the magnet24are produced, which each form a magnet element. The magnet24in the state illustrated by solid lines, has the form of a cuboid. The constellation in which the segment24is displaced in the axial direction4along the surface26relative to the segment24bis shown in a dashed manner. An extension of the magnet24is provided in the axial direction, and a compression from the diameter D to the diameter d, as specified inFIG.16in the right-hand part, is provided in a radial direction. As a result of the shown construction of the rotor, this can also be radially compressible, such that the motor is compressible either in the assembled state by joint compression of stator and rotor or is also compressible merely by radial compression of the stator in the state pulled apart axially. The motor can therefore be compressed in order to be brought to its site of use; for example, it may be implantable as a drive apparatus for a blood pump and may be displaced through a blood vessel in the compressed state within a patient body to a site of use. There, the motor can be expanded, as can a blood pump for example, and the motor in the expanded state can build up the necessary torques or the required power to drive a pump. FIG.17shows a design variant in which the winding arrangement is divided into a plurality of sub-windings27,28, which are each constructed in the form of a circular ring and, arranged axially in succession, form a hollow-cylindrical winding arrangement. If, in such an arrangement, the circular-annular sub-windings are tilted, the cross section of the winding means is thus elliptical, but is compressed in diameter in an axis29relative to the untilted arrangement. In the axis30arranged at right angles hereto, the diameter remains the same. Nevertheless, a form more favorable for the positioning of the motor may be provided with the tilted arrangement. The tilting can be reversed at any moment following the positioning of the motor. FIG.18shows a device consisting of a stator according toFIG.7having the windings7,8already described there. A magnetic rotor1″, to which a radially compressible pump rotor29is connected in such a way that it can rotate jointly with the rotor1″ about the same axis, is located in this stator. In the exemplary embodiment the pump rotor is formed from a resilient, preferably hyperelastic plastics material, which makes it possible for the pump rotor29to collapse in the event of compression of the stator and to resiliently or hyperelastically expand back again into the starting form in the event of expansion of the stator. FIG.19schematically illustrates how the stator compresses similarly toFIG.8, wherein the pump rotor29likewise assumes a compressed form. Here, the blades of the pump rotor fold around the axis of the rotor and bear against the hub of the pump rotor. In principle, the device may also be formed such that the rotor1″ together with the pump rotor29can be axially removed from the rotor in a manner corresponding toFIG.20. The pump rotor29and the stator are thus arranged axially in succession. In this state the stator and the pump rotor can be jointly compressed, which enables a further reduction of the compressed diameter compared with the embodiment inFIG.19. The pump rotor can be formed in principle in very different ways. Besides the variant formed from resilient or hyperelastic plastics material shown inFIGS.18and19, various other variants are known from the prior art, for example from U.S. Pat. Nos. 4,753,221; 5,749,855; 7,393,181; US 2009/0062597 A1; EP 2047873 A1; US 2011/0275884 A1; EP 2229965 A1; WO 2010 149393 A1; EP 2299119 A1; EP 2338540 A1; EP 2338541 A1; EP 2363157; EP 2407185 A1; EP 2407187 A1; EP 2407186 A1. | 15,544 |
11863038 | DETAILED DESCRIPTION The stator illustrated inFIG.1is a stator10of a permanent-magnet-excited rotating field machine20. The stator10is configured for rotation about an axis of rotation, from which a plurality of individual-tooth winding carriers40extend away in a radial direction50. The individual-tooth winding carriers40carry individual-tooth windings60, which are wound around the individual-tooth winding carriers40in individual turns70. The turns70are formed with a flat conductor that has, for example, a height of one millimeter and a width of four millimeters. In this case, the flat conductor is arranged such that the height thereof extends in the radial direction50, and the width thereof extends in the circumferential direction (e.g., in a direction perpendicular to the radial direction50and to a direction30parallel to the axis of rotation). The turns70are wound in contact with one another around individual-tooth winding carriers40in the radial direction50. In the radial direction50, the individual-tooth windings60have, for example, ten turns70and consequently a dimension of, for example, about 10 millimeters. The individual-tooth winding carriers40are spaced apart by, for example, about 11.3 millimeters in the circumferential direction, opening up between the individual-tooth windings a cooling channel80that is, for example, 3.3 millimeters wide in the circumferential direction and by which the individual-tooth windings60may be cooled. During the operation of the rotating field machine20, there is a flow of synthetic oil forming an electric insulator through the cooling channel80in order to cool the individual-tooth windings60. The individual-tooth windings60are electrically insulated with respect to the individual-tooth winding carriers40by slot insulation90. In the illustrative embodiment illustrated, the slot insulation90is formed by a surface insulating material (e.g., an aramid laminate). The individual turns70of the individual-tooth windings60are electrically insulated from one another by turn insulators70, thus making it possible to avoid voltage flashovers100. The stator10′ illustrated inFIG.3is a stator10′ of a permanently excited rotating field machine20′ according to the present embodiments. The stator10′ is likewise rotatable about an axis of rotation and, like the stator10described above, has individual-tooth winding carriers40′ for individual-tooth windings60′. According to the present embodiments, the winding carriers40′ each have a comb-like element110having mutually spaced recesses120, as illustrated inFIG.4. The recesses120of the comb-like element110are configured to accommodate turns70′ of the individual-tooth windings60′ and to space the turns70′ apart. Like the turns70described above, the turns70′ of the individual tooth windings60′ are formed with a flat conductor that has a height of, for example, one millimeter and a width of, for example, four millimeters. In this case, the flat conductor is likewise arranged such that the height thereof extends in the radial direction50and the width thereof extends in the circumferential direction (e.g., in a direction perpendicular to the radial direction50and to a direction30parallel to the axis of rotation). In this case, the turns70′ of the individual-tooth windings60′ are spaced apart by the comb-like element110such that the turns70′ enclose between the turns70′ a clear width of, for example, 0.3 millimeters. Owing to this additional clear width, the individual-tooth windings60′ have an extent that is, for example, a few millimeters greater in the radial direction50. Owing to the spacing of the turns70′ of the individual-tooth windings60′ in the radial direction50, the turns70′ open up radially between the turns70′ cooling channels130, through which cooling fluid in the form of synthetic oil, which forms an electric insulator, flows in order to cool the individual-tooth windings60′. Owing to the opening up of cooling channels130between the turns70′, it is possible to dispense with a cooling channel80in the circumferential direction between the individual-tooth winding carriers40′. Instead, as shown inFIGS.3and4, it is possible to provide an insulator140that requires less than, for example, 2 millimeters of the gap provided between the individual-tooth windings60′. The insulator140is formed as a dividing wall composed of the same material as the slot insulation90. The dividing wall extends in a plane (e.g., in the radial direction50and in the direction30parallel to the axis of rotation). Owing to the smaller gap in the circumferential direction provided between the individual-tooth windings60′, the individual-tooth windings60′ are spaced apart by less than, for example, 2.25 times the width of the flat conductors of the turns70′. In the illustrative embodiment, the turns70′ are not specially insulated but are electrically insulated to a sufficient extent by the spacing apart thereof and the synthetic oil flowing without bubbles through the cooling channels130. In other illustrative embodiments, not specially illustrated, there may also be insulation of the turns70′. As illustrated inFIG.5, the hybrid electric airplane400according to the present embodiments has a rotating field machine20′ according to the present embodiments, having a stator10′, for driving a propeller410and/or a power unit (not illustrated in the drawing). The elements and features recited in the appended claims may be combined in different ways to produce new claims that likewise fall within the scope of the present invention. Thus, whereas the dependent claims appended below depend from only a single independent or dependent claim, it is to be understood that these dependent claims may, alternatively, be made to depend in the alternative from any preceding or following claim, whether independent or dependent. Such new combinations are to be understood as forming a part of the present specification. While the present invention has been described above by reference to various embodiments, it should be understood that many changes and modifications can be made to the described embodiments. It is therefore intended that the foregoing description be regarded as illustrative rather than limiting, and that it be understood that all equivalents and/or combinations of embodiments are intended to be included in this description. | 6,389 |
11863039 | DESCRIPTION OF EMBODIMENTS Hereinafter, embodiments of the present invention will be described in detail by referring to the accompanying drawings. An orthogonal coordinate system (X, Y, Z) is used for describing the embodiments. The same orthogonal coordinate system (X, Y, Z) is used also in illustrations of the drawings to be described later. Hereinafter, the width, length, and height of vibration actuator10are lengths in X-direction, Y-direction, and Z-direction, respectively. In addition, descriptions will be provided assuming that the plus side in the Z-direction is “upper side” and the minus side in the Z-direction is “lower side”. Embodiment 1 <Entire Configuration of Vibration Actuator10> FIG.1is a plane-side appearance perspective view of a vibration actuator according to Embodiment 1 of the present invention,FIG.2is a bottom-surface side appearance perspective view of the vibration actuator according to Embodiment 1 of the present invention,FIG.3is a plan view of the vibration actuator according to Embodiment 1 of the present invention,FIG.4is a sectional view taken along line A-A inFIG.3, andFIG.5is an exploded perspective view of the vibration actuator according to Embodiment 1 of the present invention. Vibration actuator10illustrated inFIG.1toFIG.5achieves a vibrating function of an electronic device by being mounted on the electronic device as a vibration generation source of touch panel140(seeFIG.17) that is an example of an operation contact surface part. Vibration actuator10in Embodiment 1 is mounted on a touch panel apparatus (seeFIG.17) of a car navigation system as an electronic device and functions as a vibration presenting apparatus that presents vibration to a user of touch panel140. Note that touch panel apparatus100is an example of the vibration presenting apparatus and, in Embodiment 1, includes touch panel140as a panel that can be touched by the user via hands, fingers, and the like. Touch panel140may be a panel having a display function for displaying images and the like the user can touch or may be a configuration having no display function but simply having an operation contact surface part that can be operated by touch of the user. Vibration actuator10according to Embodiment 1 is mounted on touch panel (operation contact surface part)140(seeFIG.17) that displays images, for example. In this case, vibration actuator10is a configuration applied to touch panel apparatus100that is capable of allowing the user who touches touch panel140to perform intuitive operations by transmitting vibration in response to touch operations on the screen to the user to feel bodily sensations. Note that touch panel140of touch panel apparatus100includes a contact position output part that receives a touch operation of the user on touch panel140and outputs the contact position thereof. Vibration actuator10is joined with touch panel140, receives a driving signal from a control part (not illustrated), generates and drives vibration corresponding to the contact position outputted from touch panel140and transmits the vibration to touch panel140to directly vibrate touch panel140. That is, vibration actuator10receives the operation of the user performed on touch panel140, and is driven accordingly. Vibration actuator10includes: fixing part30that includes core assembly20formed by winding coil22around core24, and base part32; movable part40that includes magnetic yoke41; and plate-shaped elastic parts50(50-1,50-2). Plate-shaped elastic parts50(50-1,50-2) elastically support movable part40to be movable in vibrating direction with respect to fixing part30. Vibration actuator10vibrates yoke41of movable part40with core assembly20. Specifically, movable part40is vibrated with the attraction force of energized coil22and core24excited by energized coil22as well as the urging force by plate-shaped elastic parts50(50-1,50-2). Vibration actuator10is formed in a flat shape having the Z-direction as the thickness direction. Vibration actuator10vibrates movable part40in the Z-direction with respect to fixing part30, that is, by having the thickness direction as the vibrating direction to bring closer or away one of top and back surfaces spaced apart from each other in the thickness direction of vibration actuator10itself with respect to the other surface in the Z-direction. In Embodiment 1, vibration actuator10moves movable part40to −Z-direction by the attraction force of core24, and moves movable part40in +Z-direction by the urging force of plate-shaped elastic parts50(50-1,50-2). In vibration actuator10of Embodiment 1, movable part40is elastically supported by a plurality of plate-shaped elastic parts50(50-1,50-2) that are disposed along the direction orthogonal to the Z-direction at point symmetrical positions with respect to the moving center of movable part40. However, the configuration is not limited thereto. Plate-shaped elastic part50is fixed between movable part40and fixing part30, includes an elastically deformable bellows-shaped part, and elastically supports movable part40in a movable manner with respect to fixing part30in the direction opposing to at least one end out of both ends (magnetic pole parts242,244) of core24. With such configuration, how plate-shaped elastic parts50are provided is not an issue. For example, plate-shaped elastic part50may elastically support movable part40with respect to fixing part30(core assembly20) to be movable in the direction opposing to one end (magnetic pole part242or magnetic pole part244) of core24. Further, plate-shaped elastic parts50-1,50-2may be disposed line symmetrically with respect to the center (moving center) of movable part40, and two or more plate-shaped elastic parts50may be used as well. Each of plate-shaped elastic parts50-1and50-2is fixed to fixing part30at one end side and fixed to movable part40at the other end side to movably support movable part40with respect to fixing part30in the vibrating direction (Z-direction, and it is up-and-down direction herein). <Fixing Part30> As illustrated inFIG.5, fixing part30includes: core assembly20including coil22and core24; and base part32. Base part32has core assembly20fixed thereto, is connected to movable part40via plate-shaped elastic parts50(50-1,50-2), and supports movable part40to be movable in the vibrating direction. Base part32is a flat-shape member, and forms the bottom surface of vibration actuator10. Base part32includes attaching parts32ato which one end of each of plate-shaped elastic parts (50-1,50-2) is fixed to sandwich core assembly20. Each of attaching parts32ais disposed with a same space provided from core assembly20. Note that the space is a space to be a deforming area of plate-shaped elastic parts50(50-1,50-2). Attaching part32aincludes fixing holes321for fixing plate-shaped elastic parts50(50-1,50-2) and fixing holes322for fixing base part32to a base material. Fixing holes322are provided at both ends of attaching part32aby sandwiching fixing holes321. Thereby, base part32is fixed to the base member (for example, back surface plate120illustrated in FIGS.17) in a fully stable manner. Base part32in Embodiment 1 is formed by processing a sheet metal and configured such that one side part and the other side part as attaching parts32are spaced apart from each other in the width direction with bottom surface part32binterposed therebetween. Between attaching parts32a, provided is a recessed part having bottom surface part32blower in height than that of attaching parts32a. Inside the recessed part, that is, the space on the top surface side of bottom surface part32bis for securing the elastic deforming area of plate-shaped elastic parts50(50-1,50-2), and for securing a movable area of movable part40supported by plate-shaped elastic parts50(50-1,50-2). Bottom surface part32bis a rectangular shape, opening part36is formed in the center thereof, and core assembly20is located inside opening part36. Core assembly20is fixed while being partially inserted into opening part36. Specifically, split body26bof bobbin26on the lower side of core assembly20and a lower-side part of coil22are inserted inside opening part36, and core assembly20is fixed such that core24is located on bottom surface part32bon a side view. Thereby, length (thickness) in the Z-direction becomes decreased compared to a configuration where core assembly20is attached on bottom surface part32b. Further, because a part of core assembly20, that is, a part of the bottom surface side herein, is fixed while being fitted into opening part36, core assembly20is firmly fixed in a state where it is hard to come off from bottom surface part32b. Opening part36is in a shape contoured to the shape of core assembly20. Opening part36in Embodiment 1 is formed in a square shape. Thereby, entire vibration actuator10can be shaped substantially into a square shape on a plan view by disposing core assembly20and movable part40in the center of vibration actuator10. Note that opening part36may be a rectangular shape (including a square shape). Core assembly20vibrates (reciprocal linear motion in the Z-direction) yoke41of movable part40in cooperation with plate-shaped elastic parts50(50-1,50-2). Core assembly20in Embodiment 1 is formed in a rectangular plate-shaped. Magnetic pole parts242and244are disposed in both side portions of the rectangular plate-shaped spaced from each other in the longitudinal direction. Magnetic pole parts242,244are disposed to oppose to bottom surfaces of attracted surface parts46,47of movable part40with gap G (seeFIG.6) provided therebetween in the X-direction, and counter surfaces (counter surface parts)20a,20bas the upper surfaces oppose to the bottom surfaces of attracted surface parts46,47of yoke41in the vibrating direction of movable part40. Core assembly20in Embodiment 1 is formed in a rectangular plate-shaped, and includes magnetic pole parts242and244at the both side portions spaced from each other in the longitudinal direction. Magnetic pole parts242and244are disposed to oppose to attracted surface parts46and47of movable part40with gap G provided therebetween in the Z-direction. As illustrated inFIG.1andFIG.3, core assembly20is fixed to base part32with a winding axis of coil22aligned toward the opposing direction of spaced attaching parts32ain base part32. Core assembly20in Embodiment 1 is disposed in the center of base part32, specifically in the center of bottom surface part32b. Core assembly20is configured by winding coil22around circumference of core24via bobbin26. As illustrated inFIG.4, core assembly20is fixed to bottom surface part32bsuch that core24is located on the bottom surface over opening part36while being in parallel to bottom surface part32b. Core assembly20is fixed by screw68as a fastening member (seeFIG.1,FIG.3toFIG.7) in a state where coil22and the part (core main body241) to which coil22is wound are located within opening part36of base part32. Specifically, core assembly20is fixed to bottom surface part32bby fastening screw68via fixing hole28and fastening hole33(seeFIG.5) of bottom surface part32bin a state where coil22is disposed in opening part36. Core assembly20and bottom surface part32bare joined at two points on the axial center of coil22by sandwiching coil22with both side parts of opening part36spaced from each other in the X-direction and magnetic pole parts242,244via screws68. Coil22is a solenoid that is energized and generates a magnetic field at the time of driving vibration actuator10. Coil22together with core24and movable part40forms a magnetic circuit (magnetic path) that attracts and moves movable part40. Note that power is supplied to coil22from an external power source via a control part, not illustrated. For example, through supplying a driving signal to the control part, the power is supplied to coil22to drive vibration actuator10. Core24includes: core main body241around which coil22is wound; and magnetic pole parts242,244provided at both ends of core main body241excited by energizing coil22. Core24may be in any types of configuration as long as it is a configuration having the length with which the both ends can function as magnetic pole parts242,244when coil22is energized. For example, while it is possible to employ a straight-type (I-type) tabular shape, core24of Embodiment 1 is formed in an H-type tabular shape on a plan view. When formed as an I-type core, the area of surfaces (gap side surface) on attracted surface parts46,47side opposing to the both ends (magnetic pole parts) of the I-type core with gap (air gap) G provided therebetween becomes narrower. Thereby, magnetic resistance in the magnetic circuit may be increased, so that the conversion efficiency may be deteriorated. Further, there may be no space for positioning of the bobbins or may only be a small space in the longitudinal direction of the core for attaching the bobbins to the core, so that it is necessary to provide the space for positioning separately. In the meantime, because core24is the H-type, the gap side surface in the both ends of core main body241can be expanded in the front-and-rear directions (Y-directions) longer than the width of core main body241around which coil22is wound, thereby making it possible to decrease the magnetic resistance and improve the efficiency of the magnetic circuit. Further, positioning of coil22can be performed by simply fitting bobbins26between portions of magnetic pole parts242,244extended out from core main body241, so that it is unnecessary to separately provide a positioning member of bobbins26for core24. In core24, magnetic pole parts242and244are provided at each of the both ends of tabular core main body241around which coil22is wound by being projected toward the direction orthogonal to the winding axis of coil22. Core24is of a magnetic material, and formed from a silicon steel sheet, permalloy, ferrite, or the like. Further, core24may also be made of electromagnetic stainless steel, a sintered material, an MIM (metal injection mold) material, a laminated steel sheet, an electrogalvanized steel sheet (SECC), or the like. Each of magnetic pole parts242and244is provided by being projected in the Y-direction from both opening parts of coil22. Magnetic pole parts242and244are excited by energizing coil22, attracts and moves yokes41of movable part40spaced in the vibrating direction (Z-direction). Specifically, magnetic pole parts242and244attract, by a magnetic flux to be generated, attracted surface parts46and47of movable part40counter-disposed via gap G. Magnetic pole parts242and244are tabular bodies extended in the Y-direction that is the vertical direction with respect to core main body241extended in the X-direction. Magnetic pole parts242and244are lengthy in the Y-direction, so that the area of counter surfaces20aand20bopposing to yokes41are wider than the configuration formed in the both ends of core main body241. Magnetic pole parts242and244have fixing holes28formed in the center thereof in the Y-direction, and are fixed to base part32by screws68inserted into fixing holes28. Bobbin26is disposed to surround core main body241of core24. Bobbin26is formed from a resin material, for example. This makes it possible to secure electrical insulation with other metallic members (for example, core24), so that reliability as the electric circuit can be improved. By using a resin of high fluidity for the resin material, formability can be improved so that the thickness can be decreased while securing the strength of bobbin26. Through mounting split bodies26aand26bto sandwich core main body241, bobbin26is formed in a cylindrical shape that covers the periphery of core main body241. In bobbin26, a flange is provided to the both ends of the cylindrical body to regulate so that coil22comes to be located on the outer circumference of core main body241. <Movable Part40> Movable part40is disposed to oppose to core assembly20with gap G provided therebetween in the direction orthogonal to the vibrating direction (Z-direction). Movable part40is provided to be able to reciprocally vibrate in the vibrating direction with respect to core assembly20. Movable part40includes yokes41, and includes movable-body side fixing parts54of plate-shaped elastic parts50-1and50-2fixed to yokes41. Movable part40is disposed by being hanged while being spaced substantially in parallel and to be movable in the approaching/leaving directions (Z-directions) with respect to bottom surface part32bvia plate-shaped elastic parts50(50-1,50-2). Yoke41is a tabular body made of a magnetic material such as electromagnetic stainless steel, a sintered material, an MIM (metal injection mold) material, a laminated steel sheet, an electrogalvanized steel sheet (SECC), or the like. Yoke41in Embodiment 1 is formed by processing an SECC sheet. Yokes41are hanged to oppose to core assembly20with gap G (seeFIG.6) provided therebetween in the vibrating direction (Z-direction) by plate-shaped elastic parts50(50-1,50-2) fixed to respective attracted surface parts46and47spaced from each other in the X-direction. Yoke41includes: surface-part fixing part44to which an operation contact surface part (see touch panel140illustrated inFIG.17) is attached; and attracted surface parts46and47counter-disposed with respect to magnetic pole parts242and244. Yoke41in Embodiment 1 includes opening part (fixing-part side opening part)48in the center thereof. Yoke41has a rectangular frame shape. Yoke41is formed in a frame shape that surrounds opening part48with surface-part fixing part44and attracted surface parts46,47. Opening part48opposes to coil22. In Embodiment 1, opening part48is located right above coil22, and the opening shape of opening part48is formed in a shape to which coil22part of core assembly20can be inserted when yoke41moves to bottom surface part32bside. By configuring yoke41to have opening part48, the thickness of the entire vibration actuator can be decreased compared to a case having no opening part48. Further, core assembly20is located within opening part48, so that yoke41is not disposed in the vicinity of coil22. Therefore, it is possible to suppress deterioration in the conversion efficiency due to the magnetic flux leaked from coil22, so that high output can be achieved. Surface-part fixing part44includes fixing surface44athat comes in surface-contact to fix touch panel140as an example of the operation contact surface part. Fixing surface44aforms a trapezoid shape on a plan view, and surface-contacts with touch panel140that is fixed to surface-part fixing part44via fastening member such as a screw inserted into surface-part fixing hole42. Attracted surface parts46,47are attracted to magnetized magnetic pole parts242,244in core assembly20, and plate-shaped elastic parts50(50-1,50-2) are fixed thereto. Movable-body side fixing parts54of plate-shaped elastic parts50-1and50-2are fixed by being laminated, respectively, on attracted surface parts46and47. Attracted surface parts46and47are provided with notches49functioning as clearance of the heads of screws64of core assembly20when moved to bottom surface part32bside. Thereby, even when movable part40moves to bottom surface part32bside and attracted surface parts46,47approach magnetic pole parts242,244, magnetic pole parts242,244are not to be in contact with screws68that fix magnetic pole parts242,244to bottom surface part32b, so that a movable area of yoke41in the Z-direction can be secured for that. <Plate-Shaped Elastic Parts50(50-1,50-2)> Plate-shaped elastic parts50(50-1,50-2) support movable part40to be movable with respect to fixing part30. Plate-shaped elastic parts50(50-1,50-2) support the upper surface of movable part40to be the same height as that of the upper surface of fixing part30or to be on a lower surface side than the upper surface of fixing part30(upper surface of core assembly20in Embodiment 1) to be in parallel to each other. Note that plate-shaped elastic parts50-1,50-2have symmetrical shapes with respect to the center of movable part40and, in Embodiment 1, are members formed in the same manner. Plate-shaped elastic parts50are disposed such that yoke41is substantially in parallel to oppose to magnetic pole parts242,244of core24of fixing part30with gap G provided therebetween. Plate-shaped elastic parts50support the lower surface of movable part40to be movable in the vibrating direction at a position closer to bottom surface part32bside than the level substantially the same as the height level of the upper surface of core assembly20. Plate-shaped elastic part50is a plate spring including fixing-body side fixing part52, movable-body side fixing part54, and bellows-like elastic arm part56that communicates fixing-body side fixing part52with movable-body side fixing part54. Plate-shaped elastic parts50attaches movable part40while attaching fixing-body side fixing part52to the top surface of attaching part32a, attaching movable-body side fixing part54to the top surface of attracted surface parts46,47of yoke41, and having bellows-like elastic arm part56in parallel to bottom surface part32b. Fixing-body side fixing part52surface-contacts with attaching part32aby being joined and fixed by screws62, and movable-body side fixing part54surface-contacts with attracted surface parts46,47by being joined and fixed by screws64. Bellows-like elastic arm part56is an arm part having a bellows-shaped part. By having the bellows-shaped part, bellows-like elastic arm part56secures the length that allows deformation required for vibration of movable part40between fixing-body side fixing part52and movable-body side fixing part54and also on the surface orthogonal to the vibrating direction (surface formed in the X-direction and the Y-direction). Bellows-like elastic arm part56in Embodiment 1 extends in the opposing direction of fixing-body side fixing part52and movable-body side fixing part54and folds back, and the ends that are joined, respectively, to fixing-body side fixing part52and movable-body side fixing part54are formed at positions shifted in the Y-direction. Bellows-like elastic arm parts56are disposed at point-symmetrical or line-symmetrical positions with respect to the center of movable part40. Thereby, movable part40is supported from both sides by bellows-like elastic arm parts56having bellows-shaped springs, so that it is possible to disperse the stress at the time of elastic deformation. That is, plate-shaped elastic parts50can move movable part40in the vibrating direction (Z-direction) without tilting with respect to core assembly20, thereby making it possible to improve reliability of the vibrating state. Each of plate-shaped elastic parts50includes at least two or more bellows-like elastic arm parts56. Thereby, compared to a case where there is only one each of bellows-like elastic arm part56, it is possible to improve the reliability of the vibrating state by dispersing the stress at the time of elastic deformation and to improve the stability because the support for movable part40can be well-balanced. The plate spring as plate-shaped elastic part50in Embodiment 1 is formed from a magnetic material. Further, movable-body side fixing parts54of plate-shaped elastic parts50are disposed at positions opposing to both ends (magnetic pole parts242,244) of core24in the coil winding axis direction or on the upper side thereof and function as a magnetic path. In Embodiment 1, movable-body side fixing parts54are fixed by being laminated on the upper side of attracted surface parts46and47. This makes it possible to increase thickness H (seeFIG.6) of attracted surface parts46and47opposing to magnetic pole parts242,244of core assembly20as the thickness of the magnetic material. The thickness of plate-shaped elastic parts50and the thickness of yoke41are the same, so that the cross sectional area of the magnetic material portion opposing to magnetic pole parts242,244can be doubled. This makes it possible to expand the magnetic circuit, to ease the deterioration in the property of the magnetic circuit due to magnetic saturation, and to improve the output compared to a case where the plate spring is nonmagnetic. Note that vibration actuator10of Embodiment 1 may be provided with a detection part that detects push-in amount of movable part40when the operation surface part fixed by surface-part fixing part44is operated. For example, as illustrated inFIG.6, strain detection sensor70that detects strain of plate-shaped elastic parts50may be provided as a detection part. Strain detection sensor70detects strain of plate-shaped elastic parts50that are deformed when surface-part fixing part44is pushed into bottom surface part32bside. Detected strain is outputted to the control part and the like, coil22is energized to attract and move yoke41such that movable part40moves in an amount corresponding to the strain. Embodiment 1 can function without determining the moving amount of the operation contact surface part to be operated, as long as contact to the operation contact surface part can be detected. Further, a more natural sense of touch can be expressed when the push-in amount with respect to plate-shaped elastic parts50can be detected with the moving amount corresponding to the actual moving amount on the operation contact surface part. Strain detection sensor70is attached between heads of screws62and64on bellows-like elastic arm parts56of plate-shaped elastic parts50, and disposed in the so-called dead space that is an area not obstructing other members. Further, as illustrated inFIG.7, it is also possible to dispose the detection part for detecting push-in in a lower part of plate-shaped elastic parts50as the dead space. In that case, the detection sensor is electrostatic capacitance sensor80for detecting the push-in amount and disposed on bottom surface part32bopposing to plate-shaped elastic parts50. The distance with respect to plate-shaped elastic parts50displaced by being pushed in is measured. Thereby, the distance when deformed by following the push-in on the operation contact surface part can be measured. With such method using the electrostatic capacitance, it is possible to detect fluctuation in plate-shaped elastic parts50or movable part40on the lower side of plate-shaped elastic parts50. In addition, it is also possible to achieve detection of the push-in amount of the operation contact surface part and to generate vibration of movable part40by corresponding to the push-in amount while maintaining the external dimension of vibration actuator10. FIG.8is a diagram illustrating the magnetic circuit of vibration actuator10. Note thatFIG.8is a perspective view of vibration actuator10cut along line A-A ofFIG.3and, in the magnetic circuit, there are also magnetic flux flows M similar to the illustration thereof existing in the part having no such illustration. Further,FIG.9are sectional views schematically illustrating move of movable part40caused by the magnetic circuit.FIG.9Ais a diagram illustrating a state where movable part40is held by plate-shaped elastic parts50at a position spaced from core assembly20, andFIG.9Billustrates movable part40attracted and moved toward core assembly20side by a magneto motive force generated by the magnetic circuit. Specifically, when coil22is energized, core24is excited and a magnetic field is generated, thereby forming magnetic poles in both ends of core24. For example, as illustrated inFIG.8, in core24, magnetic pole part242is the N-pole, and magnetic pole part244is the S-pole. Thereby, the magnetic circuit indicated by magnetic flux flow M is formed between core assembly20and yoke41. Magnetic flux flow M in the magnetic circuit flows to attracted surface part46of opposing yoke41from magnetic pole part242, passes through surface-part fixing part44of yoke41, and reaches magnetic pole part244opposing to attracted surface part47from attracted surface part47. In Embodiment 1, plate-shaped elastic parts50are also of magnetic materials. Thereby, the magnetic flux (illustrated as magnetic flux flow M) flown to attracted surface part46passes through attracted surface part46of yoke41and movable-body side fixing part54of plate-shaped elastic parts50-1, reaching attracted surface part47and both ends of movable-body side fixing part54of plate-shaped elastic part50-2via surface-part fixing part44from both ends of attracted surface part46. Thereby, according to the principle of electromagnetic solenoid, magnetic pole parts242,244of core assembly20generate attraction force F for attracting attracted surface parts46,47of yoke41. Thereupon, attracted surface parts46,47of yoke41are attracted to both of magnetic pole parts242,244of core assembly20, coil22is inserted into opening part48of yoke41, and movable part40including yoke41moves in F-direction against the urging force of plate-shaped elastic parts50(seeFIG.9AandFIG.9B). In the meantime, when energization to coil22is stopped, the magnetic field disappears, attraction force F of core assembly20for movable part40is lost, and movable part40is moved back to the original position (moved to —F-direction) by the urging force of plate-shaped elastic parts50. By repeating such action described above, in vibration actuator10, movable part40reciprocally moves and generates vibration in the vibrating direction (Z-direction). In vibration actuator10, it is possible to increase the efficiency of the magnetic circuit and achieve high output by disposing attracted surface parts46,47of yoke41adjacent to magnetic pole parts242,244of core assembly20. Further, vibration actuator10uses no magnet, so that a low-cost configuration can be achieved. The bellows-shaped springs that are plate-shaped elastic parts50(50-1,50-2) enable dispersion of the stress, so that the reliability can be improved. Especially, because movable part40is supported by a plurality of plate-shaped elastic parts50(50-1,50-2), more effective dispersion of the stress is possible. As described, vibration actuator10is capable of providing a more direct sense of touch by the drive of up-and-down direction. By fixing core24around which coil22is wound and core assembly20to fixing part30, movable part40is supported to be movable. Thereby, it becomes unnecessary to provide a magnetic generating part in the Z-direction, and design becomes simple because the supporting structure is simple. Thus, space can be saved, so that it is possible to decrease the thickness of vibration actuator10. Hereinafter, the driving principle of vibration actuator10will simply be described. Note that it is the same for vibration actuators10A,10B of Modification Examples 1, 2 to be described later, and vibration actuators10,10A,10B can be driven by generating a resonance phenomenon with a pulse by using following motion equation and circuit equation. The actions are not resonance driven but for expressing operational feeling of mechanical switches displayed on the operation contact surface part, and it is also possible to drive the vibration actuator by generating any types of vibration without using a short pulse while the vibration actuator in Embodiment 1 is driven by inputting a short pulse via a control part, not illustrated. Examples of the mechanical switch may be a tactile switch, alternate-type switch, a momentary switch, a toggle switch, a slide switch, a rotary switch, a DIP switch, and a rocker switch. Note that movable part40in vibration actuator10performs reciprocal motions based on Expressions (1) and (2). [Expression1]md2x(t)dt2=Kfi(t)-Kspx(t)-Ddx(t)dt(1)m: Mass [kg]x(t): Displacement [m]Kf: Thrust constant [N/A]i(t): Current [A]Ksp: Spring constant [N/m]D: Damping coefficient [N/(m/s)] [Expression2]e(t)=Ri(t)+Ldi(t)dt+Kedx(t)dt(2)e(t): Voltage [V]R: Resistance [Ω]L: Inductance [H]Ke: Counter electromotive force constant [V/(rad/s)] That is, mass “m” [kg], displacement “x(t)” [m], thrust constant “Kf” [N/A], current “i(t)” [A], spring constant “Ksp” [N/m], and damping coefficient “D” [N/(m/s)] in vibration actuator10can be changed as appropriate within the range satisfying Expression (1). Also, voltage “e(t)” [V], resistance “R” [Ω], inductance “L” [H], and counter electromotive force constant “Ke” [V/(rad/s)] can be changed as appropriate within the range satisfying Expression (2). As described, the drive of vibration actuator10is determined based on mass “m” of movable part40, and spring constant Kspof metal springs as plate-shaped elastic parts50(elastic bodies; plate springs in Embodiment 1). Further, in vibration actuator10, screws62and64are used for fixing base part32and plate-shaped elastic parts50and for fixing plate-shaped elastic parts50and movable part40. Thereby, plate-shaped elastic parts50required to be firmly fixed to fixing part30and movable part40for allowing movable part40to drive can be firmly fixed mechanically in a state capable of reworking. Vibration actuator10includes fixing part30that includes: coil22; and core24around which coil22is wound and both ends thereof are projected from coil22. Further, vibration actuator10includes movable part40that: includes yokes41,41A formed from a magnetic material and disposed adjacently opposite to counter surfaces20a,20bof magnetic pole parts242,244as the both ends of core24with gap G provided therebetween in the direction crossing with the winding axis of coil22; and is capable of being fixed to the operation contact surface part that is operated by contact. Vibration actuator10includes plate-shaped elastic parts50that: are fixed between movable part40and fixing part30; and include bellows-like elastic arm parts56that are elastically deformed to elastically support movable part40with respect to fixing part30to be movable in the direction opposing to magnetic pole parts242,244. While it is preferable that a plurality of plate-shaped elastic parts50be fixed at symmetrical positions with respect to the center of movable part40, movable part40may also be support by one plate-shaped elastic part50to be vibratable with respect to fixing part30as described above. Plate-shaped elastic part50may include at least two or more arm parts that connect movable part40and fixing part30, and include bellows-like elastic arm part56. Plate-shaped elastic part50may be made of a magnetic material. In that case, each of movable-body side fixing parts (movable-body side attachment parts)54of plate-shaped elastic parts50is disposed in the winding axis direction of coil22or the direction orthogonal to the winding axis direction with respect to the both ends of core24, and forms a magnetic path together with core24when coil22is energized. Thereby, even when attached to a touch panel that is the operation contact surface part, it is possible to give a preferable operational feeling to the user at the time of operating the touch panel while achieving reduction in the thickness and the cost. Modification Example 1 FIG.10is a plane-side appearance perspective view of Modification Example 1 of the vibration actuator, andFIG.11is a bottom-surface side appearance perspective view of Modification Example 1 of the vibration actuator.FIG.12is a sectional view illustrating a configuration of main components of Modification Example 1 of the vibration actuator, andFIG.13is an exploded perspective view of Modification Example 1 of the vibration actuator. In vibration actuator10A illustrated inFIG.10toFIG.13, screws62,64, and68used in the configuration of vibration actuator10for fixing base part32and plate-shaped elastic parts50and for fixing plate-shaped elastic parts50and movable part40, respectively, are changed. Specifically, vibration actuator10A is configured by using rivets92,94, and98instead of screws62,64, and68. Each of rivets92,94, and98is formed with a head and a body without a threaded part, which is inserted into a bored member, and an end on the opposite side is riveted and plastically deformed to join the bored members with each other. The riveting may be performed by a pressing machine or a special tool, for example. Rivets92fix attaching part32aof fixing part30and plate-shaped elastic parts50, and rivets94fix plate-shaped elastic parts50and yoke41. Further, rivets98fix fixing part30to bottom surface part32bin a state where coil22of core assembly20is disposed in fastening hole33of bottom surface part32b. Thereby, plate-shaped elastic parts50can be more firmly fixed than the case of using screw62,64, and68, so that plate-shaped elastic parts50can be stably fixed to fixing part30and movable part40. Modification Example 2 FIG.14is a plane-side appearance perspective view of Modification Example 2 of the vibration actuator,FIG.15is a bottom-surface side appearance perspective view of Modification Example 2 of the vibration actuator, andFIG.16is an exploded perspective view of Modification Example 2 of the vibration actuator. Vibration actuator10B of Modification Example 2 includes movable yoke40A that is a single member formed by integrating plate-shaped elastic parts50and yoke41in the configuration of vibration actuator10. Vibration actuator10B includes: fixing part30in the configuration of vibration actuator10; and movable yoke40A that is movable with respect to fixing part30. Movable yoke40A includes: yoke41A having the same function as that of yoke41; and plate-shaped elastic parts450-1,450-2having the same function as that of plate-shaped elastic parts50(50-1,50-2). Yoke41A is formed by integrating attracted surface parts46,47of yoke41and movable-body side fixing parts54of plate-shaped elastic parts50-1,50-2into a single member. In movable yoke40A, yoke41A forms a frame shape surrounding opening part48(seeFIG.16) with surface-part fixing part44and attracted surface parts46A,47A, and plate-shaped elastic parts450-1,450-2are provided to be projected toward the X-direction from attracted surface parts46A,47A, respectively. Each of plate-shaped elastic parts450-1and450-2includes: fixing-body side fixing part452having the same function as that of fixing-body side fixing part52of plate-shaped elastic part50; and bellows-like elastic arm part456having the same function as that of bellows-like elastic arm part56. With such configuration, yoke41A and plate-shaped elastic part450can be in a same height level with respect to bottom surface part32bof fixing part30, so that the thickness (height in the Z-direction) of vibration actuator10B itself can be reduced for that. Further, the number of components can be decreased compared to that of vibration actuator10, so that manufacturing steps can be decreased. Embodiment 2 FIGS.17A and17Bare perspective views of a touch panel apparatus including vibration actuator10according to Embodiment 2 of the present invention. FIGS.17A and17Bare perspective views of touch panel apparatus100including vibration actuator10according to Embodiment 2 of the present invention.FIG.17Ais a perspective view of touch panel apparatus100including vibration actuator110according to Embodiment 2 of the present invention, andFIG.17Bis a right side view of the same apparatus. Touch panel apparatus100illustrated inFIG.17is an example of a vibration presenting apparatus. Vibration actuator110is vibration actuator10, and it is fixed to back surface plate120of touch panel140that displays images via connection pillar part160. Further, while vibration actuator10is used as vibration actuator110, vibration actuator110is not limited thereto but may also be vibration actuator10A or vibration actuator10B. In touch panel apparatus100including touch panel140, touch panel140is fixed to movable part40of vibration actuator110having fixing part30fixed to the center of back surface plate120. Note that touch panel140is an example of the operation contact surface part, and it is fixed at its back surface side to be in surface-contact with surface-part fixing part44of movable part40. Thereby, touch panel140itself integrally drives with movable part40. In touch panel140, the direction the operator touches the screen at the time of operations is the same as the vibrating direction of movable part40and movable yoke40A in vibration actuator110. As described, with touch panel apparatus100on which vibration actuator10is mounted, touch panel140is directly operated, that is, touch panel140together with movable part40is driven in the same direction as the finger touching direction. Therefore, touch panel140can be directly driven with strong vibration. Thereby, at the time of operations performed by touching an image such as a mechanical switch displayed on touch panel140, it is possible to move movable part40to give an operational feeling according to the image, such as an operational feeling felt when operating an actual mechanical switch, thereby achieving fine operability with comfortableness. With in-vehicle products and industrial devices in particular, touch panel apparatus100can be applied to an operation apparatus on which operations are inputted by having a finger or the like touch the image on the screen. In that case, touch panel apparatus100is effective as a touch display apparatus and an operation apparatus provided with the touch panel apparatus that generates vibration in response to a touch operation of the operator made on the image and returns a same operational feeling as the operational feeling at the time of touching the image such as the mechanical switch displayed on the screen. Embodiments of the present invention have been described above. Note that descriptions above are exemplifications of the preferred embodiments of the present invention, and the scope of the present invention is not limited thereto. That is, descriptions of the configurations of the apparatuses and shapes of each component are examples, and it is obvious that various changes and modifications of the examples are possible without departing from the scope of the present invention. INDUSTRIAL APPLICABILITY Even when attached to a touch panel, the vibration actuator according to the present invention exhibits the effect capable of giving a preferable operational feeling to the user at the time of operating the touch panel and reducing the thickness. For example, the vibration actuator is effective when used for moving the touch panel itself in a car navigation apparatus and the like. REFERENCE SIGNS LIST 10,10A,10B,110Vibration actuator20Core assembly22Coil24Core26Bobbin26a,26bSplit body28Fixing hole30Fixing part32Base part32aAttaching part32bBottom surface part33Fastening hole36Opening part40Movable part40A Movable yoke41,41A Yoke44Surface-part fixing part44aFixing surface46,47,46A,47A Attracted surface part48Opening part (Fixing-part side opening part)49Notch50,50-1,50-2,450-1,450-2Plate-shaped elastic part52,452Fixing-body side fixing part54Movable-body side fixing part (movable-body side attachment part)56,456Bellows-like elastic arm part62,64,68Screw70Detection sensor80Electrostatic capacitance sensor92,94,98Rivet100Touch panel apparatus120Back surface plate140Touch panel160Pillar part241Core main body242,244Magnetic pole part | 43,214 |
11863040 | While embodiments shown in these figures accomplish various aspects and objects of the inventions, it is appreciated that it may not be possible to clearly show each element and aspect of the invention in a single figure, and as such, multiple figures are presented to separately illustrate the various details of the inventions in greater clarity. Similarly, not every embodiment need accomplish all advantages of the present inventions. DETAILED DESCRIPTION Aspects of the present disclosure and certain examples, features, advantages, and details thereof, are explained more fully below with reference to the non-limiting examples illustrated in the accompanying drawings. Descriptions of well-known materials, fabrication tools, processing techniques, etc., are omitted so as not to unnecessarily obscure the relevant details. It should be understood, however, that the detailed description and the specific examples, while indicating aspects of the disclosure, are given by way of illustration only, and are not by way of limitation. Various substitutions, modifications, additions, and/or arrangements, within the spirit and/or scope of the underlying inventive concepts will be apparent to those skilled in the art from this disclosure. The details of the inventions will now be discussed in relation to the accompanying drawings so as to enable one skilled in the art to practice the present inventions. These, and other aspects and objects of the present inventions, will be better appreciated and understood when considered in conjunction with the following description and the accompanying drawings. It should be understood, however, that the following description, while indicating embodiments of the present inventions and numerous specific details thereof, is given by way of illustration and not of limitation. The drawings and following description are exemplary of various aspects of the inventions and are not intended to narrow the scope of the appended claims. Many changes and modifications may be made within the scope of the present invention without departing from the spirit thereof and the invention includes all such modifications. This invention includes all such modifications. Approximating language, as used herein throughout disclosure, may be applied to modify any quantitative representation that could permissibly vary without resulting in a change in the basic function to which it is related. Accordingly, a value modified by a term or terms, such as “about” or “substantially,” is not limited to the precise value specified. For example, these terms can refer to less than or equal to ±5%, such as less than or equal to ±2%, such as less than or equal to ±1%, such as less than or equal to ±0.5%, such as less than or equal to ±0.2%, such as less than or equal to ±0.1%, such as less than or equal to ±0.05%. In some instances, the approximating language may correspond to the precision of an instrument for measuring the value. Terminology used herein is for the purpose of describing particular examples only and is not intended to be limiting. As used herein, the singular forms “a”, “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. Furthermore, references to an “example” are not intended to be interpreted as excluding the existence of additional examples that also incorporate the recited features. Moreover, unless explicitly stated to the contrary, the terms “comprising” (and any form of “comprise,” such as “comprises” and “comprising”), “have” (and any form of “have,” such as “has” and “having”), “include” (and any form of “include,” such as “includes” and “including”), and “contain” (and any form of “contain,” such as “contains” and “containing”) are used as open-ended linking verbs. As a result, any examples that “comprises,” “has,” “includes” or “contains” one or more step or element possesses such one or more step or element, but is not limited to possessing only such one or more step or element. As used herein, the terms “may” and “may be” indicate a possibility of an occurrence within a set of circumstances; a possession of a specified property, characteristic or function; and/or qualify another verb by expressing one or more of an ability, capability, or possibility associated with the qualified verb. Accordingly, usage of “may” and “may be” indicates that a modified term is apparently appropriate, capable, or suitable for an indicated capacity, function, or usage, while taking into account that in some circumstances the modified term may sometimes not be appropriate, capable or suitable. For example, in some circumstances, an event or capacity can be expected, while in other circumstances the event or capacity cannot occur—this distinction is captured by the terms “may” and “may be.” As used herein, the term “propeller” and grammatical variants thereof includes any rotor, fan, airscrew, screw propeller, screw, turbine, or other similar spinning device, and is used synonymously herein. The term “airfoil” is used herein synonymously with “hydrofoil” and “aerofoil,” and refers to any cross-sectional shape of a wing, blade or sail that produces an aerodynamic force of thrust and/or lift when moved through a fluid (gaseous and/or liquid). As used herein, the terms blade and wing are used herein synonymously when referring to a propeller. In one aspect, the present disclosure provides electrohydrodynamic (EHD) rotary systems and related methods that generate thrust from a propeller via EHD-generated rotation of the propeller for propulsion. In some embodiments, the systems and methods provide sufficient thrust to physically move a device or apparatus housing or including the systems. For example, in some embodiments, an EHD rotary system and corresponding method may be fully incorporated into an aerial vehicle (e.g., an unmanned aerial vehicle (UAV), a rotorcraft (e.g., a drone, a helicopter, etc.), a fixed wing aircraft, an aerostat or a combination thereof), and the EHD rotary system and corresponding method may rotate at least one propeller thereof at a sufficient rotational speed and/or torque such that sufficient thrust and/or lift is provided to liftoff and/or fly of the aerial device above a ground surface. As another example, in some embodiments, an EHD rotary system and corresponding method may be fully incorporated into a land-based vehicle (e.g., a wheeled vehicle, hovercraft, boat, sled, etc.), and the EHD rotary system and corresponding method may rotate at least one propeller thereof at a sufficient rotational speed and/or torque such that sufficient thrust and/or lift is provided to translate the vehicle along a ground surface. The EHD rotary systems and related methods of the present disclosure may thereby be incorporated into any propeller-based air/fluid-flow apparatus as a replacement/substitute of or a supplement to, the power source of the apparatus (i.e., the torque/rotation generating system, such as an electric motor, combustion engine, etc.). For example, the EHD rotary systems and related methods of the present disclosure may be utilized with any fan, pump, turbine, jet, propeller, rotor or turbine. There are a multitude of practical applications based on the EHD-induced torque and/or rotational motion embodied in EHD rotary systems and related methods of the present disclosure. The EHD rotary systems and related methods of the present disclosure apply an electrode voltage to at least one rotary electrode of a rotary device that generates a relatively strong inhomogeneous or non-uniform electric field between the rotary electrode and a nearby stationary counter electrode above the corona onset, which becomes very intense near one or more relatively sharp edge of the rotary electrode. The field accelerates the free electrons in the region which leads to an avalanche process and local breakdown. More free electrons and ions are created producing local cold plasma. As the ionization region is proximate to the one or more relatively sharp edge of the rotary electrode, ions of the opposite polarity of the electrode quickly reach the electrode and are neutralized. Electrons have a much higher mobility than ions, and they clear the air-gap fast, leading to the formation of drifting space charge of opposite polarity to the emitting rotary electrode. Moving ions impart momentum to neutral molecules in a coupling mechanism, and move in the electric field towards the opposite electrode (e.g., from a rotary electrode to the counter electrode). This generates an EHD flow, also known as corona discharge, corona wind, ionic wind or electric wind (these terms are used synonymously herein). It is noted that in the corona discharges, diffusion and magnetic effects play a relatively minor role in the ion flow. In some embodiments, the EHD rotary systems and related methods of the present disclosure achieve rotational motion of a propeller (or other rotary device) via EHD thrust and torque generated from the ion flow resulting in the strong inhomogeneous or non-uniform electric field between the rotary electrode on a propeller (or other rotary device) and the nearby stationary counter electrode above the corona onset. When the ionic wind is produced, EHD thrust is generated from the action of the electric field on the space charge and as a result of momentum conservation for the air-electrode system, as discussed above. If the electric field E and spatial charge density ρ distributions in the volume v are known, the thrust magnitude, TEHD, may be calculated as TEHD=∫ρEdv. However, potentially more realistically, the EHD thrust may also be modeled as TEHD=Ikdnμ, where I is the corona current; d is the distance between the corona point and the counter electrode, k is the constant of proportionality, n is a constant coefficient and y is the average mobility of ions. In the EHD rotary systems and related methods of the present disclosure, the EHD thrust may generate torque and rotational motion relative to the axis of rotation of the propeller (or other rotary device), which may be characterized by Jp{umlaut over (θ)}+C{dot over (θ)}N=τEHD−τFf, where Jpis the moment of inertia of the propeller, {umlaut over (θ)} is the angular acceleration, C is a constant; {dot over (θ)} is the angular speed; τEHDis the torque associated to the EHD generated thrust; C{dot over (θ)}Nis the torque associated to the drag forces, N=1 for laminar flow and N=2 for turbulent flow; τFfis the torque associated to the frictional forces between the propeller and the axial support on/in which the propeller rotates. However, it is noted that thrust may not identically correspond to the EHD force due to the presence of drag and also due to contact frictional forces. During the motion of the propeller or other rotary device of the EHD rotary systems and related methods of the present disclosure, the rotational motion relative to the axis of rotation of the propeller may be characterized by Is{umlaut over (θ)}=τEHD−τFv−τFf, where Isis the moment of inertia of the propeller, {umlaut over (θ)} is the angular acceleration, τEHDis the torque associated to the EHD generated thrust, τFvis the torque associated to the viscosity forces, τFfis the torque associated to the frictional forces between the propeller and the axial support. In some embodiments, the rotational induced friction/resistance between the propeller and the axial support may be minimal. In such embodiments, the τFfmay be neglected relative to τEHDand τFv. Further, viscosity/drag forces (Fv) may depend, at least partially, on the linear speed of the propeller, profile/cross-sectional shape of the propeller, the density of the fluid in which the propeller rotates and/or the viscosity of the fluid in which the propeller rotates. In some embodiments, the fluid parameters may be considered constant, and the cross-sectional shape/profile of the propeller does not change in time, and thus the viscosity forces relationship may be reduced to Fv=Cvk, where C, k are constants for a given linear speed. Consequently, the associated torque may also depend on vk. In some embodiments of the EHD rotary systems and related methods of the present disclosure, the generated frictional and drag torque acting on the propeller or other rotary device may be small relative to the EHD generated thrust and torque for relatively low speeds or angular velocities (which may depend, at least in part, upon the diameter or size of the propeller/rotary device). Hence, in some embodiments, at the start of the rotational motion of the propeller or other rotary device, the dominant torque applied to the propeller or other rotary device may be due to the constant EHD thrust. Accordingly, a constant angular acceleration (i.e., a linear increase of the angular velocity with time) of the propeller or other rotary device may be experienced during such a time period during this region. In some embodiments of the EHD rotary systems and related methods of the present disclosure, the variation of the angular velocity (ω) of the propeller or other rotary device may linearly increase at the beginning of its rotation/motion, irrespective of the high voltage (above the corona onset) applied to the rotary electrode. The linear portion in the variation of the angular velocity ω(t) may provide acceleration α as α=θ¨=dω(t)dt. In some embodiments, the slope of the linear region linear portion in the variation of the angular velocity is the angular acceleration, and therefore the constant torque of EHD thrust may also be determined as τEHD=Isαlwhere αlis the slope of the linear portion of variation of the angular velocity ω(t). In some embodiments, the propeller or other rotary device comprises a plurality of projections extending outwardly from each blade and/or a plurality of blades thereof proximate to a back edge of the blades. In some such embodiments, the projections may extend along an axis thereof that is tangent to the rotational pathway of the projection. In another embodiment, the projections may be angled radially outward from the axis of rotation as they extend from the trailing edge of a blade. In some embodiments with the projections being oriented tangent to the rotational/angular pathway thereof, the EHD generated torque with respect to the rotational axis may be expressed as τEHD=Σi=1N(FEHD,tiRi), where FEHD,tiis the tangential thrust per projection-pair number i, R; is the effective radius corresponding to the FEHD,tiand N is the total number of projection pairs used in the propeller or other rotary device. Also, in some embodiments of the EHD rotary systems and related methods of the present disclosure, the angular velocity of the propeller or other rotary device may stabilize over time to a maximum value (i.e., to terminal angular velocity). In some embodiments, a period of nonlinear increase in angular velocity of the propeller/rotary device and a period of linear increase in angular velocity of the propeller/rotary device may occur between the period of initial motion angular rotation/motion and the terminal angular velocity. In some such embodiments, the period of nonlinear increase in angular velocity of the propeller/rotary device may comprise a transitory regime from the start of the motion of the propeller/rotary device to the steady state/terminal velocity of the propeller/rotary device. In some embodiments of the EHD rotary systems and related methods of the present disclosure, the EHD thrust generated at/on the rotary electrode of the propeller or other rotary device may be proportional or otherwise related to the corona current and/or the current applied to rotary emitter electrode (e.g., depending upon the particular configuration. However, it is noted that in some other embodiments, the EHD thrust generated at/on the rotary electrode of the propeller or other rotary device may not be proportional to the corona current and/or the current applied to rotary emitter electrode. The EHD thrust may be derived theoretically by FEHD=Ikdμ, where I is the corona current, d is the distance between the emanating point/position of the corona and the counter electrode, k is a constant of proportionality, while μ is the average mobility of ions. It is noted that this theoretical derivation may be valid for a uniform electric field, which may not be the case for embodiments of the EHD rotary systems and related methods of the present disclosure. For example, it has been shown that FEHD=Ikdnμ, with “n” being a constant smaller than unity, may be a more realistic or accurate theoretical derivation of the EHD thrust. In such a revised theoretical derivation, the thrust per unit of power may be expressed as η=FEHDIV=AV, with A being a constant when geometrical and physical conditions are not changed. As such, the EHD rotary systems and related methods of the present disclosure may comprise a tradeoff between efficiency of thrust generation and applied voltage. It is noted that corona current is generally accepted to follow a quadratic variation with the voltage above the corona onset of I=KV(V−Vo), where Vostands for corona onset voltage and K is a proportionality constant. Thereby, thrust may follow a quadratic variation with the applied voltage for constant d of FEHD=BV(V−Vo), where B is the new resulting constant. FIGS.1-5illustrate an exemplary EHD rotary system100and related method according to the present disclosure that utilizes a propeller101as the rotary device. As shown inFIGS.1and2, the propeller101includes a hub portion104and a plurality of blade portions102extending radially from the hub portion104. The propeller101is configured to rotate about an axis X-X of rotation that extends through the hub portion104. In this way, the hub portion104may physically support and arrange the blade portions102, at least partially, and allow the propeller101, as a whole, to rotate about the axis X-X. As shown inFIGS.2and5, in some embodiments the hub portion104may include a circular or cylindrical aperture or hole that is aligned with the axis X-X that is configured to accepts a shaft portion122therein, and potentially therethrough. The hub portion104may thereby rotate on (directly or indirectly) the shaft portion122about the axis X-X. The hub portion104(and thereby the propeller101as a whole) may be axially fixed along the length of the shaft portion122along the axis X-X, or may be able to axially slide or otherwise axially translate along the length of the shaft portion122along the axis X-X. In such embodiments, the EHD thrust is utilized to rotate the propeller101with respect to the shaft portion122about the axis X-X. In some other embodiments, the shaft portion122may be rigidly or fixedly coupled to the hub portion104. In such embodiments, the EHD thrust is utilized to rotate the propeller101and the shaft portion122about the axis X-X. The shaft portion122may define an axis that is aligned with, or comprises, the axis X-X of rotation of the propeller101. In the illustrated exemplary embodiment, the propeller101includes three blade portions102. However, the propeller101may include only a single blade portion102, a pair of blade portions102, or three or more blade portions102. The plurality of blade portions102may be shaped, sized and otherwise configured identically to each other, or the blade portions102may vary in at least one physical characteristic. The blades102may be formed of any relatively rigid and strong material, such as wood, plastic, polymer, foam, alloy, metal, glass, ceramic, composite, fiber reinforced composite, carbon-fiber, fiberglass, or any combination thereof. In some embodiments, the blades102may comprise an electrically insulative or non-conductive (or semiconductor) material. In some embodiments, the blades102may comprise a material and/or portion with an electrical resistivity of greater than or equal to 0 Ω·m at 20 degrees Celsius, or greater than or equal to 100 Ω·m at 20 degrees Celsius, greater than or equal to 500 Ω·m at 20 degrees Celsius. As shown inFIGS.1and2, the plurality of blade portions102extend radially from the hub portion104to tips or free ends106thereof. The plurality of blade portions102are circumferentially or angularly arranged about the axis X-X. As shown inFIGS.1and2, the plurality of blade portions102may be circumferentially or angularly spaced from each other about the axis X-X. As also shown inFIGS.1and2, each blade102includes an outer surface that includes a front leading edge105, a back trailing edge107, a top surface103that extends between the front and back edges105,107, and a bottom surface109that extends between the front and back edges105,107. It is noted that the front leading edge105and/or the back trailing edge107may comprise the junction of, or a surface extending between, the top and bottom edges103,109. The term “edge” is used herein with respect to the front and back edges105,107to refer to either configuration. The front edge105may comprise the outer edge or surface portion of a respective blade102that is positioned furthest, or travels first, in a first angular or rotational direction R1about the axis X-X. The front edge105is thereby the foremost edge of that blade102that first contacts or passes through the fluid about the propeller101as the propeller101rotates about the axis X-X in the first rotational direction R1. The back edge107may comprise the outer edge or surface portion of a respective blade102that is positioned furthest in a second angular or rotational direction R2about the axis X-X that opposes the first rotational direction R1, or travels last in the first rotational direction R1about the axis X-X. The back edge107is thereby the trailing edge of that blade102that last contacts or passes through the fluid about the propeller101as the propeller101rotates about the axis X-X in the first rotational direction R1. As shown inFIGS.3and5, the blades102of the propeller101may be oriented to include an angle of attack that converts rotational motion of the blades102about the axis X-X in the first rotational direction R1through a fluid into an aerodynamic/hydrodynamic force including a thrust force or flow component acting away from the bottom surface109of the blades102perpendicular to the first rotational direction R1(or a plane defined by the path of the blades) and/or parallel (in all directions) to the axis of rotation X-X. The thrust force component125may comprise all of the aerodynamic/hydrodynamic force or a portion of the aerodynamic/hydrodynamic force (typically a portion), depending, at least partially, upon the configuration and/or orientation of the blades102, for example. As the blades102rotate about the axis X-X in the first rotational direction R1through a fluid (e.g., a gas and/or a liquid), the blades102and the angle of attack thereof are configured to deflect and force the fluid away from the bottom side107thereof in a flow of the fluid (i.e., the thrust component force125). According to Newton's third law, as shown inFIGS.3and5, the fluid (gaseous or liquid fluid) surrounding the at least one propeller101or other rotary device must exert an equal and opposite force to the thrust force component125on the blades102, referred to herein as a lift force component127. The blades102of the propeller101may thereby be oriented to include an angle of attack and/or airfoil shape that converts rotational motion of the blades102about the axis X-X in the first rotational direction R1through a fluid into an aerodynamic/hydrodynamic force comprising the thrust force component125and an aerodynamic/hydrodynamic reaction force comprising the lift force component127acting against the bottom surface109of the blades102perpendicular to the first rotational direction R1(or a plane defined by the path of the blades) and/or parallel (in all directions) to the axis of rotation X-X (i.e., equal and opposite to the thrust component force125). The lift force component127may comprise all of the reaction aerodynamic/hydrodynamic force or a portion of the aerodynamic/hydrodynamic force, depending, at least partially, upon the configuration and/or orientation of the blades102, for example. The propeller101or other rotary device may be oriented such that the thrust force component125and the lift force component127extend vertically, horizontally or at an angle between vertical and horizontal. The desired use of the thrust force component125and the lift force component127may dictate, at least partially, the configuration of the propeller101and the orientations of the thrust force component125and the lift force component127. For example, an aerial device may include the exemplary EHD rotary system100with the propeller101configured such that the thrust force component125and the lift force component127are oriented about vertically to achieve liftoff of the aerial device via the lift force component127. As another example, an aerial device may include the exemplary EHD rotary system100with the propeller101configured such that the thrust force component125and the lift force component127are oriented about horizontal to translate the device horizontally in the air and/to achieve liftoff via airfoil-shaped wings. As yet another example, a land-based device may include the exemplary EHD rotary system100with the propeller101configured such that the thrust force component125and the lift force component127are oriented about horizontal to translate the device horizontally in across a ground surface. In a further example, a fan or ventilation device may include the exemplary EHD rotary system100with the propeller101configured such that the thrust force component125and the lift force component127are oriented with respect to a portion of the device or another apparatus or device to create a flow of a fluid (e.g., air) over or from the portion of the device or the other apparatus or device (e.g., for cooling or heating). In another example, a pump device may include the exemplary EHD rotary system100with the propeller101configured such that the thrust force component125and the lift force component127are oriented with respect to a housing to create a flow of a fluid (e.g., a liquid) through or from the housing. In some such embodiments, as shown inFIG.3, the front edge105, the back edge107, the top surface103and the bottom surface109of the blades102of the propeller101may form an airfoil shape (with an angle of attack) in cross-section that more efficiently produces the thrust force/flow125and lift force127as compared a non-airfoil cross-sectional shape. In some other embodiments, the front edge105, the back edge107, the top surface103and the bottom surface109of the blades102of the propeller101may not form an airfoil shape, but rather include an angle of attack that forms the thrust force/flow125and lift force127via rotation of the propeller101in the first rotational direction R1. For example, the top and/or bottom surfaces103,109may be flat/planar, and may potentially extend parallel to each other. The blades102of the propeller101may include any cross-sectional shape (airfoil or non-airfoil shapes) and any attack angle that is effective in producing an aerodynamic/hydrodynamic force including the thrust component125(the component perpendicular to the first rotational direction R1acting away from the bottom surface109) and the corresponding lift force component127(the component perpendicular to the first rotational direction R1acting toward and on the bottom surface109) as the blades102rotate in the first rotational direction R1. At least one blade102of the propeller101of the exemplary EHD rotary system100includes an exposed, electrically conductive rotary emitter electrode131positioned (at least partially) proximate to the back edge107thereof. For example, as shown inFIGS.2-4, at least one blade102of the propeller101of the exemplary EHD rotary system100may include an electrically conductive exposed rotary emitter electrode131comprising at least one electrically conductive member110extending radially along a blade102proximate to the back edge107thereof. The at least one conductive member110may also extend radially along the hub portion104, as also shown inFIGS.2-5. For example, the at least one conductive member110may extend from the aperture or shaft122and radially along the hub portion104and at least one blade102. As such, the at least one conductive member110may be electrically coupled with the shaft portion122. The at least one conductive member110may comprise a single integral electrically conductive member, or a plurality of electrically coupled separate and distinct electrically conductive members. The at least one conductive member110may be configured to carry electrical current and high voltage above corona onset, and generate EHD flow/corona wind in the strong electric field, and thereby EHD thrust to the blade102to rotate the propeller101about the axis X-X in the first rotational direction R1. In some embodiments, the at least one conductive member110may be formed of any electrically conductive or semiconductor material, such as one or more metal (e.g., copper, silver, gold, aluminum, steel, etc.), alloy, semiconductor (e.g., silicon, germanium, gallium arsenide, silicon carbide, ternary compounds, oxides and alloys, arsenic, selenium, tellurium, organic semiconductors (made of organic compounds), etc.) nonmetallic conductor (e.g., graphite, conductive polymers, etc.), composite, conductor, or a combination thereof. In some embodiments, the at least one conductive member110may be comprised of one or more film, paint, ink, wire, tape, bar, member (e.g., stiff or flexible member) or any combination thereof. As shown inFIG.2, in some embodiments the at least one conductive member110of the rotary emitter electrode131may extend radially from the hub portion104proximate to the back edge107of the blade102to, or proximate to, the tip106. In some other embodiments, the at least one conductive member110may not extend from the hub portion104and proximate to the tip106. For example, the at least one conductive member110may extend radially about two-thirds the radial length of the blade102, or half the radial length of the blade102, or about one-third the radial length of the blade102, or about one-quarter the radial length of the blade102. As another example, the at least one conductive member110may not extend entirely proximate to the back edge107of the blade102. For example, in some embodiments, only a portion of the at least one conductive member110may extend along, or be positioned proximate to, the back edge107of the blade102. In the exemplary embodiment shown inFIGS.2-5, the at least one conductive member110of the rotary emitter electrode131of the propeller101is affixed to, and extends over, the bottom surface109of the blades102. In other embodiments, as explained further below, the at least one conductive member110may be coupled and/or extend only along the back edge107, along the top surface103, along the top surface103and the back edge107itself, along the bottom surface109and the back edge107itself, along the top and bottom surfaces103,109and not along the back edge107itself, or along the top and bottom surfaces103,109and the back edge107itself. At least a portion of the at least one conductive member110may or may not be exposed. In the exemplary embodiment shown inFIGS.2-5, a relatively sharp or narrow outer edge portion of the at least one conductive member110proximate to the back edge107of the at least one conductive member110(and distal to the front edge105) is exposed. The remainder of the outer surfaces or portions of the at least one conductive member110are covered by the blade102, the hub portion104or an electrically insulative material113that is configured to prevent the formation of the electric field and/or EHD flow/corona wind from the underlying portions of the at least one conductive member110. For example, all of the outer surfaces of a portion of the at least one conductive member110proximate to the front edge105(and distal to the back edge107) may covered by the blade102and/or the electrically insulative material113. Similarly, all of the outer surfaces of the at least one conductive member110coupled to the hub portion104may be covered by the hub portion104and/or the electrically insulative material113. In some embodiments, the electrically insulative material113may be formed of any electrically insulative or semiconductor material, such as one or more glass, ceramic, porcelain, dielectric, composite, paper, mica, PTFE, PFA, rubber, wax, oil, asbestos, xylene, ethylbenzene, toluene, cumene, Super Corona Dope™, or and any combination thereof. In some embodiments, the electrically insulative material113may be comprised of one or more film, paint, ink, tape, bar, member (e.g., stiff or flexible member) or any combination thereof. In some embodiments, the blade102, the hub portion104and/or the insulative material113may include a relatively high dielectric strength to prevent corona formation/generation, such as a dielectric strength of at least about 3,000 V/mm (thickness), at least about 3,500 V/mm, or at least about 4,000 V/mm. In some embodiments, the blade102, the hub portion104and/or the insulative material113may include a thickness of at least about ⅓ mm, at least about ½ mm, at least about 1 mm, or at least about 1½ mm. In some embodiments, a portion of the at least one conductive member110proximate to the front edge105(and distal to the back edge107) may be embedded within the blade102. Similarly, in some embodiments, the portion of the at least one conductive member110coupled to the hub portion104may be embedded within the hub portion104. In some alternative embodiments, a portion of the at least one conductive member110proximate to the back edge107may not be exposed (e.g., covered by the blade102and/or the insulative material113). In some alternative embodiments, the entirety of the at least one conductive member110may not be exposed (e.g., completely covered by the blade102and/or the insulative material113). In some other alternative embodiments, the entirety of the outer surface portion of the at least one conductive member110of the blade102that is not coupled to and/or abutting the blade102may be exposed. As shown inFIGS.1-5, in some embodiment the electrically conductive exposed rotary emitter electrode of one or more blades102of the propeller101may comprise at least one electrically conductive projection or protrusion108electrically coupled with the at least one conductive member110. The at least one conductive member110may be configured to carry electrical current and high voltage above corona onset from the at least one conductive member110, and generate EHD flow/corona wind in the strong electric field, and thereby EHD thrust to the blade102to rotate the propeller101about the axis X-X in the first rotational direction R1. In some embodiments, the EHD flow/corona wind generated by and/or emanating from a blade102may be localized or predominately concentrated on/from one or more projection108thereof. As such, the EHD thrust to such a blade102may be dependent, at least partially, on the one or more projection108of the blade102. In some embodiments, the at least one conductive member110may be formed of any electrically conductive or semiconductor material, such as one or more metal (e.g., copper, silver, gold, aluminum, steel, etc.), alloy, semiconductor (e.g., silicon, germanium, gallium arsenide, silicon carbide, ternary compounds, oxides and alloys, arsenic, selenium, tellurium, organic semiconductors (made of organic compounds), etc.) nonmetallic conductor (e.g., graphite, conductive polymers, etc.), composite, conductor, or a combination thereof. The projections108may be coupled to the blades102and extend from the at least one conductive member110proximate to the back edge107, as shown inFIGS.1-4. A projection108may be coupled to and/or extend from only the back edge107itself, the top surface103proximate to the back edge107, the top surface103and the back edge107itself, the bottom surface109proximate to the back edge107, the bottom surface109and the back edge107itself, the top and bottom surfaces103,109and not the back edge107itself, or the top and bottom surfaces103,109and the back edge107itself. The projections108may define a relatively sharp end edge, surface or point at the tip or free end thereof. As such, the projections108may taper to the tips thereof. In some embodiments, a base portion of a projection108opposing the tip thereof may be covered by the insulative material113and/or embedded in the blade102. The projections108extend from proximate to the back edge107thereof outwardly past the back edge107in a direction opposing the first rotational direction R1(i.e., a direction extending from the front edge105to the back edge107). In the illustrated exemplary embodiment, the protrusions108extend linearly. However, the protrusions108may extend nonlinearly (e.g., arcuately, rectilinearly, or a combination thereof). The physical configuration (e.g., cross-sectional shape, path, thickness, distance pas the back edge107, etc.) may vary, and may depend, at least partially, upon the particular blade102, propeller101or use of the propeller101, for example. The projections108of the blades102of the propeller101may be aligned along a plane, which may be normal to the axis X-X of rotation and/or extend in a direction tangential to the hub portion104. In some embodiments, the one or more projections108of the blades102are aligned in a tangential or perpendicular (in all directions) direction to the axis of rotation X-X (e.g., the axis of the aperture and/or shaft portion122) to optimize the EHD torque of the propeller101produced by the EHD thrust/corona wind via the projections108. For example, the one or more projections108of the blades102may be aligned perpendicularly or relatively perpendicular to the radial direction from the axis X-X of rotation. While only one projection108is depicted per blade102of the propeller101illustrated inFIGS.1-5, one or more blades102of the propeller101may include a plurality of radially spaced projections108. It is noted that, in some embodiments, the greater the number of projections108, the greater the EHD thrust (and thereby torque of the propeller101and/or shaft122) the EHD rotary system100may be capable of producing. The one or more projections108of a blade102may be positioned radially anywhere along the blade. In some embodiments, the blades102may include at least one projection108spaced form the hub portion104at least about one-quarter the radial length of the blade102, or at least about one-third the radial length of the blade102, or at least about half the radial length of the blade102, or at least about two-thirds the radial length of the blade102. In the illustrated embodiment, each blade102includes a single projection position about two-thirds of the radial length of the blades102from the hub portion104. In some embodiments, each projection108of a blade102may be spaced from the tip106thereof and the hub portion104. For example, in some embodiments, each projection108of a blade102may be radially spaced from the tip106of the blade102by at least about 3%, or at least about 5%, or at least about 7%, or at least about 10%, or at least about 15%, or at least about 20%, or at least about 25%, or at least about 30%, or at least about 33%, or at least about 35% of the total radial length of the blade102extending from the tip portion106thereof to the hub portion104. Radially spacing each projection108of a blade102from the tip106thereof may unexpectedly increase the EHD thrust/coronal wind generated by the blade102, as compared to a projection108being positioned at or closer to the tip106. In some embodiments, each projection108of a blade102may be spaced from the hub portion104by at least about 3%, or at least about 5%, or at least about 7%, or at least about 10%, or at least about 15%, or at least about 20%, or at least about 25% of the total radial length of the blade102extending from the hub portion104to the tip106thereof. As shown inFIGS.4and5, the EHD rotary system100further includes at least one electrically conductive counter electrode114positioned proximate to the propeller101. The at least one counter electrode114is be configured to form the highly inhomogeneous electric field between the at least one rotary emitter electrode and the at least one counter electrode114when the electrical current and high voltage above corona onset is applied to the rotary emitter electrode to generate the EHD flow/corona wind in the electric field, and, thereby, effectuate the EHD thrust to the blades102to rotate the propeller101about the axis X-X in the first rotational direction R1. The at least one electrically conductive counter electrode114may positioned in any position and orientation in relatively close proximity with respect to the propeller101, and comprise any two or three dimensional shape, that facilitates or enables formation of the electric field and the EHD flow/corona wind. For example, the at least one counter electrode114may be positioned adjacent at least one side of the propeller101, such as adjacent to the top side103, the bottom side109and/or the tips106of the one or more blade102of the propeller101. The at least one propeller101or other rotary device and the at least one counter electrode114may be fixedly coupled together such that the spaced relationship therebetween is fixed. In some other embodiments, at least one propeller101or other rotary device and the at least one counter electrode114may be selectably movably coupled together such that the spaced relationship therebetween can be changes or altered. In some such embodiments, relative movement between the least one propeller101or other rotary device and the at least one counter electrode114may change the direction of the rotation of the at least one propeller101or other rotary device about the axis X-X. As another example, one or more blade102of the propeller101may be positioned, at least partially, within at least one electrically conductive counter electrode114or between two or more electrically conductive counter electrodes114(e.g., between parallel plate electrodes114) during at least a portion of its rotation about the axis of rotation X-X. For example, as shown inFIGS.4and5, the exemplary illustrated conductive counter electrode114of the EHD rotary system100comprises a ring or hollow cylindrical shape that extends about the propeller101such that the propeller101is positioned, at least partially, within an internal void or cavity115of the conductive counter electrode114. In some such embodiments, the propeller101may be located centrally within the void115of the counter electrode114and the counter electrode114may extend symmetrically about the axis X-X of rotation. Alternatively, the propeller101may be placed outside of the void115of the counter electrode114(e.g., axially outside of a ring or cylindrical counter electrode114). The at least one electrically conductive counter electrode114may comprise any two or three dimensional shape that facilitates or enables formation of the electric field and the EHD flow/corona wind. For example, the at least one electrically conductive counter electrode114may comprise a two dimensional, three dimensional, solid and/or hollow shape that is selected from the group comprising: a regular polygonal shape, a simple polygonal shape, an equilateral polygonal shape, an equiangular polygonal shape, a convex polygonal shape, a concave polygonal shape, an isogonal polygonal shape, a triangular shape, a quadrilateral shape, a tetragonal shape, a rectangular shape, a square shape, a rhombus shape, a pentagonal shape, a hexagonal shape, a heptagonal shape, an octagonal shape, a nonagonal shape, and a decagonal shape, a cylinder shape, a ring shape, and combinations thereof. It is noted, however, that shapes of the at least one electrically conductive counter electrode114that confer a relatively uniform distribution of the electric field between the rotary emitter electrode(s) and the counter electrode114may be advantageous as they may facilitate and maintain a relatively strong and constant electric field during rotation at the propeller101(e.g., during rotational movement of the at least one rotary emitter electrode of one or more blade102of the propeller101). The at least one electrically conductive counter electrode114may include at least one exposed relatively smooth surface that faces and is positioned in close proximity to the propeller101, such as in close proximity to the path of the blades102including the at least one rotary emitter electrode. The at least one electrically conductive counter electrode114may be a solid member or include a plurality of apertures or through holes (e.g., to allow the thrust component flow/force125to pass therethrough). In some embodiments, at least the exposed surface portion of the at least one electrically conductive counter electrode114proximate to the propeller101may be planar or curved (i.e. arcuate). In some embodiments, the at least one electrically conductive counter electrode114may be formed of any electrically conductive or semiconductor material effective informing the string electric field with the at least one rotary emitter electrode of the blades102of the propeller101, such as one or more metal (e.g., copper, silver, gold, aluminum, steel, etc.), alloy, semiconductor (e.g., silicon, germanium, gallium arsenide, silicon carbide, ternary compounds, oxides and alloys, arsenic, selenium, tellurium, organic semiconductors (made of organic compounds), etc.) nonmetallic conductor (e.g., graphite, conductive polymers, etc.), composite, conductor, or a combination thereof. As shown inFIG.5, the EHD rotary system100further comprises an electrical system120that supplies high voltage above corona onset to the one or more rotary emitter electrode of the propeller101(or other rotary device) to generate the strong electrical field and the EHD thrust/corona wind to effectuate rotation of the propeller101about the axis X-X and, ultimately, thereby generate the lift force component127(when the propeller101comprises the rotary device). For example, the electrical system120that supplies electric potential difference/high voltage above corona onset to the at least one projection108and/or the at least one electrically conductive member110of at least one blade102, such as via the shaft portion122and/or the portion of the at least one electrically conductive member110coupled to the hub portion104, that generates corona discharges from the at least one projection108and/or the at least one electrically conductive member110that form flows of ionic wind emanating therefrom that rotate the at least one rotary device about the axis of rotation in a first rotational direction R1. It is noted that the entirety of the EHD thrust/coronal wind (or the reactionary force thereto) may not act against the at least one blade102of the propeller101or other rotary device in the first rotational direction R1(or tangentially thereto, such as normal to the trailing edge107) (in a plane of the rotation or pathway of the at least one blade102about the axis X-X). Rather, a portion of the EHD thrust/coronal wind (or the reactionary force thereto) may be angled within respect to the first rotational direction R1(or tangentially thereto). For example, the shape/orientation of a blade102(e.g., the pitch thereof), and or the physical configuration of the at least one emitter electrode thereof, may be configured such that the EHD thrust/coronal wind (or the reactionary force thereto) comprises a thrust component that extends in the direction of the thrust force/flow component125of the propeller101and/or a reactionary lift force component that extends in the direction of the lift force component127the propeller101. In this way, the EHD thrust/coronal wind may add an additional lift force component127to the propeller101that acts against the bottom side109of the at least one blade102thereof, for example (which may assist in translating a device including the EHD flow system, such as through the air and/or over a ground surface). The electrical system120may be configured to apply a voltage (electric potential difference) to the one or more rotary emitter electrode of the propeller101of a positive polarity or a negative polarity. In some embodiments, the electrical system120may be configured to apply a voltage (electric potential difference) to the one or more rotary emitter electrode of the propeller101and to the one or more counter electrode114of opposite polarities. In some other embodiments, the one or more counter electrode114(and the electrical system120) may be grounded (to a common ground). It is noted that the application of negative verse positive polarity electric potential difference to the one or more rotary emitter electrode of the propeller101(or other rotary device) (as compared to the one or more counter electrode114) of an EHD rotary system100may generate about the same angular/rotational speed of the propeller101, but may generate more overall EHD thrust. Further, a larger negative polarity voltage as compared to a positive polarity voltage may be able to be applied to the rotary emitter electrode without breakdown/arcing. Negative polarity voltage applied to the rotary emitter electrode may thereby generate more overall EHD thrust as comparted to positive polarity voltage applied to the rotary emitter electrode. In some embodiments, the corona current in the negative polarity may be larger than in the positive polarity at the same voltage, potentially due, at least in part, to a greater ion mobility and an improved ability to generate larger momentum than in the positive polarity. The electrical system120may comprise any source of electrical energy (e.g., current and voltage) that is electrically coupled to the at least one counter electrode114and/or the one or more rotary emitter electrode of the propeller101(or other rotary device). For example, in some embodiments the electrical system120comprises a battery, a generator, a fuel cell, a solar cell, an electrical grid input line, a supercapacitor, or a combination thereof. In some embodiments, the electrical system120may apply a voltage (electric potential difference) to one or more rotary emitter electrode of the propeller101or other rotary device (e.g., to the at least one projection108and/or the at least at least one electrically conductive member110of at least one blade102) (and/or potentially to the at least one counter electrode114) of at least about +/−1 kV, or about +/−2 kV, or about +/−3 kV, or about +/−5 kV, or about +/−10 kV, or about +/−15 kV, or about +/−20 kV, or about +/−25 kV, or about +/−35 kV, or about +/−50 kV, or about +/−75 kV, or about +/−100 kV. The maximum voltage applied by the electrical system120may depend, at least partially, upon the particular configuration of the system100and the fluid encountered, and is limited to below the voltage (i.e., the breakdown voltage) that causes electrical breakdown or arcing between the or more rotary emitter electrode and the at least one counter electrode114. It is noted that, in some embodiments, the terminal angular velocity of the rotation of the propeller101or other rotary device about the axis X-X may depend, at least partially, (e.g., a fairly linear dependence) with the applied voltage. For example, as shown inFIG.5, in some embodiments the electrical system120may comprise a voltage source including a first terminal126that is electrically coupled to the one or more rotary emitter electrode of the propeller101(or other rotary device) that supplies the high voltage above corona onset thereto. As also shown inFIG.5, the voltage source may also include a second terminal128that is electrically coupled to the at least one counter electrode114. The voltage source thereby includes an electric potential difference between the first and second terminals126,128of a sufficient voltage that generates corona discharges from the at least one rotary emitter electrode that form the flows of ionic wind emanating therefrom that act to rotate the propeller101or other rotary device about the axis of rotation X-X in the first rotational direction R1. In some embodiments, the electrical system120may apply an electrical current to one or more rotary emitter electrode of the propeller101or other rotary device (e.g., to the at least one projection108and/or the at least at least one electrically conductive member110of at least one blade102) (and/or potentially to the at least one counter electrode114) within the microamps to milliamps range. For example, the electrical system120may apply a direct electrical current (DC) to the at least one conductive member110and/or the at projection108of at least one blade102of the propeller101or other rotary device. As another example, the electrical system120may apply an alternating electrical current (AC) to the at least one conductive member110and/or the at projection108of at least one blade102of the propeller101or other rotary device. As shown inFIG.5, in some embodiments, the electrical system120may include at least one high voltage resistor129in series with the one or more rotary emitter electrode of the propeller101or other rotary device (e.g., to the at least one projection108and/or the at least at least one electrically conductive member110of at least one blade102). In some embodiments (not shown), the electrical system120may include the least one high voltage resistor129in series with the counter electrode114. In some embodiments (not shown), the electrical system120may include the at least one high voltage resistor129in series with the counter electrode114and the at least one high voltage resistor129in series with the one or more rotary emitter electrode of the propeller101or other rotary device. The high voltage resistor129may allow for larger voltages to be applied to the one or more rotary emitter electrode of the propeller101or other rotary device while keeping the current relatively low or lower, but generate higher rotational speeds and EHD thrust than as compared to when the high voltage resistor129is not utilized. In use of such an EHD rotary system100, when the corona current starts flowing, a voltage drop may occur across the at least one high voltage resistor129, but the overall voltage between the one or more rotary emitter electrode of the propeller101or other rotary device and the counter electrode114may be able to be greater than as compared to the system100without the at least one high voltage resistor129, thereby potentially creating a more intense electric field which better accelerates the ions as compared to the system100without the at least one high voltage resistor129. In some embodiments, the voltage applied between the one or more rotary emitter electrode of the propeller101and the counter electrode114to effectuate EHD flow/corona discharge when the high voltage resistor129is utilized may be greater than that compared to when the high voltage resistor129is not utilized, potentially thereby creating a greater EHD flow/corona discharge (i.e., greater torque on the propeller101or other rotary device). In some embodiments, the at least one high voltage resistor129may include an electrical resistance of at least 100 kΩ. In some alternative embodiments (not shown), the EHD system may include a plurality of propellers101or other rotary devices associated with the same shaft portion122. For example, two or more propellers101may independently rotate about, or be affixed to (i.e., rotationally fixed together), a common shaft122and be radially or rotationally offset with respect to each other (i.e., the one or more blades102of the plurality of propellers101may not be angularly aligned about the axis X-X, but rather radially angularly offset about the axis X-X). The rotary emitter electrode of the one or more blades102of the plurality of propellers101may each be electrically coupled to the shaft portion122for receiving the electrical potential (and current) from the power source120via the shaft portion122. The plurality of propellers101may also be associated with a common counter electrode114, such as being positioned within the same cavity115of a counter electrode114. In some such embodiments, the plurality of propellers101may be configured to rotate in the same rotational direction R1about the axis X-X via the generated EHD flow/corona wind (and produce the thrust force/flow component125and the reactionary lift force component127in the same directions). In some other such embodiments, the plurality of propellers101may be configured to rotate in opposing rotational directions R1, R2about the axis X-X via the generated EHD flow/corona wind but produce the thrust force/flow components125and the reactionary lift force components127thereby in the same directions, which may beneficially provide angular momentum compensation. For example, one or more pairs of propellers101of the plurality of propellers101may be configured to rotate about, or be affixed to, a common shaft122and rotate in opposing rotational directions R1, R2about the axis X-X via the generated EHD flow/corona wind but produce the thrust force/flow components125and the reactionary lift force components127thereby in the same directions. In some other alternative embodiments (not shown), the EHD system may include a plurality of axially-adjacent propellers101or other rotary devices associated with the same shaft portion122, with at least one of the propellers101configured as the counter electrode114(i.e., a counter electrode propeller). The EHD system may thereby be void of a counter electrode114other than the at least one counter electrode propeller. For example, a plurality of propellers101(including the at least one counter electrode propeller) may be configured to rotate about, or may be rotationally affixed to, a common shaft122and be fixedly radially or rotationally offset with respect to each other (i.e., one or more blades102of the plurality of propellers101may not be angularly aligned about the axis X-X, but rather radially angularly offset about the axis X-X). The at least one rotary emitter electrode of the one or more blades102of the plurality of propellers101not comprising the at least one counter electrode propeller may be electrically coupled to the shaft portion122for receiving the electrical potential (and current) from the power source120via the shaft portion122. Further, the at least one counter electrode propeller may also be electrically coupled to the electrical potential from the power source120via the shaft portion122, but the at least one counter electrode propeller and the at least one rotary emitter electrode of the one or more blades102of the other propeller(s)101may not be electrically coupled (i.e., may be electrically isolated). At least one of the propellers101could thereby carry the high voltage above corona onset and the at least one counter electrode propeller could carry the ground potential or opposite polarity voltage. The electric field may thereby be generated between the at least one counter electrode propeller and the at least one rotary emitter electrode of the other propeller(s)101for the generation of the EHD flow/corona wind (and thereby rotation about the axis X-X), as described above. The plurality of propellers101may be configured such that the EHD flow/corona wind is generated between the at least one counter electrode propeller and the at least one rotary emitter electrode of the other propeller(s)101such that the plurality of propellers101(including the counter electrode propeller) are rotated about the axis X-X in the first rotational direction. FIG.6illustrates a bottom view of another exemplary propeller201for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller201may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIG.6, in some embodiments the propeller201may include a plurality of blades202that each include a rotary emitter electrode. In alternative embodiments, only one blade202may include a rotary emitter electrode. The rotary emitter electrode includes a plurality of electrically conductive projections208that extend from proximate to the back trailing edge207of the blades202out past the trailing back edge207in a direction extending from the front leading edge205to the trailing back edge207. As shown inFIG.6, each blade202includes a pair of projections208that are positioned radially distal to the hub portion204and radially proximate to the tip206, but radially spaced from the tip206. In some embodiments, the pair of projections208may be positioned radially past the midpoint of the radial length of the blade202toward the tip206. For example, at least one of the projections208may be positioned at about two-thirds the radial length of the blade202from the hub portion204. The electrically conductive radially-extending member210may extend radially from the hub portion204to at least the projection208positioned closest to the tip206. As also shown inFIG.6, the electrically conductive radially-extending member210may radially extend along the radial length of the blade202proximate to the back trailing edge207, such as on the back side209of the blade202adjacent to the back trailing edge207. The radially-extending member210may be relatively thin or narrow, such as a comprising a wire or wire-like member. As discussed above, the radially-extending member mayor may not be exposed. As also discussed above, the radially-extending member210and the projections208of each blade202are electrically coupled. FIG.7illustrates a bottom view of another exemplary propeller301for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller301may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIG.7, in some embodiments the propeller301may include a plurality of blades302that each include a rotary emitter electrode. In alternative embodiments, only one blade302may include a rotary emitter electrode. The rotary emitter electrode includes a plurality of electrically conductive projections308that extend from proximate to the back trailing edge307of the blades302out past the trailing back edge307in a direction extending from the front leading edge305to the trailing back edge307, as shown inFIG.7. As also shown inFIG.7, each blade302also includes an electrically conductive radially-extending portion310that radially extends along the radial length of the blade302proximate to the back trailing edge307, such as on the back side309of the blade302adjacent to the back trailing edge307. As shown inFIG.7, each radially-extending portion310may include a plurality of differing portions. The differing portions of the radially-extending portion310may differ in size (e.g., differing widths extending in a direction between the front leading edge305and the back trailing edge307) and/or being exposed verse insulated (e.g., via an insulating material, such as being covered by an insulating material applied over the portion(s) or by being embedded within the material of the blade302). FIG.8illustrates a bottom view of another exemplary propeller401for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller401may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIG.8, in some embodiments the propeller401may include a plurality of blades402that each include a rotary emitter electrode. In alternative embodiments, only one blade402may include a rotary emitter electrode. As shown inFIG.8, the rotary emitter electrode of each blade402includes an electrically conductive radially-extending portion410that radially extends along the radial length of the blade402proximate to the back trailing edge407, such as on the back side409of the blade402adjacent to the back trailing edge407. As also shown inFIG.8, the rotary emitter electrode of each blade402includes at least one electrically conductive projection408coupled to the radially-extending portion410thereof. The at least one electrically conductive projection408extends from proximate to the back trailing edge407of the blades402out past the trailing back edge407in a direction extending from the front leading edge405to the trailing back edge407, as shown inFIG.8. As shown inFIG.8, each blade402may include one projection408. In some other embodiments, each blade402may include a plurality of projections408. As also shown inFIG.8, the at least one projection408may be oriented on an angle as it extends out past the trailing back edge407. For example, as shown inFIG.8, the at least one projection408extends at an angle extending radially outward from the axis X-X as it extends out past the trailing back edge407(e.g., toward an outer counter electrode extending about the propeller401). As another example, the at least one projection408may extend at an angle extending radially inward toward the axis X-X as it extends out past the trailing back edge407(e.g., away from an outer counter electrode extending about the propeller401). As another example, the at least one projection408may extend at an angle extending from the top surface to the bottom surface409of the blade402, or at an angle extending from the bottom surface409to the top surface of the blade402. FIG.9illustrates a bottom view of another exemplary propeller501for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller501may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIG.9, in some embodiments the propeller501may include a plurality of blades502that each include a rotary emitter electrode. In alternative embodiments, only one blade502may include a rotary emitter electrode. As shown inFIG.9, the rotary emitter electrode of each blade502includes an electrically conductive radially-extending portion510that radially extends along the radial length of the blade502proximate to the back trailing edge507, such as on the back side509of the blade502adjacent to the back trailing edge507. The propeller501may also include an electrically insulative material513that extends over or covers the portion of the radially-extending portion510that is adjacent or facing the front leading edge505. In some embodiments, the electrically insulative material513may comprise a material that is applied or coupled over the previously-exposed outer surfaces of the portion of the radially-extending portion510that is adjacent or facing the front leading edge505. In some other embodiments, the electrically insulative material513may comprise a material of the blade502itself such that the portion of the radially-extending portion510that is adjacent or facing the front leading edge505is embedded within the blade502. As also shown inFIG.9, the portion of the radially-extending portion510that is adjacent or facing the back trailing edge507is exposed and is void of an electrically conductive projection (coupled to the radially-extending portion510thereof). In this way, the rotary emitter electrode of a blade502of the propeller501may include a partially-exposed radially-extending portion510that is relatively flat or smooth such that is it void of any projections extending therefrom past the back trailing edge507. FIGS.10and11illustrate a bottom view of another exemplary propeller601for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller601may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIGS.10and11, in some embodiments the propeller601may include a plurality of blades602that each include a rotary emitter electrode. In alternative embodiments, only one blade602may include a rotary emitter electrode. The rotary emitter electrode of each blade602includes an electrically conductive radially-extending portion610that radially extends along the radial length of the blade602proximate to the back trailing edge607, as shown inFIGS.10and11. As also shown inFIGS.10and11, the radially-extending portion610extends along the back trailing edge back trailing edge607, along the back side609of the blade602adjacent to the back trailing edge607, and along the top side603of the blade602adjacent to the back trailing edge607. In this way, the radially-extending portion610may wrap or over the back trailing edge607and partially onto/over the back side609of the blade602and the top side603of the blade602. In some other embodiments, the radially-extending portion610may not extend over the back side609of the blade602and/or the top side603of the blade602. FIG.12illustrates a bottom view of another exemplary propeller701for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller701may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIG.12, in some embodiments the propeller701may include a plurality of blades702that each include a rotary emitter electrode. In alternative embodiments, only one blade702may include a rotary emitter electrode. As shown inFIG.12, the rotary emitter electrode of each blade702includes an electrically conductive radially-extending portion710that radially extends along the radial length of the blade702proximate to the back trailing edge707, such as on the back side709of the blade702adjacent to the back trailing edge707. The rotary emitter electrode of each blade702includes a plurality of radially-spaced electrically conductive projection708coupled to the radially-extending portion710thereof. As also shown inFIG.12, each blade702includes four radially-spaced electrically conductive projections708coupled to the radially-extending portion710thereof. The plurality of projections708extend from proximate to the back trailing edge707of the blades702out past the trailing back edge707in a direction extending from the front leading edge705to the trailing back edge707. The plurality of projections708are spaced radially from each other and from the hub portion704and the tip706of the blade702. In some embodiments, the plurality of projections708may be evenly radially spaced from each other. In some other embodiments, the plurality of projections708may be unevenly radially spaced from each other. It is noted that each blade702may include fewer than four projections708or more than four projections708. FIG.13illustrates a bottom view of another exemplary propeller801for an EHD rotary system in accordance with the present disclosure, such as the EHD rotary system100described above with respect toFIGS.1-5. It is noted that the propeller801may be utilized with a shaft portion, at least one counter electrode and an electrical system of an EHD rotary system, as described above. As shown inFIG.13, in some embodiments the propeller801may include a plurality of blades802that each include a rotary emitter electrode. In alternative embodiments, only one blade802may include a rotary emitter electrode. As shown inFIG.13, the rotary emitter electrode of each blade802includes an electrically conductive radially-extending portion810that radially extends along the radial length of the blade802proximate to the back trailing edge807, such as on the back side809of the blade802adjacent to the back trailing edge807. The rotary emitter electrode of each blade802also includes a plurality of radially-spaced electrically conductive projections808coupled to the radially-extending portion810thereof that extend from proximate to the back trailing edge807of the blades802out past the trailing back edge807in a direction extending from the front leading edge805to the trailing back edge807, as also shown inFIG.13. As also shown inFIG.13, the physical configuration of at least one projection808of the plurality of projections808of one blade802may differ from at least one other projection808of the plurality of projections808of the blade802. For example, the projections808of a blade802may differ in materials, lengths (extending from the back trailing edge807), widths/diameters, tapers and/or orientations with respect to each other. In the illustrated embodiment shown inFIG.13, the projections808of each blade802differ in lengths extending from the back trailing edge807. As also shown inFIG.13, the projections808of one blade802may differ from the projections808of another blade802. For example, one blade802may include more projections808that another blade802, as shown inFIG.13. As another example, one blade802may include projections808that are differently physically configured than projections808of another blade802. As yet another example, one blade802may include projections808that are arranged/positioned (e.g., radially positioned and/or spaced) differently than projections808of another blade802. As noted above, the EHD rotary systems and methods of the present disclosure may be utilized in a variety of differing applications that make use of at least one of the rotation or torque of the rotary device, the thrust force component if a propeller it utilized, and the lift force/flow component if a propeller is utilized. For example, the EHD rotary systems and methods of the present disclosure that generate rotational motion and thrust/lift can be utilized with previously existing aerial, marine and submarine vehicles. In one embodiment, a drone or unmanned aerial vehicle (UAV)900may utilize an EHD rotary system and method of the present disclosure to provide lift off and flight of the UAV900, as shown inFIG.14. For example, one or more of the EHD rotary systems100with propellers101described above with respect toFIGS.1-5(or an alternative embodiment thereof) may be utilized to form the thrust force component125and the lift force component127to achieve lift off and flight of the UAV900. In some embodiments, a plurality of propellers102and corresponding counter electrodes114may be utilized, and attached to an UAV body or fuselage via one or more articulated joint116, operable to direct the thrust force component125, and thereby lift force component127, created via rotation of the propellers102of the UAV during liftoff and flight. In some embodiments, the UAV900well may comprise a microdrones, having a weight on the order of about 0.5 gram up to about 500 grams, for example. FIG.15illustrates another aerial vehicle that may utilize an EHD rotary system and method of the present disclosure to provide lift off and/or flight of the vehicle. As shown inFIG.15, a blimp or lighter-than-air airship1000may incorporate one or more of the EHD rotary systems100with the propellers101described above with respect toFIGS.1-5(or an alternative embodiment thereof) may be utilized to liftoff and fly the airship1000. The airship1000may make use of the Archimedes buoyant force from the lighter-than-air balloon thereof, and the EHD forces of the rotary systems100may be able to produce a sufficient lift force component127relative to the overall weight of the airship1000to achieve liftoff and flight thereof. The lift force127may also be utilized to fly or translate the airship1000through the air after liftoff. FIGS.16and17illustrate another aerial vehicle that may utilize an EHD rotary system and method of the present disclosure to provide lift off and/or flight of the vehicle. As shown inFIGS.16and17, a rotorcraft1100(e.g., a helicopter) may incorporate one or more of the EHD rotary systems100with the propellers101described above with respect toFIGS.1-5(or an alternative embodiment thereof) to provide liftoff and fly the rotorcraft1100. It is noted that the counter electrode of the EHD rotary system is not depicted inFIGS.16and17. The rotorcraft1100may make use of the EHD rotary system to rotate a first propeller101A to provide the thrust force component125, and thereby the lift force component127, to achieve liftoff and flight of the rotorcraft1100. Further, as shown inFIGS.16and17, the rotorcraft1100may also make use of the EHD rotary system to rotate a second propeller101B to further provide the thrust force component125, and thereby the lift force component127, to achieve liftoff and flight of the rotorcraft1100. In such embodiments, the first and second propellers101A,101B may be arranged and otherwise configured such that the rotational directions R1, R2thereof that create the thrust force component125and the lift force component127via the EHD thrust/corona wind thereof, are opposing or opposite angular or rotational directions. For example, the orientation/configuration of the first and second propellers101A,101B may be opposing, such that the rotational direction R1of the first propeller101A that generates the thrust force component125and the lift force component127(via the EHD thrust/corona wind thereof) is opposite the rotational direction R2of the second propeller101B that generates the thrust force component125and the lift force component127(via the EHD thrust/corona wind thereof). The respective opposing rotational directions R1, R2of the first and second propellers101A,101B may thereby inherently compensate for the opposite angular momentum/torque present in single rotorcraft designs, and may obviate a for a tail rotor (as shown inFIG.17). In such embodiments, the first and second propellers101A,101B may rotate about the shaft portion122in the opposing rotational directions R1, R2as the EHD thrust/corona wind effectuates rotation of the first and second propellers101A,101B as compared to rotation/torque of the shaft portion122as in traditional rotorcraft1100. FIG.18illustrates a land-based vehicle that may utilize an EHD rotary system and method of the present disclosure to power or translate the vehicle along/across a ground surface. As shown inFIG.18, a wheeled vehicle1100may incorporate one or more of the EHD rotary systems100with at least one propeller101described above with respect toFIGS.1-5(or an alternative embodiment thereof) to translate the wheeled vehicle1100along/across aground surface via wheels thereof. The wheeled vehicle1100may make use of the EHD forces of the EHD rotary systems100to rotate the at least one propeller101to produce a sufficient lift force component127via the at least one propeller101relative to the wheeled vehicle1100to achieve motion or translation of the wheeled vehicle1100along/across a ground surface (via the wheels thereof). FIG.19illustrates a land-or-water based vehicle that may utilize an EHD rotary system and method of the present disclosure to power or translate the vehicle along/across a ground and/or water surface. As shown inFIG.19, a hovercraft1300may incorporate one or more of the EHD rotary systems100with at least one propeller101described above with respect toFIGS.1-5(or an alternative embodiment thereof) to pressurize a bladder thereof and/or translate the hovercraft1300along/across a ground and/or water surface via the bladder. The hovercraft1300may make use of the EHD forces of the EHD rotary systems100to rotate the at least one propeller101to produce a sufficient thrust force125via the at least one propeller101to pressurize the bladder, and/or make use of the EHD forces of the EHD rotary systems100to rotate the at least one propeller101to produce a sufficient lift force component127via the at least one propeller101relative to the hovercraft1300to achieve motion or translation of the hovercraft1300along/across a ground and/or water surface (via the bladder thereof). FIG.20illustrates a water-based vehicle that may utilize an EHD rotary system and method of the present disclosure to power or translate the vehicle along/across a water surface. As shown inFIG.20, a fan boat1400may incorporate one or more of the EHD rotary systems100with at least one propeller101described above with respect toFIGS.1-5(or an alternative embodiment thereof) to translate the fan boat1400along/across a water surface via the hull of the fan boat1400. The fan boat1400may make use of the EHD forces of the EHD rotary systems100to rotate the at least one propeller101to produce a sufficient lift force component127via the at least one propeller101relative to the fan boat1400to achieve motion or translation of the fan boat1400along/across a water surface (via the hull thereof). FIG.21illustrates an EHD-driven pump that utilizes an EHD rotary system and method of the present disclosure to power the pump. As shown inFIG.20, a pump1500, such as a centrifugal pump, may incorporate an EHD rotary system according to the present disclosure to rotate an impeller or rotor1501of the pump1500within a housing or casing1550to pressurize the fluid and create an outflow of the fluid from the pump1500(i.e., translate a fluid through the pump1500). The casing1550may thereby form an enclosure that defines or contains the fluid environment that the impeller1501rotates within. As shown inFIG.21, the impeller1501may rotate about an axis of rotation X-X in a first rotational direction R1to pressurize the fluid and create an outflow of the fluid from the pump1500. The impeller1501may rotate about the axis of rotation X-X in the first rotational direction R1on/over a shaft portion1522, or the impeller1501may be fixedly coupled to a shaft portion1522that is configured to rotate about the axis of rotation X-X in the first rotational direction R1. As shown inFIG.21, the impeller1501may include a plurality of radially extending vanes1502extending outwardly from proximate to an inlet or intake aperture1552to tips1506thereof, and extend axially from a base plate portion. The inlet1552provides for an inlet flow of the fluid into the casing1550. An interior side of the casing1550includes an outlet aperture1554that provides for an outlet flow of the pressurized fluid out of the casing1550. The impeller1501(e.g., at least the back side1507of the vanes1502and the face of the base plate portion) may be formed of an electrically insulative material and/or covered or encased by an electrically insulative material. The vanes1502include front leading side surfaces1505that “push” the fluid within the casing1550radially during rotation of the impeller1502in the first rotational direction R1, and back trailing side surfaces1505that trail the leading side surfaces1505during rotation of the impeller1502in the first rotational direction R1. The back trailing side surfaces1505of at least some of the vanes1502include at least one electrically conductive rotary emitter electrode. For example, the rotary emitter electrode of the back trailing side surfaces1505of the vanes1502may include at least one electrically conductive projection1508extending outwardly therefrom, and at least one electrically conductive radially-extending member1510extending radially from at least the at least one projection1508at least partially to the shaft portion1522. The impeller transfers rotational energy thereof (generated EHD flow/corona wind, as explained below) that drives the pump1500to the fluid being pumped by accelerating the fluid outwards from the axis of rotation X-X. The velocity achieved by the impeller1502transfers into pressure when the outward movement of the fluid is confined by the casing1550. As shown inFIG.21, at least a portion of the interior sides of the casing1550extending axially about at least a portion of the impeller1501about the axis of rotation X-X includes a counter electrode1514. The counter electrode1514may be similarly configured to the counter electrode114described above. The impeller1501may be similarly configured to the propellers or other rotary devices101,201,301,401,501,601,701and/or801described above, and/or the vanes1502may be similarly configured to the blades102,202,302,402,502,602,702and/or802described above. In this way, pump1500may include an electrical system/voltage source that applies a potential difference between the at least one rotary emitter electrode of the plurality of vanes1502and the counter electrode1514that generates an electrical field and corona discharges from the at least one rotary electrode that form flows of ionic wind emanating therefrom that rotate the impeller1501about the axis of rotation X-X in the first direction R1to pressurize the fluid within the casing1550and from an outlet flow of the fluid from the outlet aperture1554, as shown inFIG.21. FIG.22illustrates an EHD-driven sensor, gauge or meter that utilizes an EHD rotary system and method of the present disclosure to rotate a rotary device and measure the rotation thereof based on changes of fluid within the sensor to obtain information about the fluid. As shown inFIG.20, a sensor1600may incorporate an EHD rotary system according to the present disclosure to rotate a rotary device1601with a plurality or radially-extending and angularly spaced blades1602. The blades1602may or may not include an angle of attack/pitch and/or form an airfoil shape to generate thrust and lift forces upon rotation about the axis X-X. As shown inFIG.22, the rotary device1601may rotate about an axis of rotation X-X in a first rotational direction R1within a casing or housing1650. The casing1650may form an enclosure that defines or contains the fluid environment in which that the rotary device1601rotates within. The rotary device1601may rotate about the axis of rotation X-X in the first rotational direction R1on/over a shaft portion1622within the casing1655, or the rotary device1601may be fixedly coupled to a shaft portion1622that is configured to rotate about the axis of rotation X-X in the first rotational direction R1within the casing1655. As shown inFIG.22, the casing1650may include an inlet aperture1652that provides for an inlet flow of the fluid into the casing1650, and an outlet aperture1654that provides for an outlet flow of the fluid out of the casing1650. As such, the interior of the casing1650, the inlet aperture1652and the outlet aperture1654may form a flowpath through the casing1650. The blades1602may include front leading edges or surfaces1605and back trailing edges or surfaces1607that trail the respective leading surfaces1605during rotation of the blades1602in the first rotational direction R1. The back trailing edges1607of at least some of the blades1602include at least one electrically conductive rotary emitter electrode. For example, the rotary emitter electrode of the back trailing edges1607of the blades1602may include at least one electrically conductive projection1608extending outwardly therefrom, and at least one electrically conductive radially-extending member1610extending radially from at least the at least one projection1608at least partially to a hub portion1604of the rotary device1601and/or the shaft portion1622. The rotary device1601(e.g., at least the trailing back edges1607) may be formed of an electrically insulative material and/or covered or encased by an electrically insulative material. As shown inFIG.22, at least a portion of the interior sides of the casing1650extending axially about the rotary device1601about the axis of rotation X-X includes a counter electrode1614. The counter electrode1614may be similarly configured to the counter electrode114described above. The rotary device1601may be similarly configured to the propellers or other rotary devices101,201,301,401,501,601,701and/or801described above, and/or the blades1602may be similarly configured to the blades102,202,302,402,502,602,702and/or802described above. In this way, the sensor1600may include an electrical system/voltage source that applies a potential difference between the at least one rotary emitter electrode of the plurality of blades1602and the counter electrode1614that generates an electrical field and corona discharges from the at least one rotary electrode that form flows of ionic wind emanating therefrom that rotate the rotary device1601about the axis of rotation X-X in the first direction R1, as shown inFIG.22. As the corona discharge from the at least one rotary emitter electrode of the plurality of blades1602is sensitive to the characteristics (e.g., pressure, humidity, particulate amount, etc.) of the fluid within the casing1650in which the rotary device1601rotates, the rotation of the rotary device1601about the axis X-X can be monitored as one or more fluids (gaseous or liquid) flow into and/or through the casing1650to determine or derive (implicitly) information about the fluid that the rotary device1601encounters. For example, the sensor1600may be used in conjunction with, or include, a tachometer, torque meter or other rotational metric measurement tool to measure a characteristic (angular speed, torque, acceleration/deceleration, etc.) of the rotation of the rotary device1601about the axis X-X to determine or derive (implicitly) information about the fluid within the casing1650that the rotary device1601encounters. FIGS.23and24illustrate an EHD-driven torque generator or motor that utilizes an EHD rotary system and method of the present disclosure to rotate a rotary device and, thereby, a shaft to provide rotational motion and force/torque to an object. As shown inFIGS.23and14, a torque generator or motor1700may incorporate an EHD rotary system according to the present disclosure to rotate a rotary device1701with a plurality or radially-extending and angularly spaced blades1702. The blades1702may or may not include an angle of attack/pitch and/or form an airfoil shape to generate thrust or lift forces upon rotation about the axis X-X. As shown inFIGS.23and24, the rotary device1701may rotate about an axis of rotation X-X in a first rotational direction R1within a casing or housing1750. The rotary device1701may rotate about the axis of rotation X-X in the first rotational direction R1on/over a shaft portion1722within the casing1755, or the rotary device1701may be fixedly coupled to a shaft portion1722that is configured to rotate about the axis of rotation X-X in the first rotational direction R1within the casing1755. The casing1750may form an enclosure that defines or contains the fluid environment in which that the rotary device1701rotates within. It is noted that the fluid environment in which that the rotary device1701rotates may effect the torque and/or rotational speed of the shaft portion1722generated by the EHD-driven torque generator or motor1700. In some embodiments, the enclosure of the casing1750comprises a gaseous environment, such as an environment comprising air, O, O2, O3, CO, CO2, Ar, NH3, H, CH4, Ne, Natural gas, N, Cl, SF6, WF6, Kr, Xe, He, water vapor, or a combination thereof. In some embodiments, the enclosure of the casing1750comprises a gaseous environment that is denser than air at the same temperature and pressure, which may generate a higher torque and/or angular speed as compared to an environment of air. In some embodiments, the enclosure of the casing1750comprises a gaseous environment that is less dense than air at the same temperature and pressure. In some embodiments, the enclosure of the casing1750comprises a liquid environment. For example, the casing1750comprise an environment comprising water, an oil, glycerin, liquid silicone, a halocarbon, hydrogen peroxide or a combination thereof. The blades1702may include front leading edges or surfaces1705and back trailing edges or surfaces1707that trail the respective leading surfaces1705during rotation of the blades1702in a first rotational direction R1, as shown inFIGS.23and24. The back trailing edges1707of at least some of the blades1702include at least one electrically conductive rotary emitter electrode. For example, the rotary emitter electrode of the back trailing edges1707of the blades1702may include at least one electrically conductive projection1708extending outwardly therefrom, and at least one electrically conductive radially-extending member1710extending radially from at least the at least one projection1708at least partially to a hub portion1704of the rotary device1701and/or the shaft portion1722, as shown inFIGS.23and24. The rotary device1701(e.g., at least the trailing back edges1707) may be formed of an electrically insulative material and/or covered or encased by an electrically insulative material. As shown inFIGS.23and24, at least a portion of the interior sides of the casing1750extending axially about the rotary device1701about the axis of rotation X-X includes a counter electrode1714. The counter electrode1714may be similarly configured to the counter electrode114described above. The rotary device1701may be similarly configured to the propellers or other rotary devices101,201,301,401,501,601,701and/or801described above, and/or the blades1702may be similarly configured to the blades102,202,302,402,502,602,702and/or802described above. In this way, the torque generator or motor1700may include an electrical system/voltage source1720that applies a potential difference between the at least one rotary emitter electrode of the plurality of blades1702and the counter electrode1714that generates an electrical field and corona discharges from the at least one rotary electrode that form flows of ionic wind emanating therefrom that rotate the rotary device1701about the axis of rotation X-X in the first direction R1, as shown inFIGS.23and24. In this way, the EHD flow/corona wind may torque and rotate the rotary device1701about the axis of rotation X-X in the first direction R1to rotate the shaft portion1722about the axis X-X in the first direction R1. As shown inFIG.23, the torque and rotational/angular motion of the shaft portion1722of the torque generator or motor1700(generated via the EHD flow/corona wind) may be applied to any mechanism, object or system1760to effectuate rotational motion thereof or otherwise utilize the torque of the shaft portion1722. FIGS.25-27illustrate another EHD-based rotary system1800and related methods according to the present disclosure. Torque and/or angular motion provided by the rotary system1800(generated via EHD flow/corona wind) may be utilized for any number of a variety of differing potential applications by any number of a variety of differing mechanisms. For example, the EHD rotary system1800and related method of the present disclosure may be utilized with any fan, pump, turbine, jet, propeller, rotor or turbine. As shown inFIGS.25-27, the EHD rotary system1800may include a rotary device1601with an outer cylindrical member or portion1802, a hub portion1804and one or more radial support members or portions1860extending between and coupling the cylindrical portion1802and the hub portion1804. The rotary device1801also includes an axis or rotation X-X about which the cylindrical portion1802, hub portion1804and one or more radial support members1860rotate. The hub portion1804may rotate about the axis of rotation X-X on/over a shaft portion (not shown), or the rotary device1601may be fixedly coupled to a shaft portion (not shown) that is configured to rotate about the axis of rotation X-X. The hub portion1804and at least one of the one or more radial support members1860may include an electrically conductive portion that electrically couples the shaft portion and the cylindrical portion1802. The cylindrical portion1802may include an axially-elongated circular outer surface1807that includes a plurality of angularly (e.g., circumferentially) and/or axially spaced or positioned electrically conductive emitter electrodes thereon, as shown inFIGS.25-27. The plurality electrically conductive emitter electrodes are electrically coupled to the shaft portion (not shown) via the electrically conductive portions of the hub portion1804and the one or more radial support members1860such that the relatively high voltage above corona onset is applied to the plurality of emitter electrodes via the shaft portion, as described above. As shown inFIGS.25-27, the plurality of electrically conductive emitter electrodes comprise a plurality of electrically conductive axially-extending members1810and a plurality of electrically conductive projections1808that extend away from the outer surface1807of the cylindrical portion1802. The axially-extending members1810and the projections1808are electrically coupled, and the axially-extending members1810are electrically coupled to the shaft portion (not shown), such as via the electrically conductive portions of the hub portion1804and the one or more radial support members1860. The plurality of electrically conductive projections1808may be oriented at least generally in a common angular or rotational direction about the axis X-X, as shown inFIGS.25-27. For example, each of the plurality of projections1808may extend away from the outer surface1807of the cylindrical portion1802in the same direction, as shown inFIGS.25-27. In some embodiments, each of the plurality of projections1808may extend tangentially away from the outer surface1807from the cylindrical portion1802in the same direction, as shown inFIGS.25-27. In some embodiments, each of the plurality of projections1808may extend away from the outer surface1807from the cylindrical portion1802on an angle that extends radially-outwardly and angularly. The axially-extending members1810may be angularly or rotationally spaced from each other, as shown inFIGS.25-27. The axially-extending members1810may extend radially along the cylindrical portion1802and include a plurality of axially-spaced projections1808extending therefrom, as shown inFIGS.25and26. In such embodiments, the axially-extending members1810may extend to, or proximate to, a first axial end of the cylindrical portion1802. The first axial end of the cylindrical portion1802may include a circumferentially extending electrically conductive contact (not shown) that electrically couples the axially-extending members1810together. In some such embodiments, the contact (not shown) may be coupled to the shaft portion (not shown), such as via the electrically conductive portions of the hub portion1804and the one or more radial support members1860. The axially-extending members1810may or may not be exposed. For example, in some embodiments, the axially-extending members1810may be covered or encased by an electrically insulative material. In some other embodiments, at least a portion of the axially-extending members1810may be exposed. In some embodiments, the cylindrical portion1802may comprise, or be covered or encased by, an electrically insulative material such that the cylindrical portion1802itself is not electrically coupled to the rotary emitter electrodes and does not interfere with the formation of an electric field between the rotary emitter electrodes (e.g., the plurality of projections1808) and a counter electrode1814. As shown inFIG.27, the rotary device1601may be positioned within a cavity1815of a cylindrical electrically conductive counter electrode1814(as discussed above). The rotary device1601is configured to rotate about the axis X-X within the cavity1815of the counter electrode1814. The counter electrode1814may be similarly configured to the counter electrode114described above. In this way, the EHD rotary system1800may include an electrical system/voltage source (not shown) that applies a potential difference between the plurality of rotary emitter electrodes (e.g., the plurality of projections1808) on the outer surface1807of the cylindrical portion1802and the counter electrode1614that generates an electrical field and corona discharges from the plurality of rotary electrodes that form flows of ionic wind emanating therefrom that rotate the rotary device1601about the axis of rotation X-X in a first direction R1, as shown inFIGS.25-27. As shown inFIG.27, in some embodiments the plurality of projections1808may include arm portions1809that extend (at least partially) radially away from the outer surface1807of the cylindrical portion1802. The plurality of projections1808may extend angularly or in a rotational direction therefrom (e.g., tangentially with respect to the rotational or angular path of the projections1808or angularly and radially outwardly), as shown inFIG.27. The arm portions1809may thereby space the plurality of projections1808from the outer surface1807of the cylindrical portion1802, such as to prevent the outer surface1807of the cylindrical portion1802from interfering with the EHD flow/corona wind emenating from the plurality of projections1808. In some alternative embodiments, as opposed to the axially-extending members1810, the rotary device1601may include a plurality of axially-spaced circumferentially-extending (or angularly-extending) members that include a plurality of circumferentially or angularly spaced electrically conductive projections1808extending therefrom. EXAMPLES Hereinafter, inventions of the present disclosure will be described in detail with reference to examples, but the Examples are expressly not meant to limit the scope of the present inventions. Commercial plastic propellers were converted to EHD propellers by equipping them with conductive electrodes on the blades. A dielectric layer was normally used to partially cover the electrode leaving the edge exposed to air at the trailing edge of the blade. An axial shaft was used to inject high voltage (HV) in the blades through conductive material running along the blades. A surrounding ground electrode (a metal cylinder) was utilized to create an intense electric field when high-voltage above corona onset was applied to the electrodes. Corona wind formation was mediated by electric field forces acting both on gaseous charges and propeller. Conservation of overall momentum lead to forces acting on the propeller blades, which created torque and eventually propeller rotation. Propellers were tested in the negative polarity, and ranged from 0.2 g and 3.5 cm diameter to 28 g and 25 cm diameter. They were able to spin and fly either off the shaft or sliding upwards on the shaft. Specifically, conductive material was aligned along the trailing edge of the blades. The material conductive was copper foil tape with double conductive adhesive of 0.035 mm thickness. Bare conductive electric paint and 46 AWG tungsten wire were also used as the conductive material of the electrode components. Metal pins were cut to about 1 cm length and used either with sharp or blunt ends (both designs supported flight). The pins were electrically connected to the copper tape or a metal wire set along at least one propeller blade. 3M™ PTFE Film Electrical Tape 61 was used for partial covering of the conductive material. However, some propellers did not include the insulation, and the propeller liftoff was also obtained with such propellers At (or towards) the end of each blade tip, a metal pin about 0.5-1 cm long (connected to the conductive material on the blade) was attached. The metal pins were oriented substantially orthogonally to the local rotational radius. The propellers were mounted on a high voltage shaft or balanced on a sharp vertical shaft, which injects high voltage (HV) into the blades through the conductive material running along them. In one alternative setup, a metal bead was placed coaxially on the HV shaft and sustained the weight of the propeller so that it could spin or slide on the vertical shaft. High voltage was applied from Glassman power supplies (+60 kV—PS/FR60P05.0, −60 kV—PS/FR60N05.0) or Gamma High Voltage Research model ES60R 20 W/DAM/OL. An intense non-uniform electric field was created on the propeller electrode to generate ionic wind in order to make the EHD propeller spin. The counter electrodes were mainly metal cylinders, but metal disks and parallel plates were also tested. Each of the tested counter electrode designs supported liftoff for some of the tested propellers. Different counter electrode materials were also tested. Similar sizes of copper, aluminum, and steel cylinders did not result in significant rotational speed changes in the experiments. Some of the propellers were placed coaxially inside a grounded cylindrical electrode, and some propellers were positioned outside the cylindrical electrode (e.g., beneath, above and sideways). Each of the positions of the cylindrical electrode supported propeller liftoff, although some required larger voltage values. A Photron high speed camera FASTCAM SA-X2 1000K-M4—Monochrome 1000K with 64 GB memory was used for recording the rotational motion of the propellers. The frame rates ranged from 500 to 30,000 fps. Determination of the rotational speeds were manually performed by analyzing individual video frames or using a digital laser tachometer DT-2234C+, with 0.05% accuracy. Different EHD propellers and counter electrodes were tested to voltages up to −60 kV. A first tested propeller was 27.8 g, 25 cm in diameter, and included copper tape on the trailing edges of the blades and two pins per blade. Flight of the first propeller was induced at −52 kV, 4.7 cm below a centered 61 cm diameter ground disk counter electrode. A second tested propeller was 0.46 g, 4.6 cm in diameter, and included conductive ink on the trailing edges of the blades. Flight of the second propeller was induced at −28.9 kV in a 10.5 cm diameter, 11 cm height copper cylinder counter electrode. A third tested propeller was 8 g, 12.5 cm in diameter, and included copper tape on the blades and one pin per blade. Flight of the third propeller was induced at −60 kV, 4.7 cm below a centered 61 cm diameter ground disk counter electrode. A fourth tested propeller was 0.58 g, 4 cm in diameter, and included copper tape and one pin on the trailing edges of the blades. Flight of the fourth propeller was induced at −32 kV in a 10.5 cm diameter, 11.2 cm height copper cylinder counter electrode. Another set of propellers were tested for liftoff regimes. A first tested propeller was 0.582 g, 4 cm in diameter, and included three blades with copper tape and one pin per blade; the applied voltage at takeoff was −25.8 kV, and the test was performed in a 10.5 cm diameter, 11 cm height grounded copper cylinder counter electrode. A second tested propeller was 8 g, 12.5 cm in diameter, and included six blades with two opposite blades equipped each with copper tape and one pin; the applied voltage at takeoff was −36 kV, and the test was performed in the 10.5 cm diameter, 11 cm height grounded copper cylinder counter electrode. Rotational speed characteristics for a 44 g, 25 cm radius, five-blade propeller designed with copper tape along the blade trailing edges and placed axially above and parallel to a 61 cm diameter ground metal disk counter electrode were also tested. The distance between the ground disk and the parallel propeller plane was 3 cm, and the distance to ground disk was 6 cm. A linear variation of the rotational speed with applied voltage was confirmed by a regression coefficient. In the case of a propeller set axially, parallel, and above a disk ground electrode, the induced rotational speed was proportional to the applied voltage. The vertical thrust and rotational speed was a function of applied voltage for an only partially optimized propeller-cylinder system (negative polarity). Positive polarity was also tested, but the voltage range for induced rotation (after propeller rotation started and before breakdown was reached) was only 2 kV. A linear range of the terminal rotational speed with applied voltage (at voltages much larger than the corona onset) was apparent in some tests conducted in air, CO2, and SF6. Testing a six-blade propeller showed that current increased linearly with the number of active blades (equipped with copper tape). A similar linear increase with the number of EHD-active blades (except for the case of a single coper tape blade) was apparent for the vertical thrust of a tested propeller. Thrust measurements were performed in the negative polarity. The distance between high voltage electrode and the scale allowed for minimum interference so that the scale would always return to zero when the voltage was turned off. In some experiments, the rotational speed increased as a propeller was lowered in the cylinder axis of the cylindrical counter electrode while keeping the voltage constant. When the length of the copper tape on the blades was extended from the center of the blade along a radius, the rotational speed appeared to increase linearly. When the copper tape (or wire) was translated across the width of the blade, while keeping its length and voltage constant, the rotational speed decreased towards the central positioning and ceases there. When the electrode was translated past the central position on the blade, the propeller reversed direction of rotation. The corresponding current had a minimum on the central position. In some experiments, the angular speed of the propellers varied with the variation of the copper tape width on the blade for constant voltage applied while one side of the tape was aligned on the trailing edge of the blade. Testing was performed using insulating electrical tape on the non-trailing edge of the copper tape; also, with no insulation. Both situations indicated the presence of a maximum for the rotational speed. When the copper tape fully covered the width of a blade of a propeller, the speed reduced significantly in the insulated case and dropped to zero if no insulation is present—indicating that the ionization region was much contained within the thickness of the insulating tape which disrupted the EHD flow and thrust. It also pointed to the EHD flow being the actuator in the system. Direct flow visualizations around the blade showed that no propeller rotation was present before EHD flow was initiated. For comparison, a 46 AWG wire electrode (the same length as the copper tape) was used on some blades. Propeller rotation was also controlled by the design of the counter electrode. The inner side of a glass cylinder counter electrode was only partly covered with vertical aluminum foil strips, and an EHD propeller was tested inside it at constant voltage. Full metal coverage of the cylinder provided maximum rotational speed, while current was maximum when coverage area was about 60%. Propeller flight was achieved only in certain situations, which showed that the resultant force on the propeller was significantly influenced by the counter electrode design. The current voltage characteristics of a propeller for both negative and positive polarities along with the corresponding rotational speed was also tested. Breakdown occurred at much smaller voltages for positive than for the negative polarity. Various other propeller setups showed similar outcomes, and they led to higher rotational speed achievable in the negative polarity in the tests. Negative polarity was thereof particularly studied. In some experiments, propellers were tested at and below atmospheric pressure (Po) using the same EHD propeller-cylinder system each time. At each pressure, maximum voltage (before breakdown occurs) was applied. The lowest pressure at which propeller flight was still obtained was between 0.7 PO and 0.8 PO. A quasi-linear variation of rotational speed with pressure was apparent, thereby predicting much larger achievable EHD forces at pressures above PO. Propeller performance in other gases was also tested at atmospheric pressure. The maximum achievable rotational speed at PO (with the used power supply) clustered around 28 rot/s for in air, carbon dioxide (CO2), and sulfur hexafluoride (SF6). The corresponding voltages were −60 kV in SF6, −27 kV in air, and −31.5 kV in CO2. As the specific gravity for CO2 is larger than 1.5 and for SF6 is larger than 5, it was apparent that more EHD force was generated for CO2 than in air, and much more in SF6 at the specified parameters. Testing at PO for krypton (Kr), nitrogen (N), and argon (Ar) showed very low values for propeller rotational speed, whereas testing in helium (He) resulted in insufficient EHD torque to compensate for friction. In addition, no propeller rotation was achieved below PO for Kr, N, Ar, and He. In some experiments, a comparison of propeller performance in gases at atmospheric pressure were tested, as described above. An EHD propeller of 2.3 g, 12.6 cm in diameter was placed axially on a high voltage metal rod and 2.5 cm inside an 18 cm high, 14 cm-inner diameter ground cylinder counter electrode. The system was placed in a glass enclosure equipped with a three-way “open/close/open” valve connected to a rotary vacuum pump, pressure gage, and gas supply pipe. Pressure inside the glass bell was controllable in 0.1 PO increments. Data was collected at the highest voltage achievable before breakdown or the power supply tripped. All liquid gases used has 5.0 purity and were obtained from Linde Gas, Romania. The propeller in SF6 had a voltage of 60 kV, a current of 0.33 mA, a rotational speed of 27 rot/s, a gas density of 6.17 kg/m{circumflex over ( )}3, a gas specific gravity of 5.12, and flight of the propeller was achieved. The propeller in CO2 had a voltage of 27 kV, a current of 0.34 mA, a rotational speed of 29 rot/s, a gas density of 1.81 kg/m{circumflex over ( )}3, a gas specific gravity of 1.51, and flight of the propeller was achieved. The propeller in air had a voltage of 27 kV, a current of 0.34 mA, a rotational speed of 29 rot/s, a gas density of 1.2 kg/m{circumflex over ( )}3, a gas specific gravity of 1, and flight of the propeller was not achieved. The propeller in Kr had a voltage of 15 kV, a current of 0.06 mA, a rotational speed of 6 rot/s, a gas density of 3.74 kg/m{circumflex over ( )}3, a gas specific gravity of 3.11, and flight of the propeller was not achieved. The propeller in N had a voltage of 7.5 kV, a current of 0.17 mA, a rotational speed of 2 rot/s, a gas density of 1.16 kg/m{circumflex over ( )}3, a gas specific gravity of 0.96, and flight of the propeller was not achieved. The propeller in Ar had a voltage of 8 kV, a current of 0.07 mA, a rotational speed of 1.8 rot/s, a gas density of 1.64 kg/m{circumflex over ( )}3, a gas specific gravity of 1.36, and flight of the propeller was not achieved. The propeller in He had a voltage of 6 kV, a current of 0.2 mA, a rotational speed of 0 rot/s, a gas density of 0.163 kg/m{circumflex over ( )}3, a gas specific gravity of 0.14, and flight of the propeller was not achieved. The tested EHD systems and propellers were also compared to classic EHD thrusters. Classic EHD thrusters accelerate ions one direction and thrust is obtained in the opposite direction. With the rotational EHD thrusters described herein, in some embodiments most of the ions may accelerate within a rotational plane of the propeller, while the propeller-counter-electrode system converts the thrust to a direction orthogonal to it. The 90 degree shift is obtained at the expense of frictional losses in the propeller shaft and due to the gaseous media flow through the system. For some tested rotational EHD devices of the present disclosure, using power supply readings for voltage and current at propeller liftoff in air, lead to estimated lower limits for thrust to power ratio of 5.4 N/kW at −43 kV and thrust density of 3.73 N/m2 for attested 28 g, 25 cm in diameter four-blade propeller in a cylinder counter electrode of 30.5 cm in diameter. Also, 8.3 N/kW (the top value obtained in air) and 1.57 N/m2 at −19.5 kV were lower estimates at liftoff for a tested 2.3 g, 12.6 cm diameter two-blade propeller in a cylinder counter electrode of 13.5 cm in diameter. The values were competitive to values reported for an ionic plane, which had a sustained flight at about 5 N/kW and a designed/desired thrust density of 3 N/m{circumflex over ( )}2. In addition, lower-limits estimated from a tested 2.3 g-propeller liftoff at atmospheric pressure gave 14.3 N/kW and 1.57 N/m2 at −21 kV in CO2 and 30.48 N/kW and 1.57 N/m2 at −21 kV in SF6. It is noted that thrust to power ratio is a quantity that can change dramatically with the voltage applied and also with slight modifications in the electrode design or positioning. The estimates were calculated at the propeller liftoff/flight (i.e., EHD thrust is at least the weight of the propeller). The EHD rotational systems were only partly optimized. The EHD forces were spatially concentrated towards the end terminal of the HV electrode emitters placed towards the blade ends. Although relatively small, the EHD forces could generate significant torque through the lever arm. Rotational kinetic energy is incrementally accumulated and stored in the propeller rotation. This is phenomenon differs from classic EHD thrusters which do not integrate energy for liftoff. The EHD rotational systems of the present disclosure appeared to be versatile with respect to thrust direction and control. Rotational direction was controlled by the position of the emitter electrodes on the blades. Further, in some embodiments, the propellers or other rotary devices may include mobile or repositionable emitters to control the direction of rotation of the propeller/rotary device. As another benefit, a pair of left and right propellers of the present disclosure could be configured to spin on the same axis and create thrust while keeping zero the total angular momentum. As such, pairs of left-right spinning propellers on the same axis could b utilized for an EHD aerial vehicle (e.g., a drone) with no need for additional angular momentum compensation used in classical helicopters. Still further, classical EHD thrusters may use multiple stages to enhance thrust. The EHD rotational systems of the present disclosure can achieve a similar effect with additional blades or multiple propellers placed on the same axis. In contrast to classic thrusters, the EHD rotational systems of the present disclosure appear to result in more homogeneous thrust (e.g., potentially due to rapid rotation of the emitter propeller electrodes) as compared to classical EHD thrusters, potentially providing a smoother vertical thrust (potentially even in non-uniform distributions of the electric field) as compared to classical EHD thrusters. It is to be understood that the above description is intended to be illustrative, and not restrictive. For example, the above-described examples (and/or aspects thereof) may be used in combination with each other. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the various examples without departing from their scope. For example, the material, conductivity, resistivity, shape, orientation and position of the rotary electrode(s) and the counter electrode(s) may vary without departing form the spirit and scope of the inventions. As another example, the composition, configuration or dielectric strength of the insulating material or portions may vary without departing form the spirit and scope of the inventions. As yet another example, the pitch, shape, size and number of the blades of the propellers or other rotary devices may vary without departing form the spirit and scope of the inventions. As a further example, the number of propellers or other rotary devices used in conjunction with a single ground electrode may vary without departing form the spirit and scope of the inventions. As another example, the amount of voltage and/or current applied to the rotary emitter electrodes may vary without departing form the spirit and scope of the inventions. While dimensions and types of materials may be described herein, they are intended to define parameters of some of the various examples, and they are by no means limiting to all examples and are merely exemplary. Many other examples will be apparent to those of skill in the art upon reviewing the above description. The scope of the various examples should, therefore, be determined with reference to the appended claims, along with the full scope of equivalents to which such claims are entitled. In the appended claims, the terms “including” and “in which” are used as the plain-English equivalents of the respective terms “comprising” and “wherein.” Moreover, in the following claims, the terms “first,” “second,” and “third,” etc. are used merely as referee labels, and are not intended to impose numerical, structural or other requirements on their objects. Forms of term “based on” herein encompass relationships where an element is partially based on as well as relationships where an element is entirely based on. Forms of the term “defined” encompass relationships where an element is partially defined as well as relationships where an element is entirely defined. Further, the limitations of the following claims are not written in means-plus-function format and are not intended to be interpreted based on 35 U.S.C. § 112, sixth paragraph, unless and until such claim limitations expressly use the phrase “means for” followed by a statement of function void of further structure. It is to be understood that not necessarily all such objects or advantages described above may be achieved in accordance with any particular example. Thus, for example, those skilled in the art will recognize that the devices, systems and methods described herein may be embodied or carried out in a manner that achieves or optimizes one advantage or group of advantages as taught herein without necessarily achieving other objects or advantages as may be taught or suggested herein. While the disclosure has been described in detail in connection with only a limited number of examples, it should be readily understood that the disclosure is not limited to such disclosed examples. Rather, this disclosure can be modified to incorporate any number of variations, alterations, substitutions or equivalent arrangements not heretofore described, but which are commensurate with the spirit and scope of the disclosure. Additionally, while various examples have been described, it is to be understood that aspects of the disclosure may include only one example or some of the described examples. Also, while some examples are described as having a certain number of elements, it will be understood that the examples can be practiced with less than or greater than the certain number of elements. It should be appreciated that all combinations of the foregoing concepts and additional concepts discussed in greater detail below (provided such concepts are not mutually inconsistent) are contemplated as being part of the inventive subject matter disclosed herein. In particular, all combinations of claimed subject matter appearing at the end of this disclosure are contemplated as being part of the inventive subject matter disclosed herein. | 123,957 |
11863041 | DETAILED DESCRIPTION OF THE INVENTION This invention is based in part on the disclosed novel strategy that applies a combined electrical and MHD force to rapidly move charged molecules within cleared tissue samples. This invention is particularly useful for (1) removing endogenous heterogenous molecules that obstruct light-based imaging techniques from tissue samples, and (2) introducing exogenous charged molecules (e.g., antibodies, dyes) into tissue. Magnetohydrodynamic forces are used to efficiently remove lipids form hydrogel-infused tissue and reliably produce transparent samples for fluorescence microscopy. MHD forces are generated when a magnetic field and an electric field are applied in perpendicular directions and propel charged molecules (e.g., lipid filled micelles or antibodies) at the intersection of these fields in the third orthogonal direction (FIG.1). MHD forces increase the force pushing the dissolved lipids out of the tissue and simultaneously generate flow in the detergent solution. Thus, MHD-accelerated tissue clearing provides efficient thermal regulation without the need for an external pump. MHD-accelerated tissue clearing provides direct optical access to fluorescent markers inside a large tissue sample and can also be modified to accelerate the penetration of charged molecules into tissue samples. MHD-accelerated labeling rapidly introduces targeted primary antibodies and fluorescently labeled secondary antibodies into volumes of brain tissue up to 0.22 cm3(medio-lateral: 6 mm, ventro-dorsal: 6 mm, antero-posterior: 6 mm) while maintaining antigen specificity. These MHD-accelerated strategies work in both vertebrate (shown for mouse and zebrafish) and invertebrate (shown for the nudibranch molluskBerghia stephanieae) species and allow an intact tissue sample to be rendered transparent, labeled with antibodies, and prepared for imaging in as few as 4 days. The effects of the MHD conjugation on clearing and labeling of intact tissue may result from several complementary mechanisms. There is a marked increase in the flow rate through during both MHD-accelerated clearing and MHD-accelerated labeling. Because MHD forces act directly on charged molecules, buffer, lipids, and antibodies should all move within the sample more rapidly in MHD as compared with passive staining or with only an electric field. Thus, the rapid staining obtained through this protocol can be due to the direct action of MHD forces on antibodies. Alternatively, antibody penetration can be accelerated as a result of the increased speed of buffer flow, within the tissue sample, in the MHD condition. Without intending to be bound by the theories, these they are not mutually exclusive and likely both contribute to the efficacy of MHD-based approaches. Key advantages of the disclosed devices and methods include: (1) the system is easy to build and cost-effective; (2) adaptable for use with a variety of tissues; (3) no need for an exogenous pump because the MHD force induces bulk flow around the sample to dissipate heat produced from the electrodes; (4) a built-in safeguard against tissue damage; (5) efficient clearing of biological tissue rendering it transparent while preserving endogenous proteins, and (6) efficient biomarker labeling (e.g., antibodies and vital stains) to rapidly introduce these labels into large tissue samples. This allows subsequent anatomical investigation with fluorescence microscopy. Proteins, including fluorophores (e.g., GFP) can then be imaged with subcellular resolution in large (>1 cm3) samples. Taken together, these advantages allow MHD-based approaches to remove and introduce molecules into tissue samples quickly. The efficacy of the technique has been demonstrated by clearing and labeling zebrafish, mouse, and sea slug tissue and for multiple antibodies. Taken with the linear rate of antibody penetration observed with longer durations of active labeling, the invention can be adapted for labeling larger samples with minor optimization of the strength and orientation of electrical and magnetic fields. In one aspect, the invention generally relates to a magnetohydrodynamic system or device. The device comprises: at least one channel having space therein for holding an electrically conducting solution and a sample emerged therein; at least two electrodes creating an electric field within the channel; and a magnetic system creating a magnetic field within the channel and perpendicular to the electric field. The electric field and the magnetic field jointly induce charged particles in the sample and/or the electrically conducting solution to flow in a direction perpendicular to both the electric field and the magnetic field. In certain embodiments, the device has one channel. In certain embodiments, the device has two channels. In certain embodiments of the magnetohydrodynamic system or device, the at least one channel has a longitudinal dimension. In certain embodiments, the at least two electrodes are configured such that the electric field is perpendicular to the longitudinal dimension. In certain embodiments, the magnetic system is configured such that the magnetic field is perpendicular to the longitudinal dimension. In certain embodiments, both the electric field and the magnetic field are perpendicular to the longitudinal dimension. In certain embodiments, the at least one channel is cylindrically shaped. In certain embodiments, the at least one channel has a space in the range from about 1 mL to about 100 mL (e.g., from about 1 mL to about 75 mL, from about 1 mL to about 50 mL, from about 1 mL to about 25 mL, from about 1 mL to about 10 mL, from about 1 mL to about 5 mL, from about 5 mL to about 100 mL, from about 10 mL to about 100 mL, from about 25 mL to about 100 mL, from about 50 mL to about 100 mL) in volume for holding the sample. In certain embodiments, the at least one channel has a space sufficient for holding a sample of up to 10 mL in volume. In certain embodiments, the channel has a space greater than about 2 mL in volume for holding the sample. The channel may be made of any suitable material. In certain embodiments, the channel is made of a non-conductive material selected from nylon or acrylic. In certain embodiments, the electrically conducting solution comprises a polar solvent and an electrolyte. Any suitable polar solvent may be used. Suitable solvents allow for ample current, maintain the buffered pH, and minimize the loss of fluorescence. In certain embodiments, an aqueous solution with a high concentration of ions and detergent with a basic pH is used, achieve pH to 8.5 with NaOH. For example, the aqueous solution may include a detergent or surfactant such as Triton™ X-100, sodium dodecyl sulfate, or Tween-20 and have a pH around 8.5. Any suitable electrolyte may be used. In certain embodiments, the electrolyte is selected from the group consisting of salts (e.g., NaCl), NaOH, LiOH, H3BO3providing suitable pH buffering. In certain embodiments, the electrolyte is selected from the group consisting of NaOH, LiOH, and H3BO3. In certain embodiments, the electrically conducting solution comprises a borate-based buffer. In certain embodiments, the borate-based buffer comprises sodium borate. In certain embodiments, the sample is a tissue sample. Any suitable samples may be treated or analyzed using the devices of the invention. In certain embodiments, the sample is a tissue sample (e.g., a soft tissue sample). In certain embodiments, the tissue sample is a diseased (e.g., a cancerous tissue sample). In certain embodiments, the tissue sample is an intact tissue sample. In certain embodiments, the tissue sample is selected from soft biological samples (e.g., brain, lungs, muscle tissues). Any charged particles may be induced to move directionally. In certain embodiments, the electric field and the magnetic field jointly are capable of clearing endogenous charged particles from the sample. In certain embodiments, the electric field and the magnetic field jointly are capable of pushing exogenous charged particles into the sample. In certain embodiments, the cleared endogenous charged particles are selected from lipids, nucleic acids (e.g., DNAs and RNAs), and proteins. In certain embodiments, the exogenous charged particles are selected from lipid-filled micelles, nucleic acids (e.g., DNAs and RNAs), proteins, antibodies and chemical dyes. In certain embodiments, the charged particles are antibodies. In certain embodiments, the antibodies are fluorescently labeled (e.g. fluorescent conjugated IgG). In certain embodiments, the charged particles are nucleic acids. In certain embodiments, the charged particles are dyes (e.g. methylene blue). In another aspect, the invention generally relates to a method for moving charged particles in a biological sample. The method comprises: providing a sample submerged in a channel having an electrically conducting solution; applying an electric field within the channel; simultaneously applying a magnetic field within the channel and perpendicular to the electric field; and inducing the charged particles in the sample to flow in a direction perpendicular to both the electric field and the magnetic field. In certain embodiments of the method, the electrically conducting solution comprises a polar solvent and an electrolyte. Any suitable polar solvent may be used in the method. In certain embodiments, the polar solvent is water. Any suitable electrolyte may be used in the method. In certain embodiments, the electrolyte is selected from the group consisting of salts (e.g., NaCl), NaOH, LiOH, H3BO3providing suitable pH buffering. In certain embodiments, the electrolyte is selected from the group consisting of NaOH, LiOH, and H3BO3. In certain embodiments, the electrically conducting solution comprises a borate-based buffer. In certain embodiments, the borate-based buffer comprises sodium borate. In certain embodiments of the method, the sample is a tissue sample. Any suitable samples may be treated or analyzed using the devices of the invention. In certain embodiments, the sample is a tissue sample (e.g., a soft tissue sample). In certain embodiments, the tissue sample is a diseased (e.g., a cancerous tissue sample). In certain embodiments, the tissue sample is an intact tissue sample. In certain embodiments, the tissue sample is selected from soft biological samples (e.g., brain, lungs, and muscle tissues). Any charged particles may be induced to move directionally. In certain embodiments, the electric field and the magnetic field jointly are capable of clearing endogenous charged particles from the sample. In certain embodiments, the electric field and the magnetic field jointly are capable of pushing exogenous charged particles into the sample. In certain embodiments, the cleared endogenous charged particles are selected from lipids, nucleic acids (e.g., DNAs and RNAs), and proteins. In certain embodiments, the exogenous charged particles are selected from lipid-filled micelles, nucleic acids (e.g., DNAs and RNAs), proteins, antibodies and chemical dyes. In certain embodiments, the charged particles are antibodies. In certain embodiments, the charged particles are nanobodies. In certain embodiments, the antibodies or nanobodies are fluorescently labeled (e.g. fluorescent conjugated IgG). In certain embodiments, the charged particles are nucleic acids. In certain embodiments, the charged particles are dyes (e.g. methylene blue). In certain embodiments, the sample has a volume in the range from about 0.1 mL to about 10 mL (e.g., about 0.1 mL to about 5 mL, about 0.1 mL to about 2 mL, about 0.1 mL to about 1 mL, about 0.1 mL to about 0.5 mL, about 0.5 mL to about 10 mL, about 1 mL to about 10 mL, about 2 mL to about 10 mL, about 2 mL to about 5 mL). As discussed herein, the MHD-based strategy employs an electrical force and a conjugated magnetic field to move charged molecules into and out of the tissue. The two complementary forces increase the total force produced on the charged particles, which allows more powerful as well as controlled removal of lipids and other particles from, as well as introduction of biological particles (such as labeled antibodies) into, large tissue samples. The application of an MHD force induces bulk flow around the sample to dissipate heat produced from the electrodes, which eliminates the need for an exogenous pump. Coupling of buffer flow directly to the clearing force creates a built-in kill-switch, where thermal damage associated, for example, with pump malfunction is impossible as the flow necessary to clear tissue comes directly from the MHD force itself. Thus, the risk of damage to, often precious, tissue samples is dramatically reduced with MHD-assisted clearing. Experiments demonstrate an important distinction between tissue opacity and the ability to resolve fine structures with fluorescence microscopy. While tissue samples quickly became transparent, the ability to resolve cells microscopically, was improved with previous exposure to the clearing solution. Together, these strategies facilitate structural investigation of large tissue samples with fluorescence microscopy. Both the MHD clearing device and the MHD antibody staining device described here are cost-effective. Moreover, the simple modular design of these devices makes modification easy, allowing the MHD strategy to be quickly applied to different types of tissues. This MHD-based labeling strategy allowed rapid introduction of exogenous labels into large tissue samples using a reasonable volume (e.g., 4.5 mL) and concentration of antibody in solution (e.g., 1:500 primary; 1:200 secondary). Using the MEM labeling protocol, a single tissue sample can be maximized to yield more useful data than a comparable transgenic sample. By employing multiple stains, a researcher can maximize the data collected from each tissue, allowing more precious samples to be annotated for multiple molecular features. This allows direct comparison of molecular profiles of individual samples at the cellular level. The MHD force may be augmented to exert as much strength as possible on the tissue. Unnecessary projections of the electric field in the clearing device have been minimized by blocking all, but the necessary electric field path with non-conductive materials (e.g., nylon, acrylic). This ensures that the full electric field strength is projected through the tissue and eliminates other possible routes between the electrodes that could siphon off field strength. Additional forces can be produced, as shown in the following equation. Magnetohydrodynamic Force: F=qE+qv×BF=Force on charged particleq=Charge on the particleE=Strength of electric fieldv=Diffusive velocity of the particleB=Strength of magnetic field The magnetic field may be isolated with non-ferromagnetic materials. By isolating the magnetic field, the magnetic field strength projected through the tissue can be maximized. This allows increased MHD force across the tissue and improve labeling efficacy. The techniques outlined here represent an advance for visualizing large, intact tissue samples. The approach outlined here provides a reliable and durable strategy to efficiently clear, and subsequently label, intact tissue. The directionality of the label may also be addressed. As demonstrated in staining experiments there is a clear directionality of label through a sample. While this was helpful in determining the ability of fluorophores to penetrate the sample, in tissue, this could cause uneven labeling of a sample. Switching electrode polarity and/or magnetic polarity can be used to adjust and improve both the speed of penetration and consistency of labeling. The MHD force is produced inside of the tissue, allowing consistent application of force regardless of tissue depth. Electrical and magnetic forces penetrate through the tissue as they are both waveform forces. This means that the force produced on the electromobile species in the tissue is consistent regardless of tissue depth. Other methods produce a force only on the surface of the tissue, causing the force to degrade as the species move into the tissue. The disclosed strategy minimizes damage caused by prolonged exposure to a high voltage electric field. Prolonged exposure to high-voltage electric fields can cause tissue damage. The conjugation of the electric field with a magnetic field increases the force produced by the tissue proportional to the strength of the magnetic field. This means that a lower electrical current can be used to achieve the same force, minimizing tissue damage. The disclosed devices are designed to be sturdy and durable. With no moving parts and full encapsulation of all vulnerable components, the device does not break down often. Other devices require substantial maintenance to function, and often break down. The disclosed devices can also be fabricated at lower costs than the currently available alternative devices. Alternative clearing and antibody labeling techniques cost tens of thousands of dollars. Both the clearing and antibody staining devices cost less than $500 to construct. Unlike many common clearing and labeling alternatives, the disclosed devices do not require large work area or complex setups. Both devices described are only several cubic inches in volume. This both minimizes the volume of label required and makes temperature control simple. Yet another advantage of the devices is the fast action afforded by the unique system designs allowing both rapid clearing and staining. For example, the disclosed device allows for complete optical clarity of hydrogel infused tissue in 15 hours without passive staining and 12 hours with the addition of a 2-day passive staining step. Common clearing methods take, on average, 3 days to achieve complete optical clarity. Complete labeling of a large, intact tissue sample is also achieved in as few as 12 hours. Furthermore, antibody in the labeling device is cycled from one end of the tissue to the other through the attached tubing. This maximizes the binding potential of the antibody solution by ensuring that any antibodies that remained unbound through a single pass across the tissue will have additional opportunities to bind. EXAMPLES Methods: Tissue Fixation and Hydrogel Polymerization Mice were euthanized and perfused with 0.01 M phosphate buffered saline (PBS) followed by 4% paraformaldehyde (PFA) in 0.01 M PBS. Tissue was then post-fixed in 4% PFA at 4° C. overnight. Next, the tissue was placed in a hydrogel solution (4% acrylamide, 4% PFA, 0.05% bis acrylamide, and 0.25% VA-044 initiator suspended in 0.01 M PBS) at 4° C. overnight. Oxygen was flushed out of hydrogel-infused tissues with at least 1 L nitrogen gas and then the samples were polymerized by incubating them at 37° C. overnight. Excess hydrogel was removed from the surface and tissue samples were transferred to PBS to flush hydrogel monomers. Adult zebrafish were euthanized in 0.2 mg/mL tricaine mesylate (MS-222), decapitated, and the heads placed in 4% paraformaldehyde overnight. Heads were then placed in PBS and brains were carefully dissected, incubated in hydrogel at 4° C. overnight, and processed as above. Adult nudibranchs (Berghia stephanieae) were anaesthetized in cold 4.5% magnesium chloride in artificial sea water for 20 minutes, pinned to a sylgard-lined dish, and fixed in 4% paraformaldehyde in sea water overnight at 4° C. Whole animals were washed with PBS and then incubated in hydrogel at 4° C. overnight and processed as above. Active Tissue Delipidation (Clearing) Tissue samples were incubated in SDS-clearing solution (10 mM sodium dodecyl sulfate in 0.1 M borate buffer, pH 8.5) for 2 days at 37° C. unless otherwise noted (FIG.1A). Samples were then transferred to the MHD-accelerated clearing chamber, consisting of two interlocking cell-strainers (ThermoFisher; catalog #: 87791). The tissue chamber was placed into the central chamber of the MHD-accelerated clearing device (FIG.1B,1C). This holds the tissue at the intersection of the electrical and magnetic fields. The clearing chamber was submerged in a large (5 L) bath of clearing solution at 37° C. and 30 VDC (0.25 Amps) was applied across the tissue for several hours (typically 16 hours for mouse brain tissue and 2 hours for intact zebrafish brains;FIG.1D). Antibody Labeling Delipidated tissue was placed inside of a 2-inch length of 0.25-inch diameter dialysis tubing (SpectraPor® 1; Spectrum). After equilibration the dialysis tubing was positioned in the central channel of the MHD chamber so that the tissue was at the intersection of electrical and magnetic fields (FIG.4). Magnets (Applied magnets; NB057-6-N52) were placed on the top and bottom of the MHD labeling device creating a central chamberFIG.4B). The ends of the dialysis tubing were connected to 9.5 mm diameter vinyl tubing (ThermoFisher: S504591) using 0.25-inch Leur lock barbs (Cole-Parmer; UX-45501-20) to create a torus-shaped chamber allowing the antibody solution to circulate continuously and provide an even and continuous source of antibody to the tissue sample (FIG.4C). Antibody solution (4.5 mL; 0.1 M borate buffer titrated to pH 9.5 with 0.1 M LiOH, 1% heparin, 0.1% Triton™ X-100; 1:500 primary antibody) was transferred into the dialysis tubing using a 5 mL syringe. The labeling chamber was submerged in a 1 L tub containing 0.1M Borate Buffer pH 9.5/0.1% Triton X-100 solution. A 5 mL syringe filled with the buffer solution was attached to the circulation line to maintain constant pressure inside of the dialysis tube. 60 volts DC (˜0.2 Amps) was applied across the electrodes for 15 minutes, followed by 3 minutes of inactivity repeatedly for 12 hours to drive antibodies into the tissue sample. The system was held at 37° C. (FIG.4D) and protected from ambient light to minimize bleaching of fluorophores throughout the procedure. Following each round of MHD-accelerated labeling, the antibody solution was replaced with a wash solution (0.1 M borate buffer titrated to pH 9.5 with 0.1 M LiOH, 1% heparin, 0.1% Triton™ X-100) and the tissue was exposed to 6-hours of “active washing” using the same voltage settings. Labeled tissue was then washed in 0.01 M PBS for at least 12 hours. Refractive Index Matching and Light Sheet Microscopy The tissue was transferred to Optiview™ refractive index matching solution and incubated at 37° C. for at least 12 hours to achieve optical clarity through RI matching (FIG.1). Samples were imaged at 5× or 20× magnification with a lightsheet microscope adapted for a 1.45 RI imaging solution (Zeiss Z1™). Measures of Clearing Efficacy 36 mouse brains were embedded in hydrogel, cleared using the MHD-accelerated clearing protocol, and assessed for transparency. The tissue was divided into two groups: one that was pretreated by passively dilapidating in SDS clearing solution for two days at 37° C. (n=18), and a second that was placed in a 0.1 M borate buffered solution at 37° C. (n=18) for the same time as the pretreatment. Tissue samples from each condition (n=3) were then actively delipidated using the MHD-accelerated clearing system for 2, 6, 8, 12, or 15 hours (FIG.2). After washing in 0.01M PBS, the tissue was equilibrated in Optiview™ for 48 hours at 37° C. Transparency was determined by the percentage of light transmitted through the tissue and the maximum depth from the external surface at which the morphology of neural processes (including primary dendrites and axons) could be resolved. Light transmission was measured using a wide-spectrum light-source and calibrated photodiode. Data from each condition was fit with a saturating exponential curve in MATLAB. Genetically encoded fluorophores: Tissue was collected from mice that transgenetically expresses Cre-recombinase under the control of the aromatase promoter. Animals were then injected with a modified G-deleted rabies virus expressing GFP in the medial amygdala to identify synaptically coupled neurons. Optimization of MHD-Accelerated Immunohistochemistry 0.22 cm3cubes of hydrogel were incubated with FITC-conjugated antibodies (1:200; Jackson Immunoresearch) with MHD assistance, with an equivalent electric field, or passively for 1 hour at 37° C. (n=1). The distance of antibody penetration from the nearest surface was measured using lightsheet microscopy as described above. Results: Tissue Clearing/Delipidation The MHD-accelerated clearing technique rendered whole adult mouse brains transparent in as few as 15 hours. Lipids were actively removed from tissue samples using MHD-accelerated clearing and incubated these samples in a RI matching solution (FIG.2A). The MHD-accelerated clearing protocol removed lipids from an adult mouse brain without passive incubation in clearing solution in 15 hours (FIG.2A,2B), however, pretreatment with passive incubation in SDS-containing clearing solution (37° C.) for two days prior to MHD-accelerated clearing improved optical transparency in terms of both light transmission and effective clarity. Pretreatment also reduced the amount of time required to effectively clear tissue using MHD-accelerated clearing (FIG.2A) and typically produced better samples for imaging (FIG.2B). Because passive delipidation was gentler on tissue and reduced the time of MHD clearing required to achieve optimal transparency, all subsequent tissue samples were passively cleared prior to active clearing. MHD-accelerated clearing reliably produced tissue samples with excellent optical transparency while also preserving genetically encoded fluorescent proteins (FIG.3). An intact adult mouse brain was prepared using MHD-accelerated clearing and imaged on a Zeiss Z1™ lightsheet microscope. Sparse GFP cells are easily identified even in even in the center of the brain (FIG.3B,3C,3D). Whole brain images resolve tissue architecture throughout the brain with subcellular resolution (FIG.3C,3D). Higher magnification shows that fine processes, such as dendrites and axons, can be easily identified and analyzed (FIG.3D). MHD-Accelerated Histology Samples were incubated in an antibody solution inside dialysis tubing at the center of intersecting electrical and magnetic fields where the MHD force was strongest (FIG.4). Confining the tissue sample inside dialysis tubing reduced the volume of antibody required for labeling and protected the tissue sample and antibody solution from direct exposure to the electrodes. Vinyl tubing allowed continuous circulation of the antibody during the MHD-accelerated labeling process (FIG.4C). Initial tests with a 6 mm3hydrogel block incubated with antibody (mouse anti-rat; FITC-conjugated; 1:200 dilution; Jackson Immunoresearch; 60 volts; 1 hour) revealed two perpendicular penetration fronts with roughly 50% improvement for antibody penetration under MHD over that observed in samples exposed to electrical force only (153.8% of electrical only). The passive control showed less than 1% penetration, as compared with electrical only, over the same time (0.92% of electrical only). MHD-accelerated labeling improves antibody penetration and allowed labeling of intact tissue samples. An intact adult nudibranch (Berghia stephanieae) (medio-lateral: 1.3 mm, dorso-ventral: 1.5 mm, anterio-posterior: 2 cm) that had been delipidated using the MHD-accelerated clearing device was incubated with an anti-serotonin (5-HT; Immunostar; 1:500) antibody followed by a fluorescent secondary (488 nm conjugated; ThermoFisher; 1:200). Passive antibody labeling failed to effectively reach the brain (FIG.5A); however, MHD-accelerated antibody labeling drove antibodies throughout the sample and revealed 5-HT expressing cell bodies and projections (FIG.5B). Intact zebrafish brains (medio-lateral: 3 mm; dorso-ventral: 3 mm; anterio-posterior: 6 mm) were passively delipidated in SDS for 7 days and then incubated with anti-acetylated tubulin antibodies (Immunostar; 1:500) to identify neural fibers (FIG.5C,5D). Control tissue samples (no MHD force applied) showed only nominal penetration along the outer edge of the tissue and weak binding in the upper layer of the optic tectum (FIG.5C). In contrast, MHD-accelerated labeling showed robust labeling of neural tracts throughout the brain (FIG.5D). To test MHD-accelerated labeling in mammalian tissue, an anti-oxytocin (OT) antibody was applied to a cube of mouse brain (medio-lateral: 6 mm, ventro-dorsal: 6 mm, antero-posterior: 6 mm) containing the PVN (1:500 primary; 1:200 secondary). Antibodies did not effectively penetrate the control tissue sample without MHD (FIG.5E). However, OT-expressing cells were clearly visible in the PVN of tissue prepared with the MHD-accelerated labeling device (FIG.5F). OT-expressing neurons sent processes towards the third ventricle and, consistent with OT neuron morphology, these processes were visible several mm into the tissue from the nearest edge. Moreover, it was possible to visualize subcellular components such as axonal varicosities of MHD-labeled OT neurons (FIG.5F). To confirm the specificity of antibody binding is maintained in MHD-accelerated labeling, an anti-vasopressin antibody in mice was used that expressed tdTomato in vasopressin-expressing neurons (FIG.5G-5I). Tissue was generated by crossing the Ai9 Rosa26:LSL:tdTomato reporter line and a line where Cre recombinase is expressed under the control of the arginine vasopressin (AVP) promoter. This produced tissue where the fluorescent reporter tdTomato was expressed under the control of the AVP promoter. By labeling this tissue with MHD accelerated labeling, specific colabeling of the genetically encoded fluorophores and the anti-AVP antibody was observed (FIG.5G-5I). To compare the effects of MHD force versus an electric only field on solution motility, we measured the velocity of dyed sodium alginate spheres for voltages ranging from 0 to 30 volts (FIG.6). The MHD produced a much more movement in the solution and was statistically significant from the electric field only condition for all conditions except 0 volts. Applicant's disclosure is described herein in preferred embodiments with reference to the Figures, in which like numbers represent the same or similar elements. Reference throughout this specification to “one embodiment,” “an embodiment,” or similar language means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment of the present invention. Thus, appearances of the phrases “in one embodiment,” “in an embodiment,” and similar language throughout this specification may, but do not necessarily, all refer to the same embodiment. The described features, structures, or characteristics of Applicant's disclosure may be combined in any suitable manner in one or more embodiments. In the description, herein, numerous specific details are recited to provide a thorough understanding of embodiments of the invention. One skilled in the relevant art will recognize, however, that Applicant's composition and/or method may be practiced without one or more of the specific details, or with other methods, components, materials, and so forth. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the disclosure. In this specification and the appended claims, the singular forms “a,” “an,” and “the” include plural reference, unless the context clearly dictates otherwise. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art. Although any methods and materials similar or equivalent to those described herein can also be used in the practice or testing of the present disclosure, the preferred methods and materials are now described. Methods recited herein may be carried out in any order that is logically possible, in addition to a particular order disclosed. INCORPORATION BY REFERENCE References and citations to other documents, such as patents, patent applications, patent publications, journals, books, papers, web contents, have been made in this disclosure. All such documents are hereby incorporated herein by reference in their entirety for all purposes. Any material, or portion thereof, that is said to be incorporated by reference herein, but which conflicts with existing definitions, statements, or other disclosure material explicitly set forth herein is only incorporated to the extent that no conflict arises between that incorporated material and the present disclosure material. In the event of a conflict, the conflict is to be resolved in favor of the present disclosure as the preferred disclosure. EQUIVALENTS The representative examples are intended to help illustrate the invention, and are not intended to, nor should they be construed to, limit the scope of the invention. Indeed, various modifications of the invention and many further embodiments thereof, in addition to those shown and described herein, will become apparent to those skilled in the art from the full contents of this document, including the examples and the references to the scientific and patent literature included herein. The examples contain important additional information, exemplification and guidance that can be adapted to the practice of this invention in its various embodiments and equivalents thereof. | 33,629 |
11863042 | DETAILED DESCRIPTION OF EMBODIMENTS The invention will be now described herein with reference to illustrative embodiments. Those skilled in the art will recognize that many alternative embodiments can be accomplished using the teachings of the present invention and that the invention is not limited to the embodiments illustrated for explanatory purposed. Hereinafter, an embodiment of the present invention will be described with reference toFIGS.1to8B. In all the drawings, the similar components are denoted by the same reference numerals, and the description thereof will not be repeated. As illustrated in any one ofFIGS.1to6, a power transmission device100according to the present embodiment includes an annular transmission unit10having an annular first magnetic core31and a transmission coil61, and an annular reception unit20having an annular second magnetic core33and a reception coil62. The transmission unit10and the reception unit20are arranged to face each other. The power transmission device100transmits power from the transmission unit10to the reception unit20. As illustrated inFIGS.4and6, the transmission unit10and the reception unit20are relatively rotatable about a rotation axis95passing through a cavity11inside the transmission unit10and a cavity21inside the reception unit20, the first magnetic core31has a structure divided into a plurality of first split cores32in a circumferential direction, and the second magnetic core33has a structure divided into a plurality of second split cores34in the circumferential direction. According to the present embodiment, the first magnetic core31has the structure divided into the plurality of first split cores32, and the second magnetic core33has the structure divided into the plurality of second split cores34. That is, desired power transmission efficiency can be achieved by an aggregate of the first split cores32having individual dimensions smaller than those of the first magnetic core31and an aggregate of the second split cores34having individual dimensions smaller than those of the second magnetic core33. Therefore, as compared with a case where the entire first magnetic core31and the entire second magnetic core33are integrally molded, the structural strength (fracture resistance) and durability of the first magnetic core31and the second magnetic core33can be sufficiently secured, and the ease of manufacturing the first magnetic core31and the second magnetic core33is also improved. The transmission unit10and the reception unit20face each other with a reference plane130(seeFIGS.1and6), which is a virtual plane, interposed therebetween. More specifically, the reception unit20is disposed in a non-contact manner with the transmission unit10and is disposed in proximity to the transmission unit10. The rotation axis95is, for example, orthogonal to the reference plane130, and is a virtual axis passing through the center of the annular transmission unit10and the center of the annular reception unit20. In the present embodiment, the reception unit20rotates relative to the transmission unit10about the rotation axis95. Note that the relative rotation of the reception unit20with respect to the transmission unit10is freely set in each of the counterclockwise direction and the clockwise direction in the present embodiment. Hereinafter, in order to simplify the description, the positional relationship between the components will be described on the assumption that the rotation axis95extends in the vertical direction (top-bottom direction). Therefore, in the following description, it is assumed that a direction orthogonal to the rotation axis95is a horizontal direction. In the vertical direction, a side on which the transmission unit10is disposed is referred to as a lower side (bottom side), and a side on which the reception unit20is disposed is referred to as an upper side (top side). In a plane orthogonal to the rotation axis95, a direction passing through the rotation axis95is referred to as a radial direction. Further, in the radial direction, a side close to the rotation axis95is referred to as a radial inner side, and a side away from the rotation axis95is referred to as a radial outer side. The circumferential direction is a direction around the rotation axis95. The positional relationship of the respective units of the power transmission device100has been described in a state where the respective units of the power transmission device100are assembled to each other to manufacture the power transmission device100unless otherwise specified. However, the direction of the rotation axis95when the power transmission device100is used is not limited to the vertical direction. Here, as an example, the power transmission device100is used by being attached to a steering wheel (steering) of a vehicle such as an automobile and a portion around the steering wheel, and supplies power to various loads (not shown in the drawings) mounted on the steering wheel. That is, the reception coil62of the reception unit20is electrically connected to the loads, and supplies the power transmitted from the transmission coil61of the transmission unit10to the loads. In the present embodiment, the transmission unit10and the reception unit20are disposed around a steering shaft110which is made of metal. The steering shaft110includes a shaft115formed in a cylindrical shape with the vertical direction as an axial direction, and a connecting member120connected to an upper end portion of the shaft115. Here, to say that the steering shaft110is made of metal means that at least one of the shaft115and the connecting member120is made of metal, and preferably both the shaft115and the connecting member120are made of metal. The steering shaft110is inserted into the cavity11of the transmission unit10and the cavity21of the reception unit20, for example, and the outer peripheral surface110aof the steering shaft110faces the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20. The steering shaft110is disposed coaxially with the rotation axis95. The steering wheel is connected to, for example, a distal end portion (upper end portion) of the steering shaft110. Further, the steering shaft110is held by a base (not shown in the drawings) in a state of being rotatable about the axis of the steering shaft110. In the configuration of the power transmission device100, the transmission unit10is provided on the base, and the reception unit20is provided on the steering wheel. Therefore, when the driver of the vehicle rotates the steering wheel, the loads and the reception unit20rotate with the steering wheel, but the transmission unit10provided on the base does not rotate. The objects to which the power transmission device100is attached are not limited to the steering wheel of the vehicle and the portion around the steering wheel, and the power transmission device100may be attached to other devices. Examples of the other devices include an amusement device such as a game machine and a device having a steering wheel, such as a simulator. However, each of the other devices may be a device that does not have a steering wheel but has two parts that are relatively rotatable, or may be a device in which two parts are relatively rotatably connected without using the steering shaft110. In addition, the power transmission device100may be provided as, for example, a steering component having a configuration in which the reception unit20is incorporated in a steering wheel in advance. As illustrated inFIGS.1and6, the transmission unit10and the reception unit20are formed vertically symmetrically with respect to the reference plane130. More specifically, as illustrated inFIG.6, the reception unit20is disposed above the transmission unit10. In the following, first, of the transmission unit10and the reception unit20, the reception unit20will be described in detail. As illustrated inFIGS.3,4, and6, in the present embodiment, the second magnetic core33is formed in, for example, an annular shape in plan view. More specifically, the second magnetic core33includes a plate-shaped portion37having an annular shape in plan view centered on the rotation axis95, an inner peripheral wall portion35protruding downward from an inner peripheral edge portion of the plate-shaped portion37, and an outer peripheral wall portion36protruding downward from an outer peripheral edge portion of the plate-shaped portion37. Each of the upper surface and the lower surface of the plate-like portion37is formed flat, for example, and is arranged horizontally. Each of the inner peripheral wall portion35and the outer peripheral wall portion36is formed in, for example, a cylindrical shape with the vertical direction as the axial direction, and protrudes from the lower surface of the plate-shaped portion37. In the present embodiment, the plate-shaped portion37, the inner peripheral wall portion35, and the outer peripheral wall portion36are arranged concentrically around the rotation axis95in plan view. The outer diameter of the plate-shaped portion37is set to be equal to the outer diameter of the outer peripheral wall portion36, and the inner diameter of the plate-shaped portion37is set to be equal to the inner diameter of the inner peripheral wall portion35. The height position of the lower end surface of the inner peripheral wall portion35and the height position of the lower end surface of the outer peripheral wall portion36are set to the same height position. In the present embodiment, the inner peripheral wall portion35, the outer peripheral wall portion36, and the plate-shaped portion37constitute a groove portion38having an annular shape in plan view, and the groove portion38is opened downward, for example. In addition, one surface (hereinafter, it may be referred to as a first surface33a) of the second magnetic core33is constituted by the lower end surface of the inner peripheral wall portion35and the lower end surface of the outer peripheral wall portion36, and a surface (hereinafter, it may be referred to as a second surface33b) of the second magnetic core33opposite to the one surface33ais constituted by the upper surface of the plate-shaped portion37. As described above, the second magnetic core33has the structure divided into the plurality of second split cores34. As illustrated inFIGS.3and4, each of the second split cores34is formed in, for example, a shape obtained by dividing the second magnetic core33into six equal parts in the circumferential direction. That is, the planar shape of each of the second split cores34is formed in a fan shape having a central angle of 60 degrees, and the second magnetic core33is constituted by an aggregate of the six second split cores34. However, the number of second split cores34included in the second magnetic core33is not limited to the above-described example, and may be at least two or more. Each of the second split cores34includes, for example, a plate-shaped portion34aformed in a shape obtained by dividing an annular ring centered on the rotation axis95into six equal parts, that is, a fan shape having a central angle of 60 degrees in plan view, an inner peripheral wall portion34bprotruding downward from an inner peripheral edge of the plate-shaped portion34a, and an outer peripheral wall portion34cprotruding downward from an outer peripheral edge of the plate-shaped portion34a. An aggregate of the plate-shaped portions34aconstitutes the plate-shaped portion37of the second magnetic core33. An aggregate of the inner peripheral wall portions34bconstitutes the inner peripheral wall portion35of the second magnetic core33, and an aggregate of the outer peripheral wall portions34cconstitutes the outer peripheral wall portion36of the second magnetic core33. Therefore, a combined portion of an aggregate of the lower end surfaces of the inner peripheral wall portions34band an aggregate of the lower end surfaces of the outer peripheral wall portions34cconstitute the first surface31aof the second magnetic core33. An aggregate of the upper surfaces of the plate-shaped portions34aconstitutes the second surface33bof the second magnetic core33. Here, a gap is preferably formed between the second split cores34in the circumferential direction, but the gap may not be formed. In addition, in the present embodiment, a pair of notch-shaped portions39is formed in the outer peripheral wall portion34cof each of the second split cores34, and a second engagement protrusion56described later is engaged with each of the notch-shaped portions39. Each of the notch-shaped portions39is formed, for example, at one portion in the circumferential direction of the outer peripheral wall portion34c. Each of the notch-shaped portions39is formed, for example, from the upper end to the lower end of the outer peripheral wall portion34c, and penetrates the outer peripheral wall portion34cin the thickness direction (radial direction). Here, in the present embodiment, the reception unit20includes a resin-made second holder member51that holds the second magnetic core33. As illustrated inFIGS.2and6, the second holder member51includes, for example, second upright wall portions57disposed along the outer peripheral surface33cof the second magnetic core33, and the second upright wall portions57are disposed at a plurality of locations in the circumferential direction. More specifically, the second holder member51has, for example, a second annular portion54formed in a substantially annular shape centered on the rotation axis95in plan view. Each of the upper surface and the lower surface of the second annular portion54is, for example, formed flat and arranged horizontally. Each of the second upright wall portions57stands upward from, for example, an outer peripheral edge portion of the upper surface of the second annular portion54. For example, the second upright wall portions57are formed in the same size and shape and are intermittently disposed in the circumferential direction. Each of the second upright wall portions57is formed in, for example, an arc shape centered on the rotation axis95in plan view. The outer peripheral surface of each of the second upright wall portions57is on the same cylindrical surface as the outer peripheral surface of the second annular portion54. The height position of the upper end of each of the second upright wall portions57excluding the second engagement claw portion58, which will be described later, is set to a height position substantially equivalent to the height position of the second surface33bof the second magnetic core33. Here, in the present embodiment, each of the second upright wall portions57has, for example, a second engagement claw portion58that engages with the second surface33bof the second magnetic core33. More specifically, the common second engagement claw portion58is engaged with two adjacent second split cores34among the plurality of second split cores34of the second magnetic core33. More specifically, each of the second engagement claw portions58is formed, for example, at the upper end portion of each second upright wall portion57. Each of the second engagement claw portions58protrudes inward in the radial direction from the upper end of the portion of each of the second upright wall portions57excluding the second engagement claw portion58, and extends in the circumferential direction. In the radial direction, the tip of each of the second engagement claw portions58is disposed inside the inner peripheral surface of the corresponding second upright wall portion57excluding the second engagement claw portion58. The dimension of each of the second engagement claw portions58in the circumferential direction is set to be smaller than the dimension of each of the second upright wall portions57excluding the second engagement claw portion58in the circumferential direction. The upper surface and the lower surface of each of the second engagement claw portions58are formed substantially flat and arranged horizontally. However, for example, a chamfered portion is formed in the distal end side portion (distal end in the protruding direction) of each of the second engagement claw portions58. The upper portion of each of the chamfered portions has, for example, a C-chamfered shape. As a result, each of the second engagement claw portions58can be smoothly engaged with the second surface33bof the second magnetic core33. The lower portion of each of the chamfered portions has, for example, a rounded chamfered shape. As a result, the contact of the second engagement claw portions58with the second magnetic core33can be made soft. In a state where the second engagement claw portions58are engaged with the second surface33bof the second magnetic core33, the lower surfaces of the second engagement claw portions58are in contact with the upper surface of the plate-shaped portion37of the second magnetic core33. Furthermore, the second holder member51includes, for example, a plurality of second outer peripheral edge portions52arranged along the outer peripheral surface33cof the second magnetic core33. Each of the second outer peripheral edge portions52is arranged between two second upright wall portions57adjacent to each other in the circumferential direction among the second upright wall portions57. More specifically, the second outer peripheral edge portions52and the second upright wall portions57are alternately arranged in the circumferential direction. The second outer peripheral edge portions52are formed to have the same size and shape, for example. Each second outer peripheral edge portion52is formed in, for example, an arc shape centered on the rotation axis95in plan view, and slightly stands upward from the outer peripheral edge portion of the upper surface of the second annular portion54. The outer peripheral surface of each of the second outer peripheral edge portions52is disposed on the same cylindrical surface as the outer peripheral surface of the second annular portion54and the outer peripheral surface of each of the second upright wall portions57. The thickness dimension (dimension in the radial direction) of each second outer peripheral edge portion52is set to a dimension equivalent to the thickness dimension (dimension in the radial direction) of each second upright wall portion57. The height position of the upper end surface of each of the second outer peripheral edge portions52is, for example, lower than the height position of the upper end surface of each of the second upright wall portions57. The second outer peripheral edge portions52and the second upright wall portions57are disposed apart from each other in the circumferential direction, and a slit portion54b(seeFIG.4) is formed in each of gaps between the second outer peripheral edge portions52and the second upright wall portions57in the circumferential direction. The slit portion54bextends, for example, in the radial direction. The slit portion54bvertically penetrates the second annular portion54and is opened outward in the radial direction. Further, the second holder member51has an opening portion57a(seeFIG.3) formed along the inner peripheral surface of each of the second upright wall portions57. The opening portion57aextends, for example, in the circumferential direction and penetrates the second annular portion54in the vertical direction. In the circumferential direction, the length dimension of the opening portion57ais, for example, smaller than the length dimension of each of the second upright wall portions57. According to the above configuration, since the second holder member51can be easily elastically deformed in the radial direction, the second magnetic core33can be suitably accommodated inside the second holder member51. More specifically, when the second magnetic core33is attached to the second holder member51, the second holder member51is easily elastically deformed (expanded in diameter) toward the radial outer side, and after the attachment of the second magnetic core33is completed, the second holder member51is satisfactorily elastically restored (reduced in diameter) toward the radial inner side. The number of second upright wall portions57is 6, for example. On the other hand, the number of second outer peripheral edge portions52is 5, for example. As illustrated inFIGS.3and4, in the present embodiment, the second outer peripheral edge portion52is not disposed at one of gap between the two second upright wall portions57adjacent to each other in the circumferential direction, and instead, a second terminal holding portion72described later is disposed at the gap. Here, in the present embodiment, the second holder member51has, for example, the second engagement protrusions56that engage with the second split cores34, and positional displacements of the second split cores34in the circumferential direction are restricted by the second engagement protrusions56. More specifically, the second engagement protrusions56are formed to have the same size and shape, for example. Each of the second engagement protrusions56protrudes upward from the upper surface of the second annular portion54in the vicinity of the outer peripheral edge portion of the upper surface of the second annular portion54. Each of the second engagement protrusions56is formed in, for example, a substantially rectangular parallelepiped shape whose dimension in the radial direction is longer than the dimension in the circumferential direction. The width dimension of each of the second engagement protrusions56is set to be substantially constant in the circumferential direction. One surface in the radial direction of each of the second engagement protrusions56is connected to the inner peripheral surface of the second outer peripheral edge portion52. As a result, even if the position of the second split core34is displaced in the radial direction, the second engagement protrusions56can be satisfactorily engaged with the notch-shaped portions39of the second split core34. Further, the height position of the upper end of each of the second engagement protrusions56is preferably higher than the height position of the upper end surface of each of the second outer peripheral edge portions52, for example. As a result, the engagement of the second engagement protrusions56with the second magnetic core33is improved. In the present embodiment, as an example, individual second engagement protrusions56are connected to both end portions (both end portions in the circumferential direction) of one second outer peripheral edge portion52. Therefore, the number of second engagement protrusions56included in the second holder member51is 10, for example. Here, as described above, a pair of notch-shaped portions39is formed in each second split core34, and each second engagement protrusion56is engaged with the corresponding notch-shaped portion39. However, in the present embodiment, the number of second engagement protrusions56included in the second holder member51is 10, whereas the number of second split cores34included in the second magnetic core33is 6. Therefore, the second engagement protrusions56are not engaged with the notch-shaped portions39of any one of the six second split cores34, and instead, extended wiring portions66(described in detail later) of the reception coil62are extended to the outside of the second magnetic core33through the notch-shaped portions39. In addition, the second holder member51includes, for example, a second inner peripheral edge portion53disposed along the inner peripheral surface of the second magnetic core33, and a plurality of upright portions53astanding upward from the upper end of the second inner peripheral edge portion53. The second inner peripheral edge portion53slightly stands upward from the inner peripheral edge portion of the upper surface of the second annular portion54, for example, and is formed in a circular shape. For example, the upright portions53aare intermittently disposed in the circumferential direction. Each upright portion53ais formed in, for example, an arc shape centered on the rotation axis95in plan view. Each of the inner peripheral surface of the second inner peripheral edge portion53and the inner peripheral surface of each upright portion53ais disposed, for example, on the same cylindrical surface as the inner peripheral surface of the second annular portion54. The height position of the upper end of each upright portion53ais, for example, substantially equal to or higher than the height position of the second surface33bof the second magnetic core33. Since the upright portions53aare formed, when the steering shaft110is inserted into the cavity11of the transmission unit10, the steering shaft110is guided by the upright portions53aand smoothly inserted into the power transmission device100. Further, the steering shaft110and the second magnetic core33can be prevented from interfering with each other by the upright portions53a. In the present embodiment, the number of upright portions53aof the second holder member51is 6, for example. Further, the second holder member51includes, for example, a second terminal holding portion72that holds a terminal portion80described later. More specifically, the second terminal holding portion72includes, for example, an arc-shaped fourth upright wall portion75standing upward from the upper surface of the outer peripheral edge portion of the second annular portion54, and a flat plate-shaped second plate-shaped portion76protruding outward in the radial direction from the upper edge portion of the fourth upright wall portion75. The fourth upright wall portion75is formed in, for example, an arc shape centered on the rotation axis95in plan view. In addition, notch-shaped portions77(seeFIG.4) are formed between the fourth upright wall portion75and the second upright wall portions57in the circumferential direction. In the present embodiment, the outer peripheral surface of the fourth upright wall portion75is disposed on the same cylindrical surface as the outer peripheral surfaces of the second outer peripheral edge portions52. The thickness dimension (dimension in the radial direction) of the fourth upright wall portion75is set to, for example, a dimension equivalent to the thickness dimension (dimension in the radial direction) of each second outer peripheral edge portion52. The second plate-shaped portion76is formed in, for example, a substantially rectangular shape in plan view. The upper surface and the lower surface of the second plate-shaped portion76are arranged horizontally, for example. The height position of the upper surface of the second plate-shaped portion76is set to be equal to the height position of the upper end of the portion of each of the second upright wall portions57excluding the second engagement claw portion58, for example. The inner peripheral surface of the fourth upright wall portion75is disposed, for example, along the outer peripheral surface of any one of the six second split cores34. More specifically, among the six second split cores34, the second split core34with which the second engagement protrusion56is not engaged is arranged side by side with the fourth upright wall portion75in the radial direction. As described above, the transmission unit10is formed to be vertically symmetric with the reception unit20with respect to the reference plane130. Therefore, similarly to the second magnetic core33, the first magnetic core31of the transmission unit10includes a plate-shaped portion37, an inner peripheral wall portion35, and an outer peripheral wall portion36. One surface (hereinafter, it may be referred to as a first surface31a) of the first magnetic core31is constituted by the upper end surface of the inner peripheral wall portion35of the first magnetic core31and the upper end surface of the outer peripheral wall portion36of the first magnetic core31, and a surface (hereinafter, it may be referred to as a second surface31b) of the first magnetic core31opposite to the one surface (the first surface31a) of the first magnetic core31is constituted by the lower surface of the plate-shaped portion37of the first magnetic core31. Similarly to the second split cores34, each of the first split cores32is formed in, for example, a shape obtained by dividing the first magnetic core31into six equal parts in the circumferential direction, and includes a plate-shaped portion32asimilar to the plate-shaped portion34a, an inner peripheral wall portion32bsimilar to the inner peripheral wall portion34b, and an outer peripheral wall portion32csimilar to the outer peripheral wall portion34c(seeFIG.5). Similarly to the second split cores34, a pair of notch-shaped portions39is formed in the outer peripheral wall portion32cof each of the first split cores32, and each first engagement protrusion46(described later) of a first holder member41(described later) is engaged with the notch-shaped portions39of the corresponding first split cores32. In addition, the first engagement protrusions46are not engaged with the notch-shaped portions39of any one of the six first split cores32, and instead, extended wiring portions66(described in detail later) of the transmission coil61are extended to the outside of the first magnetic core31through the notch-shaped portions39. Further, the transmission unit10includes a first holder member41similar to the second holder member51. Therefore, the first holder member41includes a first annular portion44similar to the second annular portion54, first engagement protrusions46similar to the second engagement protrusions56, first upright wall portions47similar to the second upright wall portions57, first outer peripheral edge portions42similar to the second outer peripheral edge portions52, a first inner peripheral edge portion43similar to the second inner peripheral edge portion53, and upright portions43asimilar to the second upright portions53a. The first holder member41further includes a first terminal holding portion71that holds a terminal portion80. Similarly to the second terminal holding portion72, the first terminal holding portion71has, for example, a third upright wall portion73similar to the fourth upright wall portion75and a first plate-shaped portion74similar to the second plate-shaped portion76. The inner peripheral surface of the third upright wall portion73is disposed, for example, along the outer peripheral surface of any one of the six first split cores32of the first magnetic core31. More specifically, among the six first split cores32, the first split core32with which the above-described first engagement protrusion46is not engaged is arranged side by side with the third upright wall portion73in the radial direction. As described above, the transmission unit10includes the first holder member41made of resin that holds the first magnetic core31, the reception unit20includes the second holder member51made of resin that holds the second magnetic core33, the first holder member41has an annular shape and is arranged along one surface (the first surface31a) of the first magnetic core31in the axial direction, and the second holder member51has an annular shape and is arranged along the one surface (first surface33a) of the second magnetic core33in the axial direction. As a result, since the separation distance between the first magnetic core31and the second magnetic core33can be kept constant, more stable characteristics of the power transmission device100can be achieved. As described above, the first magnetic core31has the structure divided into the plurality of first split cores32, and the second magnetic core33has the structure divided into the plurality of second split cores34. As a result, for example, when a temperature changes in the environment around the power transmission device100, even if thermal stress occurs in the first magnetic core31due to thermal deformation of the first holder member41due to a difference in thermal expansion coefficient between the first magnetic core31and the first holder member41, the thermal stress can be reduced by enlarging or reducing gaps between the first split cores32. Similarly, even if thermal stress occurs in the second magnetic core33due to thermal deformation of the second holder member51, the thermal stress can be reduced by enlarging or reducing gaps between the second split cores34. Therefore, more stable characteristics of the power transmission device100can be achieved. In addition, it is possible to improve the fracture resistance of the first magnetic core31and the second magnetic core33. In addition, the first holder member41includes the first engagement protrusions46that engage with the first split cores32, and the first engagement protrusions46restrict positional displacements of the first split cores32in the circumferential direction. The second holder member51includes, for example, the second engagement protrusions56that engage with the second split cores34, and the second engagement protrusions56restrict positional displacements of the second split cores34in the circumferential direction. In addition, the first holder member41has, for example, the first upright wall portions47arranged along the outer peripheral surface31cof the first magnetic core31. The first upright wall portions47are arranged at a plurality of locations in the circumferential direction. The second holder member51has, for example, the second upright wall portions57arranged along the outer peripheral surface33cof the second magnetic core33, and the second upright wall portions57are arranged at a plurality of locations in the circumferential direction. In addition, each of the first upright wall portions47has a first engagement claw portion48that engages with the surface (second surface31b) opposite to the one surface (first surface31a) of the first magnetic core31, and each of the second upright wall portions57has, for example, a second engagement claw portion58that engages with the surface (second surface33b) opposite to the one surface (first surface33a) of the second magnetic core33. That is, the first holder member41has the first upright wall portions47arranged along the outer peripheral surface31cof the first magnetic core31, and each of the first upright wall portions47has the first engagement claw portion48that engages with the surface (the second surface31b) opposite to the one surface (the first surface31a) of the first magnetic core31. The second holder member51has the second upright wall portions57arranged along the outer peripheral surface33cof the second magnetic core33, and each of the second upright wall portions57has the second engagement claw portion58that engages with the surface (the second surface33b) opposite to the one surface (the first surface33a) of the second magnetic core33. As a result, the first magnetic core31can be favorably held by the first holder member41, and the second magnetic core33can be favorably held by the second holder member51. Further, the common first engagement claw portion48is engaged with two first split cores32adjacent to each other among the plurality of first split cores32of the first magnetic core31, and the common second engagement claw portion58is engaged with two second split cores34adjacent to each other among the plurality of second split cores34of the second magnetic core33. As a result, a state in which each of the first split cores32is held by the first holder member41can be favorably maintained by a smaller number of the first engagement claw portions48, and a state in which each of the second split cores34is held by the second holder member51can be favorably maintained by a smaller number of the second engagement claw portions58. In addition, the height positions (positions in the vertical direction) of the upper surfaces of the second split cores34adjacent to each other can be aligned. Similarly, the height positions (positions in the vertical direction) of the lower surfaces of the first split cores32adjacent to each other can be aligned. In the present embodiment, the inner diameter of each of the first magnetic core31and the second magnetic core33is larger than the outer diameter of each of the first inner peripheral edge portion43and the second inner peripheral edge portion53. The outer diameter of each of the first magnetic core31and the second magnetic core33is smaller than the inner diameter of each of the first outer peripheral edge portions42and the second outer peripheral edge portions52. The height position of the upper end of each of the second inner peripheral edge portion53and the second outer peripheral edge portions52is lower than the height position of the upper surface of the plate-shaped portion37of the second magnetic core33. The height position of the lower end of each of the first inner peripheral edge portion43and the first outer peripheral edge portions42is higher than the height position of the lower surface of the plate-shaped portion37of the first magnetic core31. As illustrated inFIG.6, the second magnetic core33is disposed between the second outer peripheral edge portions52and the second inner peripheral edge portion53. More specifically, the outer peripheral surface33cof the second magnetic core33is disposed along the inner peripheral surfaces of the second outer peripheral edge portions52, and the inner peripheral surface of the second magnetic core33is disposed along the outer peripheral surfaces of the second inner peripheral edge portions53. That is, the second magnetic core33is arranged concentrically with the second outer peripheral edge portions52and the second inner peripheral edge portion53. The first surface33aof the second magnetic core33is in surface contact with the upper surface of the second annular portion54. Similarly, the first magnetic core31is disposed between the first outer peripheral edge portions42and the first inner peripheral edge portion43. More specifically, the outer peripheral surface31cof the first magnetic core31is disposed along the inner peripheral surfaces of the first outer peripheral edge portions42, and the inner peripheral surface of the first magnetic core31is disposed along the outer peripheral surface of the first inner peripheral edge portion43. That is, the first magnetic core31is arranged concentrically with the first outer peripheral edge portions42and the first inner peripheral edge portion43. The first surface31aof the first magnetic core31is in surface contact with the lower surface of the first annular portion44. Furthermore, in the present embodiment, the cavity11inside the transmission unit10is defined by the inner peripheral surface10aof the transmission unit10, and the inner peripheral surface10aof the transmission unit10is constituted by the inner peripheral surface of the first annular portion44, the inner peripheral surfaces of the first inner peripheral edge portions43, and the inner peripheral surfaces of the upright portions43a. Similarly, the cavity21inside the reception unit20is defined by the inner peripheral surface20aof the reception unit20, and the inner peripheral surface20aof the reception unit20is constituted by the inner peripheral surface of the second annular portion54, the inner peripheral surfaces of the second inner peripheral edge portions53, and the inner peripheral surfaces of the upright portions53a. Further, the outer peripheral surface of the transmission unit10is constituted by the outer peripheral surface of the first annular portion44, the outer peripheral surfaces of the first upright wall portions47, and the outer peripheral surfaces of the first outer peripheral edge portions42, and the outer peripheral surface of the reception unit20is constituted by the outer peripheral surface of the second annular portion54, the outer peripheral surfaces of the second upright wall portions57, and the outer peripheral surfaces of the second outer peripheral edge portions52. Here, the thickness dimension (dimension in the radial direction) of each of the first inner peripheral edge portion43and the upright portions43ais smaller than the thickness dimension (dimension in the radial direction) of each of the first outer peripheral edge portions42, for example. Similarly, the thickness dimension (dimension in the radial direction) of each of the second inner peripheral edge portion53and the upright portions53ais smaller than the thickness dimension (dimension in the radial direction) of the second outer peripheral edge portion52, for example. As a result, it is possible to sufficiently secure the diameters of the cavities11and21without changing each of the outer diameter of the transmission unit10and the outer diameter of the reception unit20. In the present embodiment, each of the transmission coil61and the reception coil62is made of, for example, an insulation-coated metal wire. Each of the transmission coil61and the reception coil62includes a winding portion65formed by winding the wire around the inner peripheral wall portion35, and a pair of extended wiring portions66constituted by both end portions of the wire. The winding portion65of the reception coil62is accommodated inside the groove portion38of the second magnetic core33, and the opening of the groove portion38is closed by the second annular portion54. Similarly, the winding portion65of the transmission coil61is accommodated inside the groove portion38of the first magnetic core31, and the opening of the groove portion38is closed by the first annular portion44. Each extended wiring portion66of the reception coil62is extended to the outside of the outer peripheral wall portion36through, for example, the notch-shaped portion39of the second split core34disposed to face the above-described second terminal holding portion72. Similarly, each extended wiring portion66of the transmission coil61is extended to the outside of the outer peripheral wall portion36through, for example, the notch-shaped portion39of the first split core32disposed to face the above-described first terminal holding portion71. Here, the reception coil62and the second magnetic core33may be fixed to each other by, for example, a double-sided adhesive tape (not shown in the drawings). The double-sided fixing tape is formed in, for example, an annular shape substantially the same as the plate-shaped portion37of the second magnetic core33in plan view. It is also preferable that each of the second split cores34is disposed in a circumferential shape along the fixing tape. With this configuration, it is possible to suppress relative displacement between the second split cores34. Similarly, the transmission coil61and the first magnetic core31may be fixed to each other by, for example, a double-sided adhesive tape (not shown in the drawings). In the present embodiment, as illustrated inFIGS.1,2,3, and4, the power transmission device100includes, for example, a first terminal portion81, a second terminal portion82, a third terminal portion83, and a fourth terminal portion84as the terminal portion80. Each of the first terminal portion81and the second terminal portion82is held by the first terminal holding portion71of the first holder member41and is electrically connected to the transmission coil61. Each of the third terminal portion83and the fourth terminal portion84is held by the second terminal holding portion72of the second holder member51and is electrically connected to the reception coil62. The first to fourth terminal portions81to84are formed by, for example, bending a long plate-like metal member (at, for example, it is bent 90 degrees). More specifically, the third terminal portion83has an L shape in front view and has a vertical portion extending in the vertical direction and a horizontal portion extending in the horizontal direction from the lower end of the vertical portion. The vertical portion is an external terminal85externally connected when the power transmission device100is mounted, and the horizontal portion is a crimp terminal86to which the extended wiring portion66of the reception coil62is fixed. The lower portion of the external terminal85of the third terminal portion83is embedded in the second plate-shaped portion76of the second terminal holding portion72, and the upper portion of the external terminal85is exposed to the outside from the upper surface of the second plate-shaped portion76. On the other hand, the crimp terminal86extends horizontally along the lower surface of the second plate-shaped portion76(Approximately extending in the tangential direction of the outer circumference of the second annular portion54in a plan view). Furthermore, in the present embodiment, a groove portion (not shown in the drawings) having a shape corresponding to the crimp terminal86of the third terminal portion83is formed on the lower surface of the second plate-shaped portion76. A portion of the crimp terminal86of the third terminal portion83is accommodated in the groove portion and is arranged horizontally along the bottom surface of the groove portion. Further, the tip portion (the end portion on the side opposite to the external terminal85side) of the crimp terminal86constitutes a fitting portion87curved in a substantially cylindrical shape whose axial direction is horizontal and is orthogonal to the extending direction of the crimp terminal86. By closing the distal end of the fitting portion87and the proximal end of the fitting portion87in a state where one extended wiring portion66of the pair of extended wiring portions66of the reception coil62is inserted into the inner cavity of the fitting portion87, the extended wiring portion66can be fitted to the fitting portion87. That is, by swaging the fitting portion87, the extended wiring portion66can be fixed to the crimp terminal86in a state of being inserted into the inner cavity of the fitting portion87. In this state, for example, welding fixation or soldering fixation is further performed. For example, the fourth terminal portion84is formed in a symmetrical shape with the third terminal portion83with respect to a plane including a radial direction of the second holder member51toward from the rotation axis95to the second terminal holding portion72and the rotation axis95. Therefore, similarly to the third terminal portion83, the fourth terminal portion84includes an external terminal85, a crimp terminal86, and a fitting portion87. The other extended wiring portion66(the extended wiring portion66that is not connected to the third terminal portion83) of the pair of extended wiring portions66of the reception coil62is fixed to the crimp terminal86at the fitting portion87, similarly to the third terminal portion83. A groove portion (not shown in the drawings) having a shape corresponding to the crimp terminal86of the fourth terminal portion84is formed on the lower surface of the second plate-shaped portion76. A portion of the crimp terminal86of the fourth terminal portion84is accommodated in the groove portion and is arranged horizontally along the bottom surface of the groove portion. For example, the first terminal portion81is formed in a vertically symmetrical shape with the third terminal portion83. Therefore, similarly to the third terminal portion83, the first terminal portion81includes an external terminal85, a crimp terminal86, and a fitting portion87. Similarly to the third terminal portion83, one extended wiring portion66of the pair of extended wiring portions66of the transmission coil61is fixed to the crimp terminal86at the fitting portion87. Similarly, for example, the second terminal portion82is formed in a vertically symmetrical shape with the fourth terminal portion84. Therefore, the second terminal portion82includes an external terminal85, a crimp terminal86, and a fitting portion87. Similarly to the third terminal portion83, the other extended wiring portion66(the extended wiring portion66that is not connected to the first terminal portion81) of the pair of extended wiring portions66of the transmission coil61is fixed to the crimp terminal86at the fitting portion87. Each of the first split cores32and the second split cores34is integrally formed of a magnetic material as a whole. Each of the first holder member41and the second holder member51is integrally molded of, for example, an insulating material such as resin as a whole. In the present embodiment, the transmission unit10is connected to a power supply (not shown in the drawings), and a current is applied to the transmission coil61from the power supply. Since a magnetic field is generated around the transmission coil61by applying a current to the transmission coil61, an induced electromotive force is generated in the reception coil62. That is, in the power transmission device100, power is transmitted from the transmission coil61of the transmission unit10to the reception coil62of the reception unit20by the electromagnetic induction method. Here, in the present embodiment, the power transmission device100further includes a magnetic seal90. The magnetic seal90is disposed along the outer peripheral surface110aof the steering shaft110from the gap between the outer peripheral surface110aof the steering shaft110and the inner peripheral surface10aof the transmission unit10to the gap between the outer peripheral surface110aof the steering shaft110and the inner peripheral surface20aof the reception unit20. As a result, since the magnetic seal90is disposed around the steering shaft110, even when the steering shaft110is made of a metal material, the occurrence of an eddy current in the surface layer of the outer peripheral surface110aof the steering shaft110can be suppressed, and the power transmission efficiency of the power transmission device100can be improved. More specifically, in the power transmission device100, the power transmission efficiency is indicated by a product of a coupling degree between the transmission coil61and the reception coil62and a quality factor (Q factor (Q value)) of each of the transmission coil61and the reception coil62. Therefore, in order to improve the power transmission efficiency, it is required to increase the Q factor. According to the above-described configuration, since the resistance value of the transmission coil61can be further reduced by suppressing the occurrence of the eddy current, the Q factor of the transmission coil61can be increased. Furthermore, since the occurrence of an eddy current in the surface layer of the outer peripheral surface110aof the steering shaft110is suppressed by the magnetic seal90, even if each of the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20is arranged closer to the steering shaft110, high power transmission efficiency of the power transmission device100can be achieved. Therefore, the power transmission device100can be further downsized. Here,FIGS.8A and8Billustrate an example of measured values of each characteristic of the transmission coil61measured using the power transmission device100according to the present embodiment, andFIGS.9A and9Billustrate an example of measured values of each characteristic of the transmission coil61measured using a power transmission device100according to a first modification (power transmission device100not including the magnetic seal90) described later. Among these values, FIGS.8A and9A illustrate, in tables, a measured value (L [uH] inFIGS.8A and9A) of the inductance of the transmission coil61, a measured value (R [mohm] [Ω] inFIGS.8A and9A) of the resistance of the transmission coil61, and a measured value (Q inFIGS.8A and9A) of the Q factor of the transmission coil61at each frequency [kHz] of the power supply.FIG.8Billustrates a profile obtained by plotting the measured values [mohm] [Ω] of the resistance of the transmission coil61inFIG.8Aand a profile obtained by plotting the measured values of the Q factor of the transmission coil61inFIG.8A.FIG.9Billustrates a profile obtained by plotting the measured values [mohm] [Ω] of the resistance of the transmission coil61inFIG.9Aand a profile obtained by plotting the measured values of the Q factor of the transmission coil61inFIG.9A. In addition, inFIGS.8B and9B, the vertical axis on the right side represents the Q factor of the transmission coil61, the vertical axis on the left side represents the resistance value [mohm], and the horizontal axis represents the frequency [kHz]. As illustrated inFIGS.8A and8B, as compared with the case where the power transmission device100according to the first modification described later is used (seeFIGS.9A and9B), when the power transmission device100according to the present embodiment is used, the measured value of the resistance of the transmission coil61further decreases, while the measured value of the Q factor of the transmission coil61further increases. More specifically, for example, when the frequency of the power supply is 100 kHz, an example of the measured value of the resistance of the transmission coil61measured using the power transmission device100according to the present embodiment is 107 mohm, and an example of the Q factor of the transmission coil61is 100. On the other hand, for example, when the frequency of the power supply is 100 kHz, an example of the measured value of the resistance of the transmission coil61measured using the power transmission device100without the magnetic seal90is 200 mohm, and an example of the Q factor of the transmission coil61is 39. As described above, by disposing the magnetic seal90around the steering shaft110, the Q factor of the transmission coil61can be increased, so that the power transmission efficiency of the power transmission device100can be improved. As illustrated inFIG.7, the magnetic seal90includes, for example, a sheet-like base material91, a plurality of magnetic bodies92disposed on one surface of the base material91, and an adhesive layer93formed on the other surface of the base material91.FIG.7is a cross-sectional view taken along the radial direction. The base material91is made of, for example, a soft resin material. Each of the plurality of magnetic bodies92is formed in, for example, a thin plate shape, and the shape of each magnetic body92is not particularly limited, but may be, for example, a substantially rectangular shape. For example, the plurality of magnetic bodies92is arranged in an array on substantially the entire one surface of the base material91. The plurality of magnetic bodies92is, for example, sintered ferrite having high magnetic permeability. Therefore, the magnetic seal90can satisfactorily suppress leakage of magnetic flux generated around the transmission coil61. The magnetic seal90is fixed to the outer peripheral surface110aof the steering shaft110via the adhesive layer93, for example. More specifically, the opposite surface of the base material91on which the adhesive layer93is formed is disposed along the outer peripheral surface110aof the steering shaft110. In addition, the one surface of the base material91on which the plurality of magnetic bodies92is disposed is disposed along each of the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20. For example, the magnetic seal90is fixed to the steering shaft110via the adhesive layer93in a state of being wound along the outer peripheral surface110aof the steering shaft110. Here, when the magnetic seal90is wound around the steering shaft110, since the plurality of magnetic bodies92is arranged in an array, the magnetic seal90can be easily deformed into a shape along the outer peripheral surface110aof the steering shaft110. However, the magnetic seal90may be disposed, for example, along each of the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20. In this case, the opposite surface of the base material91on which the adhesive layer93is formed is disposed along each of the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20. Here, for example, the steering shaft110has the shaft115formed in a cylindrical shape as described above, and the connecting member120(seeFIGS.1and2and the like) is attached to the upper end portion of the shaft115. For example, the steering wheel is attached to the connecting member120and is connected to the upper end portion of the shaft115via the connecting member120. Note thatFIGS.1and6illustrate a state in which the connecting member120is inserted through the cavity11of the transmission unit10and the cavity21of the reception unit20. When a rotation operation is performed on the steering wheel, the shaft115rotates about the axis of the shaft115along with the steering wheel and the connecting member120. More specifically, the connecting member120includes a columnar portion121formed in a columnar shape with the vertical direction as the axial direction, a through hole123penetrating the columnar portion121in the vertical direction, and a flange portion124protruding from the upper edge of the columnar portion121to the periphery. In the present embodiment, the steering wheel is attached to the connecting member120by fitting a part of the steering wheel into the through hole123of the connecting member120. By fitting a lower portion of the columnar portion121into the shaft115, the connecting member120is attached to the upper end portion of the shaft115. More specifically, the lower portion of the columnar portion121of the connecting member120constitutes a stepped fitting portion122, and the outer diameter of the connecting member120decreases downward in two stages. The connecting member120and the steering wheel is attached to the shaft115by fitting the fitting portion122into the inner cavity of the upper end portion of the shaft115. The upper portion of the columnar portion121is positioned above the upper end portion of the shaft115, and the outer peripheral surface of the upper portion faces the inner peripheral surface20aof the reception unit20. The outer peripheral surface of the upper portion of the columnar portion121is disposed on the same cylindrical surface as the outer peripheral surface of the shaft115. The columnar portion121of the connecting member120is inserted into the cavity11of the transmission unit10and the cavity21of the reception unit20. As illustrated inFIG.6, in the present embodiment, for example, the magnetic seal90is arranged circumferentially along the outer peripheral surface of the upper portion of the columnar portion121and the outer peripheral surface of the upper end portion of the shaft115so as to cover a portion from the upper portion of the columnar portion121to the upper end portion of the shaft115. As a result, since the magnetic seal90collectively wraps the upper portion of the columnar portion121and the upper end portion of the shaft115, a state in which the connecting member120is attached to the upper end portion of the shaft115can be favorably maintained. As described above, the outer peripheral surface110aof the steering shaft110is disposed along the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20. More specifically, the upper outer peripheral surface of the columnar portion121is disposed along the inner peripheral surface20aof the reception unit20, and the outer peripheral surface of the upper end portion of the shaft115is disposed along the inner peripheral surface10aof the transmission unit10. Therefore, the outer peripheral surface of the magnetic seal90is disposed along the inner peripheral surface10aof the transmission unit10and is disposed along the inner peripheral surface20aof the reception unit20. In the present embodiment, a gap is formed between the outer peripheral surface of the magnetic seal90and each of the inner peripheral surface10aof the transmission unit10and the inner peripheral surface20aof the reception unit20. In the present embodiment, the magnetic seal90is wound, for example, one or more turns around the outer peripheral surface110aof the steering shaft110. First Modification Next,FIGS.9A and9Billustrate the first modification of the embodiment. A power transmission device100according to the present modification is different from the power transmission device100according to the above-described embodiment in that the magnetic seal90is not provided, and is configured similarly to the power transmission device100according to the above-described embodiment in other points. As described above, when the frequency of the power supply is 100 kHz, an example of the measured value of the resistance of the transmission coil measured using the power transmission device100without the magnetic seal90is 200 mohm, and an example of the Q factor of the transmission coil is 39. Second Modification Next, a second modification of the embodiment will be described with reference toFIG.10. A power transmission device100according to the present modification is different from the power transmission devices100according to the above-described embodiment and the first modification in the following points, and is configured similarly to the power transmission devices100according to the above-described embodiment and the first modification in other points. In the present modification, the first holder member41has first protrusions (not shown in the drawings), each of which is arranged between two first split cores32adjacent to each other in the circumferential direction among the first split cores32, and the second holder member51has second protrusions59(seeFIG.10), each of which is arranged between two second split cores34adjacent to each other in the circumferential direction among the second split cores34. This makes it possible to suppress displacement of the first split cores32in the circumferential direction. In addition, it is possible to suppress displacement of the second split cores34in the circumferential direction. In the present modification, as an example, the second holder member51includes second protrusions59instead of the second engagement protrusions56. Each notch-shaped portion39is formed between two first split cores32adjacent to each other among the first split cores32, and the second protrusions59are engaged with the notch-shaped portions39. More specifically, each of the second protrusions59is, for example, a protrusion extending toward the radial inner side from the inner peripheral surface of each of the second upright wall portions57, unlike the second engagement protrusions56. Further, notch-shaped portions39aand39bare formed at both ends of each of the second split cores34in the circumferential direction. For example, the notch-shaped portion39apenetrates the second split core34in the vertical direction and is opened toward one side in the circumferential direction, and the notch-shaped portion39bpenetrates the second split core34in the vertical direction and is opened toward the other side in the circumferential direction. Of two second split cores34adjacent to each other among the second split cores34, the notch-shaped portion39aof one second split core34and the notch-shaped portion39bof the other second split core34are combined to form a notch-shaped portion39. Similarly to the second holder member51, the first holder member41has, as an example, first protrusions instead of the first engagement protrusions46. Each notch-shaped portion39is formed between two first split cores32adjacent to each other among the first split cores32, and the first protrusions are engaged with the notch-shaped portions39. Also in the present modification, the number of second upright wall portions57included in the second holder member51is, for example, 6. Therefore, the number of second protrusions59included in the second holder member51is also 6, for example. However, the number of second protrusions59is not particularly limited. Similarly, the number of first upright wall portions47included in the first holder member41is 6, for example. Therefore, the number of first protrusions included in the first holder member41is also 6, for example. However, the number of first protrusions is not particularly limited. Although each embodiment has been described above with reference to the drawings, these are examples of the present invention, and various configurations other than the above description can be adopted. For example, in the above description, the example has been described in which the system for transmitting power from the transmission unit10to the reception unit20is an electromagnetic induction system, but the present invention is not limited to this example, and a magnetic field resonance system may be used. Furthermore, for example, in the above description, the example has been described in which the first upright wall portions47are arranged at the plurality of locations in the circumferential direction, and the second upright wall portions57are arranged at the plurality of locations in the circumferential direction. However, in the present invention, for example, the first upright wall portions47may be formed in a circling shape along the outer peripheral edge of the first annular portion44, and the second upright wall portions57may be formed in a circling shape along the outer peripheral edge of the second annular portion54. The present embodiment includes the following technical ideas. (1) A power transmission device including an annular transmission unit having an annular first magnetic core and a transmission coil; and an annular reception unit having an annular second magnetic core and a reception coil, wherein the transmission unit and the reception unit are arranged to face each other, the power transmission device transmits power from the transmission unit to the reception unit,the transmission unit and the reception unit are relatively rotatable about a rotation axis passing through a cavity inside the transmission unit and a cavity inside the reception unit,the first magnetic core has a structure divided into a plurality of first split cores in a circumferential direction, andthe second magnetic core has a structure divided into a plurality of second split cores in the circumferential direction. (2) The power transmission device according to (1), whereinthe transmission unit includes a first holder member made of resin and holding the first magnetic core,the reception unit includes a second holder member made of resin and holding the second magnetic core,the first holder member has an annular shape arranged along one surface of the first magnetic core in an axial direction, andthe second holder member has an annular shape arranged along one surface of the second magnetic core in the axial direction. (3) The power transmission device according to (2), whereinthe first holder member includes a first engagement protrusion that engages with the first split core, and positional displacement of the first split core in the circumferential direction is restricted by the first engagement protrusion,the second holder member includes a second engagement protrusion that engages with the second split core, and positional displacement of the second split core in the circumferential direction is restricted by the second engagement protrusion. (4) The power transmission device according to (2) or (3), whereinthe first holder member includes a first upright wall portion disposed along an outer peripheral surface of the first magnetic core, and the first upright wall portion is disposed at each of a plurality of locations in the circumferential direction,the second holder member includes a second upright wall portion disposed along an outer peripheral surface of the second magnetic core, and the second upright wall portion is disposed at each of a plurality of locations in the circumferential direction. (5) The power transmission device according to (4), whereinthe first upright wall portion includes a first engagement claw portion that engages with a surface of the first magnetic core opposite to the one surface of the first magnetic core, andthe second upright wall portion includes a second engagement claw portion that engages with a surface of the second magnetic core opposite to the one surface of the second magnetic core. (6) The power transmission device according to (2) or (3), whereinthe first holder member includes a first upright wall portion disposed along an outer peripheral surface of the first magnetic core,the first upright wall portion includes a first engagement claw portion that engages with a surface of the first magnetic core opposite to the one surface of the first magnetic core,the second holder member includes a second upright wall portion disposed along an outer peripheral surface of the second magnetic core, andthe second upright wall portion includes a second engagement claw portion that engages with a surface of the second magnetic core opposite to the one surface of the second magnetic core. (7) The power transmission device according to (5) or (6), whereinthe common first engagement claw portion is engaged with two first split cores adjacent to each other among the plurality of first split cores of the first magnetic core, andthe common second engagement claw portion is engaged with two second split cores adjacent to each other among the plurality of second split cores of the second magnetic core. (8) The power transmission device according to any one of (1) to (7), whereinthe transmission unit and the reception unit are disposed around a metal steering shaft,the power transmission device further includes a magnetic seal, andthe magnetic seal is disposed along an outer peripheral surface of the steering shaft from a gap between an outer peripheral surface of the steering shaft and an inner peripheral surface of the transmission unit to a gap between the outer peripheral surface of the steering shaft and an inner peripheral surface of the reception unit. | 70,283 |
11863043 | DETAILED DESCRIPTION In the following description, a vertical direction is defined and described based on a positional relationship when a drive device according to an embodiment is mounted on a vehicle positioned on a horizontal road surface. That is, a relative positional relationship with respect to the vertical direction described in the following embodiments needs to be satisfied at least when the drive device is mounted on a vehicle positioned on a horizontal road surface. In the drawings, an xyz coordinate system is shown appropriately as a three-dimensional orthogonal coordinate system. In the XYZ coordinate system, a Z-axis direction corresponds to the vertical direction. An arrow in the Z-axis is directed toward a side (+Z side) that is an upper side in the vertical direction, and a side (−Z side) opposite to the side toward which the arrow in the Z-axis is directed is a lower side in the vertical direction. In the following description, the upper side and the lower side in the vertical direction will be referred to simply as the “upper side” and the “lower side”, respectively. An X-axis direction is orthogonal to the Z-axis direction and corresponds to a front-rear direction of the vehicle on which the drive device is mounted. In the following embodiments, a side (+X side) toward which an arrow in the X-axis is directed is a front side in the vehicle, and a side (−X side) opposite to the side toward which the arrow in the X-axis is directed is a rear side in the vehicle. A Y-axis direction is orthogonal to both the X-axis direction and the Z-axis direction and corresponds to a left-right direction of the vehicle, i.e., a vehicle lateral direction. In the following embodiments, a side (+Y side) toward which an arrow in the Y-axis is directed is a left side in the vehicle, and a side (−Y side) opposite to the side toward which the arrow in the Y-axis is directed is a right side in the vehicle. The front-rear direction and the left-right direction are each a horizontal direction orthogonal to the vertical direction. A positional relationship in the front-rear direction is not limited to the positional relationship of the following embodiments. The side (+X side) toward which the arrow in the X-axis is directed may be the rear side in the vehicle, and the side (−X side) opposite to the side toward which the arrow in the X-axis is directed may be the front side in the vehicle. In this case, the side (+Y side) toward which the arrow in the Y-axis is directed is the right side in the vehicle, and the side (−Y side) opposite to the side toward which the arrow in the Y-axis is directed is the left side in the vehicle. In the present specification, a “parallel direction” includes a substantially parallel direction, and an “orthogonal direction” includes a substantially orthogonal direction. A central axis J illustrated in the drawings as appropriate is a virtual axis extending in a direction intersecting the vertical direction. More specifically, the central axis J extends in the Y-axis direction orthogonal to the vertical direction, i.e., in the left-right direction of the vehicle. In description below, unless otherwise particularly stated, a direction parallel to the central axis J is simply referred to as the “axial direction”, a radial direction about the central axis J is simply referred to as the “radial direction”, and a circumferential direction about the central axis J, i.e., a direction around the central axis J is simply referred to as the “circumferential direction”. In the following embodiments, the right side (−Y side) is referred to as a “first axial side”, and the left side (+Y side) is referred to as a “second axial side”. FIG.1illustrates a drive device100of the present embodiment that is mounted on a vehicle and rotates an axle64. The vehicle on which the drive device100is mounted is a vehicle including a motor as a power source, such as a hybrid vehicle (HEV), a plug-in hybrid vehicle (PHV), or an electric vehicle (EV). As illustrated inFIG.1, the drive device100includes a rotary electric machine10and a transmission device60. The transmission device60is connected to the rotary electric machine10, and transmits rotation of the rotary electric machine10, i.e., rotation of a rotor30described later, to the axle64of the vehicle. The transmission device60of the present embodiment includes a gear housing61, a speed reducer62connected to the rotary electric machine10, and a differential gear63connected to the speed reducer62. The gear housing61internally accommodates the speed reducer62, the differential gear63, and oil O. The oil O is stored in a lower region in the gear housing61. The oil O circulates in a refrigerant flow path90described later. The oil O is used as a refrigerant for cooling the rotary electric machine10. The oil O is also used as lubricating oil for the speed reducer62and the differential gear63. As the oil O, for example, an oil equivalent to an automatic transmission fluid (ATF) having a relatively low viscosity is preferably used to function as a refrigerant and a lubricating oil. The differential gear63includes a ring gear63a. The ring gear63areceives torque output from the rotary electric machine10and transmitted through the speed reducer62. The ring gear63ahas a lower end portion immersed in the oil O stored in the gear housing61. When the ring gear63arotates, the oil O is scraped up. The oil O scraped up is supplied to, for example, the speed reducer62and the differential gear63as a lubricating oil. The rotary electric machine10drives the drive device100. The rotary electric machine10is located, for example, on the first axial side (−Y side) from the transmission device60. In the present embodiment, the rotary electric machine10is a motor. The rotary electric machine10includes a motor housing20, a rotor30having a shaft31, bearings34and35that rotatably support the rotor30, a stator40, a resolver50, a nozzle member70, and a current shunter80. The bearings34and35are each a ball bearing, for example. The motor housing20internally accommodates the rotor30and the stator40. The motor housing20is connected to the gear housing61on the first axial side (−Y side). The motor housing20has a body21, a partition wall22, and a lid23. The body21and the partition wall22are each, for example, a part of a single member. The lid23is separate from, for example, the body21and the partition wall22. The body21is in a cylindrical shape that surrounds the central axis J and opens toward the first axial side (−Y side). The partition wall22is connected to an end portion of the body21on the second axial side (+Y side). The partition wall22axially partitions the inside of the motor housing20and the inside of the gear housing61. The partition wall22has a partition wall opening22athat allows the inside of the motor housing20to communicate with the inside of the gear housing61. The partition wall22holds a bearing34. The lid23is fixed to an end portion of the body21on the first axial side. The lid23closes an opening of the body21on the first axial side. The lid23holds the bearing35. As illustrated inFIG.2, the lid23has a hole23frecessed from its surface on the second axial side (+Y side) toward the first axial side (−Y side). The hole23fhas a bottom on the first axial side and opens toward the second axial side. In the present embodiment, the hole23fis a circular hole about the central axis J. Providing the hole23fprovides a bottom wall23aand a peripheral wall23bto the lid23. That is, the motor housing20includes the bottom wall23aand the peripheral wall23b. The bottom wall23ais the bottom of the hole23f. The bottom wall23ais located on the first axial side (−Y side) from an open end portion31dof the shaft31. The peripheral wall23bprotrudes from a radially outer peripheral edge of the bottom wall23atoward the second axial side (+Y side). The peripheral wall23bsurrounds the open end portion31dof the shaft31. The peripheral wall23bhas an inner peripheral surface that is an inner peripheral surface of the hole23f. In the present embodiment, the inner peripheral surface of the peripheral wall23bhas a cylindrical shape about the central axis J. The peripheral wall23bincludes a first wall portion23c, a second wall portion23d, and a third wall portion23e. The first wall portion23cis connected to a radially outer peripheral edge of the bottom wall23a. The second wall portion23dis connected to the first wall portion23con the second axial side (+Y side). The second wall portion23dhas a larger inner diameter than the first wall portion23c. The second wall portion23dhas a larger axial dimension than the first wall portion23c. The third wall portion23eis connected to the second wall portion23don the second axial side. The third wall portion23ehas a larger inner diameter than the second wall portion23d. The third wall portion23ehas a larger axial dimension than the second wall portion23d. Radially inside the third wall portion23e, the bearing35is held. The bearing35includes an outer ring fitted into the third wall portion23eradially inward. In the present embodiment, the inner peripheral surface of the peripheral wall23bhas a first stepped portion24aand a second stepped portion24b. The first stepped portion24ais provided axially between an inner peripheral surface of the first wall portion23cand an inner peripheral surface of the second wall portion23d. The first stepped portion24ahas a first shoulder surface24cfacing the second axial side (+Y side). The first shoulder surface24cis in an annular shape about the central axis J. The first shoulder surface24cis a flat surface orthogonal to the axial direction. The second stepped portion24bis provided axially between the inner peripheral surface of the second wall portion23dand an inner peripheral surface of the third wall portion23e. The second stepped portion24bhas a second shoulder surface24dfacing the second axial side. The second shoulder surface24dis in an annular shape about the central axis J. The second shoulder surface24dis a flat surface orthogonal to the axial direction. The bearing35held in the third wall portion23eis in contact with the second shoulder surface24d. Thus, the bearing35can be suitably positioned in the axial direction with respect to the motor housing20. More specifically, the outer ring of the bearing35is in contact with the second shoulder surface24dfrom the second axial side. The lid23has a surface on the second axial side (+Y side) that is provided with a resolver holding portion25. In the present embodiment, the resolver holding portion25is composed of a plurality of protruding wall portions25aprotruding toward the second axial side. The plurality of protruding wall portions25ais provided on a peripheral edge portion around the hole23fin a surface of the lid23on the second axial side. The plurality of protruding wall portions25ais disposed surrounding the shaft31. As illustrated inFIG.1, the rotor30includes the shaft31and a rotor body32. Although not illustrated, the rotor body32includes a rotor core, and a rotor magnet fixed to the rotor core. Torque of the rotor30is transmitted to the transmission device60. The shaft31is rotatable about the central axis J. The shaft31is rotatably supported by the bearings34and35. The shaft31is a hollow shaft. The shaft31has a cylindrical shape about the central axis J and extends axially. The shaft31is provided with a hole33that allows the inside of the shaft31to communicate with the outside of the shaft31. The shaft31extends across the inside of the motor housing20and the inside of the gear housing61. The shaft31has an end portion on the second axial side (+Y side) that protrudes into the inside of the gear housing61. The shaft31is connected at the end portion on the second axial side to the speed reducer62. The shaft31is open on both sides in the axial direction. As illustrated inFIG.2, the shaft31has the open end portion31dthat opens on the first axial side (−Y side). The shaft31includes a first shaft portion31a, a second shaft portion31b, and a third shaft portion31c. The second shaft portion31bis connected to the first shaft portion31aon the first axial side. The second shaft portion31bhas a smaller outer diameter than the first shaft portion31a. The third shaft portion31cis connected to the second shaft portion31bon the first axial side. The third shaft portion31chas a smaller outer diameter than the second shaft portion31b. The third shaft portion31chas a smaller axial dimension than the second shaft portion31b. The third shaft portion31chas an end portion on the first axial side that is an end portion of the shaft31on the first axial side, and that is the open end portion31d. The first shaft portion31a, the second shaft portion31b, and the third shaft portion31care equal in inner diameter to each other. Between an outer peripheral surface of the first shaft portion31aand an outer peripheral surface of the second shaft portion31b, a stepped portion having a shoulder surface facing the first axial side (−Y side) is provided. Between the outer peripheral surface of the second shaft portion31band an outer peripheral surface of the third shaft portion31c, a stepped portion having a shoulder surface facing the first axial side (−Y side) is provided. A portion of the second shaft portion31bon the first axial side (−Y side) and the third shaft portion31care located radially inside the peripheral wall23b. More specifically, the portion of the second shaft portion31bon the first axial side (−Y side) is located radially inside the third wall portion23e. The third shaft portion31cis located radially inside the second wall portion23dand the first wall portion23c. In the present embodiment, the open end portion31dis located radially inside the first wall portion23c. The outer peripheral surface of the second shaft portion31band the outer peripheral surface of the third shaft portion31care disposed away from the inner peripheral surface of the peripheral wall23bin a radially inward direction. The open end portion31dis disposed on the second axial side (+Y side) away from the bottom wall23a. As illustrated inFIG.1, the stator40faces the rotor30across a gap in the radial direction. More specifically, the stator40is located radially outward of the rotor30. The stator40is fixed inside the motor housing20. The stator40includes the stator core41and a coil assembly42. The stator core41is in an annular shape surrounding the central axis J of the rotary electric machine10. The stator core41is located radially outside the rotor30. The stator core41surrounds the rotor30. The stator core41is composed of, for example, plate members such as electromagnetic steel plates stacked in the axial direction. Although not illustrated, the stator core41includes a core back in a cylindrical shape extending in the axial direction, and a plurality of teeth extending to an inner side from the core back in the radial direction. The coil assembly42includes multiple coils42cattached to the stator core41along the circumferential direction. The multiple coils42care mounted on the respective teeth of the stator core41with respective insulators (not illustrated) interposed therebetween. The coil assembly42includes coil ends42aand42bthat protrude axially from the stator core41. The resolver50can detect rotation of the rotor30. The resolver50is accommodated inside the motor housing20. The resolver50includes a resolver rotor51and a resolver stator52. The resolver rotor51is fixed to the shaft31. The resolver rotor51is in an annular shape surrounding the shaft31. In the present embodiment, the resolver rotor51is in an annular shape about the central axis J. As illustrated inFIG.2, the resolver rotor51surrounds an end portion of the second shaft portion31bon the second axial side (+Y side) in the present embodiment. The resolver rotor51is in a plate shape in which a plate surface faces the axial direction. The resolver rotor51has a surface on the second axial side that is in contact with the shoulder surface of the stepped portion provided between the first shaft portion31aand the second shaft portion31bin the axial direction. The resolver rotor51protrudes radially outward from the outer peripheral surface of the first shaft portion31a. The resolver rotor51is disposed on the second axial side apart from the bearing35. The resolver stator52is located radially outside the resolver rotor51. The resolver stator52is in an annular shape surrounding the resolver rotor51. The resolver stator52is held by the resolver holding portion25. Although not illustrated, the resolver stator52includes a coil. When the resolver rotor51rotates together with the shaft31, induced voltage corresponding to a circumferential position of the resolver rotor51is generated in the coil of the resolver stator52. The resolver50can detect rotation of the resolver rotor51and the shaft31based on change in the induced voltage generated in the coil of the resolver stator52. This enables the resolver50to detect rotation of the rotor30. The current shunter80is located radially inside the peripheral wall23b. The current shunter80is in an annular shape surrounding the shaft31. In the present embodiment, the current shunter80is in an annular shape about the central axis J. The current shunter80surrounds the third shaft portion31c. In the present embodiment, the current shunter80is fitted inside the second wall portion23din the radial direction. The current shunter80is located on the second axial side (+Y side) from the open end portion31d. That is, the open end portion31dis located on the first axial side (−Y side) from the current shunter80. The current shunter80is located on the first axial side (−Y side) from the bearing35. This allows the bearing35to be located between the resolver rotor51and the current shunter80in the axial direction. In the present embodiment, the current shunter80faces the bearing35across a gap. An axial distance between the current shunter80and the bearing35is smaller than an axial distance between the bearing35and the resolver rotor51. As illustrated inFIG.3, the current shunter80includes a base81in an annular shape about the central axis J, and a brush82provided on a radially inner edge of the base81over the entire circumference. As illustrated inFIG.2, the base81is fitted inside the second wall portion23din the radial direction. The base81is fixed to the second wall portion23dwith, for example, an adhesive. As a result, the current shunter80is fixed to the motor housing20. A method for fixing the current shunter80to the motor housing20is not particularly limited. The current shunter80may be fixed to the motor housing20by press fitting, for example. The base81has a surface on the first axial side (−Y side) in which a radially outer edge portion is in contact with the first shoulder surface24c. As a result, the current shunter80is in contact with the first shoulder surface24c. Thus, the current shunter80can be suitably positioned in the axial direction with respect to the motor housing20. The base81is in electrical contact with the peripheral wall23b. As a result, the current shunter80is in electrical contact with the motor housing20. In the present specification, the text, “an object is in electrical contact with another object”, means that an electric current can flow between the object and the other object. The brush82is in an annular shape surrounding the shaft31. More specifically, the brush82is in an annular shape about the central axis J to surround the third shaft portion31c. In the present embodiment, the brush82is composed of a plurality of conductive fibers protruding radially inward from the radially inner edge of the base81. The fibers constituting the brush82are, for example, microfibers. The brush82is electrically connected to the base81. The brush82has a radially inner edge in electrical contact with the outer peripheral surface of the third shaft portion31c. As a result, the current shunter80is in electrical contact with the shaft31. In the present embodiment, the shaft31rotates with the third shaft portion31chaving the outer peripheral surface that is rubbed against the radially inner edge of the brush82. In this way, the shaft31and the motor housing20are electrically connected through the current shunter80. This enables a current generated in the shaft31to flow from the peripheral wall23bto the motor housing20through the brush82and the base81in this order. As a result, the current can be prevented from flowing from the shaft31to the bearings34and35that rotatably support the shaft31. Thus, electrolytic corrosion can be prevented from occurring in the bearings34and35. In the present embodiment, the current shunter80is excellent in oil resistance. That is, the current shunter80is unlikely to undergo change due to contact with the oil O. It is conceivable that the oil resistance is evaluated by an immersion test in the oil O. In this case, the oil resistance is evaluated by change in weight and change in strength after immersion for a predetermined time. The evaluation of change in weight includes, for example, evaluation in terms of corrosion and swelling. The nozzle member10is used for feeding the oil O as a fluid into the inside of the shaft31. The nozzle member70is formed by performing machining, such as press working, on a plate member made of metal, for example. The nozzle member70is disposed inside the peripheral wall23b. The nozzle member70includes a feeding tubular part71and a flange portion72. The feeding tubular part71extends in the axial direction. In the present embodiment, the feeding tubular part71is in a cylindrical shape about the central axis J. The feeding tubular part71is open on both sides in the axial direction. At least a part of the feeding tubular part71is inserted into inside the shaft31from the open end portion31d. In the present embodiment, the entire feeding tubular part71except for its end portion on the first axial side (−Y side) is inserted into inside the shaft31. The end portion of the feeding tubular part71on the first axial side is located on the first axial side from the shaft31. In the present embodiment, the feeding tubular part71has an outer peripheral surface disposed away from an inner peripheral surface of the shaft31in the radially inward direction. This enables preventing the feeding tubular part71from rubbing against the shaft31. As a result, the feeding tubular part71can be prevented from being worn. The feeding tubular part71includes a large diameter portion71a, a small diameter portion71b, and a connection portion71c. As illustrated inFIGS.2to4, the feeding tubular part71is formed in a funnel shape in the present embodiment by the large diameter portion71a, the small diameter portion71b, and the connection portion71c. As illustrated inFIG.2, the large diameter portion71ais located on the first axial side (−Y side) in the feeding tubular part71. The large diameter portion71ahas an end portion on the first axial side that is an end portion of the feeding tubular part71on the first axial side. The large diameter portion71aincludes an inserted portion71dinserted into the shaft31and an enlarged diameter portion71econnected to the inserted portion71don the first axial side. In the present embodiment, the inserted portion71dis located radially inside the third shaft portion31c. The inserted portion71dhas an outer peripheral surface disposed away from an inner peripheral surface of the third shaft portion31cin the radially inward direction. The inserted portion71dhas inner and outer diameters that are each uniform throughout the axial direction. The enlarged diameter portion71ehas inner and outer diameters that each increase toward the first axial side from the inserted portion71d. As a result, the end portion of the feeding tubular part71on the first axial side in the present embodiment has an inner diameter increasing toward the first axial side. The enlarged diameter portion71ehas an inner peripheral surface that is a tapered surface with an inner diameter decreasing linearly toward the second axial side (+Y side). The enlarged diameter portion71ehas an outer peripheral surface that is a tapered surface with an outer diameter decreasing linearly toward the second axial side. In the present embodiment, the enlarged diameter portion71ehas an end portion on the first axial side (−Y side) that is located radially outside an inner peripheral surface of the open end portion31dand that is located radially inside an outer peripheral surface of the open end portion31d. The end portion of the enlarged diameter portion71eon the first axial side is located on the first axial side away from the open end portion31d. The enlarged diameter portion71ehas a smaller axial dimension than the inserted portion71d. The small diameter portion71bis located on the second axial side (+Y side) in the feeding tubular part71. The small diameter portion71bis connected to the large diameter portion71aon the second axial side. In the present embodiment, the small diameter portion71bis connected to the large diameter portion71awith the connection portion71c. The small diameter portion71bhas an end portion on the second axial side that is an end portion of the feeding tubular part71on the second axial side. The small diameter portion71bhas a smaller inner diameter than the large diameter portion71a. The inner diameter of the small diameter portion71bis, for example, half or less of the inner diameter of the large diameter portion71a. The small diameter portion71bhas a smaller outer diameter than the large diameter portion71a. The outer diameter of the small diameter portion71bis, for example, half or less of the outer diameter of the large diameter portion71a. The small diameter portion71bhas a larger axial dimension than the large diameter portion71a. The entire small diameter portion71bis inserted into inside the shaft31. In the present embodiment, the small diameter portion71bis located radially inside the second shaft portion31b. The small diameter portion71bhas an outer peripheral surface disposed away from the inner peripheral surface of the second shaft portion31bin the radially inward direction. A radial distance between the outer peripheral surface of the small diameter portion71band the inner peripheral surface of the shaft31is larger than a radial distance between an outer peripheral surface of the large diameter portion71aand the inner peripheral surface of the shaft31. In the present embodiment, the end portion of the small diameter portion71bon the second axial side (+Y side) is located radially inside the resolver rotor51. The connection portion71cenlarges in the radial direction and connects the end portion of the large diameter portion71aon the second axial side (+Y side) to the end portion of the small diameter portion71bon the first axial side (−Y side). In the present embodiment, the connection portion71cextends toward the second axial side from its radially outer side to its radially inner side. The connection portion71chas a connection surface71ffacing the first axial side. The connection surface71fis in an annular shape about the central axis J. The connection surface71fconnects the inner peripheral surface of the large diameter portion71ato the inner peripheral surface of the small diameter portion71b. More specifically, the connection surface71fconnects an end portion of an inner peripheral surface of the inserted portion71don the second axial side to an end portion of the inner peripheral surface of the small diameter portion71bon the first axial side. The connection surface71fextends toward the second axial side from the inner peripheral surface of the large diameter portion71atoward the inner peripheral surface of the small diameter portion71b. The connection surface71fis a tapered surface with an inner diameter decreasing linearly toward the second axial side. The connection surface71fhas an inclination with respect to the axial direction that is larger than an inclination of the inner peripheral surface of the enlarged diameter portion71ewith respect to the axial direction. The flange portion72protrudes radially outward from the feeding tubular part71. In the present embodiment, the flange portion12protrudes radially outward from the feeding tubular part71on the first axial side (−Y side). The flange portion72is in an annular shape surrounding the central axis J. In the present embodiment, the flange portion72is in an annular shape about the central axis J. The flange portion72is located between the current shunter80and the bottom wall23ain the axial direction. This allows the current shunter80to be located between the bearing35and the flange portion72in the axial direction. The flange portion72is disposed facing the bottom wall23aon the second axial side (+Y side). The flange portion72is disposed facing the current shunter80on the first axial side. The flange portion72includes an annular portion72aand a tubular portion72b. The annular portion72aprotrudes radially outward from the feeding tubular part71. In the present embodiment, the annular portion72aprotrudes radially outward from the end portion of the enlarged diameter portion71eon the first axial side (−Y side). The annular portion72ais in an annular shape about the central axis J. The annular portion72ais in a plate shape in which a plate surface faces the axial direction. The annular portion72aincludes an inner annular portion72cand an outer annular portion12d. The inner annular portion72cis a radially inner portion of the annular portion72a. The inner annular portion72chas a radially inner edge connected to the end portion of the enlarged diameter portion71eon the first axial side (−Y side). The inner annular portion72chas a radially outer edge located radially outside the outer peripheral surface of the second shaft portion31b. The inner annular portion72chas a surface on the first axial side that is a flat surface72econstituting a part of a surface of the flange portion72on the first axial side. The flat surface72eis orthogonal to the axial direction. As illustrated inFIG.3, the flat surface72eis in an annular shape about the central axis J. The outer annular portion72dis a radially outer portion of the annular portion72a. The outer annular portion72dis connected to the inner annular portion72con a radially outer side. The outer annular portion72dextends toward the second axial side (+Y side) from a radially outer edge of the inner annular portion72ctoward the radially outer side. The outer annular portion72dhas a surface on the first axial side (−Y side) that is an inclined surface72fconstituting a part of the surface of the flange portion12on the first axial side. That is, the surface of the flange portion72on the first axial side includes the inclined surface72f. In the present embodiment, the surface of the flange portion72on the first axial side is composed of the flat surface72eand the inclined surface72f. The inclined surface72fextends radially outward toward the second axial side. The inclined surface72fis in an annular shape about the central axis J. The inclined surface72fis a tapered surface with an outer diameter decreasing linearly toward the first axial side. As illustrated inFIG.2, the tubular portion72bprotrudes from a radially outer edge of the annular portion72atoward the second axial side (+Y side). The tubular portion72bis in a cylindrical shape about the central axis J. The tubular portion72bis fitted radially inside the first wall portion23cin a clearance fit. This allows the flange portion72to be fitted inside the peripheral wall portion23bin the present embodiment. Thus, the nozzle member70can be positioned in the radial direction with respect to the motor housing20. The tubular portion72bprotruding in the axial direction from the radially outer edge of the annular portion72ais provided in the present embodiment, so that the nozzle member70can be suitably positioned in the radial direction with respect to the motor housing20by fitting the tubular portion72binside the peripheral wall23b. The tubular portion12bhas an end portion on the second axial side that is located on the second axial side from the open end portion31d. The end portion of the tubular portion72bon the second axial side surrounds the open end portion31d. That is, the open end portion31dis located radially inside the tubular portion72bin the present embodiment. The tubular portion72bis disposed facing the current shunter80in the axial direction. In the example ofFIG.2, the end portion of the tubular portion72bon the second axial side is in contact with the surface of the base81on the first axial side (−Y side). The flange portion72has an axial dimension L1that is smaller than an axial dimension L2between the bottom wall23aand the current shunter80. Thus, the flange portion72is disposed axially away from at least one of the bottom wall23aand the current shunter80. In the example ofFIG.2, the flange portion72is disposed on the second axial side (+Y side) away from the bottom wall23aand is in contact with the current shunter80. In the present embodiment, the nozzle member70is axially movable within a range in which the flange portion72is axially movable between the bottom wall23aand the current shunter80. The axial dimension L1of the flange portion72is an axial distance between the flat surface72eand the end portion of tubular portion72bon the second axial side. The axial distance L2between the bottom wall23aand the current shunter80is an axial distance between a surface of the bottom wall23aon the second axial side and the surface of the base81on the first axial side (−Y side). In the present embodiment, the axial distance L2between the bottom wall23aand the current shunter80is equal to an axial dimension of the first wall portion23c. In the present embodiment, a gap G is provided between the flange portion72and the bottom wall23ain the axial direction. The gap G includes a gap G1between the flat surface72eand the surface of the bottom wall23aon the second axial side (+Y side), and a gap G2between the inclined surface72fand the surface of the bottom wall23aon the second axial side. The gap G2is larger than the gap G1. When the nozzle member70moves toward the first axial side (−Y side) from the position illustrated inFIG.2and the flat surface72ecomes into contact with the surface of the bottom wall23aon the second axial side, only the gap G2is provided between the flange portion72and the bottom wall23ain the axial direction. As described above, providing the inclined surface72fenables the gap G2to be provided between the flange portion72and the bottom wall23ain the axial direction even when the flange portion72is in contact with the bottom wall23a. In the present embodiment, the flange portion72has at least one feed hole73for feeding the oil O as a fluid to the bearing35. Thus, the oil O as a lubricant can be fed to the bearing35through the feed hole73. In the present embodiment, the feed hole73passes through the flange portion72in the axial direction. In the present embodiment, the feed hole73is provided in the outer annular portion72d. The feed hole73opens in the inclined surface72f. The feed hole73opens in the gap G between the flange portion72and the bottom wall portion23a. More specifically, the feed hole73opens in the gap G2between the inclined surface72fand the surface of the bottom wall portion23aon the second axial side (+Y side). The feed hole73opens between the current shunter80and the flange portion12. As illustrated inFIGS.3and4, the feed hole73in the present embodiment is a circular hole. Multiple feed holes73are provided at intervals in the circumferential direction. The multiple feed holes73are disposed at equal intervals along the circumferential direction. For example, four feed holes13are provided. As illustrated inFIG.1, the drive device100in the present embodiment is provided with the refrigerant flow path90through which the oil O as a refrigerant circulates. The refrigerant flow path90is provided throughout from the inside of the motor housing20to the inside of the gear housing61. The refrigerant flow path90allows the oil O stored in the gear housing61to be fed to the rotary electric machine10and to return to the inside of the gear housing61again. The refrigerant flow path90is provided with a pump96, a cooler97, and the refrigerant feed part95. In the following description, an upstream side in a flow direction of the oil O in the refrigerant flow path90is simply referred to as an “upstream side”, and a downstream side in the flow direction of the oil O in the refrigerant flow path90is simply referred to as a “downstream side”. The refrigerant flow path90includes a gear-side flow path portion91, a connection flow path portion92, and a rotary-electric-machine-side flow path portion93. The gear-side flow path portion91includes a first portion91aand a second portion91b. The first portion91aand the second portion91bare provided, for example, in a wall portion of the gear housing61. The first portion91aallows a portion with the oil O stored, inside the gear housing61, to communicate with the pump96. The second portion91ballows the pump96to communicate with the cooler97. The connection flow path portion92is provided from in a wall portion of the gear housing61to in a wall portion of the motor housing20. The connection flow path portion92allows the gear-side flow path portion91to communicate with the rotary-electric-machine-side flow path portion93. More specifically, the connection flow path portion92allows the cooler97to communicate with a third flow path portion93cdescribed later. The rotary-electric-machine-side flow path portion93is provided in the rotary electric machine10. The rotary-electric-machine-side flow path portion93includes a first flow path portion93a, a second flow path portion93b, and a third flow path portion93c. That is, the rotary electric machine10includes the first flow path portion93a, the second flow path portion93b, and the third flow path portion93c. The first flow path portion93aand the third flow path portion93care each provided in a wall portion of the motor housing20. The second flow path portion93bincludes a housing flow path portion93dprovided in a wall portion of the motor housing20, and the refrigerant feed part95. In the present embodiment, the first flow path portion93a, the third flow path portion93c, and the housing flow path portion93dare provided in the lid23. The third flow path portion93ccommunicates with the first flow path portion93aand the second flow path portion93b. In the present embodiment, the first flow path portion93aand the second flow path portion93bbranch from the third flow path portion93c. The first flow path portion93aallows the oil O as a fluid to be fed into inside the peripheral wall23b. The first flow path93ahas an end portion on the upstream side that communicates with an end portion of the third flow path93con the downstream side. The first flow path portion93ahas an end portion on the downstream side that opens to the inside of the peripheral wall23b. Although not illustrated, the end portion of the first flow path portion93aon the downstream side opens, for example, in an end portion of the inner peripheral surface of the peripheral wall23bon the first axial side (−Y side). The second flow path portion93ballows the oil O as a fluid to be fed to the stator40. The second flow path93bhas an end portion upstream from the housing flow path93d, the end portion communicating with an end portion of the third flow path93con the downstream side. The housing flow path portion93dhas an end portion on the downstream side that communicates with an end portion of the refrigerant feed part95on the upstream side. In the present embodiment, the refrigerant feed part95is in a tubular shape extending in the axial direction. In other words, the refrigerant feed part95is a pipe extending in the axial direction in the present embodiment. The refrigerant feed part95has axially opposite end portions supported by the motor housing20. The refrigerant feed part95has the end portion on the second axial side (+Y side) that is supported by, for example, the partition wall22. The refrigerant feed part95has the end portion on the first axial side (−Y side) that is supported by, for example, the lid23. The refrigerant feed part95is located radially outside the stator40. In the present embodiment, the refrigerant feed part95is located above the stator40. In the present embodiment, the oil O in the refrigerant feed part95flows in a direction from the first axial side toward the second axial side. That is, the oil O in the refrigerant feed part95flows in the direction in which the first axial side is an upstream side and the second axial side is a downstream side. The refrigerant feed part95has a feed port95afor feeding the oil O as a refrigerant to the stator40. In the present embodiment, the feed port95ais an injection port that injects partially the oil O having flowed into the refrigerant feed part95to the outside of the refrigerant feed part95. Multiple feed ports95aare provided. When the pump96is driven, the oil O stored in the gear housing61is sucked up through the first portion91aand flows into the cooler97through the second portion91b. The oil O having flowed into the cooler97is cooled in the cooler97, and then flows through the connection flow path portion92and flows into the rotary-electric-machine-side flow path portion93from the third flow path portion93c. The oil O having flowed into the third flow path portion93cbranches into the first flow path portion93aand the second flow path portion93b. The oil O having flowed into the first flow path portion93aflows into inside the peripheral wall23b. In the present embodiment, the oil O from the first flow path portion93aflows into the gap G between the flange portion72and the bottom wall23ain the axial direction. More specifically, the oil O from the first flow path portion93aflows into the gap G2between the inclined surface72fand the bottom wall23a. As illustrated inFIG.2, the oil O having flowed into inside the peripheral wall23bpartially passes through the inside of the feeding tubular part71of the nozzle member70and flows into inside the shaft31. As described above, providing the first flow path portion93aenables the oil O to be fed from the inside of the peripheral wall23binto the shaft31in the present embodiment. In the present embodiment, the oil O having flowed into the gap G2flows into the feeding tubular part11from the end portion of feeding tubular part71on the first axial side (−Y side) through the gap G1. Here, the axial dimension L1of the flange portion72is smaller than the axial distance L2between the bottom wall23aand the current shunter80in the present embodiment, so that the gap G1can be suitably generated. As a result, even when the oil O is fed from the first flow path portion93ato the gap G2as in the present embodiment, the oil O can be easily fed into the feeding tubular part71through the gap G1. In the present embodiment, the inner diameter of the end portion of the feeding tubular part71on the first axial side in the present embodiment increases toward the first axial side. This enables the oil O to easily flow into the feeding tubular part71from the end portion of the feeding tubular part71on the first axial side. As a result, the oil O can be more easily fed into the feeding tubular part71. As illustrated inFIG.1, the oil O having flowed into the shaft31from the nozzle member70passes through the inside of the rotor body32from the hole33and scatters to the stator40. As illustrated inFIG.2, the oil O having flowed into inside the peripheral wall23bpartially passes through the feed hole73from the first axial side (−Y side) toward the second axial side (+Y side) to be fed to the bearing35. Here, the feed hole73opens in the gap G between the flange portion72and the bottom wall portion23ain the present embodiment. Thus, the oil O having flowed into the gap G can easily pass through the feed hole73. This enables the oil O to be easily fed to the bearing35through the feed hole73. In the present embodiment, the feed hole73opens in the inclined surface72f. This enables the oil O to easily flow into the feed hole73through the gap G2between the inclined surface72fand the bottom wall23ain the axial direction even when the flange portion72is in contact with the bottom wall23a. Thus, the oil O can be more easily fed to the bearing35through the feed hole73. The oil O to be fed to the bearing35passes through the feed hole73and then passes through a radial gap between the current shunter80and the shaft31, for example, to reach the bearing35. The oil O having flowed into the peripheral wall23bmay partially flow into the shaft31through a radial gap between the shaft31and the feeding tubular part71after passing through the feed hole73from the first axial side toward the second axial side. As illustrated inFIG.1, the oil O having flowed into the second flow path portion93bflows into the inside of the refrigerant feed part95through the housing flow path portion93d. The oil O having flowed into the refrigerant feed part95is injected from the feed port95aand fed to the stator40. As described above, providing the first flow path portion93aand the second flow path portion93b, which branch from the third flow path portion93c, enables the oil O fed from the inside of the gear housing61to be suitably and easily fed into the shaft31through the inside of the peripheral wall23band to be fed to the stator40from the refrigerant feed part95. In the present embodiment, the oil O scooped up by the ring gear63apartially enters a reservoir98provided in the gear housing61. The oil O having entered the reservoir98flows into the shaft31from its end portion on the second axial side (+Y side). The oil O having flowed into the shaft31from the reservoir98passes through the inside of the rotor body32from the hole33and scatters to the stator40. The oil O fed to the stator40from the feed port95aand the oil O fed to the stator40from the inside of the shaft31take heat from the stator40. The oil O having cooled the stator40falls downward to accumulate in a lower region in the motor housing20. The oil O accumulated in the lower region in the motor housing20returns to the inside of the gear housing61through the partition wall opening22aprovided in the partition wall22. As described above, the refrigerant flow path90allows the oil O stored in the gear housing61to be fed to the rotor30and the stator40. According to the present embodiment, the flange portion72is located between the current shunter80fixed to the motor housing20and the bottom wall23ain the axial direction. Thus, the flange portion72can be pressed by the bottom wall23afrom the first axial side, and the flange portion72can be pressed by the current shunter80from the second axial side. That is, the current shunter80can prevent the nozzle member70from moving in the axial direction with respect to the motor housing20. This enables the nozzle member70to be attached to the motor housing20by fixing the current shunter80to the motor housing20. Thus, a fixing member for fixing the nozzle member70to the motor housing20is not required in addition to the current shunter80, so that the number of parts of the rotary electric machine10can be reduced. As a result, the number of parts of the drive device100can be reduced. The number of parts of the rotary electric machine10and the number of parts of the drive device100can be reduced, so that man-hours and time required for assembling the rotary electric machine10and the drive device100can be reduced. According to the present embodiment, the current shunter80is located between the bearing35and the flange portion72in the axial direction. Thus, even when the current shunter80comes off from the motor housing20, the bearing35can prevent the current shunter80from moving toward the second axial side. As a result, even when the current shunter80comes off from the motor housing20, the current shunter80can prevent the flange portion72from moving toward the second axial side. According to the present embodiment, the bearing35is located between the resolver rotor51and the current shunter80in the axial direction. Thus, the bearing35can be disposed at a position closer to the current shunter80than when the resolver rotor51is provided between the bearing35and the current shunter80in the axial direction. This enables the oil O to easily reach the bearing35from the feed hole73in a structure in which the oil O is fed to the bearing35from the feed hole73of the nozzle member10as in the present embodiment. According to the present embodiment, the open end portion31dis located on the first axial side (−Y side) from the current shunter80and located radially inside the tubular portion72b. This enables preventing the entire rotary electric machine10from increasing in size in the axial direction by extending the shaft31toward the first axial side from the current shunter80to allow the current shunter80to easily come into contact with the shaft31, and disposing a portion of the shaft31, which is extended toward the first axial side from the current shunter80, in the tubular portion72b. According to the present embodiment, the feeding tubular part71includes the large diameter portion71aand the small diameter portion71bthat is connected to the large diameter portion71aon the second axial side (+Y side) and that has a smaller inner diameter than the large diameter portion71a. When the feeding tubular part71is provided with the small diameter portion71bto allow a part of the feeding tubular part71to be reduced in diameter as described above, the oil O can be prevented from flowing excessively from the feeding tubular part71into the shaft31. This enables the oil O having flowed into the peripheral wall23bto be partially and easily fed to the bearing35from the feed hole73. This also enables preventing an excessive increase in the total amount of the oil O flowing from the first flow path portion93ainto the peripheral wall23b. Thus, an excessive increase in the amount of the oil O branching from the third flow path portion93cto the first flow path portion93acan be prevented, so that a decrease in the amount of the oil O branching from the third flow path portion93cto the second flow path portion93bcan be prevented. This enables the oil O to be suitably fed to the stator40from the second flow path portion93b. Additionally, the oil O can be stored in the large diameter portion71a, and the oil O stored in the large diameter portion71acan be sequentially and stably fed into the shaft31from the small diameter portion71b. According to the present embodiment, the connection portion71cconnecting the end portion of the large diameter portion71aon the second axial side (+Y side) and the end portion of the small diameter portion71bon the first axial side (−Y side) has the connection surface71fthat extends toward the second axial side from the inner peripheral surface of the large diameter portion71atoward the inner peripheral surface of the small diameter portion71b. This enables the oil O having flowed into the large diameter portion71ato easily flow into the small diameter portion71balong the connection surface71f. As a result, the oil O can be easily fed into the shaft31from the small diameter portion71b. As illustrated inFIG.5, a rotary electric machine210of a drive device200of the present embodiment includes a first flow path portion293athat opens in a radial central portion of a bottom wall223a. Unlike the first embodiment, a peripheral wall223bis not provided with a first wall portion23c. In the present embodiment, a second wall portion23dis connected to the bottom wall223a. A nozzle member270includes a feeding tubular part271in which a large diameter portion271ahas an inner diameter and an outer diameter that are each uniform throughout the axial direction. A flange portion272protrudes radially outward from an end portion of the large diameter portion271aon the first axial side (−Y side). In the present embodiment, the flange portion272is in a plate shape having a flat plate surface orthogonal to the axial direction. Unlike the flange portion72of the first embodiment, the flange portion272does not have a tubular portion72b. The flange portion272has a surface on the first axial side that is in contact with a surface of the bottom wall223aon the second axial side (+Y side). A current shunter280includes a base281that has a surface on the first axial side (−Y side), the surface being in contact with a surface of the flange portion272on the second axial side (+Y side). This allows the flange portion272to be in contact with both the bottom wall223aand the current shunter280in the axial direction. Thus, the current shunter280can more suitably suppress movement of the nozzle member270toward the second axial side. As described above, the current shunter280enables the nozzle member270to be more stably attached to a motor housing220in the present embodiment. Additionally, the nozzle member270can be prevented from rattling in the axial direction. Thus, a noise can be prevented from being generated by the rotary electric machine210. In the present embodiment, the current shunter280comes into contact with the surface of the flange portion2/2on the second axial side to be positioned axially with respect to the motor housing220. Unlike the first embodiment, no gap is provided between the flange portion272and the bottom wall223ain the axial direction. Unlike the flange portion72of the first embodiment, the flange portion272does not have a feed hole73. The flange portion272has a radially outer edge that is away from an inner peripheral surface of the peripheral wall223bin the radially inward direction. In the present embodiment, an open end portion231dof a third shaft portion231cof a shaft231is located on the second axial side (+Y side) from an end portion of the current shunter280on the first axial side (−Y side). The open end portion231dis disposed facing the flange portion272on the second axial side across a gap. Other configurations of the rotary electric machine210can be made similarly to other configurations of the rotary electric machine10of the first embodiment. Other configurations of the drive device200can be made similarly to other configurations of the drive device100of the first embodiment. The present invention is not limited to the above-described embodiment, and other structures and other methods may be employed within the scope of the technical idea of the present invention. The current shunter may be any type of current shunter as long as it is in electrical contact with a shaft and a housing of a rotary electric machine to allow a current flowing through the shaft to flow to the housing. The nozzle member may have any shape as long as it has a feeding tubular part and a flange portion. The nozzle member may include a tubular portion protruding toward the first axial side from a radially outer edge of an annular portion. The feeding tubular part may have an inner diameter that is uniform throughout the axial direction. The feeding tubular part may have an outer peripheral surface that is in contact with an inner peripheral surface of the shaft. Any kind of fluid may be used as the fluid fed into inside the shaft from the nozzle member. The fluid may be an insulating liquid or water. When the fluid is water, the surface of the stator may be subjected to an insulation treatment. The structure and method in which the fluid is fed into the peripheral wall are not particularly limited. Placement of the resolver and the bearing is not particularly limited. The rotary electric machine to which the present invention is applied is not limited to a motor, and may be a generator. The rotary electric machine is not limited in application. For example, the rotary electric machine may be mounted on a vehicle for uses other than rotating an axle, or may be mounted on an apparatus other than the vehicle. The rotary electric machine is not particularly limited in attitude when being used. The rotary electric machine may have the central axis extending in the vertical direction. The structures and methods described above in the present specification can be appropriately combined within a range consistent with each other. Features of the above-described preferred embodiments and the modifications thereof may be combined appropriately as long as no conflict arises. While preferred embodiments of the present disclosure have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the present disclosure. The scope of the present disclosure, therefore, is to be determined solely by the following claims. | 57,156 |
11863044 | DETAILED DESCRIPTION OF THE EMBODIMENTS In order to provide a clear understanding of the objects, features, and advantages of the embodiments, the following are detailed and complete descriptions to the technological solutions adopted in the embodiments. Obviously, the descriptions are part of the whole embodiments. The other embodiments which are not processed creatively by technicians of ordinary skills in the field are under the protection of this disclosure. The same is given with reference to the drawings and specific embodiments. It should be noted that non-conflicting embodiments in the disclosure and the features in the embodiments may be combined with each other without conflict. In the following description, numerous specific details are set forth in order to provide a full understanding of the disclosure. The disclosure may be practiced otherwise than as described herein. The following specific embodiments are not to limit the scope of the disclosure. Unless defined otherwise, all technical and scientific terms herein have the same meaning as used in the field of the art as generally understood. The terms used in the disclosure are to describe particular embodiments and are not intended to limit the disclosure. The disclosure, referencing the accompanying drawings, is illustrated by way of examples and not by way of limitation. It should be noted that references to “an” or “one” embodiment in this disclosure are not necessarily to the same embodiment, and such references mean “at least one.” In an ultrahigh speed wind motor structure in the related technology, two bearings need to be installed on two end cover parts, which will cause tolerance accumulation, so consistency of the coaxiality of two bearing chambers cannot be ensured, if the error is great, the two bearings do not work on the same axis, and the motor does not on the same axis when running at a high speed, the service life of the bearing cannot be ensured, finally the bearing is burnt, and the service life of the motor is shortened. As shown inFIG.1toFIG.5, this embodiment provides a motor for a high-speed electric hair dryer, so that the running speed allowed by the motor is higher, the motor noise is less, the service life of the motor is longer, the motor has better radiation and is more reliable. The motor for the high-speed electric hair dryer includes: an air duct shell1, a stator mechanism2, a rotor mechanism3, blades4and a concave motor bracket5, where the blades4and the stator mechanism2are all installed in the air duct shell1, and the stator mechanism2may be integrally molded by adopting injection in an iron core mold. The rotor mechanism3includes a rotating shaft31, a first bearing32, a second bearing33and a magnetic ring34, the magnetic ring34is fixedly sleeved on a middle of the rotating shaft31, two ends of the rotating shaft31are respectively inserted from inner holes of the first bearing32and the second bearing33, the stator mechanism2is fixed in the concave motor bracket5, installation holes51for installing the first bearing32and the second bearing33are formed in two ends of the concave motor bracket5, a semicircular machine cover6is arranged on the concave motor bracket5, the magnetic ring34is arranged in the stator mechanism2, the concave motor bracket5is fixed in the air duct shell1, one end of the rotating shaft31passing through the first bearing32is connected to the blades4, and the blades4are installed in the air duct shell1. Referring toFIG.5, the motor bracket5can include a side wall501, a first installing portion502, a second installing portion503, a first connection portion504and a second connection portion505. A cross-section of the side wall501is arced shaped, and an accommodate space506for accommodating the stator mechanism2is defined by the side wall501. the first installing portion502includes a first installation hole51afor installing the first bearing32, the second installing portion503includes a second installation hole51bfor installing the second bearing33. The first connection portion504is connected between one end of the side wall501and the first installing portion502, the second connection portion505is connected between the other end of the side wall501and the second installing portion503. The side wall501, the first installing portion502, the second installing portion503, the first connection portion504and the second connection portion505are integrated in one piece by a molding process. The semicircular machine cover6is configured to connect the side wall501to fix the stator mechanism2in the accommodate space506, and a cross-section of the semicircular machine cover6is arced shaped. As shown inFIG.2, the stator mechanism2includes a stator core21, a stator winding22, a conductive column23and a PCB board24, the stator winding22is fixedly arranged in the stator core21, the stator core21is fixed in the concave motor bracket5, and the PCB board24is connected to the stator winding22through the conductive column23. As shown inFIG.3, the stator mechanism3further includes a first positioning sleeve35and a second positioning sleeve36, which are all sleeved on the rotating shaft31, and the magnetic ring34is arranged between the first positioning sleeve35and the second positioning sleeve36; and a spring37is arranged between the first positioning sleeve35and the first spring32, and the spring37is sleeved on the rotating shaft31. The magnetic ring34and the positioning sleeves are adhered to the rotating shaft31through an adhesive, and the spring37pre-presses an inner ring of the bearing to ensure the reliability of the bearing during high-speed running. A cylindrical structure is formed by the concave motor bracket5and the semicircular machine cover6, and two ends of the cylindrical structure are all provided with ventilating ports, so the cylindrical structure belongs a semi-opening structure. When the motor runs, airflow enters the motor through the ventilating ports on the shell, so as to take away an internal temperature, meanwhile the airflow is radiated through an air duct, the motor has a double-cooling effect, so that a high temperature radiating temperature, generated by the motor during the running process, is completely solved. The shape of each blade4is an Archimedes spiral curved surface. A plurality of air guide plates are uniformly arranged in the air duct shell1, and preferably, 7 or 9 air guide plates may be provided. Beneficial effects: the two bearings can be installed at the two ends of the motor stator and the motor rotor through the concave motor bracket5, and molded by processing, in order to ensure the coaxiality of the two installation holes51. The rotating speed allowed by the motor is higher, the service life is longer, with a strong rigidity, the motor running is relatively stable, and the motor has less vibration and less noise. The motor main body is separated from the air duct, the motor shell is also separating, and the two ends of the motor shell are provided with the ventilating ports, so the motor shell belongs the semi-opening structure. When the motor runs, the airflow enters the motor through the ventilating ports on the shell, so as to take away the internal temperature, meanwhile, the airflow is radiated through the air duct, and the motor has a double-cooling effect, so that the high temperature radiating temperature, generated by the motor during the running process, is completely solved. In the description of the present disclosure, It is to be understood that, The terms “center”, “longitudinal”, “transverse”, “upper”, “lower”, “front”, “rear”, “left”, “right”, “vertical”, “horizontal”, “top”, “bottom”, “inner”, “outer”, “clockwise”, “counterclockwise”, and the like indicate azimuth or positional relationships based on the azimuth or positional relationships shown in the drawings, For purposes of convenience only of describing the present disclosure and simplifying the description, Rather than indicating or implying that the indicated device or element must have a particular orientation, be constructed and operated in a particular orientation, therefore, not to be construed as limiting the present disclosure; in addition, The terms “first” and “second” are used for descriptive purposes only, While not to be construed as indicating or implying relative importance or implicitly specifying the number of technical features indicated thereby, features defining “first,” “second,” and “second” may explicitly or implicitly include one or more of the described features. In the description of the present disclosure, “multiple” means two or more unless explicitly specified otherwise. Unless otherwise indicate D, “multiple” means two or more. In addition, the terms “mounted”, “disposed”, “provided”, “connected”, “connected”, and “socket” are to be construed broadly to mean, for example, a fixed connection, a detachable connection, or an integral construction; It may be a mechanical connection, or an electrical connection; The specific meaning of the above-mentioned terms in the present disclosure will be understood by those of ordinary skill in the art as the case may be, either directly, or indirectly, via an intermediate medium, or internal communication between two devices, elements, or components. The specific meanings of these terms in the present disclosure will be understood by those of ordinary skill in the art as the case may be. Finally, it should be noted that above embodiments are merely used for illustrating the technical solutions of the disclosure, rather than limiting the disclosure; though the disclosure is illustrated in detail with reference to the aforementioned embodiments, it should be understood by those of ordinary skill in the art that modifications may still be made on the technical solutions disclosed in the aforementioned respective embodiments, or equivalent substitutions may be made to a part of technical features thereof; and these modifications or substitutions do not make the essence of the corresponding technical solutions depart from the spirit and scope of the technical solutions of the respective embodiments of the disclosure. | 10,202 |
11863045 | MODE(S) FOR CARRYING OUT INVENTION The following details an embodiment of the present invention with reference to the drawings. However, the present invention is not limited to the embodiment, but includes various modifications and applications belonging to technical conception of the present invention. The following briefly describes configuration of a steering device as an example of device to which the present invention is applied, with reference toFIG.1, prior to description of the embodiment of the present invention. First, the following describes a steering device for steering front wheels of an automotive vehicle. Steering device1is configured as shown inFIG.1. A steering shaft2is connected to a steering wheel not shown, and includes a lower end formed with a pinion not shown, wherein the pinion is in mesh with a rack not shown, wherein the rack extends in a vehicle body lateral direction. The rack includes ends linked to respective tie rods3for steering the front wheels leftward and rightward, and is housed by a rack housing4. A rubber boot5is provided between rack housing4and each tie rod3. An electric power steering device6is provided for producing an assist torque while the steering wheel is being turned. Specifically, electric power steering device6includes a torque sensor7, an electric motor section8, and an electronic control section or unit (ECU)9, wherein torque sensor7is structured to sense a direction of rotation of steering shaft2, and a rotating torque applied to steering shaft2, wherein electric motor section8is structured to apply a steering assist force to the rack via a gear10depending on a sensed value from torque sensor7, and wherein electronic control section9is configured to control an electric motor arranged in electric motor section8. Electric motor section8of electric power steering device6is connected to gear10by bolts not shown at three spots of an outer peripheral part of an output shaft side of electric motor section8. Electronic control section9is arranged at a side of electric motor section8opposite to an output shaft of electric motor section8. Electric power steering device6operates as follows. As the steering wheel is turned to rotate steering shaft2in one direction, torque sensor7then senses the direction of rotation of steering shaft2, and the rotating torque applied to steering shaft2. A control circuit part calculates a quantity of operation of the electric motor, based on a sensed value from torque sensor7. Power switching elements of a power conversion circuit part are controlled to drive the electric motor based on the calculated quantity of operation, so that the output shaft of the electric motor is rotated to drive the steering shaft2in the same direction as the direction of operation of the steering wheel. The rotation of the output shaft of the electric motor is transferred to the rack via the pinion and gear10, thereby steering the automotive vehicle. Further description is omitted because its configuration and operation are well known. As discussed above, in the electric power steering device, fixing bolt69for fixing the connector assembly64and the mounting board67of control circuit part63to fixing part68is required to be a fixing bolt having a locking function for preventing loosening. However, a fixing bolt having a locking function is expensive per unit. Accordingly, the use of such a fixing bolt inevitably raises the unit price of a product, and thereby causes a problem of adversely affecting the competitiveness of the product. In view of the foregoing background, according to the present embodiment, an electric power steering device is proposed which is configured as follows. According to the present embodiment: an electric power steering device includes: a motor housing including an end face part opposite to an output part of a rotating shaft of an electric motor; an electronic control part arranged at the end face part of the motor housing; a connector assembly arranged on a side of the electronic control part opposite to a side of the electronic control part that faces the end face part of the motor housing; an annular seal accommodating portion filled with a liquid sealing agent; a fixing member structured to fix the connector assembly to a fixing part of the end face part of the motor housing; a liquid sealing agent guiding passage formed with the annular seal accommodating portion, and structured to allow the liquid sealing agent, which is filled in the annular seal accommodating portion, to flow to the fixing member; and a metal cover structured to cover the electronic control part from outside, wherein the metal cover includes: a bottom part including: an exposure hole through which an external terminal forming part is exposed to outside; and an annular reinforcing projecting portion formed at an edge of the exposure hole, and accommodated in the annular seal accommodating portion; and a lateral peripheral part angled from the bottom part, and forming an opening through which the lateral peripheral part is fixed to the end face part of the motor housing. According to the foregoing, the feature that the annular seal accommodating portion is formed with the guiding passage for allowing the liquid sealing agent to flow toward the fixing member, serves to fix the fixing member and the connector assembly to each other by the liquid sealing agent. This serves to suppress loosening of the fixing member without any fixing bolt having a locking function and being expensive per unit, and thereby lower the unit price of the product. Furthermore, the liquid sealing agent filled between the annular reinforcing projecting portion and the annular seal accommodating portion ensures liquid tightness. The following details specific configuration of the electric power steering device according to the embodiment of the present invention with reference toFIGS.2to13. FIG.2shows whole configuration of the electric power steering device according to the present embodiment.FIG.3shows components of the electric power steering device shown inFIG.2in disassembled state as viewed diagonally.FIGS.4to9show states of assembling when the components are assembled in an assembling order. FIG.10is a sectional view of the electronic control section of the electric power steering device.FIG.11is a sectional view of the metal cover.FIG.12shows the metal cover as viewed diagonally from above.FIG.13is a sectional view of the electronic control section of the electric power steering device when the metal cover is attached. The following description refers to these drawings as appropriate. As shown inFIG.2, the electric power steering device includes electric motor section8and electronic control section9. Electric motor section8includes a motor housing11and an electric motor not shown. Motor housing11includes a cylindrical part made of an aluminum-based metal such as aluminum or an aluminum alloy. The electric motor is housed in motor housing11. Electronic control section9includes a metal cover12, and an electronic control assembly not shown housed in metal cover12. Metal cover12is made of an aluminum-based metal such as aluminum or an aluminum alloy, or an iron-based metal, and is arranged at a side of motor housing11opposite to the output shaft in the axial direction. Motor housing11and metal cover12are fixed to each other by swaging in their fixing regions each extending circumferentially in the periphery of the end face part facing each other. Metal cover12includes an accommodation space inside thereof, which accommodates the electronic control assembly. The electronic control assembly includes a power supply circuit part for supplying electric power as required, and a power conversion circuit part having power switching elements such as MOSFETs or IGBTs for driving and controlling the electric motor of electric motor section8, and a control circuit part for controlling the power switching elements. Output terminals of the power switching elements and input terminals of a coil of the electric motor are connected electrically via a bus bar. At an end face of metal cover12opposite to motor housing11, a part of a connector assembly13is exposed through an exposure hole42formed in metal cover12. Connector assembly13is fixed by fixing members to fixing parts formed in an end face of motor housing11, wherein the fixing members are fixing bolts. The part of connector assembly13includes an external terminal forming part13A for power supply, an external terminal forming part13B for sensors, and an external terminal forming part13C for sending a state of control to external devices. The electronic control assembly housed in metal cover12is supplied with electric power from a power supply via the external terminal forming part13A made of synthetic resin, and is supplied with sensing signals indicative of operating states from sensors and others via the external terminal forming part13B, and sends a present control state of the electric power steering device via the external terminal forming part13C. FIG.3shows electric power steering device6in an exploded perspective view. Inside of motor housing11, a side yoke not shown is fitted, wherein the side yoke has an annular shape and is made of iron. The electric motor not shown is mounted inside of the side yoke. The electric motor includes an output part14structured to apply a steering assist force to the rack via the gear. Description of specific configuration of the electric motor is omitted because it is well known. Motor housing11is made of an aluminum alloy, thereby serving as a heat sink member for dissipating heat to outside atmosphere, wherein the heat is generated by the electric motor, the power conversion circuit part and the power supply circuit part described below. The electric motor and motor housing11form the electric motor section8. Electronic control part EC is attached to an end face part15of motor housing11opposite to the output part14of the electric motor section8. Electronic control part EC is composed of power conversion circuit part16, power supply circuit part17, control circuit part18, and connector assembly13. The end face part15of motor housing11is formed integrally with motor housing11, but may be formed separately from motor housing11and bolted or welded to motor housing11. Power conversion circuit part16, power supply circuit part17, and control circuit part18are configured to be redundant and form a main electronic control system and an auxiliary electronic control system. Normally, the main electronic control system is employed to drive and control the electric motor, and when an abnormality or failure occurs in the main electronic control system, the control is switched from the main electronic control system to the auxiliary electronic control system so that the auxiliary electronic control system drives and controls the electric motor. Accordingly, as detailed below, heat of the main electronic control system is normally transferred to motor housing11. When the main electronic control system is failed or abnormal, operation of the main electronic control system is stopped and the auxiliary electronic control system is activated so that heat of the auxiliary electronic control system is transferred to motor housing11. However, although not adopted by the present embodiment, there is an alternative configuration that both of the main and auxiliary electronic control systems are simultaneously employed to form a normal electronic control system, and when one of the main and auxiliary electronic control systems is failed or abnormal, only the other electronic control system is employed to drive and control the electric motor with half of full performance. This ensures a power steering function, although the performance of the electric motor is only half. Accordingly, the heat of the main electronic control system and the auxiliary electronic control system is normally transferred to motor housing11. Power conversion circuit part16, power supply circuit part17, control circuit part18, and connector assembly13, which form the electronic control part EC, are arranged in this order away from end face part15of motor housing11. Control circuit part18is configured to generate control signals for driving the switching elements of power conversion circuit part16, and includes a microcomputer and a peripheral circuit. Power supply circuit part17is configured to supply electric power to drive the control circuit part18, and supply electric power to power conversion circuit part16, and includes capacitors, coils, switching elements, and others. Power conversion circuit part16is configured to regulate electric power flowing through the coil of the electric motor, and includes switching elements and others forming three-phase upper and lower arms. In electronic control part EC, power conversion circuit part16and power supply circuit part17generate more quantities of heat than others. The generated heat of power conversion circuit part16and power supply circuit part17is dissipated via motor housing11made of the aluminum alloy. This configuration is detailed below with reference toFIGS.4to9. Connector assembly13, which is made of synthetic resin, is arranged between control circuit part18and metal cover12, and is connected to a vehicle battery (power supply) and external control devices not shown. Connector assembly13is also connected to power conversion circuit part16, power supply circuit part17, and control circuit part18. Metal cover12functions to house and seal liquid-tightly the power conversion circuit part16, power supply circuit part17, and control circuit part18. In the present embodiment, metal cover12is fixed to motor housing11by swaging. Metal cover12includes a lateral peripheral part43, and a bottom part44formed by bending from one end of lateral peripheral part43. Bottom part44is formed with exposure hole42through which external terminal forming parts13A,13B,13C of connector assembly13are exposed to the outside. Metal cover12includes an open end37opposite to bottom part44, wherein open end37is engaged with end face part15of motor housing11. The following describes configuration of the components and a process of assembling the components with reference toFIGS.4to9.FIG.4shows an exterior view of motor housing11, andFIG.5shows its axial sectional view. As shown inFIGS.4and5, motor housing11is cylindrically shaped and includes a lateral peripheral part11A, end face part15, and a cover19. The end face part15closes a first end of lateral peripheral part11A, whereas the cover19closes a second end of lateral peripheral part11A. In the present embodiment, lateral peripheral part11A and end face part15are formed integrally such that motor housing11has a cylindrical shape having a bottom. The cover19serves a covering function to close the second end of lateral peripheral part11A after the electric motor is mounted inside the lateral peripheral part11A. The peripheral surface of end face part15is formed with an annular step portion35whose diameter is reduced inward in the radial direction, wherein open end37of metal cover12shown inFIG.9is engaged with step portion35. The fixation between the side wall of end face part15and open end37of the metal cover12is implemented by so-called swaging fixation. As shown inFIG.5, a stator21is fitted inside the lateral peripheral part11A of motor housing11, wherein stator21is formed by winding the coil20around an iron core. A rotor22is rotatably mounted inside the stator21, wherein a permanent magnet is embedded in rotor22. A rotating shaft23is fixed to rotor22. One end of rotating shaft23forms the output part14, whereas the other end of rotating shaft23forms a rotation-sensing target part24serving as a target for sensing the rotational phase and speed of rotating shaft23. Rotation-sensing target part24is provided with the permanent magnet, extending through a through hole25formed in end face part15, and projecting to the outside. The rotational phase and speed of rotating shaft23is sensed by a magnetic sensor such as a GMR element or the like not shown. Referring back toFIG.4, the surface of end face part15opposite to the output part14of rotating shaft23is formed with a power conversion heat dissipation region15A for power conversion circuit part16(seeFIG.3) and a power supply heat dissipation region15B for power supply circuit part17(seeFIG.3). Four corners of end face part15are each formed integrally with a board/connector-fixing projecting part26extending perpendicularly from end face part15. Each board/connector-fixing projecting part26is formed with a threaded hole26S inside. Board/connector-fixing projecting parts26are structured to fix control circuit part18described below and connector assembly13. Each board-fixing projecting part26projecting from power conversion heat dissipation region15A described below is formed with a board-receiving part27having the same height as power supply heat dissipation region15B described below in the axial direction. Each board-receiving part27is structured to mount and fix a glass epoxy board31of power supply circuit part17described below. The flat area forming the end face part15and extending in the radial direction and perpendicular to rotating shaft23is divided into two regions, namely, power conversion heat dissipation region15A and power supply heat dissipation region15B. Power conversion circuit part16, which is composed of switching elements such as MOSFETs, is attached to power conversion heat dissipation region15A. Power supply circuit part17is attached to power supply heat dissipation region15B. In the present embodiment, the area of power conversion heat dissipation region15A is set larger than that of power supply heat dissipation region15B. This serves to ensure more space for mounting the power conversion circuit part16, because the redundant system is employed. A step is provided between power conversion heat dissipation region15A and power supply heat dissipation region15B such that power conversion heat dissipation region15A and power supply heat dissipation region15B have different heights in the axial direction (the direction in which rotating shaft23extends). Namely, power supply heat dissipation region15B is formed with an outward step away with respect to power conversion heat dissipation region15A in the axial direction of rotating shaft23of the electric motor. This step is set to have a height enough to prevent interference between power conversion circuit part16and power supply circuit part17when power supply circuit part17is assembled after power conversion circuit part16is assembled. Power conversion heat dissipation region15A is formed with three heat dissipation projecting parts28. Heat dissipation projecting parts28are configured to mount power conversion circuit part16thereon, wherein power conversion circuit part16is configured to be redundant as described below. Each heat dissipation projecting part28projects away from the electric motor in the direction of rotating shaft23of the electric motor. Power supply heat dissipation region15B is generally flat and is configured to mount power supply circuit part17thereon, where power supply circuit part17is described below. Accordingly, each heat dissipation projecting part28serves as a heat dissipation portion to transfer heat from power conversion circuit part16to end face part15, whereas power supply heat dissipation region15B serves as a heat dissipation portion to transfer heat from power supply circuit part17to end face part15. Each heat dissipation projecting part28may be omitted so that power conversion heat dissipation region15A serves as a heat dissipation portion to transfer heat from power conversion circuit part16to end face part15. However, in the present embodiment, a metal board of power conversion circuit part16is securely fixed to heat dissipation projecting part28by friction stir welding. At end face part15of motor housing11according to the present embodiment described above, the axial size can be made compact because there is no heat sink member. Moreover, since motor housing11has a sufficient thermal capacity, the heat generated in power supply circuit part17and power conversion circuit part16can be dissipated to the outside effectively. FIG.6shows a state where power conversion circuit part16is placed on heat dissipation projecting parts28(seeFIG.4). As shown inFIG.6, power conversion circuit part16, which is configured to be redundant, is placed on heat dissipation projecting parts28(seeFIG.4) formed in power conversion heat dissipation region15A. The switching elements constituting the power conversion circuit part16are mounted on the metal board which is made of an aluminum-based metal in this example, promoting heat dissipation. The metal board is welded to heat dissipation projecting part28by friction stir welding. The metal board may be replaced with a glass epoxy board. In this case, heat dissipation can be enhanced by setting the thickness of the glass epoxy board as thin as possible. In this way, the metal board is securely fixed to heat dissipation projecting part28(seeFIG.4), to allow generated heat of the switching elements to be transferred to heat dissipation projecting part28(seeFIG.4) effectively. The heat transferred to heat dissipation projecting part28(seeFIG.4) is dissipated to power conversion heat dissipation region15A, and then to lateral peripheral part11A of motor housing11, and finally to the outside. As described above, power conversion circuit part16is prevented from interfering with power supply circuit part17described below, because the height of power conversion circuit part16is shorter than that of power supply heat dissipation region15B in the axial direction. As described above, power conversion circuit part16is placed on heat dissipation projecting part28formed in the power conversion heat dissipation region15A. Therefore, the heat generated by the switching elements of power conversion circuit part16can be efficiently transferred to heat dissipation projecting part28. Furthermore, the heat transferred to heat dissipation projecting part28is dissipated to power conversion heat dissipation region15A, and transferred to lateral peripheral part11A of motor housing11, and dissipated to the outside. FIG.7shows a state where power supply circuit part17is placed over power conversion circuit part16. As shown inFIG.7, power supply heat dissipation region15B is covered by power supply circuit part17. Capacitors29and coils30of power supply circuit part17are placed on glass epoxy board31. Power supply circuit part17is also configured to be redundant and include power supply circuits each of which is composed of capacitors29and coil30and arranged symmetrically with each other as shown inFIG.7. Electric elements such as capacitors other than the switching elements of power conversion circuit part16are mounted on glass epoxy board31. The surface of glass epoxy board31facing the power supply heat dissipation region15B (seeFIG.6) is fixed to end face part15, in contact with power supply heat dissipation region15B. As shown inFIG.7, this fixing is implemented by putting a fixing bolt not shown into a threaded hole27S formed in each board-receiving part27of board-fixing projecting part26, and also putting a fixing bolt not shown into a threaded hole formed in power supply heat dissipation region15B (seeFIG.6). The configuration that power supply circuit part17is based on glass epoxy board31allows the components of power supply circuit part17to be mounted on both sides of the power supply circuit part17. The surface of glass epoxy board31facing the power supply heat dissipation region15B (seeFIG.6) is provided with a rotational phase and speed sensing part composed of the GMR element and a sensing circuit not shown. This sensor is configured to sense the rotational phase and speed of rotating shaft23(seeFIG.5) in cooperation with rotation-sensing target part24(seeFIG.5) that is provided at rotating shaft23. The configuration that glass epoxy board31is fixed to power supply heat dissipation region15B (seeFIG.6), in contact with power supply heat dissipation region15B as described above, allows the generated heat of power supply circuit part17to be transferred to power supply heat dissipation region15B effectively. The heat transferred to power supply heat dissipation region15B (seeFIG.6) is transferred and spread into lateral peripheral part11A of motor housing11, and then dissipated to the outside. In order to enhance the thermal conductivity, an adhesive agent or dissipation grease or dissipation sheet having a high thermal conductivity may be disposed between glass epoxy board31and power supply heat dissipation region15B (seeFIG.6). As described above, power supply circuit part17is placed on power supply heat dissipation region15B. The surface of glass epoxy board31facing the power supply heat dissipation region15B, on which the circuit elements of power supply circuit part17are mounted, is fixed to end face part15, in contact with power supply heat dissipation region15B. Therefore, the heat generated in power supply circuit part17can be effectively transferred to power supply heat dissipation region15B. The heat transferred to power supply heat dissipation region15B is transferred and spread into lateral peripheral part11A of motor housing11, and dissipated to the outside. FIG.8shows a state where control circuit part18is placed over the power supply circuit part17. As shown inFIG.8, control circuit part18is arranged over power supply circuit part17. Microcomputers32and peripheral circuits33constituting the control circuit part18are placed on a glass epoxy board34as a mounting board. Control circuit part18is also configured to be redundant and include control circuits each of which is composed of microcomputer32and peripheral circuits33and arranged symmetrically with each other as shown inFIG.8. Microcomputers32and peripheral circuits33may be placed on the surface of glass epoxy board34facing the power supply circuit part17. As shown inFIG.8, glass epoxy board34is fixed by fixing bolts not shown through the threaded holes formed in the top portions of board-fixing projecting parts26(seeFIG.7), wherein glass epoxy board34is sandwiched between board-fixing projecting parts26and connector assembly13. The space between glass epoxy board31of power supply circuit part17(seeFIG.7) and glass epoxy board34of control circuit part18is used for arrangement of capacitors29and coils30of power supply circuit part17shown inFIG.7. Next,FIG.9shows a state in which connector assembly13is placed from above the control circuit part18. As shown inFIG.9, connector assembly13is placed over the control circuit part18. Then, connector assembly13is fixed by putting fixing bolts36into threaded holes26S each formed in the top of board-fixing projecting part26, sandwiching the glass epoxy board34of control circuit part18. In this state, as shown inFIG.3, connector assembly13is electrically connected to power conversion circuit part16, power supply circuit part17, and control circuit part18. Connector assembly13is formed with an annular seal accommodating portion45around external terminal forming parts13A,13B,13C, wherein annular seal accommodating portion45is recess-shaped in its axial sectional view. Annular seal accommodating portion45is filled with a liquid sealing agent, where an annular reinforcing projecting portion46is formed at exposure hole42of metal cover12as described below and accommodated in annular seal accommodating portion45. Accordingly, annular seal accommodating portion45, the liquid sealing agent, and annular reinforcing projecting portion46ensures liquid tightness between metal cover12and connector assembly13. Furthermore, open end37of metal cover12is engaged with step portion35of motor housing11, and is fixed by swaging to motor housing11in the fixing region extending circumferentially. As described above, the annular step portion35of the motor housing, which is formed in the peripheral surface of end face part15, and the open end37of metal cover12are engaged with each other by so-called spigot engagement or spigot fitting. FIG.10shows an enlarged sectional view of the electronic control section and its surroundings of the electric power steering device according to the present embodiment. At end face part15of motor housing11, power conversion circuit part16, power supply circuit part17, and control circuit part18, which constitute the electronic control part, are layered with each other in this order away from end face part15. Connector assembly13is arranged on the side of the control circuit part18opposite to the side facing the power supply circuit part17. However, the order of arrangement of power conversion circuit part16, power supply circuit part17, and control circuit part18may be arbitrarily selected, if the electronic control part is arranged between connector assembly13and end face part15. A plurality of swaging fixation portions not shown are formed in the side surface of step portion35of end face part15to which opening end37of metal cover12is fixed. The swaging fixation portion includes a swaging recess that is a swaging groove or recess formed in a fixation wall extending in the axial direction from the annular step portion35toward connector assembly13, wherein step portion35is formed in end face part15of motor housing11. Metal cover12is fixed by swaging the wall of metal cover12into the swaging recess with a pressing tool for pressing and plastically deforming the wall of metal cover12. Furthermore, the opening end37of metal cover12, and the wall surface extending from the annular step portion35to the fixation wall form an annular space G, which is filled with a liquid sealing agent for liquid tightness without any gap. This forms a seal region for liquid tightness between the end face part15and the open end37of the metal cover12, and prevents entrance of moisture and others by this seal region. This serves to prevent entrance of moisture and others into the swaging fixation portion, and thereby suppress corrosion of the swaging fixation portion, and enhance the mechanical reliability. This further serves to prevent entrance of moisture and others into electronic control section9, and thereby enhance the electrical reliability as well. In the present embodiment, swaging recesses each having a predetermined length are arranged at places as required (three places). The wall of metal cover12pressed in each swaging recess serves to suppress movement of metal cover12in the rotational direction and in the axial direction of rotating shaft23with respect to motor housing11. In the region of connector assembly13surrounding the external terminal forming parts13A,13B,13C, annular seal accommodating portion45is formed continuously. Annular seal accommodating portion45is formed in a groove shape in the surface47facing the bottom part44in a state where connector assembly13faces bottom part44of metal cover12. The cross section of the groove may have a rectangular shape, a semicircular shape, or an arc shape. Annular seal accommodating portion45is an annular groove having an opening facing away from the end face part15of motor housing11in the axial direction of rotating shaft23. Therefore, the annular groove is formed along the axis of rotating shaft23. Furthermore, annular seal accommodating portion45is filled with a liquid sealing agent48for ensuring liquid tightness, when metal cover12is assembled. Furthermore, the depth of the groove forming the annular seal accommodating portion45is determined by the length of annular reinforcing projecting portion46formed at the edge of exposure hole42of metal cover12as described below. As shown inFIG.9, metal cover12includes a lateral peripheral part43and a bottom part44, wherein bottom part44is formed by bending from one end of lateral peripheral part43, and wherein bottom part44is formed with exposure hole42through which external terminal forming parts13A,13B, and13C of connector assembly13are exposed to the outside. Furthermore, opening end37is formed on the side opposite to bottom part44, and is engaged with the end face part15of motor housing11. Exposure hole42is formed near the center of bottom part44for exposing the external terminal forming parts13A,13B, and13C of connector assembly13to the outside. Furthermore, at the peripheral edge of exposure hole42(in other words, at the inner periphery of bottom part44forming the exposure hole42), annular reinforcing projecting portion46(seeFIG.10) is formed by bending toward the inside of metal cover12. Annular reinforcing projecting portion46may be formed by drawing the metal cover12. Annular reinforcing projecting portion46has a shape extending in the axial direction of rotating shaft23with metal cover12mounted to end face part15, so that annular reinforcing projecting portion46corresponds in position to the groove of annular seal accommodating portion45. Namely, annular reinforcing projecting portion46is substantially parallel to the side walls forming the groove of annular seal accommodating portion45. The axial length of annular reinforcing projecting portion46is determined such that annular reinforcing projecting portion46is out of intimate contact with the bottom face of the groove forming the annular seal accommodating portion45. This ensures a margin for axial dimensional control of annular reinforcing projecting portion46and annular seal accommodating portion45, and further allows a liquid sealing agent to be interposed between annular reinforcing projecting portion46and the inner wall surfaces of annular seal accommodating portion45, ensuring a function of liquid tightness. Before swaging and fixing the metal cover12to the end face part15of motor housing11, the inside of annular seal accommodating portion45formed in connector assembly13is filled with liquid sealing agent48. Before being cured, liquid sealing agent48has fluidity and can flow relatively easily due to gravity or its own weight. When metal cover12is assembled so as to cover the electronic control section9, annular reinforcing projecting portion46formed in bottom part44of metal cover12is inserted and accommodated in annular seal accommodating portion45formed in the connector assembly13. In this state, by pressing the swaging tool onto the swaging recess of metal cover12, the wall of metal cover12is plastically deformed and fixed by swaging to the swaging fixation portion. During this operation, an external force due to swaging may act on metal cover12to deform exposure hole42and its vicinity of metal cover12. However, in the present embodiment, the formation of annular reinforcing projecting portion46serves to enhance the mechanical strength of exposure hole42and its vicinity, and suppress deformation of exposure hole42and its vicinity and deformation of metal cover12. As shown inFIG.10, annular reinforcing projecting portion46is accommodated in annular seal accommodating portion45with a predetermined gap to each inner wall surface of annular seal accommodating portion45, wherein the gap is filled with liquid sealing agent48. This ensures sufficient liquid tightness in this place. The outer surface of liquid sealing agent48filled inside the annular reinforcing projecting portion46exceeds the surface of bottom part44of metal cover12by a thickness (t). Accordingly, no dent is formed between liquid sealing agent48and the wall surface of annular reinforcing projecting portion46housed in annular seal accommodating portion45, allowing moisture to flow from the outside toward the bottom part44, and thereby suppress corrosion of annular reinforcing projecting portion46. If it is exposed to salt water or the like in particular, it promotes corrosion. However, this configuration serves to suppress such corrosion. The further configuration that annular reinforcing projecting portion46is formed at the peripheral edge of exposure hole42formed in bottom part44of metal cover12, serves to increase the mechanical strength of the periphery of exposure hole42. Accordingly, even if an external force is applied to metal cover12in the process of assembling the metal cover or in the process of actual use, it is possible to suppress deformation of the vicinity of the peripheral edge of exposure hole42and deformation of metal cover12. For example, as described above, when metal cover12is fixed by swaging to the end face part15of motor housing11, the swaging operation may cause an external force to act on exposure hole42of metal cover12so as to deform the peripheral region of exposure hole42. However, the formation of annular reinforcing projecting portion46serves to suppress deformation of the peripheral region of exposure hole42. In this way, the feature that annular reinforcing projecting portion46is formed at the edge of exposure hole42formed in bottom part44of metal cover12, and accommodated in annular seal accommodating portion45formed around external terminal forming parts13A,13B,13C of connector assembly13, serves to enhance the mechanical strength of the surrounding region of exposure hole42, and thereby suppress deformation of metal cover12. In the present embodiment, annular reinforcing projecting portion46is formed by drawing the metal cover12, but annular reinforcing projecting portion46may be provided by preparing a separate reinforcing projecting portion, and fixing it to bottom part44of metal cover12by welding, forging, etc. Furthermore, although annular reinforcing projecting portion46is formed simultaneously with formation of exposure hole42, it is also possible to form an annular reinforcing projecting portion, which has a shape corresponding to annular seal accommodating portion45, in the peripheral region around exposure hole42. The following describes configuration of liquid sealing agent guiding passages with reference toFIGS.11and12.FIG.11shows a view from a side facing the connector assembly, andFIG.12shows a view from diagonally above, when the metal cover is removed.FIG.11shows a state before the liquid sealing agent is filled. InFIGS.11and12, the electronic control unit is arranged on end face part15of motor housing11, where the power conversion circuit part not shown, power supply circuit part17, control circuit part18, and connector assembly13are arranged in this order away from end face part15. Glass epoxy board31of power supply circuit part17is fixed to the step portion of end face part15by the fixing bolts. Furthermore, glass epoxy board34of control circuit part18and connector assembly13are jointly fastened and fixed by fixing bolts36to board-fixing projecting parts26each projecting from end face part15. Connector assembly13is formed with annular seal accommodating portion45around the external terminal forming parts of connector assembly13, wherein the inside of annular seal accommodating portion45is filled with liquid sealing agent48. Annular seal accommodating portion45is formed to have a generally continuous groove-shaped recess45G in order to store liquid sealing agent48. For example, annular recess45G is defined inside an annular wall that is raised from a surface of a peripheral portion around a screwing region50in which an insertion hole is formed for insertion of fixing bolt36. As shown inFIG.9, external terminal forming parts13A to13C of connector assembly13are exposed to the outside through exposure hole42formed in bottom part44of metal cover12. Namely, exposure hole42for exposing the external terminal forming parts13A to13C of connector assembly13to the outside is formed in bottom part44formed on one end side of lateral peripheral part43of metal cover12. Exposure hole42is formed by punching the bottom part44, and is surrounded by a peripheral portion formed with annular reinforcing projecting portion46angled inside the metal cover.FIG.13shows the state before the liquid sealing agent is filled. Annular reinforcing projecting portion46formed at the peripheral edge of exposure hole42has the same shape as annular seal accommodating portion45so as to be accommodated in annular recess45G of annular seal accommodating portion45. Annular seal accommodating portion45, liquid sealing agent48, and annular reinforcing projecting portion46ensure a function of liquid tightness. Annular recess45G forming the annular seal accommodating portion45has wall parts each of which is adjacent to fixing bolt36and is partly removed to form a liquid sealing agent guiding passage49. Each liquid sealing agent guiding passage49is provided in correspondence to a respective one of fixing bolts36. It is desirable that liquid sealing agent guiding passage49is provided in the wall part of annular recess45G closest to the peripheral surface of the head part of fixing bolt36. As described above, liquid sealing agent guiding passage49has a function of fluidly connecting the vicinity of the head part of fixing bolt36to annular recess45G forming the annular seal accommodating portion45. Fixing bolt36is inserted in the insertion hole formed in a corresponding one of screwing regions50formed at four corners of face47facing the connector assembly13, and is screwed to board-fixing projecting part26. As shown inFIG.10, as viewed with the electronic control unit on the upper side, screwing region50is located below annular recess45G of annular seal accommodating portion45, and in other words, is located closer to end face part15of motor housing11than annular recess45G of annular seal accommodating portion45. Accordingly, due to gravity and its own weight, liquid sealing agent48filled in annular recess45G forming the annular seal accommodating portion45flows out through the liquid sealing agent guiding passage49and reaches the screwing region50. Then, as shown inFIGS.10and12, liquid sealing agent48has fluidity before hardening treatment, and therefore covers the head part of fixing bolt36entirely or partly. After hardened by a subsequent hardening treatment, liquid sealing agent48forms a locking part51for fixing the whole of the surface of screwing region50and the head part of fixing bolt36, or fixing parts thereof to each other. Since liquid sealing agent48has adhesiveness after cured, screwing region50and liquid sealing agent48serve to prevent loosening by preventing rotation of fixing bolt36even if fixing bolt36is about to rotate in a loosening direction due to vibration or the like. In the embodiment, liquid sealing agent guiding passage49is formed by partly removing the wall part of annular recess45G forming the annular seal accommodating portion45. However, liquid sealing agent guiding passage49may be implemented by forming a through hole in the wall part of annular recess45G, and allowing the liquid sealing agent48to flow through the through hole to the vicinity of fixing bolt36. Liquid sealing agent48for liquid tightness, which is filled between metal cover12and connector assembly13, is implemented by an adhesive synthetic resin. In the present embodiment, liquid sealing agent48is implemented by a silicone-rubber-based elastic adhesive that is cured at room temperature or is cured by heating. This silicone-rubber-based elastic adhesive has a property of absorbing stress such as external vibration and impact, and is less subject to stress concentration on the adhesive interface. When an electric power steering device is subject to vibration, impact, etc., peeling may occur at the adhesive interface and the function of liquid tightness may be lost. However, the use of a silicone-rubber-based elastic adhesive serves to reduce the risk of loss of the function of liquid tightness. Furthermore, in the present embodiment, the feature that the sealing is implemented by adhesive liquid sealing agent48allows the use of a liquid-tightening O-ring to be omitted. Liquid sealing agent48may be implemented by a liquid gasket (FIPG: Formed In Place Gasket) having an adhesive function, and may be implemented by a material that can be cured at room temperature or by heating. By this configuration, it is possible to prevent moisture from entering the inside through the vicinity of abutment between annular reinforcing projecting portion46at exposure hole42formed in bottom part44of metal cover12and annular seal accommodating portion45of connector assembly13. Although the fixing members are implemented by bolts in the embodiment, the fixing members are not limited to bolts, but may be implemented by rivets, of course. As described above, the present invention is characterized by including: a motor housing structured to house an electric motor, wherein the motor housing includes an end face part opposite to an output part of a rotating shaft of the electric motor, and wherein the electric motor is structured to drive a controlled object of a mechanical system; an electronic control part arranged at the end face part of the motor housing; a connector assembly arranged on a side of the electronic control part opposite to a side of the electronic control part that faces the end face part of the motor housing; an annular seal accommodating portion formed around an external terminal forming part of the connector assembly, and filled with a liquid sealing agent; a fixing member structured to fix the connector assembly to a fixing part of the end face part of the motor housing; a liquid sealing agent guiding passage formed with the annular seal accommodating portion, and structured to allow the liquid sealing agent, which is filled in the annular seal accommodating portion, to flow to the fixing member; and a metal cover structured to cover the electronic control part from outside, wherein the metal cover includes: a bottom part including: an exposure hole through which the external terminal forming part is exposed to outside; and an annular reinforcing projecting portion formed at an edge of the exposure hole, and accommodated in the annular seal accommodating portion; and a lateral peripheral part angled from the bottom part, and forming an opening through which the lateral peripheral part is fixed to the end face part of the motor housing. According to the foregoing, the feature that the annular seal accommodating portion is formed with the guiding passage for allowing the liquid sealing agent to flow toward the fixing member, serves to fix the fixing member and the connector assembly to each other by the liquid sealing agent. This serves to suppress loosening of the fixing member without any fixing bolt having a locking function and being expensive per unit, and thereby lower the unit price of the product. Furthermore, the liquid sealing agent filled between the annular reinforcing projecting portion and the annular seal accommodating portion ensures liquid tightness. The present invention is not limited to the embodiment described above, but includes various modified embodiments. The described embodiment is detailed merely for easy understanding of the present invention, and the present invention is not limited to a form including all of the features described above, for example. Part of features of one of the embodiments may be replaced with features of another one of the embodiments. Features of one of the embodiments may be additionally provided with features of another one of the embodiments. Part of features of each of the embodiments may be additionally provided with other features or removed or replaced. DESCRIPTION OF SYMBOLS 6. . . Electric Power Steering Device8. . . Electric Motor Section9. . . Electronic Control Section11. . . Motor Housing12. . . Metal Cover13. . . Connector Assembly14. . . Output Part15. . . End Face Part15A . . . Power Conversion Heat Dissipation Region15B . . . Power Supply Heat Dissipation Region16. . . Power Conversion Circuit Part17. . . Power Supply Circuit Part18. . . Control Circuit Part19. . . End Face Part20. . . Coil21. . . Stator22. . . Rotor23. . . Rotating Shaft24. . . Rotation-Sensing Target Part25. . . Through Hole26. . . Board-Fixing Projecting Part27. . . Board-Receiving Part28. . . Heat Dissipation Projecting Part29. . . Capacitor30. . . Coil31. . . Glass Epoxy Board32. . . Microcomputer33. . . Peripheral Circuit34. . . Glass Epoxy Board35. . . Step Portion36. . . Fixing Bolt37. . . Open End38. . . Swaging Fixation Portion39. . . Fixation Wall40. . . Swaging Recess42. . . Exposure Hole43. . . Lateral Peripheral Part44. . . Bottom Part45. . . Annular Seal Accommodating Portion46. . . Annular Reinforcing Projecting Portion48. . . Liquid Sealing Agent | 48,820 |
11863046 | Corresponding reference numerals indicate corresponding parts throughout the several views of the drawings. DETAILED DESCRIPTION The patent application file contains at least one drawing executed in color. Copies of the patent application with color drawings will be provided by the office upon request and payment of the necessary fee. InFIGS.1and2, an exemplary electric drive unit constructed in accordance with the teachings of the present disclosure is generally indicated by reference numeral10. Components, aspects, features and functions of the electric drive unit10that are not expressly described herein or shown (partly or fully) in the accompanying drawings, could be configured or function in a manner that is similar to the components, aspects, features and/or functions of electric drive units that are described in co-pending U.S. patent application Ser. No. 16/751,596 filed Jan. 24, 2020, U.S. patent application Ser. No. 16/865,912 filed May 4, 2020, U.S. patent application Ser. No. 17/128,288 filed Dec. 21, 2020, International Patent Application No. PCT/US2020/029925 filed Apr. 24, 2020, International Patent Application No. PCT/US2020/062541 filed Nov. 30, 2020, and/or U.S. Provisional Patent Application No. 63/159,511 filed Mar. 11, 2021, the disclosures of which are incorporated by reference as if fully set forth in detail herein. In brief, the electric drive unit10includes a housing assembly12, a motor assembly14, a transmission16, a differential assembly18, a pair of output shafts20, which are rotatable about an output axis22, and a lubrication and cooling system24. The housing assembly12can define one or more cavities (not specifically shown) in which the motor assembly14, the transmission16, the differential assembly18, and the output shafts20can be at least partly housed. In the example shown, the housing assembly12includes a gearbox cover30, a gearbox32, a motor housing34, a motor housing cover36and an end cover38. The gearbox cover30and the gearbox32abut one another and form a cavity into which the transmission16and the differential assembly18are received, while the gearbox32, the motor housing34and the motor housing cover36abut one another to form a cavity into which the motor assembly14is received. With specific reference toFIG.2, the motor assembly14comprises an electric motor40, and a motor control unit42that includes an inverter44. The electric motor40can be a multi-phase electric motor and includes a stator46and a rotor48that is rotatable about a motor output axis50. The stator has a stator core and a plurality of field windings that are wound about the stator core. Each of the field windings is associated with a corresponding phase of electrical power. The rotor48includes a motor output shaft52. With reference toFIGS.2and3, the transmission16can be configured in any desired manner to transmit rotary power between the motor output shaft52and a differential input member60of the differential assembly18. The transmission16could comprise one or more fixed reductions of any desired type, or could be configured as a multi-speed transmission having two or more alternately engagable reductions (and optionally one or more fixed reductions). The fixed or multi-speed reductions could be configured in any desired manner to permit the rotational axis of the motor output shaft to be oriented relative to the output axis22in a desired manner (e.g., parallel and offset, coincident, transverse, perpendicular, skewed). In the example illustrated inFIGS.3and4, the transmission16is a single-speed, multi-stage transmission employing a plurality of helical gears. The transmission16comprises a transmission input gear62, which is coupled for rotation with the motor output shaft52, a pair of compound gears64and a transmission output gear66. Each of the compound gears64is rotatable about an axis68that is parallel to and offset from the motor output axis50and can include a first reduction gear70, which can be meshingly engaged to the transmission input gear62, and a second reduction gear72that is coupled to the first reduction gear70for rotation therewith and meshingly engaged to the transmission output gear66. With reference toFIGS.5and6, each of the compound gears64can be configured such that the second reduction gear72is integrally and unitarily formed with a shaft member80and the first reduction gear70is rotationally coupled to the shaft member80in a desired manner, such as by laser welding. It will be appreciated, however, that the shaft member80could be integrally and unitarily formed with the first reduction gear70or could be a discrete component to which both the first and second reduction gears70and72are rotationally coupled. The shaft member80can extend axially outwardly from each of the first and second reduction gears70and72. In the particular example provided, the shaft member80and the second reduction gear72are identical components in each of the compound gears64. It will be appreciated, however, that the shaft member80and the second reduction gear72could be unique for each of the compound gears64, or that the compound gears64(i.e., the first and second reduction gears70and72and the shaft member80in the example provided) could be identical. Returning toFIGS.3and4, the first reduction gears70can be disposed in counterphase with respect to one another. In this regard, one of the first reduction gears70can be positioned such that one of its teeth is received between and centered between two adjacent teeth on the transmission input gear62, and one of the teeth of the transmission input gear62is disposed between two adjacent teeth on the other one of the first reduction gears70. It will be appreciated, however, that the other phasing could be employed, and that the teeth of both of the first reduction gears70could be in-phase with one another. The second reduction gears72can be disposed in-phase with one another. In this regard, one of the teeth on a first one of the second reduction gears72is received and centered between a first pair of adjacent teeth on the transmission output gear66, while at the same time one of teeth on the second one of the second reduction gears72is received and centered between a second pair of adjacent teeth on the transmission output gear66. It will be appreciated, however, that other phasing could be employed, that the teeth of the second reduction gears72could be in a counter-phase orientation with one another, and that the phasing of the second reduction gears72can be the same or different from the phasing of the first reduction gears70. With reference toFIGS.4,7and8, each compound gear64can be supported for rotation relative to the housing assembly12by a first bearing82and a second bearing84. In addition to the capability of handling and transmitting radially-directed forces between the compound gear64and the housing assembly12, the first bearing82can be a type of ball bearing that is configured to handle and transmit forces that are directed axially along the rotational axis68of the compound gear64when rotary power is transmitted through the transmission16. For example, the first bearing82could be a type of angular contact bearing or a deep groove ball bearing. The first bearing82can be received in a bore86formed in the gearbox cover30and a snap ring88or other type of engagement could be received into a counterbore, which is concentric with the bore86, and secured to the outer bearing race of the first bearing82. The snap ring88can abut a shoulder formed in an axial end of the counterbore in the gearbox cover30to inhibit axial movement of the first bearing82in a first axial direction along the rotational axis68of the compound gear64. The housing assembly12can further comprise a bearing cover90that can be mounted to the gearbox cover30to close the bores86in the gearbox cover30and shroud the first bearings82. If desired, pads or bosses92(FIG.9) can be formed onto the bearing cover90to radially overlap and axially abut the outer bearing races of the first bearings82to further support and stabilize the first bearings82. Additionally, or alternatively, passages94(FIG.9) can be provided in the bearing cover90to permit lubrication passing through the first bearings82to drain to an oil drain channel96in the gearbox cover30that permits the lubrication to drain to a desired area, such as a sump98(FIG.2). With reference toFIGS.4and8, the second bearing84can be a type of bearing that is configured to at least substantially or exclusively handle and transmit radially-directed forces between the compound gear64and the housing assembly12. Stated another way, the second bearing84can be a type of bearing that has an insufficient ability to handle or transmit forces that are directed axially along the rotational axis68of the compound gear64when rotary power is transmitted through the transmission16. In the example shown, the second bearing84is a roller bearing that employs cylindrically-shaped rollers between an inner bearing race and an outer bearing race. The second bearing84can be received into a bore99formed into the gearcase32. If desired, the inner bearing race could be formed onto the shaft member80to which the first and second reduction gears70and72are rotationally coupled. With renewed reference toFIGS.3and4, the rotational axes68of the compound gears64can be disposed relative to the motor output axis50in any desired manner to satisfy or accommodate criteria such as the overall speed or gear reduction of the transmission16, the size of the envelope into which the electric drive unit10(FIG.1) can be packaged and/or the extent to which the loading of the teeth of the first reduction gears70is equalized. In the example shown, the rotational axes68of the compound gears64and the motor output axis50are disposed in a plane P and an even number of teeth are formed on the transmission input gear62. Configuration in this manner can help to balance the loads that are transmitted between each of the first reduction gears70and the transmission input gear62. With reference toFIGS.10and11, the differential assembly18can further include a pair of differential output members100and can employ any desired means for permitting speed differentiation between the differential output members. In the example provided, the differential input member60is a differential case and the speed differentiation means comprises a differential gearset102in which the differential output members100are bevel side gears. It will be appreciated, however, that the speed differentiation means could employ a different type of differential gearset (e.g., a planetary gear arrangement in which the gears of the planetary arrangement are formed as spur or helical gears and one or more of the differential output members is a sun gear or a planet carrier of the planetary arrangement, or a helical gear arrangement in which the side gears are helical gears) or could employ one or more sets of clutches (e.g., friction clutches). InFIGS.10and12, the housing assembly12is illustrated to include a differential bearing cover110that is fixedly but removably coupled to the gearbox cover30. The differential bearing cover110defines an output shaft bore112that is disposed concentrically about the output axis22. A bearing, such as a tapered roller bearing114, can be disposed on a trunnion116formed on the differential input member60and an inner bearing race of the tapered roller bearing114can be abutted axially against a shoulder that is formed on the differential input member60. An outer bearing race of the tapered roller bearing114can be formed into a bearing bore118formed in the gearbox cover30. The differential bearing cover110can include a piloting portion120, which can be disposed in the bearing bore118, and a flange122that can extend radially outwardly from the piloting portion120. Threaded fasteners124can be disposed through the flange122and can be threaded into holes formed in the gearbox cover30. A suitable seal or gasket can be disposed between the gearbox cover30and the differential bearing cover110to form a seal that inhibits the egress of lubrication from the housing assembly12through the joint that is formed by the gearbox cover30and the differential bearing cover110. In the example shown, the seal comprises an O-ring128that is received over the piloting portion120and abutted against the flange122. The O-ring128is seated between a chamfer that is formed about the bearing bore118and the differential bearing cover110and is configured to seal an annular groove having a triangular cross-sectional shape that is defined by the chamfer, the outer diameter of the piloting portion120and an axial end of the flange122. A shim130can be received axially between the outer bearing race of the tapered roller bearing114and the piloting portion120. The shim130can be “select fit” (i.e., specifically selected for its particular thickness from a set of shims having different thicknesses) so as to aid in preloading the tapered roller bearing114to a desired preload when the differential bearing cover110is installed to the gearbox cover30. A shaft seal134can be received into the output shaft bore112. A corresponding one of the output shafts20can be received through the output shaft bore112and through the trunnion116on the differential input member60and can be non-rotatably coupled to a corresponding one of the differential output members100. In the example provided, the output shafts20have an externally splined segment that matingly engages an internally splined segment formed onto the corresponding one of the differential output members100. One or more grooves138can be formed in the axial end140of the piloting portion120to permit oil to drain through the piloting portion120to an oil drain channel146in the gearbox cover30that permits the lubrication to drain to a desired area, such as a sump98(FIG.2). FIGS.14and15illustrate the front and rear sides of the gearbox cover30in more detail. A first cooling pipe bracket150can be fixedly coupled to the inside surface of the gearbox cover30. In the example provided, the first cooling pipe bracket150is integrally and unitarily formed with the gearbox cover30. FIG.16illustrates the front side of the gearbox32in more detail. An stator oil return aperture154is formed through the gearbox32and permits fluid exiting the stator46(FIG.2) to return to the sump98. A coolant pipe aperture156is formed through the gearbox32and is configured to transmit a flow of fluid therethrough that is adapted for cooling the rotor48(FIG.2). A plurality of lubricant feed apertures160can be formed into the gearbox32and can be configured to feed a flow of fluid into each of the bores99and a bore158that are configured to receive the second bearings84(FIG.4), which support the compound gears64(FIG.4), and a tapered roller bearing114a(FIG.11) that supports the differential input member60(FIG.11). The lubricant feed apertures160intersect various lubricant feed galleries162that are formed through the gearbox32. Second and third coolant pipe brackets164and166, respectively, are fixedly coupled to the gearbox32. In the example provided, the second and third coolant pipe brackets164and166are unitarily and integrally formed with the gearbox32. A coolant pump intake port168is formed in the gearbox32and fluidly couples the sump98to a pump bore170that is also formed in the gearbox32and which is configured to receive a pump (not shown) that is part of the lubrication and cooling system24. An output shaft bore172is formed through the gearbox32and is configured to receive a corresponding one of the output shafts20(FIG.2) therethrough. InFIG.17, a portion of the gearbox32is shown in section view to better illustrate the pump bore170and one of the lubricant feed galleries162. The particular lubricant feed gallery162shown happens to intersect the pump bore17. Accordingly, it will be appreciated that a pump can be mounted in the pump bore170and can discharge fluid to the lubricant feed gallery162that intersects the pump bore17. InFIGS.18through20, a coolant pipe200is depicted connecting the coolant pipe aperture156to a central passage202in the rotor48. The coolant pipe200is configured to introduce a flow of a coolant fluid into the rotor48. The flow of coolant fluid is provided by the pump that is mounted to the gearbox32and is transmitted to the coolant pipe aperture156through various galleries (including gallery206in the motor housing34) that are formed in the housing assembly12. Clamps208can fixedly secure the coolant pipe200to the second and third coolant pipe brackets164and166on the gearbox32, while the first coolant pipe bracket150on the gearbox cover30can abut the coolant pipe200on a side of the coolant pipe200that is opposite the rotor48to inhibit withdrawal of the coolant pipe200from the rotor48. In the particular example provided, the coolant pipe200extends into the rotor48by a distance that equals or exceeds seven times the inside diameter of the coolant pipe200. However, it will be appreciated that the distance by which the coolant pipe200extends into the rotor48could be sized differently. InFIG.21, a trough220is mounted to the gearcase cover30and the gearcase32. The transmission output gear66rotates in the trough220to reduce or eliminate the churning of fluid in the sump98. The trough220can be formed of a suitable plastic material and can include a plurality of posts222that are received (e.g., slip fit or press fit) into apertures formed in the gearcase cover30and the gearcase32. With reference toFIGS.22and23, the motor housing cover36defines a cavity230for housing a rotational position sensor232and a portion of a wire harness234that couples the rotational position sensor232to a control board (not shown) of the motor control unit42(FIG.2). The motor housing cover36can also define a fluid feed gallery240, a fluid drain gallery242and an output shaft bore244. The motor housing cover36is sealingly coupled to the motor housing34to fluidly couple an inlet250of the fluid feed gallery240to an outlet port252from a heat exchanger254of the lubrication and cooling system24. The fluid feed gallery240defines a plurality of gallery outlet ports256that are configured to route fluid from the fluid feed gallery240into the motor control unit42(FIG.2) and the electric motor40(FIG.2) as will be described in more detail below. The fluid drain gallery242can be employed to route a lubricating and cooling fluid from a rotor shaft bearing (not shown) to a gallery (not shown) in the motor housing34to return the fluid to the sump98(FIG.2). The output shaft bore244can be sized to receive an output shaft seal258therein. A corresponding one of the output shafts20can extend through the motor housing cover36and can be sealingly engaged to the output shaft seal258. The end cover38can be sealingly coupled to the motor housing cover36to seal each of the cavity230, the fluid feed gallery240, and the fluid drain gallery242. InFIGS.24and25, the motor housing cover36can include a pilot rib260that can be received into a bore262formed in the motor housing34to aid in locating the motor housing cover36to the motor housing34in a desired manner. The pilot rib262can be shaped as a segment of a circle that is centered about the motor output axis50. A bearing holder264can be shown to be fixedly coupled (e.g., unitarily and integrally formed with) the motor housing cover36and disposed concentrically about the gallery outlet ports256such that the gallery outlet ports256pass axially through a portion of the bearing holder264. The bearing holder264can define a rotor shaft bore270, a bearing counterbore272, sensor mount counterbore274, a first external seal groove276, a second external seal groove278and a fluid distribution groove280. The rotor shaft bore270can be formed through the bearing holder264and can be sized to receive an end of the rotor48therein. The bearing counterbore272can be disposed in the distal end of the bearing holder264, can be concentric with the rotor shaft bore270, and is sized to receive a bearing284that is mounted on the rotor48. The bearing284is configured to support the end of the rotor48for rotation about the motor output axis50relative to the housing assembly12. The sensor mount counterbore274is formed the proximal end of the bearing holder264, can be concentric with the rotor shaft bore270, and intersects the cavity230in the motor housing cover36. The sensor mount counterbore274is sized to receive a sensor mount as will be discussed in more detail below. The first external seal groove270and the second external seal groove272are spaced axially apart from one another along the motor output axis50and are each configured to receive a respective seal (not shown), such as an O-ring seal, therein. The seal in the first external seal groove270is configured to form a fluid-tight seal between the bearing holder264and an inverter mount290of the inverter44, while the seal in the second external seal groove272is configured to form a fluid-tight seal between the bearing holder264and a cap294that covers the windings296of the stator46. The fluid distribution groove280is disposed along the motor output axis50between the first and second external seal grooves270and272. The fluid distribution groove280extends over a portion of the circumference of the bearing holder264and intersects the gallery outlet ports256. Accordingly, fluid discharged from the fluid feed gallery240(FIG.23) in the motor housing cover36through the gallery outlet ports256passes into the fluid distribution groove280and is directed radially outwardly and circumferentially about an interior perimeter of the inverter44. If desired, the downstream end of the fluid distribution groove280can be shaped in a manner that aids the outward progression of fluid toward the inverter44. In the example provided, the downstream end of the fluid distribution groove280is frustoconically shaped. Fluid passes through a plurality of heat sinks and power semiconductors (e.g., MOSFET's, IGBT's) in the inverter44to remove heat from the inverter44. Fluid discharged from the inverter44is routed against and around the cap294that covers the windings296of the stator46before it is routed into a plurality of cooling channels300that are formed in a body302of the stator46. InFIG.26, a relatively small hole310is formed radially through the bearing holder264at a location that intersects or is somewhat downstream from the fluid distribution groove280. The hole310intersects the bearing counterbore272and is configured to supply a lubricating fluid to the bearing284. A Belleville spring washer312can be disposed between an internal shoulder on the bearing holder264that defines the proximal end of the bearing counterbore272and the bearing284. The Belville spring washer312acts as a spacer that is configured to ensure that the flow of fluid exiting the hole310is not and does not become blocked by the outer bearing race of the bearing284. InFIGS.27through29, a well320and a drain gallery322are formed in the bearing holder264. The well320intersects and extends radially outwardly from the bearing counterbore272. The well320extends in an axial direction so as to be somewhat wider than the bearing284. The drain gallery322connects the well320to the fluid drain gallery242(FIG.23) in the motor housing cover36(FIG.23). Accordingly, fluid supplied to the bearing284for its lubrication and/or cooling can be discharged to the well320and will drain through the drain gallery322to the fluid drain gallery242(FIG.23) and then to the sump98(FIG.2). InFIGS.29and30, a floating seal330is mounted to the motor output shaft52. The floating seal330is disposed radially between the motor output shaft52and the bearing holder264, as well as axially on the motor output shaft52between a first shoulder and a second shoulder. The bearing284is abutted against the second shoulder. The floating seal330has an annular body332and a circumferentially extending seal member334. The annular body332is sized to slip fit over the portion of the motor output shaft58that is disposed between the first and second shoulders. The annular body332can define a first set of projections336, which face toward the first shoulder, and a second set of projections338that face toward the bearing284. The first set of projections336can be spaced circumferentially apart from one another and can be configured to minimize the surface area over which the annular body332and the first shoulder are potentially able to contact one another. The second set of projections338can be spaced circumferentially apart from one another and can be configured to minimize the surface area over which the annular body332and the outer bearing race of the bearing284are potentially able to contact one another. Additionally, the second set of projections338permit lubrication that has passed through the bearing284to drain through the second set of projections338into the well320. The circumferentially extending seal member334can be sealingly engaged to the annular body332and to the circumferentially extending surface of the bearing counterbore272. In some instances, it may be possible to configure the bearing284as a greased bearing to eliminate the need for supplying fluid to and draining fluid from the bearing284to thereby reduce the complexity of various components, such as by omitting features, such as the hole310(FIG.26) in the bearing holder264(FIG.26) and/or the fluid drain gallery242(FIG.23) in the motor housing cover36(FIG.23), and/or the well320(FIG.27) and the drain gallery322(FIG.27) in the bearing holder264(FIG.27), and/or to eliminate various components, such as the floating seal330. The (greased) bearing284could be a sealed bearing, such as a bearing that uses non-contacting seals. Optionally, the bearing (not shown) that supports the opposite end of the rotor48(FIG.2) could also be a greased bearing and can be a sealed bearing (e.g., a bearing using non-contacting seals) or alternatively could be a bearing with a single seal in which the seal is a non-contacting seal and faces toward the rotor48(FIG.2) while the open or unsealed side of the bearing faces the transmission16(FIG.2). With reference toFIGS.31through33, the rotational position sensor232can be disposed on a circuit board that can be mounted to a plug or sensor mount350that is fitted into the bearing holder264. The sensor mount350can have a body352and a plurality of fingers354that extend from and are resiliently (flexibly) coupled to the body352. A recess358can be formed in the body352on a side of the body352that is opposite the fingers354. A seal groove360can be disposed about the perimeter of the body352. A plurality of posts362can extend from an axial end of the body352the side of the body352into which the recess358is formed. The circuit board232cbof the rotational position sensor232can be abutted to the axial end of the body352such that the posts362extend through holes (not specifically shown) in the circuit board232cband various components of the rotational position sensor232are received into the recess358. The posts362can be employed to heat-stake the circuit board232cbto the body352to thereby fixedly couple the rotational position sensor232to the sensor mount350. The circuit board232cbfurther includes a wire harness socket232whsthat is configured to receive a connector on the wire harness234. A suitable seal370, such as an O-ring, is received into the seal groove360. The sensor mount350is inserted into the interior of the bearing holder264such that the seal370is sealingly engaged to the interior surface of the sensor mount counterbore274and the body352of the sensor mount350. If desired, a chamfer can be formed on the bearing holder264to aid in compressing the seal370when the sensor mount350is inserted into the bearing holder264. The body352of the sensor mount350can abut an internal shoulder that is formed on the bearing holder264where the rotor shaft bore270intersects the sensor mount counterbore274. The fingers354frictionally engage the interior surface of the rotor shaft bore270to inhibit axial movement of the sensor mount350along the motor output axis50. A suitable sensor target376can be mounted to an axial end of the motor output shaft52. In the example provided, the rotational position sensor232is a TMR sensor, and the sensor target376is a diametrically poled magnet that is secured to the motor output shaft52via a suitable adhesive. The wire harness234can be mounted to the wire harness socket232whsto electrically couple the wire harness234to the rotational position sensor232. With reference toFIGS.34and35, the motor control unit42includes a field capacitor380that is received into a capacitor cavity382that is formed in the motor housing34. The capacitor cavity382is surrounded by a mounting flange384against which a cover386is secured to close the open end of the capacitor cavity382. The motor housing34includes a heat sink388that is configured to abut the field capacitor380when the field capacitor380is installed in the capacitor cavity382. If desired, the heat sink388could be unitarily and integrally formed with the remainder of the motor housing34, or could be a discrete component that is assembled to the remainder of the motor housing34. Mass-reduction features, such as slots390can be formed into the heat sink388if desired. Threaded fasteners (not shown) can be employed to secure the field capacitor380to the heat sink388. If desired, a thermally-conductive paste or foil can be disposed between the field capacitor380and the heat sink388. InFIG.36, a stator bore400and an output shaft bore402are formed in the motor housing34. The stator bore400is sized to receive the body302of the stator46therein and the central or longitudinal axis of the stator bore400is the motor output axis50. The output shaft bore402is sized to receive a corresponding one of the output shafts20therein and the central or longitudinal axis of the output shaft bore402is the output axis22. In the example provided, the stator bore400and the output shaft bore402are spaced apart from one another so that they do not intersect one another. If desired, however, the stator bore400and the output shaft bore402could intersect one another to reduce the overall volume of the electric drive unit10(FIG.1). Stated another way, and assuming that the diameters of the stator bore400and the output shaft bore402were the same as those in the example illustrated inFIG.36, the stator bore400could be configured to intersect the output shaft bore402such that the distance between the motor output axis50and the output axis22is smaller than the dimension D that is depicted inFIG.36(i.e., the distance D is smaller than the radius of the stator bore400). With reference toFIGS.37through39, fluid that drain through the fluid drain gallery234(FIG.23) in the motor housing cover36(FIG.23) is directed axially through the motor housing cover36(FIG.23) into a gallery430that is formed in the motor housing34. The gallery430intersects the output shaft bore402. The output shaft bore402is tapered, having a smaller diameter end proximate the axial end of the motor housing34that is adjacent the motor housing cover36(FIG.2), and a larger diameter end proximate the axial end of the motor housing34that is adjacent the gearbox32. Configuration in this manner aids the draining of fluid through the output shaft bore402to the sump98(FIG.2), as well as ensures that sloshing fluid in the sump98(FIG.2) cannot be pushed from the sump98(FIG.2) to the rotor48(FIG.2) via the output shaft bore402, the gallery430and the fluid drain gallery234(FIG.23). With reference toFIG.40, the electrical connection between the electric motor40and the inverter44is shown in more detail. In addition to the inverter mount290, the inverter44comprises a plurality of power semiconductors450, a plurality of busbars (e.g., positive busbar452, ground busbar454, and a plurality of phase busbars456), and an inverter circuit board458. The inverter44controls the frequency of power supplied to the electric motor40. More specifically, the inverter44employs the power semiconductors450(e.g., MOSFET's, IGBT's) to control the switching of DC electricity to create three AC electric outputs, each being associated with a given phase of the windings296of the stator46. Each phase of the windings296is fixedly and electrically coupled to a bridge member594on an associated one of the phase busbars456in the inverter44. With additional reference toFIGS.41and42, a phase terminal470is electrically coupled to each phase of the windings296. In the example provided, an end (not shown) of the phase of the windings296is received into a winding aperture472in the phase terminal470. In the example shown, the winding aperture472is transverse to the longitudinal axis of the phase terminal470and the end of the phase of the windings296is soldered to the phase terminal470. The phase terminal470can further include an anti-rotation feature, such as knurling474, a seal groove476and a connecting feature478. The seal groove476can be configured to receive an associated seal, such as an O-ring, that can form a seal between the phase terminal470and the inverter mount290. The connecting feature478aids in fixedly and electrically coupling the phase terminal470to an associated one of the phase busbars456. In the example provided, the connecting feature478is a threaded aperture that is configured to receive a threaded fastener480that is inserted through the bridge member594on an associated one of the phase busbars456to thereby fixedly and electrically couple the phase terminal470to the bridge member594. The cap294can be formed by overmolding a material over the windings296. The material that is used to form the cap294is an electrically insulating material but also has relatively good thermally conductive properties. The phase terminals470are partly encased in the material that forms the cap294. More specifically, the portion of the phase terminals470that includes the winding aperture472and the knurling474is encased in the material that forms the cap294. The knurling474and the material that forms the cap294cooperate to resist relative rotation between the phase terminal470and the cap294. With reference toFIGS.43and44, the inverter mount290is illustrated in more detail. The inverter mount290can include a base500, a plurality of terminal receptacles502, a plurality of sensor receptacles504, a first side wall506and a second side wall508. The base500can have a generally annular configuration. A first axial side or face of the base500can have a central portion that is somewhat thicker than an outer portion that is disposed radially outwardly of the central portion. A second, opposite side or face of the base500can be flat. The base500can define a plurality of semiconductor mounts510that can be formed into the central portion on the first face of the base500. Each of the semiconductor mounts510can define a semiconductor recess512and a plurality of semiconductor terminal apertures514. The semiconductor mounts510can be disposed in any desired arrangement, but in the particular example provided, the semiconductor mounts510are disposed in a ring-shaped arrangement. The semiconductor terminal apertures514are disposed in each the semiconductor recess512and are formed through the base500. Each of the terminal receptacles502can have a first portion, which is located on the portion of the base500that is disposed radially outwardly of the central portion and which extend axially away from the first face of the base500, and a second portion that extends axially away from the second face of the base500. In the example shown, each of the terminal receptacles502is a generally tubular structure that is disposed through the outer portion of the base500. The terminal receptacles502can be spaced circumferentially apart from one another. Each of the sensor receptacles504can extend from the second face of the base500and can intersect an associated one of the terminal receptacles502. The first and second sidewalls506and508can be fixedly coupled to the base500and can encircle the outer perimeter and the inner perimeter, respectfully, of the base500. The first side wall506can extend from the first face of the base500by a relatively large distance and from the second face of the base500by a relatively short distance. The second side wall508can extend from the second face of the base500by a relatively large distance and from the first face of the base by a relatively small distance. A seal groove516is formed about the first side wall506and is configured to receive a seal518(FIG.40) therein that sealingly engages the first side wall506and the motor housing34(FIG.40. A seal that is mounted in the first external seal groove276(FIG.25) in the bearing holder264(FIG.25) is sealingly engaged to the bearing holder264(FIG.25) and the radially inner surface of the second side wall508. InFIG.45, a heat sink520is illustrated as being fixedly coupled to one of the power semiconductors450to form a heat-sinked power semiconductor assembly522. The power semiconductor450includes a plurality of pins or terminals524. The heat sink520can be formed of a suitable thermally conductive material and can be electrically coupled to an associated one of the terminals524. As a non-limiting example, the heat sink520could be formed of a metal material, such as aluminum, brass, bronze or copper. The heat sink520can define a plurality of fins526that can be employed to discharge heat into a flow of fluid passing through the fins526. In the particular example illustrated, the fins526comprise rod-like projections having an oval cross-sectional shape, but it will be appreciated that the fins526can be formed in any desired manner and can have any desired cross-sectional shape (e.g., circular, rectangular, diamond). The heat sink520could be unitarily and integrally formed, and if desired, the fins526can be formed with draft (i.e., taper along their longitudinal axis). Alternatively, the fins526could be discrete components that are assembled/fixedly coupled to a base of the heat sink520. Each of the heat sinks520can be disposed between a pair of the power semiconductors450(i.e., each heat sink520is physically coupled to one of the power semiconductors450and is adjacent, but circumferentially spaced from, another power semiconductor450). Each heat sink520can be configured such that the height of the fins526(i.e., a distance that the fins526extend from the power semiconductor450of a given heat-sinked power semiconductor assembly522) on a radially inward end of the heat sink520can be shorter than the height of the fins526on a radially outward end of the heat sink520. Accordingly, the height of the heat sink520of a heat-sinked power semiconductor assembly522can taper between the radially inward end of the heat-sinked power semiconductor assembly522and the radially outer end of the heat-sinked power semiconductor assembly522. With reference toFIGS.43,45and46, each of the heat-sinked power semiconductor assembly522can be mounted in a respective one of the semiconductor mounts510on the inverter mount290such that each of the power semiconductors450is received into a corresponding one of the semiconductor recesses512and the terminals524on each of the power semiconductors450are received through semiconductor terminal apertures514. With reference toFIGS.47and48, the positive busbar452includes a first busbar portion550and a second busbar portion552that are fixedly and electrically coupled to one another. The first and second busbar portions550and552can be generally similar in their construction and as such, only the first busbar portion550will be described in detail herein. The first busbar portion550can be formed of an electrically conductive material, such as copper, and can include an annular body560, a conductor link562and a plurality of fingers564. The annular body560is sized to be received within the first side wall506(FIG.44) in the inverter mount290(FIG.44). The inner circumference of the annular body560can be disposed radially outwardly of the semiconductor terminal apertures514(FIG.44). A plurality of apertures570can be formed through the annular body560that permit the annular body560to be received over the terminal receptacles502(FIG.44) and the sensor receptacles504(FIG.44). The conductor link562can extend radially outwardly from the annular body560and can be configured to extend over the inverter mount290(FIG.46) where it can be disposed at a location where it can be electrically coupled to a capacitor (not shown). Each of the fingers564can having a first portion572and a second portion574. The first portion572can be generally L-shaped, having a leg that extends radially inwardly from the inner circumferential edge of the annular body560, and an arm that extends away from the leg in a first circumferential direction. The second portion574can extend from the distal end of the arm in a direction that is perpendicular to the annular body560. It should be noted that the arms of the first portion572aof the fingers564of the second busbar portion552extends away from an associated one of the legs in a second, opposite circumferential direction so that the arms of the first portions572and572aface one another. It will be appreciated that the fingers564are configured to electrically connect to a first set of the terminals524(FIG.45) on a first set of the power semiconductors450(FIG.46). The ground busbar454is illustrated inFIG.49and has a configuration that is generally similar to that of the positive busbar452(FIG.47) except for the locations of the fingers564′ and the radial length of the legs of the first portions572′,572a′ of the fingers564′. In this regard, the fingers564′ are configured to electrically connect to a second set of the terminals524(FIG.45) on a second set of the power semiconductors450(FIG.46). With reference toFIGS.50and51, each phase busbar456can be formed in a manner that is similar to that of the positive busbar452(FIG.47) except that the phase busbars456do not include a conductor link, each phase busbar456has first and second busbar portions550″ and552″ that each have an annular segment-shaped body560″, and the phase busbars456each include sets of different fingers564″,564a″, an end finger590, and a bridge592. In the example provided, the fingers564a″ are relatively longer than the fingers564″, and the quantity of sets of the fingers564″ is equal to the quantity of sets of fingers564a″ on each of the phase busbars456. The end finger590is configured to extend outwardly from the annular segment-shaped body560″ of the phase busbar456. The bridge592has a bridge member594that is parallel to but spaced apart from the first busbar portion550″. A sensor slot596is formed through one side of the bridge592, while a fastener aperture598is formed through the bridge member594. The bridge592can be fixedly coupled to the first busbar portion550″ in any desired manner, such as projection welding. With reference toFIGS.40,44,45and52, the positive busbar452can be received within the first side wall506in the inverter mount290such that the positive busbar452is abutted against the second side of the base500and each adjacent pair of the arms574,574a(FIG.48) is engaged to opposite sides of one of the terminals524on a respective one of the power semiconductors450. A first electrically insulating member can be disposed over the positive busbar452and the ground busbar454can be positioned within the first side wall506in the inverter mount290such that the ground busbar454is abutted against the first electrically insulating member and each adjacent pair of the arms of the fingers564′ (FIG.49) is engaged to opposite sides of one of the terminals524on a respective one of the power semiconductors450. A second electrically insulating member can be disposed over the ground busbar454and each of the phase busbars456can be positioned within the first side wall506and abutted against the second insulator. Each adjacent pair of the arms of the fingers564″ and564a″ (FIG.50) of the phase busbar456is engaged to opposite sides of one of the terminals524on a respective one of the power semiconductors450. The positive busbar452, the ground busbar454and each of the phase busbars456can be oriented relative to the inverter mount290such that the terminal receptacles502and an adjacent one of the sensor receptacles504are received through each of the apertures570. Each of the terminals can be soldered, sintered or welded to the set of fingers that connect the terminal to an associated one of the busbars to thereby mechanically and electrically couple the terminals and busbars. With reference toFIGS.52through54, it will be appreciated that assembly of the several busbars to the power semiconductors in this manner will position the distal ends of each of the fingers and end fingers in abutment with an associated one of the terminals524on an associated one of the power semiconductors450. Thereafter, the fingers and end fingers can be fixedly and electrically coupled to the terminals524, for example by resistance welding or resistance soldering adjacent pairs of them to the terminals524. InFIGS.55and56, the motor control unit42(FIG.2) includes a plurality of current sensors600, each of which being associated with a corresponding set of the terminal receptacles502, the sensor receptacles504, and the bridges592on the phase busbars456. Each of the current sensors comprises a sensor602and a plurality of C-shaped sensor laminations604. The sensor602can be an eddy current sensor and can be received in the sensor receptacle504. The sensor laminations604are abutted against one another and are received about the terminal receptacle502between the first busbar portion550″ and the bridge member594. The open ends of the sensor laminations604are disposed on opposite sides of the sensor receptacle504(i.e., a radially inner side and a radially outer side). With additional reference toFIG.40, the terminals524of the power semiconductors450and the terminals of the sensor602are received into the inverter circuit board458and can be electrically coupled to other componentry of the inverter circuit board458in a desired manner. The phase terminals470are received into the terminal receptacles502and the seal that is disposed in the seal groove476in each of the phase terminals470forms a seal between the phase terminal470and the inverter mount290that inhibits the flow of fluid therethrough. The threaded fastener480is received through the fastener aperture598in the bridge member594and is threadably engaged to the connecting feature478to fixedly and electrically couple the phase terminal470to the phase busbar456. During operation of the electric motor40, current passing from the bridge member594of the phase busbar456to the phase terminal470will generate a magnetic field that will correspondingly generate eddy currents in the sensor laminations604. The sensor602is configured to sense the eddy currents in the sensor laminations604and responsively generate a sensor signal that is indicative of a magnitude of eddy currents in the sensor laminations. Significantly, the current sensor602is positioned in as close a proximity to the interface between the phase busbar456and the phase terminal470as is possible. FIGS.57and58depict a second cap620that is coupled to (e.g., overmolded over) the windings296on an end of the motor40that extends into a cavity formed by the gearbox32. The second cap620can be generally similar to the cap294(FIG.25). An annular oil diverter622is mounted about the second cap620and aids in directing fluid discharged from the cooling channels300(FIG.25) to flow against the outer circumferential surface of the second cap620as well as an axial end of the second cap620that is spaced apart from the body302of the stator46. A cylindrical projection626can be formed on the gearcase32and can extend into the second cap620. A seal628, such as an O-ring, can be received into a groove formed in the cylindrical projection626and can sealingly engage the inside circumferential surface of the second cap620. A bore630can be formed through the axial end of the cylindrical projection626and can intersect a bearing bore632that houses a bearing634that supports the motor output shaft52. The bore630can be sized to receive the motor output shaft50in a non-contacting manner. If desired, a non-contact seal, such as one of the non-contact seals shown inFIG.59, can be mounted in the bore630to resist migration of fluid through the axial end of the cylindrical projection626. With reference toFIGS.60through63, an alternately configured inverter is shown. With reference toFIG.60, the field capacitor380bin this example is received in the motor housing34and has an annular configuration. The inverter44is received radially within the field capacitor380b. If desired, the field windings296can be encapsulated in an encapsulant material700that can form an annular chamber into which the field capacitor380bcan be received. The encapsulant material700can be cohesively bonded to the motor housing34. Optionally, the encapsulant material700can be sealingly engaged with a seal member to form a seal between the encapsulant material and another component. For example, an O-ring702that is mounted to the motor cover36bcan form a seal between the motor cover36band the encapsulant material700, while another O-ring704, which is mounted to the bearing holder264, can form a seal between the bearing holder264and an inside diametrical surface of an annular flange706that is formed by the encapsulant material700. With reference toFIGS.61and62, a cooling channel710is formed in the motor cover36band receives a flow of coolant/lubricant that is fed between a pair of seal members702and712that are mounted to the motor cover36band engaged to the encapsulant material700and the motor housing34, respectively. Fluid is discharged from the cooling channel710into the fluid distribution groove280in the bearing holder264. It will be appreciated that the fluid in the fluid distribution groove280can travel through the heat-sinked power semiconductor assemblies522, about the field windings296and into the cooling channels300in the stator46. With reference toFIGS.63and64of the drawings, another exemplary electric drive unit (EDU) constructed in accordance with the teachings of the present disclosure is generally indicated by reference numeral1010. Except as described herein, the EDU1010can be configured in a manner that is similar to that of the electric drive unit that is illustrated and described in detail in U.S. Provisional Patent Application No. 63/161,164 filed Mar. 15, 2021, the disclosure of which is incorporated by reference as if fully set forth in detail herein. The EDU1010includes a housing1012and a motor assembly1014that is received in the housing1012and which has an electric motor1016and a motor control unit1018that includes an inverter1020. The electric motor1016includes a rotor1024, a motor output shaft1026and a stator1030. The rotor1024is rotatable relative to the stator1030about a motor axis34. The motor output shaft1026is coupled to the rotor1024for rotation therewith. With reference toFIGS.64through67, the stator1030includes a stator core1040, a plurality of windings1042, a plurality of phase terminals1044and a cap1046. The windings1042are wound about the stator core1040and are segregated into several phases. Each phase terminal1044is mechanically and electrically coupled to an associated phase of the windings1042. In the example provided, an end1048of a phase of the windings1042is received into a winding aperture1052in a respective one of the phase terminals1044. In the example shown, the winding aperture1052is transverse to the longitudinal axis of the phase terminal1044and the end of the phase of the windings1042is physically and electrically coupled (e.g., soldered) to the phase terminal1044. The phase terminal1044can further include an anti-rotation feature, such as knurling1056, a seal groove1058and a connecting feature1060. The seal groove1058can be configured to receive an associated seal, such as an O-ring1062, that can form a seal between the phase terminal1044and the inverter1020. The connecting feature1060aids in fixedly and electrically coupling the phase terminal1044to the inverter1020. In the example provided, the connecting feature1060is a threaded aperture that is configured to receive a threaded fastener1066. The cap1046can be a discrete component that can be formed in a suitable process, such as injection molding, and can be fitted over the windings1042. The cap1046defines a plurality of pockets1070, each of which being disposed about a respective one of the phase terminals1044. The material that is used to form the cap1046is an electrically insulating material but also has relatively good thermally conductive properties. A suitable material, such as an epoxy material1072, can be injected between the windings1042and the cap1046and can fill the pockets1070to a desired extent to seal between the interior surface of each of the pockets1070and an associated one of the phase terminals1044. It will be appreciated that the phase terminals1044are partly encased in the epoxy material1072. More specifically, the knurling1056on each phase terminal1044is encased in the epoxy material1072. The knurling1056and the epoxy material1072cooperate to resist relative rotation between the phase terminal1044and the cap1046. With reference toFIGS.65,66and68, a portion of the inverter1020is shown in more detail. The inverter1020includes an inverter mount1080, a plurality of power semiconductors1082, a plurality of busbars (e.g., positive busbar1090, ground busbar1092, and a plurality of phase busbars1094), a plurality of insulating layers (not specifically shown), and an inverter circuit board1096. The inverter1020controls the frequency of power supplied to the electric motor1016. More specifically, the inverter1020employs the power semiconductors1082, which can be MOSFET's or IGBT's, for example, to control the switching of DC electricity to create three AC electric outputs, with each AC electric output being associated with a given phase of the windings1042of the stator1030. Each phase of the windings1042is fixedly and electrically coupled to a bridge member1212on an associated one of the phase busbars1094in the inverter1020. The inverter mount1080can include a base1120, a plurality of terminal receptacles1122, a plurality of sensor receptacles1124, a first side wall1126and a second side wall1128. The base1120can have a generally annular configuration. A first axial side or face of the base1120can have a radially outer portion that is somewhat thicker than a central portion that is disposed radially inwardly of the radially outer portion. A second, opposite side or face of the base1120can be flat. The base1120can define a plurality of semiconductor mounts (not specifically shown) that can be formed into the radially outer portion on the first face of the base1120. Each of the semiconductor mounts can define a plurality of semiconductor terminal apertures (not specifically shown). The semiconductor mounts can be disposed in any desired arrangement, but in the particular example provided, the semiconductor mounts are disposed in a ring-shaped arrangement about the outer perimeter of the base1120. Each of the terminal receptacles1122defines an aperture, which is formed through the base1120, and can have a first portion, which is located on the central portion of the base1120and which extends axially away from the first face of the base1120, and a second portion that extends axially away from the second face of the base1120. In the example shown, each of the terminal receptacles1122is a generally tubular structure that is disposed on the central portion of the base1120. The terminal receptacles1122can be spaced circumferentially apart from one another. Each of the sensor receptacles1124can extend from the second face of the base1120and can be disposed about an associated one of the terminal receptacles1122. The first and second sidewalls1126and1128can be fixedly coupled to the base1120and can encircle the outer perimeter and the inner perimeter, respectfully, of the base1120. The first side wall1126can extend from the first face of the base1120by a first distance and from the second face of the base1120by a second, relatively shorter distance. The second side wall1128can extend from the first face of the base1120by a third distance that can be relatively larger than the first distance. A first seal groove1132is formed about the first side wall1126and is configured to receive a first seal1134therein that sealingly engages the first side wall1126and the housing1012. A second seal groove1136is formed about the second side wall1128and is configured to receive a second seal1138therein that sealingly engages the second side wall1128and the cap1046. With reference toFIG.69, each of the power semiconductors1082has a plurality of pins or terminals1150and is fixedly coupled to a respective heat sink1152to form a heat-sinked power semiconductor assembly1154. The heat sink1152can be formed of a suitable thermally conductive material and can be electrically coupled to an associated one of the terminals1150. As a non-limiting example, the heat sink1152could be formed of a metal material, such as aluminum, brass, bronze or copper. The heat sink1152can define a plurality of fins1156that can be employed to discharge heat into a flow of fluid passing through the fins1156. The fins1156are schematically illustrated in the particular example provided. Returning toFIGS.66and68, each of the heat-sinked power semiconductor assemblies1154can be mounted in a respective one of the semiconductor mounts on the inverter mount1080such that each of the power semiconductors1082is received into a corresponding one of the power semiconductor recesses and the terminals1150on each of the power semiconductors1082are received through semiconductor terminal apertures in the inverter mount1080. Configuration in the manner illustrated positions the power semiconductors1082radially outwardly of the windings1042. With reference toFIGS.66,68and70, the busbars and the insulating layers are stacked to form a busbar assembly in which an insulating layer is disposed between the positive busbar1090and the ground busbar1092, and an insulating layer is disposed between the ground busbar1092and the phase busbars1094. The insulating layers are formed of an electrically insulating material and electrically insulate axially adjacent busbars from one another. Each of the positive and ground busbars1090and1092and each of the phase busbars1094includes a first busbar portion1160a,1160band1160c, respectively, and a second busbar portion1162a,1162band1162c, respectively, that are fixedly and electrically coupled to one another. Each of the second busbar portions1162a,1162band1162ccan be generally similar in their construction to their associated first busbar portion1160a,1160band1162c, respectively, and as such, only the first busbar portions1160a,1160band1160cwill be described in detail herein. Each of the first busbar portions1160a,1160band1160ccan be formed of an electrically conductive material, such as copper, and can include a body1170a,1170band1170c, respectively, and a set of fingers1174a,1174band1174c, respectively. The first busbar portions1160aand1160bof the positive busbar1090and the ground busbar1092also include a conductor link1176aand1176b, respectively. The bodies1170aand1170bof the positive busbar1090and the ground busbar1092can have an annular shape, while the body1170cof each phase busbar1094can be shaped as an annular segment. The bodies1170a,1170band1170care sized to be received over the second side wall1128of the inverter mount1080and within the first side wall1126of the inverter mount1080. The outer circumference of the annular bodies1170a,1170band1170ccan be disposed radially inwardly of the semiconductor terminal apertures in the inverter mount1080. The annular bodies1170aand1170bcan define a central aperture1180, a plurality of terminal apertures1182and a cooling standpipe aperture1184that are formed through the annular body1170. The terminal apertures1182can be received over the terminal receptacles1122(FIG.65) and the sensor receptacles1124(FIG.65), while the cooling standpipe aperture1184can be received over a cooling standpipe1200that is integrally formed with the inverter mount1080. The cooling standpipe1200is configured to direct coolant through the inverter mount1080to a coolant chamber1202(FIG.65) that is disposed between the inverter mount1080and the cap1046. Heat from the power semiconductor assemblies1154and the windings1042can be transmitted to the coolant in the coolant chamber1202(FIG.65) to thereby cool the inverter1020and the electric motor1016(FIG.64). The conductor links1176aand117bcan extend radially outwardly from the annular body1170aand1170band can be configured to extend over the inverter mount1080and disposed at locations where they can be electrically coupled to an associated terminal (not shown) of a capacitor (not shown). A bridge1210can be coupled to the radially inner side of the body1170cof each of the phase busbars1094. The bridge1210has a bridge member1212that is parallel to but spaced apart from the body1170c. A terminal aperture is formed through the bridge member1212. The bridge1210can be fixedly coupled to the body1170cin any desired manner, but in the particular example provided the bridge1210is integrally and unitarily formed with the body1170c. Each of the sets of fingers1174a,1174band1174ccomprises a plurality of fingers that are configured to mechanically and electrically couple a terminal1150(FIG.69) of an associated one of the power semiconductors1082to an associated one of the positive, ground and phase busbars1090,1092and1094. With reference toFIGS.71and72, each of the fingers (e.g., fingers1230) in a set of fingers (e.g., fingers1174c) can having a first portion1250, which can extend radially outwardly from the body (e.g., body1170c), and at least one second portion1252that is configured to be mechanically and electrically coupled to a respective terminal1150(FIG.69) on an associated one of the power semiconductors1082(FIG.69). In the example shown, each of first portions1250is disposed in a plane in which the body (e.g., body1170c) is disposed, each of the first portions1250is spaced circumferentially apart about the outside perimeter of the body (e.g., body1170c), a pair of second portions1252is coupled to each of the first portions1250, and each of the second portions1252is generally L-shaped, having a leg1254that extends from a respective one of the first portions1250and an arm1256that extends from the leg1254in a direction that is perpendicular to the leg1254. With reference toFIGS.69through72, the sets of fingers1174a,1174b,1174con the first portions1160a,1160band1160cof the positive, ground and phase busbars90,92and94can be generally similar in their configuration. The relative length of the first portions1250and the positioning of the second portions1252is configured so that each of the sets of fingers1174a,1174band1174cis configured to engage a respective set of terminals on the power semiconductors1082. The set of fingers1174aof the first portion1160aof the positive busbar1090are configured to be mechanically and electrically coupled to a first one of the terminals1150on the power semiconductors1082, the set of fingers1174bof the first portion1160bof the ground busbar1092are configured to be mechanically and electrically coupled to a second one of the terminals1150on the power semiconductors1082, and the sets of fingers1174cof the first portion1160cof the phase busbars1094are configured to be mechanically and electrically coupled to a third one and a fourth one of the terminals1150on the power semiconductors1082. It will be appreciated that the sets of fingers1174a,1174band1174care staggered both in a circumferential direction and in a radial direction to avoid direct electrical coupling between two or more of the busbars. Each of the second busbar portions1162a,1162band1162ccan be generally similar to its associated first busbar portion1160a,1160band1160c, respectively, except for the configuration of the sets of fingers1174a′,1174b′ and1174c′. More specifically, the fingers of each of the sets of fingers1174a′,1174b′ and1174c′ are offset from the fingers of the sets of fingers1174a,1174band1174cin a circumferential direction and a second portion1252′ of each of the fingers of the sets of fingers1174a′,1174b′ and1174c′ faces a second portion1252of an associated one of the fingers of the sets of fingers1174a,1174band1174c. Construction in this manner permits each terminal1150on each power semiconductor1082to be mechanically and electrically coupled to a second portion (e.g., second portion1252) of a finger on a first busbar portion (e.g., first busbar portion1160c) and to a second portion (e.g., second portion1252′) on a finger on a second busbar portion (e.g., second busbar portion1162c). With reference toFIGS.65,66and68, the positive busbar1090can be received within the first side wall1126in the inverter mount1080such that the positive busbar1090is abutted against the second side of the base1120and each adjacent pair of fingers of the sets of fingers1174aand1174a′ is engaged to a first one of the terminals1150on a respective one of the power semiconductors1082. A first electrically insulating member can be disposed over the positive busbar1090. The ground busbar1092can be received within the first side wall1126in the inverter mount1080such that the ground busbar1092is abutted against a side of the first electrically insulating member that is opposite the positive busbar1090and each adjacent pair of fingers of the sets of fingers1174band1174b′ is engaged to a second one of the terminals1150on a respective one of the power semiconductors1082. A second electrically insulating member can be disposed over the ground busbar1092. Each of the phase busbars1094can be positioned within the first side wall1126and abutted against the second insulator on a side of the second insulator that is opposite the ground busbar1092and each adjacent pair of fingers of the sets of fingers1174cand1174c′ is engaged to either third or fourth one of the terminals1150on a respective one of the power semiconductors1082. The positive busbar1090, the ground busbar1092and each of the phase busbars1094can be oriented relative to the inverter mount1080such that the terminal receptacles1122and an adjacent one of the sensor receptacles1124are received through each of the terminal apertures1182and the cooling standpipe aperture1184is received over the cooling standpipe1200. It will be appreciated that assembly of the several busbars to the power semiconductors in this manner will position the distal ends (i.e., arms1256(FIG.71)) of each adjacent pair of the fingers in abutment with an associated one of the terminals1150on an associated one of the power semiconductors1082. Thereafter, the distal ends of the fingers can be fixedly and electrically coupled to the terminals1150, for example by resistance welding or resistance soldering adjacent pairs of them to the terminals1150. InFIGS.65and66, the motor control unit42(FIG.64) includes a plurality of current sensors1300, each of which being associated with a corresponding set of the terminal receptacles1122, the sensor receptacles1124, and the bridges1210on the phase busbars1094. Each of the current sensors1300comprises a sensor1302and a plurality of C-shaped sensor laminations1304. The sensor1302can be an eddy current sensor and can be received in the sensor receptacle1124. The sensor laminations1304are abutted against one another and are received about an associated one of the terminal receptacles1122and axially between the windings1042and the bridge member1212on an associated one of the phase busbars1094. Accordingly, the open ends of the C-shaped sensor laminations1304can straddle the portion of the sensor receptacle1124where the sensor1302is located. The terminals1150of the power semiconductors1082and the terminals of the sensor1302are received into the inverter circuit board1096and can be electrically coupled to other componentry of the inverter circuit board1096in a desired manner. The phase terminals1044are received into the terminal receptacles1122and the seal62that is disposed in the seal groove1058in each of the phase terminals1044forms a seal between the phase terminal1044and the inverter mount1080that inhibits the flow of fluid therethrough. The threaded fastener66is received through the fastener aperture in the bridge member1212and is threadably engaged to the connecting feature1060to fixedly and electrically couple the phase terminal1044to the phase busbar1094. During operation of the electric motor40(FIG.64), current passing from the bridge member1212of the phase busbar1094to the phase terminal1044will generate a magnetic field that will correspondingly generate eddy currents in the sensor laminations1304. The sensor1302is configured to sense the eddy currents in the sensor laminations1304and responsively generate a sensor signal that is indicative of a magnitude of eddy currents in the sensor laminations. Significantly, the current sensor1302is positioned in as close a proximity to the interface between the phase busbar1094and the phase terminal1044as is possible. When configured in this manner, the phase terminals1044are disposed radially inwardly of the positive and ground busbars1090and1092, and the current sensors1300are disposed radially inwardly of the power semiconductors1082. While the inverter1020has been illustrated and described as including power semiconductors1082that are disposed about an outer perimeter of the several busbars, it will be appreciated that the teachings of the present disclosure have application to inverters that are configured somewhat differently, such as a configuration where the power semiconductors are disposed about an inner perimeter of the several busbars. With reference toFIGS.73and74, an exemplary phase busbar1094ais illustrated to include first and second busbar portions1160c-1and1162c-1, respectively, that each include sets of fingers1174c-1and1174c-1′, respectively that are disposed about an inside perimeter of the bodies1170c-1and1170c-1′, respectively, of the first and second busbar portions1160c-1and1162c-1. The foregoing description of the embodiments has been provided for purposes of illustration and description. It is not intended to be exhaustive or to limit the disclosure. Individual elements or features of a particular embodiment are generally not limited to that particular embodiment, but, where applicable, are interchangeable and can be used in a selected embodiment, even if not specifically shown or described. The same may also be varied in many ways. Such variations are not to be regarded as a departure from the disclosure, and all such modifications are intended to be included within the scope of the disclosure. | 71,445 |
11863047 | DETAILED DESCRIPTION OF EMBODIMENTS Embodiments of the present teachings are explained below, with reference to the drawings. Explanation of a Representative Lawn Mower FIG.1is a center, longitudinal, cross-sectional view that shows a rechargeable lawn mower1, which is one example of an electric work machine according to the present teachings, andFIG.2is an enlarged view of a motor unit portion thereof. The lawn mower1comprises: a base (deck)2, which extends in a rear-front direction and has an open lower surface; a main body3, which is coupled to a center upper side of the base2; and a handle4, which extends from the base2rearward and diagonally upward. The base2has two pairs of wheels5,5, one pair forward and one pair rearward, and can be moved forward and rearward by using the handle4. Downward of the handle4, a rear cover6and a grass-collection basket (grass catcher)7are provided on a rear portion of the base2. A switch lever8is provided on a rear end of the handle4; and forward thereof, a lock-OFF button9is provided that, in a normal state, locks the operation of the switch lever8. Pressing the lock-OFF button9unlocks the switch lever8, so that it becomes possible to pull the switch lever8. The main body3comprises a main-body housing (cowling)10, which has a lower end tubular part11that opens downward and protrudes into the base2. A battery-mount part12, into which one or more battery packs13that serve as a power supply for the lawn mower1can be inserted from a rear upper side, is formed on an upper part of the main-body housing10in an inclined manner such that it is lower in the front. The battery-mount part12is openable and closable by a battery cover14. In addition, on a front part of the main-body housing10, a controller15comprising a control circuit board (not shown) is supported such that it extends vertically in an up-down direction. Rearward thereof, a motor unit16is provided downward of the battery-mount part12. A rotary shaft25of a brushless motor21, which is described below, protrudes downward from the motor unit16, and a spindle17is coaxially coupled to a lower end of the rotary shaft25. The spindle17protrudes downward from the tubular part11into the base2, and a horizontal, plate-shaped cutting blade20is orthogonally attached to a lower end of the spindle17by using an inner flange18and a bolt19. The spindle17is one non-limiting example of an output part according to the present teachings. As shown inFIGS.2-4, the motor unit16comprises the brushless motor21and a motor case22, which holds the brushless motor21. The brushless motor21is an inner-rotor type that comprises a tubular-shaped stator23and a rotor24, which passes through the interior of the stator23and has the rotary shaft25at its axial center. The motor case22comprises an upper case26and a lower case27, which hold the stator23from above and below and axially support the rotary shaft25. The lower case27of the motor case22is joined to a mounting base28, which is provided on an upper side of the tubular part11. A motor cover29, which covers the motor unit16from above, is provided upward of the mounting base28. The upper case26and lower case27are typically made of metal. On the other side of the motor unit16, a bearing retainer30, which axially supports the spindle17via a bearing31, is joined, from below by a plurality of screws32, to a lower side of the mounting base28. A lower end of the spindle17passes through the bearing retainer30and also passes through a baffle plate33, which is screwed onto a lower end of the tubular part11, so as to protrude into the interior (grass cutting) space defined by the base2. A tube part34mates with the lower end of the spindle17, and is provided on the inner flange18, on which the cutting blade20is mounted. A centrifugal fan35is provided on the outer circumference of the tube part34. Explanation of a Representative Stator As shown inFIGS.4-7, the stator23of the brushless motor21comprises a stator core40composed of a plurality of steel plates40alaminated (stacked) in an axial direction (FIG.8), and a plurality of (here, twelve) teeth41protrude inwardly. An upper insulator42and a lower insulator43, which are made of resin or polymer, serve as electrically insulating members, and are integrally formed on both the upper and lower ends of the stator core40. An insulation part44, which is also made of resin or polymer, is continuous with and integrally formed with the upper and lower insulators42,43, i.e. insulating parts42-44are made from the same integral piece of resin or polymer. The insulation part44covers an inner-circumferential surface of the stator core40and an outer-circumferential surface of the teeth41, except for the protruding end surfaces of the teeth41. Coils45are respectively wound around each tooth41through (adjacent) the insulation part44. A short-circuiting member46, which is electrically connected to the wires that form the coils45and that forms (defines) a three-phase connection, and a sensor circuit board47, which detects the rotational position of the rotor24, are joined (attached) to the upper insulator42. Further details concerning the upper and lower insulators42,43, the short-circuiting member46, and the sensor circuit board47are provided below. Three ridges48A,48A,48B are formed on a circumferential surface of the stator core40such that they are equispaced in the circumferential direction. Among these, the ridges48A,48A each have a tapered transverse-cross-sectional shape in which the width in the circumferential direction becomes small toward the outer side of the stator core40in the radial direction. On the other hand, the ridge48B is not tapered, but rather has a quadrilateral, transverse-cross-sectional shape in which its width in the circumferential direction does not vary along the radial direction. Slight rounds (curved surfaces), which expand outward along the circumferential direction, are provided on radially outward end surfaces of the ridge48B. In addition, a through hole49is formed in each of the ridges48A,48B. The ridges48A,48B are formed such that they overlap projections50A,50B formed on each of the steel plates40aas shown inFIG.8. Among these, the projections50A are each formed with a taper in which the width in the circumferential direction becomes small toward the outer side of the stator core40in the radial direction. On the other hand, the projection50B is not tapered but rather has a quadrilateral shape in which the width in the circumferential direction does not vary along the radial direction. Pass-through holes51are formed in each of the projections50A,50B. Slight rounds (curved surfaces), which expand outward along the circumferential direction, are provided on radially outward end edges of the projection50B. In addition, a notch52is formed between each adjacent pair of the projections50A,50B, and a grooves53(FIG.7) for positioning during manufacture of the stator is formed between each adjacent pair of the ridges48A,48B. The ridges48A are tapered in this way to avoid interference with a mold at the time that the upper and lower insulators42,43and the insulation part44are integrally formed by insert molding. That is, as shown by a chain double-dashed line inFIG.8, when the upper and lower insulators42,43and the insulation part44are to be integrally formed on (joined to) the stator core40by using left and right split molds54,54, the two tapered ridges48A,48A are positioned, one on the left and one on the right, with respect to the split molds54,54, and the ridges48A,48A are set such that they do not interfere, owing to their tapered surfaces, with the split molds54,54, which move in the left and right directions inFIG.8. In the present embodiment, although the angle θ of the taper with respect to the movement direction is set to 3°, but the angle θ of the taper with respect to the movement direction may be set, e.g., within a range of 1°-10°. In addition, because rounds that expand (widen) outward also are provided on the end surfaces of the ridge48B, interference with the left and right split molds54,54is prevented. Referring now toFIGS.2and5-7, the rotary shaft25of the rotor24passes through the axial center of a circular-cylindrical-shaped rotor core55, which is composed of a lamination of a plurality of steel plates in the axial direction. The rotor core55and the rotary shaft25are integrally joined (connected, held) together by a resin56. A bevel part57(FIG.4) is formed on the lower end of the rotary shaft25. The resin56may also be referred to as a resin insert, a resin sleeve, a resin bushing, a polymer insert, a polymer sleeve, a polymer bushing, etc., or any variations or combinations thereof. The resin56is thus not limited to naturally occurring resins and may comprise natural and/or synthetic polymers. Furthermore, the basic requirement of the resin56is that it is interposed between the rotor core55and rotary shaft25, which are both typically made of a metal, and serves to connect or hold them together, so that the rotor shaft25rotates integrally with the rotor core55, i.e. they are rotationally-fixed. The resin56also is preferably design to prevent slippage of the rotary shaft25relative to the rotor core55in or along the axial direction of the rotary shaft25. The resin or polymer that constitutes the resin56optionally may be reinforced with fibers, e.g., glass fibers, carbon fibers, etc., and preferably exhibits electrical insulation properties. For example, the resin56preferably has a resistivity of 1×1010Ω·m or higher, more preferably 1×1012Ω·m or higher, and/or a conductivity of 1×10−10σ or less, more preferably 1×10−12σ or less. As shown inFIG.5, a diamond knurl25aor other type of knurling or gripping pattern may be formed on the outer circumference of the rotary shaft25along all or only a portion of the length of the rotary shaft25that contacts the resin56. The knurl25aprovides an unevenness or roughness that is formed, e.g., in a lattice or grid shape on the outer-circumferential surface of the rotary shaft25. The knurl25aacts as a rotation-impeding part and/or a slippage-impeding part, because the knurl25aenables the resin56to securely grip the rotary shaft25, so that no rotation of the rotary shaft25relative to the resin56is possible and/or so that no axial slippage/movement of the rotary shaft25relative to the resin56is possible. The knurl25ashown inFIG.5is merely one example of an uneven surface that can perform one or both of the rotation-impeding function and/or the axial slippage-impeding function, and additional examples will be provided below. In addition, multiple (here, eight) magnet holes58are concentrically formed in a circumferential-edge part of the rotor core55such that they pass through in the axial direction thereof. Plate-shaped permanent magnets59are embedded (inserted) in the magnet holes58. By forming through holes in the steel plates40a, except at the upper and lower ends, that are aligned in the axial direction, spaces (cutouts, voids)60are formed radially inward of the permanent magnets59, which reduces the weight of the rotor24. Explanation of a Representative Motor Case The upper case26and the lower case27of the motor case22are each circular-cup-shaped and cover an upper part and a lower part, respectively, of the stator23. The upper case26is formed of a nonmagnetic material, e.g., a metal such as an aluminum alloy. As shown inFIGS.3-5, fins65for dissipating heat are provided, from the upper surface of the outer circumference down along the side surface of the upper case26and extending in the up-down direction, at prescribed spacings in the circumferential direction. In addition, an upper-bearing retaining part66is formed at the center of the upper surface of the upper case26. A bearing68is held in the upper-bearing retaining part66by an insulating cap67, which is made of resin or polymer, and rotatably supports an upper end of the rotary shaft25. A pass-through hole69is formed at the center of the upper-bearing retaining part66and is closed up by a resin or polymer cap70. The resin or polymer of the insulating cap67and the resin cap70may be selected from any of the resins described above with regard to resin56, which description is equally applicable to the insulating cap67and resin cap70. Furthermore, three screw-boss parts71A,71A,71B, which project radially outward, are formed on the circumferential surface of the upper case26such that they extend in the up-down direction and are equispaced in the circumferential direction. The screw-boss parts71A,71B correspond to the ridges48A,48B of the stator core40. In particular, the lower ends of the screw-boss parts71A,71B are open and have either a tapered shape in transverse cross section or a quadrilateral shape in transverse cross section that mates with the respective ridges48A,48B. A slit72, which extends upward from a lower end of the upper case26, is formed in or on the circumferential surface of the upper case26between the ridges48A,48B. Turning now to the lower case27, it comprises a circular-shaped end surface part73, in which a lower-bearing retaining part74is formed at the center, the same as in the upper case26. A tubular part75rises upward from the outer circumference of the end surface part73. A bearing76is held by the lower-bearing retaining part74and supports the rotary shaft25, which passes through the lower-bearing retaining part74. Bosses77for fastening screws to the mounting base28are formed on the outer circumference of the tubular part75so as to point downward at four locations equispaced in the circumferential direction. In addition, a resin layer78is formed on (joined to) an inner surface of the end surface part73(except for the lower-bearing retaining part74), an inner circumference of the tubular part75, and an outer circumference of the tubular part75(except for on the bosses77), such that the resin layer78continuously covers from the inner surface of the end surface part73to the inner circumference and then to the outer circumference of the tubular part75. Boss parts79A,79B have shapes the same as the corresponding screw-boss parts71A,71B of the upper case26, and are formed, extending in the axial direction, at locations of the resin layer78corresponding to the ridges48A,48B of the stator core40. Through holes, which have tapered shapes in transverse cross section or quadrilateral shapes in transverse cross section and mate with the respective ridges48A,48B, are formed in the upper ends of the boss parts79A,79B. Furthermore, recessed grooves80are formed continuously on the lower sides of the boss parts79A,79B. Again, the resin of the resin layer78may be selected from any of the resins or polymers described above with regard to resin56, which description is equally applicable to the resin layer78. Thus, the upper case26of the motor case22is placed onto the upper portion the stator23by aligning the ridges48A,48B of the stator core40with the respective screw-boss parts71A,71B of the upper case26and then inserting the ridges48A,48B into the corresponding screw-boss parts71A,71B. The upper bearing68, which is joined to the upper end of the rotary shaft25of the rotor24, is held by the upper-bearing retaining part66. On the other side, the lower case27placed on the lower portion of the stator23by aligning the ridges48A,48B of the stator core40with the boss parts79A,79B of the lower case27and then inserting the ridges48A,48B into the corresponding boss parts79,79B. The lower bearing76, which is joined to the lower end of the rotary shaft25, is held by the lower-bearing retaining part74. In this assembled state, screws81are inserted, from below, into the boss parts79A,79B of the lower cases27, then passed through the ridges48A,48B, and are screwed into the screw-boss parts71A,71B of the upper case26. As a result, the brushless motor21is covered by the upper case26and the lower case27, except for an intermediate portion of the outer circumference of the stator core40, and thereby the motor unit16is obtained. In this state, the brushless motor21contains a basic-insulation member (the upper and lower insulators42,43and the insulation part44, which are integrally formed), which is interposed between the stator core40, which has a metal interior, and the coils45, which are energized (supplied with current) during operation of the brushless motor21. In addition thereto, supplementary electrical insulation is provided by: (i) the resin56, which serves as an insulating member on the rotary-shaft side and is interposed between the rotary shaft25and the rotor core55, (ii) the insulating cap67, which is interposed between the upper case26and the rotary shaft25, and (iii) the resin layer78, which is interposed between the tubular part75of the lower case27and the stator core40. Therefore, the space between the stator core40and the rotary shaft25is double insulated. In addition, by providing an adjustable gap in the up-down direction between the upper case26and the lower case27, assembly of the motor unit16is not negatively influenced even if the dimension (length) of the stator23in the axial direction changes. When the motor unit16is placed, with the rotary shaft25facing downward, on the mounting base28and screws are screwed into the bosses77from below the mounting base28, the motor unit16is fixed to the mounting base28. Concentric arcuate ribs73a(FIGS.2,5), which mate with the tubular part11to position the motor unit16, are formed on the lower surface of the end surface part73of the lower case27. In the present embodiment, when the motor cover29is put on, the motor unit16is mostly covered while a center portion of the upper case26that includes the upper-bearing retaining part66is exposed. In this state, the fins65of the upper case26are proximate to the inner surface of the motor cover29. Furthermore, the stator23of the brushless motor21is impeded (blocked) from rotating relative to the motor case22by the screws81, which pass through the ridges48A,48B, and also by the screw-boss parts71A,71B of the upper case26and the boss parts79A,79B of the lower case27, which respectively mate with (engage) the ridges48A,48B. Explanation of Representative Upper and Lower Insulators Referring now toFIGS.9and10, the upper insulator42is a ring body that is integrally formed on (joined to) an upper-side end surface of the stator core40. Twelve terminal-holding parts85, which respectively hold fusing terminals99provided on the short-circuiting member46, are provided on an upper surface of the upper insulator42equispaced in the circumferential direction. In each of the terminal-holding parts85, an inner-wall part86on the inner-circumference side and an outer-wall part87on the outer-circumference side extend vertically and are spaced apart radially by a spacing (distance) that substantially corresponds to the diameter of wires115. A mating groove88, which mates with its corresponding fusing terminal99, is formed between the inner-wall part86and the outer-wall part87at the center in the circumferential direction. In addition, stop bosses89for joining to (attaching) the short-circuiting member46protrude from the upper surface of the upper insulator42at five locations, i.e., at locations at which they contact the base of every other tooth41. As shown inFIG.11, the lower insulator43is a ring body that is integrally formed on (joined to) the lower-side end surface of the stator core40. Twelve vertically-extending guide walls90are provided along the circumferential direction on the lower surface of the lower insulator43at locations slightly shifted in the circumferential direction from the bases of the teeth41. Explanation of a Representative Short-Circuiting Member and Sensor Circuit Board Still referring toFIGS.9-10, the short-circuiting member46includes a ring body made of resin, polymer, etc. and has a circumference that is smaller than the circumference of the upper insulator42. Five mating bosses95, which are quadrilateral-tube-shaped and respectively mate, from above, with the stop bosses89of the upper insulator42, and three ribs96, which respectively engage with the grooves53of the stator core40, protrude from the outer circumference of the short-circuiting member46. In addition, the short-circuiting member46is formed in steps such that its thickness in the axial direction becomes smaller in steps, starting from the upper surface, from the outer circumference toward the inner circumference. Furthermore, as shown inFIGS.12-14, a first metal fitting97U having the maximum diameter and that is located in an outer-circumferential portion having the greatest wall thickness, a second metal fitting97W having an intermediate diameter and that is located in an intermediate-wall-thickness portion on the inner side thereof, and a third metal fitting97V having the minimum diameter and that is located in an inner-circumference portion on the inner side thereof are concentrically disposed in the thickness portions and insert molded. The letters U, W, and V appended to the metal fittings indicated the corresponding phases of the three-phase current: U phase, W phase, and V phase. Each of the first to third metal fittings97U-97V is a strip-shaped, curved plate having a substantially C shape in plan view. Protruding pieces98protrude radially outward at four locations, namely, at both ends and at locations point symmetric with the two ends of the metal fittings97U-97V. One of the fusing terminals99is formed at the tip of each protruding piece98by first bending it downward, then folding it upward, and further bending it outward. A welding part101for spot welding a power-supply line100U is formed at the base of the protruding piece98on (at) one end of the first metal fitting97U. In addition, welding parts101for spot welding power-supply lines100W,100V are formed on (at) the bases of the protruding pieces98of the second and third metal fittings97W,97V on (at) the ends on the side opposite that of the first metal fitting97U. In the state in which the first to third metal fittings97U-97V are disposed and insert-molded inside the resin ring body of the short-circuiting member46, starting from above, in the order of the first metal fitting97U, the second metal fitting97W, and the third metal fitting97V, such that their phases are shifted by a prescribed angle in the circumferential direction, the fusing terminals99respectively protrude from the outer-circumferential surface of the short-circuiting member46without contacting each other and are substantially equispaced in the circumferential direction. Pass-through holes102, which respectively expose the welding parts101of the metal fittings97U-97V, are formed in the short-circuiting member46such that the pass-through holes102are offset by prescribed spacings in one portion in the circumferential direction. The power-supply lines100U-100V are respectively spot welded to the welding parts101. A notch103for drawing the power-supply lines100U-100V to the outer side is formed between two of the welding parts101such that only the lower side of the short-circuiting member46is connected to (via) the notch103. In addition, support pieces104, which comprise mount bosses105for mounting the sensor circuit board47, radially inwardly protrude from the inner circumference of the short-circuiting member46at point-symmetric positions. Support pieces106(FIG.13), which support the outer circumference of the sensor circuit board47, radially inwardly protrude from the inner circumference of the short-circuiting member46between the support pieces104. As shown inFIG.10, the sensor circuit board47has an arcuate strip shape that extends around the inner side of the short-circuiting member46. Mating holes107, which respectively mate with the mount bosses105of the support pieces104, are formed on both circumferential ends of the sensor circuit board47. Because the mount bosses105mate with the mating holes107and the outer circumference of the sensor circuit board47is supported by the support pieces106, the sensor circuit board47is held on the inner-circumference side of the short-circuiting member46. Rotation-detection devices108(FIG.11), such as Hall-effect devices, which detect the magnetic fields of the permanent magnets59provided on the rotor24, are installed on a back surface of the sensor circuit board47. Signal lines109, which are connected to the sensor circuit board47, and the power-supply lines100U-100V are drawn through the notch103of the short-circuiting member46to the outer side. This drawn-out position corresponds to the slit72provided in the upper case26of the motor case22. When the five mating bosses95on the outer circumference are mated with the stop bosses89, which are provided on the upper surface of the upper insulator42, and are screwed to the stop bosses89from above using screws91(FIG.10), and when the tips of the three ribs96are engaged with the grooves53of the stator core40, and the fusing terminals99are caused to be held by the terminal-holding parts85of the upper insulator42, the short-circuiting member46is joined, together with the sensor circuit board47, to the stator23. In particular, because the ribs96engage with and hold fast to the grooves53at three locations, the ribs96function as anchors that stably support the short-circuiting member46. The power-supply lines100U-100V and the signal lines109are drawn out from the slit72, which is provided in the upper case26, to the exterior through a sleeve-shaped gasket82(FIG.3), which is fitted into the slit72. Explanation of a Representative Coil-Forming Method Twelve of the coils45herein are formed at the same time, using three winding nozzles, by starting windings, using a single wire115as shown inFIG.15(however, when distinguishing wires, the symbols A-C are appended, as in115A,115B,115C, and the same applies to other portions hereinbelow) on three of the teeth41located at 120° spacings, and winding, in order, the four teeth41adjacent in the circumferential direction of the stator23. For example, with regard to the wire115A shown inFIG.15, after a start end116A has been initially latched (attached) to the corresponding fusing terminal99, the coils45are formed, in order, on the teeth41adjacent in the clockwise direction. The winding direction at this time is the counterclockwise direction, facing the teeth41. In addition, a crossover wire117A after forming each coil45returns to the upper insulator42side (wiring-connection side) and latches (attaches) to the fusing terminal99between two of the teeth41,41. Furthermore, after the fourth coil45has been formed, as shown inFIG.11, the wire115A is first drawn out to the lower insulator43side (opposite wiring-connection side) and wound from the outer side of the guide wall90at the base of the tooth41being wound, after which the wire115A once again returns to the upper insulator42side, is latched (attached) to the fusing terminal99to which a start end116B of the separate adjacent wire115B is latched, and becomes a terminal end118A, as shown inFIG.15. Thereupon, the orientation of the start end116B of the separate wire115B and the orientation of the terminal end118A are made to coincide and can be simultaneously cut at the portion at which they are completely surrounded. This applies likewise for a terminal end118B of the wire115B and the start end116A of the wire115C, as well as a terminal end118C of the wire115C and the start end116A of the wire115A. The first to third metal fittings97U-97V of the short-circuiting member46are disposed such that their phases are shifted in the circumferential direction one coil45at a time. As shown inFIG.16, crossover wires117A-C, which are disposed between the twelve coils45, are each fused with respect to three adjacent coils45. InFIG.16, to make it easy to distinguish the crossover wires117A-C to which the metal fittings97U-97V are fused, linear hatching is applied to the first metal fitting97U, cross hatching is applied to the second metal fitting97W, and dots are applied to the third metal fitting97V. Thus, the three coils45adjacent in the circumferential direction are configured as a delta connection of U (W-U), V (U-V), W (V-W) phases by the first to third metal fittings97U-97V of the three phases. This is because four sets are sequentially arranged in parallel by the first to third metal fittings97U-97V; and the three-phase circuit herein is formed as shown inFIG.17. This is equivalent to a delta connection in which the four coils U1-U4, V1-V4, W1-W4of each of the U, V, and W phases are connected in parallel. Operation of the Representative Lawn Mower In the lawn mower1configured as described above, when the switch lever8is unlocked by pressing the lock-OFF button9and the switch lever8is pulled, a main switch turns ON and an ON signal is transmitted from the battery pack13to the control circuit board of the controller15. A microcontroller of the control circuit board acquires the rotational state of the rotor24based on detection signals obtained from the rotation-detection devices108of the sensor circuit board47, turns ON/OFF switching devices, which are provided on the control circuit board, in accordance with the acquired rotational state, and supplies electric current, in order, to the coils45, for each phase, of the stator23, and thereby rotates the rotor24. Thus, when the rotary shaft25rotates and causes the spindle17to rotate together with the cutting blade20, and when the base2is pushed using the handle4, it becomes possible to cut grass with the cutting blade20while the lawn mower1travels via the wheels5. At this time, the stator23of the brushless motor21is impeded (blocked) from rotating relative to the motor case22, which is joined to the mounting base28, by the screws81that pass through the ridges48A,48B. Therefore, any effects caused by manufacturing tolerances are small and it becomes possible to impede (block) relative rotation of the brushless motor21and the motor case22with good accuracy. In addition, high strength is also obtained. In particular, because the screws81pass directly through the stator core40, flexure tends not to occur on the outer side of the stator core40, as compared with a structure that couples upper and lower cases of a motor case using screws that do not pass through the stator core. Furthermore, because the wires115that form the coils45do not cross one another, scraping of the wires115caused by contacting each other tends not to occur, and therefore durability is also increased. Furthermore, in the rotor24, because the diamond knurl25ais provided on the outer circumference of the rotary shaft25, the bite or grip of the resin56is increased, thereby reducing or minimizing slippage in the rotational direction and the axial direction. Consequently, the integration (secure attachment) of the rotary shaft25and the rotor core55is maintained even if the load on the rotary shaft25produced by the rotation of the cutting blade20becomes large. Advantages of the Representative Stator Core In the lawn mower1of the above-mentioned embodiment, screw members (the screws81) pass through the stator core40of the brushless motor21and furthermore, the through holes49are formed to impede (block) the relative rotation of the stator23. Therefore, it is possible to impede (block) relative rotation of the stator23using the stator core40, which has high accuracy and high strength, instead of by using the upper and lower insulators42,43to impede (block) relative rotation. Thereby, the stator23can be impeded (blocked) from rotating relative to the motor case22with high accuracy and high strength. In addition, because cover members (the upper case26and the lower case27) are provided on both ends of the stator23in the axial direction in the present embodiment, and because the rotation-impeding parts (the ridges48A,48B), which mate with the upper case26and the lower case27, are provided on the outer circumference of the stator core40, rotation of the stator23relative to the motor case22can be impeded (blocked), using the stator core40, with high accuracy and high strength. Furthermore, because protruding parts (the ridges48A,48B) for impeding relative rotation by engaging with the cover members (the upper case26and the lower case27) are formed on the outer circumference of the stator core40, and because the through holes49, which are provided for the screws81to pass through, are formed in the ridges48A,48B, rotation of the stator23relative to the motor case22can be impeded (blocked), using the ridges48A,48B and the screws81, with high accuracy and high strength. Furthermore, because the ridges48A,48A have a tapered shape in transverse cross-section such that the width in the circumferential direction gradually narrows toward the outer side in the radial direction of the stator core40, interference with the split molds54can be prevented when the upper and lower insulators42,43and the insulation part44are being integrally formed. It is noted that, in the stator core, the number, shape, or the like of the ridges that impede (block) relative rotation of the stator is not limited to the above-mentioned embodiment. For example, the number of ridges can be increased or decreased, the transverse-cross-sectional shape can be modified where appropriate, or the like. In another modified example, the ridges do not necessarily have to be provided across the entire up-down length of the stator core and may instead be provided across a distance shorter than the overall length, such as an upper-end side, a lower-end side, an intermediate region, or the like. Relative rotation can also be impeded (blocked) by passing the screw members through the stator core, without providing the ridges. In addition, the rotation-impeding function is not limited to being effected by the ridges, the through holes, and the like. For example, as shown inFIG.18, axially-extending recesses61may be formed on (in) the outer-circumferential surface of the stator core40and the screws81may be mated (engaged) in the recesses61in order to impede rotation of the stator23relative to the motor case22. In this modified embodiment too, the stator23can be impeded (block) from rotating using the stator core40, which has high accuracy and high strength. Furthermore, the rotation-impeding function can also be effected by using the through holes and the recesses in combination. The screw members may be bolts. The target of the rotation impeding is also not limited to the motor case; that is, if there is no motor case, then rotation of the stator relative to the motor housing or the like may be effected using the through holes, the recesses, or the like. It should be noted that, although an electric work machine, in which a motor is fixed to a housing via a motor case was explained, the present teachings can also be applied to electric work machines in which a motor is fixed directly to the housing by fastening screws through through holes provided in the stator core. In such embodiments as well, the ridges may be mated with the housing. In addition, if rotation is impeded (blocked) by mating the ridges with the housing, it is also possible to fix the brushless motor by sandwiching the brushless motor between the half housings. In such an embodiment, the through holes may be omitted. Furthermore, in the above-described embodiment, although the motor case is formed from (comprises) the upper case and the lower case, the upper-lower arrangement is a positional relationship strictly for the sake of convenience, and there is no problem with respect to the electric work machine even if two half cases are arranged in the left-right direction, the forward-rearward direction, a diagonal direction, or the like. Furthermore, any one of the cover members alone, such as the upper case, may be fixed to the housing side. In such an embodiment, because the other cover member can be omitted, ease of assembly is improved. In addition, fins for heat dissipation may be provided on the stator core. Moreover, the heat-dissipating properties can also be improved by connecting structures to the motor case that have high heat-transfer (heat conductivity) properties. Advantages of the Representative Three-Phase Coil Connection Method In the lawn mower1of the above-mentioned embodiment, because the three phases of the coils45of the brushless motor21are delta connected, with each phase having four coils in parallel, the wire diameter of the wires115A-115C can be made narrower than in a star connection, even given the same output, and thereby winding characteristics during manufacture can be improved. In addition, because the winding nozzles can be narrowed, dead space can be reduced, which ultimately leads to an increase in output power. Here in particular, because the delta connection is formed by short circuiting, using the plurality of sheet-metal members (the first to third metal fittings97U-97V) mounted on the upper insulator42, the crossover wires117A-C between the coils45wound on the teeth41adjacent in the circumferential direction of the stator core40through the upper insulator42, a wiring connection becomes possible in which complex crossover wires, cross wires, and the like are not created. Thereby, productivity becomes high and, moreover, the risk of scraping of the wires115caused by contacting each other can be reduced. In addition, there are twelve of the coils45for the three phases, each of the three wires115is used to continuously form four coils45adjacent in the circumferential direction of the stator core40. Furthermore, a start end116A, B of one wire115and a terminal end118A, B, C of another wire115adjacent in the circumferential direction are electrically connected to each of the first to third metal fittings97U-97V with the same orientation relative to the short-circuiting member46. Therefore, the start ends116A-C and the terminal ends118of the wires115can be cut simultaneously, whereby productivity is further improved. It is noted that in the three-phase, coil-connection method, the wire-winding method is not limited to the above-described embodiment, and there is no problem even if the coils are formed with one, two, four, or six wires using one, two, four, or six winding nozzles. If there is one winding nozzle, then all twelve teeth are wound with one wire (12×1); if there are two winding nozzles, then six teeth are wound with two wires (6×2). In addition, if there are four winding nozzles, then three teeth are wound with four wires (3×4); and if there are six winding nozzles, then two teeth are wound with six wires (2×6). If there is one winding nozzle, then it takes time to wind the wire on the teeth; however, because the number of the winding nozzles is small, the equipment is compact and equipment expenses can also be kept low. As the number of the winding nozzles increases, the time needed to wind the wires on the teeth decreases; however, the equipment increases in size and equipment expenses also increase. In the above-mentioned embodiment, three of the winding nozzles are used because that it provides an advantageous balance between time and equipment for winding the wires. However, if more importance is attached to the advantages of equipment, then one or two of the winding nozzles should be used. On the other hand, if more importance is attached to reducing time requirements, then four or six of the winding nozzles should be used. In addition, each phase is not limited four in parallel; five or more in parallel may be used. Furthermore, the shape of the sheet-metal members is also not limited to the first to third metal fittings of the above-mentioned embodiment. For example, the width may be increased, some of the first to third metal fittings may be made to overlap in the axial direction without contacting, without being disposed concentrically, or the like. Advantages of the Representative Diamond Knurl of the Rotary Shaft In the lawn mower1of the above-mentioned embodiment, the slip torque (grip) between the rotary shaft25and the rotor core55can be increased by the provision of the diamond knurl25a, which constitutes the rotation-impeding part in the rotational direction relative to the resin56and the slippage-impeding part in the axial direction relative to the resin56, on the outer circumference of the rotary shaft25. Here in particular, by using the diamond knurl25a, slip torque (grip) in the rotational direction and slip torque (grip) in the axial direction can be improved at the same time. It is noted that the rotation-impeding part and the slippage-impeding part are not limited to a shape that acts upon both, as in the diamond knurl, and it is also possible to provide only one of the rotation-impeding part and the slippage-impeding part. For example, as the rotation-impeding part, a straight knurl25bas shown inFIG.19Amay be provided. In addition, the grip (unevenness) on the rotary shaft25is not limited to knurling, and it is also possible to impede rotation by making the transverse-cross-sectional shape of the rotary shaft25into a shape other than a circular shape.FIGS.19Band C show, as differently shaped parts, examples in which a bevel part25cextending in the axial direction is formed at one or two locations, andFIG.19Dshows an example in which a V-shaped groove25dextending in the axial direction is formed. However, the number of bevel parts may be formed at three or more locations, and the number of the grooves also may be increased. The groove may have a shape other than a V shape. Furthermore, the differently shaped part is not limited to the bevel part, the groove, or the like, and the transverse-cross-sectional shape may be a quadrilateral shape, a polygonal shape, or the like and may be an elliptical shape, an oval shape, or the like. On the other hand, as the slippage-impeding part, as shown inFIG.19E, ring grooves25eextending in the circumferential direction of the rotary shaft25can also be formed at prescribed spacings in the axial direction. There may be one or three or more of the ring grooves. Moreover, the grooves may be provided only partially in the circumferential direction, i.e. in a ring (annular) shape. Furthermore, the rotation-impeding part and the slippage-impeding part can also be combined. For example, if the ring groove is combined with the straight knurl or if the ring groove is combined with a differently shaped part configured as a bevel part, a groove, or the like, then the effect of both rotation impeding and slippage impeding are obtained. Advantages of the Representative Insulating Means Between the Stator Core and the Rotary Shaft The lawn mower1of the above-described embodiment comprises: the main-body housing10(an exterior housing); the motor case22(an interior case) fixed inside the main-body housing10; the brushless motor21housed inside the motor case22and comprising the stator23having the stator core40, the coils45, and the upper and lower insulators42,43, and the rotor24disposed inward of the stator23and having the rotary shaft25; and the spindle17(an output part) driven by the rotary shaft25. The motor case22holds the stator23and axially supports the rotary shaft25via bearings68,76. The insulating cap67and the resin layer78(which are each an insulating means) provide electrical insulation between the stator core40and the rotary shaft25. Therefore, even if the brushless motor21is housed in the (metal) motor case22inside the main-body housing10, double insulation becomes possible in which the conducting pathway from the (metal) stator core40through the (metal) motor case22to the (metal) rotary shaft25can be effectively insulated. Here in particular, because the resin56(the insulating member on the rotary-shaft side) is interposed between the rotor core55and the rotary shaft25, which are provided in the rotor24, more effective insulation becomes possible. In addition, because the insulating means may include the insulating cap67(bearing-side insulating member), which is provided at a portion of the motor case22that rotatably supports the rotary shaft25via the bearing68, and/or the resin layer78(stator-side insulating member), which is provided between the motor case22and the stator core40, the insulating means can be formed effectively. Furthermore, if the resin layer78is integrally formed on (joined to) the lower case27of the motor case22, it becomes possible to form the resin layer78in a simple manner at the time of manufacturing the lower case27. It is noted that the insulating means according to the present teachings is not limited to the insulating cap67and the resin layer78of the above-described embodiment. For example, as the bearing-side insulating member, a resin material (electrically insulating resin or polymer material) may be interposed between the bearing and the rotary shaft. In addition or in the alternative, as the stator-side insulating member, an electrically insulating resin or polymer layer may be formed on the outer side of the rotor core. The bearing-side insulating member can also be integrally formed with (on) the motor case. In addition or in the alternative, either one or both of the upper case and the lower case of the motor case can also be made of resin or polymer. Furthermore, the bearing-side insulating member and/or the stator-side insulating member can also be used in combination. Either one of the bearing-side insulating member and the stator-side insulating member can also be omitted. In another embodiment of the present teachings, a compressor120is schematically shown inFIG.20. In the compressor120, the stator23of the brushless motor21is supported on the outer side of a housing121via a stator-support member122, which has a bottomed-tube shape. Furthermore, the rotary shaft25extends into the housing121and is connected to an output part (not shown), which may be e.g., a piston, rotary screw, vane, scroll, etc. In addition, a first insulating member123is interposed between the stator-support member122and the stator core40and a second insulating member124is interposed between the rotary shaft25and the rotor core55, thereby providing double insulation between the stator core40and the rotary shaft25on one side and the housing121on the other (here, although not shown, basic insulation is also implemented by interposing a basic-insulation member between the coils and the stator core, as in the above-described embodiment). It is noted that the housing121and the stator-support member122herein may be integral. In addition or in the alternative, the rotation-impeding part and/or the slippage-impeding part, such as a diamond knurl, also may be fabricated on the rotary shaft25. Finally, it should be noted the present teachings are applicable to other types of electric work machines, i.e. other than a lawn mower, a compressor, or the like. For example, the present teachings may be suitably applied to gardening tools (e.g., outdoor power equipment), such as electrically powered chain saws, hedge trimmers, mowing machines, blowers, and the like, as well as to other types of power tools, such as angle drills, grinders, hammers, hammer drills, circular saws, reciprocating saws (“recipro saws”), and the like. Thus, in such embodiments, the output part of the electric work machine may be e.g., a saw chain, shears, a blower fan, a tool chuck, a tool holder, a disk (e.g., a grinding, sanding or polishing disk), a cutting blade, a piston, etc. Representative, non-limiting examples of the present invention were described above in detail with reference to the attached drawings. This detailed description is merely intended to teach a person of skill in the art further details for practicing preferred aspects of the present teachings and is not intended to limit the scope of the invention. Furthermore, each of the additional features and teachings disclosed above may be utilized separately or in conjunction with other features and teachings to provide improved brushless motors and electric work machines that utilize such brushless motors. Moreover, combinations of features and steps disclosed in the above detailed description may not be necessary to practice the invention in the broadest sense, and are instead taught merely to particularly describe representative examples of the invention. Furthermore, various features of the above-described representative examples, as well as the various independent and dependent claims below, may be combined in ways that are not specifically and explicitly enumerated in order to provide additional useful embodiments of the present teachings. All features disclosed in the description and/or the claims are intended to be disclosed separately and independently from each other for the purpose of original written disclosure, as well as for the purpose of restricting the claimed subject matter, independent of the compositions of the features in the embodiments and/or the claims. In addition, all value ranges or indications of groups of entities are intended to disclose every possible intermediate value or intermediate entity for the purpose of original written disclosure, as well as for the purpose of restricting the claimed subject matter. EXPLANATION OF THE REFERENCE NUMBERS 1Lawn mower2Base3Main body4Handle10Main-body housing15Controller16Motor unit17Spindle20Cutting blade21Brushless motor22Motor case23Stator24Rotor25Rotary shaft25aDiamond knurl26Upper case27Lower case40Stator core40aSteel plate41Tooth42Upper insulator43Lower insulator45Coil46Short-circuiting member47Sensor circuit board48A,48B Ridge50A,50B Projection54Split mold55Rotor core56Resin (representative insulating member)65Fin67Insulating cap (representative insulating means)71A,71B Screw-boss part78Resin layer (representative insulating means)79A,79B Boss part81Screw85Terminal-holding part97U-97V First to third metal fittings99Fusing terminal115Wire120Compressor122Stator-support member123,124Insulating member | 50,173 |
11863048 | It should be understood that the appended drawings are not necessarily to scale, presenting a somewhat simplified representation of various preferred features illustrative of the basic principles of embodiments of the disclosure. The specific design features of embodiments of the present disclosure as disclosed herein, including, for example, specific dimensions, orientations, locations, and shapes will be determined in part by the particular intended application and use environment. In the figures, reference numbers refer to the same or equivalent sections of the present disclosure throughout the several figures of the drawings. DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS Hereinafter, preferred embodiments of the present disclosure will be described in detail with reference to the accompanying drawings. Specific structures or functions described in the embodiments of the present disclosure are merely for illustrative purposes. Embodiments according to the concept of the present disclosure may be implemented in various forms, and it should be understood that they should not be construed as being limited to the embodiments described in the present specification, but include all modifications, equivalents, or substitutes included in the spirit and scope of the present disclosure. It will be understood that, although the terms “first,” “second,” etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another element. For instance, a first element discussed below could be termed a second element without departing from the teachings of the present invention. Similarly, the second element could also be termed the first element. It will be understood that when an element is referred to as being “coupled” or “connected” to another element, it can be directly coupled or connected to the other element or intervening elements may be present therebetween. In contrast, it should be understood that when an element is referred to as being “directly coupled” or “directly connected” to another element, there are no intervening elements present. Other expressions that explain the relationship between elements, such as “between,” “directly between,” “adjacent to,” or “directly adjacent to,” should be construed in the same way. Like reference numerals denote like components throughout the specification. In the meantime, the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting. As used herein, the singular forms “a,” “an,” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprise,” “include,” “have,” etc., when used in this specification, specify the presence of stated components, steps, operations, and/or elements, but do not preclude the presence or addition of one or more other components, steps, operations, and/or elements thereof. Hereinafter, embodiments of the present disclosure will be described in detail with reference to the accompanying drawings. As illustrated inFIGS.2and3, a rotor according to an embodiment of the present disclosure includes a rotor core10, a plurality of permanent magnets20, and one or more balancing structures30. The permanent magnets20are interior mounted and inserted into the rotor core10. In particular, an embodiment of the present disclosure provides the rotor for an interior permanent magnet synchronous motor. Each permanent magnet20is inserted at a preset position of the rotor core10. The permanent magnets20may be disposed at a preset interval and positioned along the circumference of the rotor core10. In particular, the permanent magnets20are mounted on an outer circumference portion of the rotor core10. The balancing structure30is provided on the rotor core10in advance. In other words, the balancing structure30may be formed in advance before the presence of mass imbalance in the rotor core10is determined. For example, the balancing structure30may be included in a mold of the rotor core10. The part of the balancing structure30may be easily removed when it is identified that the mass imbalance exists. When the balancing structure30is formed as described above, only the corresponding portion may be easily removed without modifying a neighboring shape upon balancing operation. The balancing structure30may be provided on an inner circumference portion of the rotor core10. A certain gap40is provided between the balancing structure30and the rotor core10. According to an implementation example of an embodiment of the present disclosure, the balancing structure30includes a first portion32and a second portion34. The first portion32is formed to be spaced apart from the rotor core10by the gap40. As a non-limiting example, a cross section of the first portion32may have a circular shape. In other words, the first portion32having the circular cross section extends in the axial direction of the rotor core10. The second portion34connects the first portion32to the rotor core10. The first portion32extends the gap40to connect the first portion32to the rotor core10. According to an implementation example of an embodiment of the present disclosure, the gap40is substantially similar to a C shape. According to some embodiments of the present disclosure, the balancing structure30includes the first portion32having the circular cross section and the second portion34having a rectangular cross section that has a cross section smaller than that of the first portion32. When the balancing structure30has such a shape, the balancing structure30, that is, the first portion32may be very easily separated from the rotor core10, and it is also possible to largely increase a balancing speed. According to an implementation example of an embodiment of the present disclosure, the balancing structure30is formed integrally with the rotor core10. Therefore, according to embodiments of the present disclosure, the end plate required for the balancing may be omitted, thereby reducing the weight and size of the motor. The balancing structure30is formed at a position with a low magnetic flux density. In particular, the balancing structure30may be provided around the inner circumference portion of the rotor core10. It is confirmed that the portion between the permanent magnet and the inner circumference of the rotor core10has a low magnetic flux density. According to embodiments of the present disclosure, the balancing structure30is formed on the portion with the low magnetic flux density, thereby not interfering with the flow of a magnetic path and barely affecting the output. According to an implementation example of an embodiment of the present disclosure, the rotor core10includes a plurality of balancing structures30. According to embodiments of the present disclosure, the plurality of balancing structures30may be provided. The plurality of balancing structures30may be spaced apart from each other at predefined intervals along the inner circumference portion of the rotor core10. As a result, it is possible to remove the part of the balancing structure30at a position where mass imbalance exists, thereby resolving the imbalance. According to embodiments of the present disclosure, the balancing structure30may be provided on the entire circumference of the inner circumference portion of the rotor core10to increase the possibility that a position where the balancing is required matches with the position of the balancing structure30. Therefore, it is possible to resolve the imbalance of the motor easily and more accurately. According to embodiments of the present disclosure, the balancing structure30is provided on the inner circumference portion of the rotor core10regardless of weight reduction holes generally formed in the motor for reducing the weight of the motor. However, the balancing structure30may also result in reducing the weight of the motor. The rotor including the specific shape, number, and placement of the balancing structure30according to embodiments of the present disclosure may obtain the effective balancing results even while facilitating the balancing operation. According to an implementation example of an embodiment of the present disclosure, the first portion32and the second portion34extend outward in a radial direction (R) of the rotor core10. In other words, the first portion32and the second portion34may be formed outward in the radial direction of the rotor core10around the rotation axis of the rotor core10. Referring toFIG.4, as described above, the rotor according to embodiments of the present disclosure does not include the end plate. Since the balancing against the mass imbalance is performed by the balancing structure previously formed on the rotor core10, it is not necessary to provide the end plate. As a result, according to embodiments of the present disclosure, it is possible to reduce the cost, weight, and size of the motor. As illustrated inFIG.5, a method for correcting a rotation imbalance of the rotor according to an embodiment of the present disclosure may be executed by a series of steps. In step S100, the balancing structure30is formed on the rotor core10. The balancing structure30is provided on the inner circumference portion of the rotor core10where the magnetic flux density is small, or between the permanent magnet20and the inner circumference of the rotor core10. Further, a plurality of balancing structures30may be formed along the circumference of the rotor core10. The rotor core10is formed in a circular projection shape as described above. By removing the part of the balancing structure or the first portion32, which is the circular portion, upon balancing, it is possible to perform the balancing very simply and easily. In step S200, a mass imbalance point of the rotor core10is identified (S200). The imbalance point may be determined by the generally known method. In step S300, the part of the balancing structure30at the corresponding imbalance point is removed. According to an embodiment of the present disclosure, the minus balancing that reduces the mass may be performed, thereby also helping to reduce the weight of the motor. The present disclosure described above is not limited to the aforementioned exemplary embodiments and the accompanying drawings, and it will be apparent to those skilled in the art to which the present disclosure pertains that various substitutions, modifications, and changes thereof may be made without departing from the technical spirit of the present disclosure. | 10,711 |
11863049 | Reference number in the drawings:10. center shaft;11. wire passing channel;12. through hole;20. rotating housing;21. housing;211. fixing column;22. rear cover;221. penetrating hole;201. accommodating cavity;202. fixing grooves;203. positioning embedding groove;204. snapping groove;30. stator;40. driver;50. permanent magnet;51. sleeve;60. fan blades;61, base plate;62. blade;601. elastic snapping portion;70. bearing. DETAILED DESCRIPTION Referring toFIGS.1-4, a specific structure of a fan is shown in one optional embodiment of the present disclosure. The fan includes a center shaft10, a rotating housing20, a stator30, a driver40, a permanent magnet50, and a plurality of fan blades60. The center shaft10is of a hollow structure. A wire passing channel11allowing power lines to pass through is defined in the center shaft10, A through hole12communicated with the wire passing channel11is defined on one side of the center shaft10. In the embodiment, the center shaft10is cylindrical. The wire passing channel11passes through two end surfaces of the center shaft10. Two ends of the center shaft10are respectively connected to a fixing component and a lamp component, where the fixing component and the lamp component are not shown in the drawings. The rotating housing20is rotatably sleeved on an outer side of the center shaft10. An accommodating cavity201is formed in the rotating housing20. In the embodiment, the rotating housing20is rotatably disposed on the center shaft10through bearings70. Each of the bearings70is a ball bearing. Specifically, the rotating housing20includes a housing21and a rear cover22, The rear cover22is fixed to a rear side of the housing21. The housing21and the rear cover22are enclosed to form the accommodating cavity201. Fixing grooves202are respectively defined in an inner side of a middle of the housing21and an inner side of a middle of the rear cover22. The bearings70are one-to-one fixed in the fixing grooves202. A fixing column211is disposed in the housing21. A fixing hole is defined in the fixing column211, where the fixing hole is not shown in the drawings. A penetrating hole221is defined on the rear cover22, The penetrating hole221directly faces the fixing hole. A fixing screw passes through the penetrating hole221to connect to the fixing hole. Two ends of the center shaft10respectively extend out of a front side of the housing21and a rear side of the rear cover22. In addition, the rotating housing20is cylindrical. The stator30is fixed on the center shaft10and is disposed in the accommodating cavity201. The driver40is fixed to the stator30and is disposed in the accommodating cavity201. The driver40is electrically connected to the stator30. The driver40is disposed beside the through hole12. In the embodiment, the driver40is in a disc shape. The driver40is sleeved on the outer side of the center shaft10. The permanent magnet50is fixed to an inner side wall of the accommodating cavity201and is disposed on an outer periphery of the stator30. In the embodiment, the permanent magnet50is a circular sleeve. The permanent magnet is disposed in a sleeve51. The sleeve51is tightly matched and fixed with an inner wall of the accommodating cavity201. The plurality of fan blades60are disposed on the rotating housing20and rotate along with the rotating housing20. In the embodiment, the plurality of fan blades60are disposed on an outer wall surface of the rotating housing20and are evenly distributed at intervals. The number of the plurality of fan blades60is 3+N, which is not limited thereto. The plurality of fan blades60and the rotating housing20are integrally formed and connected. Of course, the plurality of fan blades60are fixedly clamped with the rotating housing20. When the plurality of fan blades60are fixedly clamped with the rotating housing20, the specific structures are as follows: Positioning embedding grooves203are defined on an outer wall of the rotating housing20. A snapping groove204is concavely defined on a bottom surface of each of the positioning embedding grooves203. Each of the plurality of fan blades60comprises a base plate61and a blade62. The base plate61and the blade62of each of the plurality of fan blades60are integrally formed and connected. Each base plate61is matched with a corresponding positioning embedding groove203and is embedded in the corresponding positioning embedding groove203for positioning. An elastic snapping portion601is disposed on and protruded from each base plate61. Each elastic snapping portion601is fixedly snapped on a corresponding snapping groove204. Each of the positioning embedding grooves203is arc-shaped; and each base plate61is arc-shaped. Each snapping groove204is located in a center position of a corresponding positioning embedding groove203. Each elastic snapping portion601is located in a center position of a corresponding base plate61. During assembly, each base plate61is inserted into the corresponding positioning embedding groove203, and after each base plate61is inserted in place, each elastic snapping portion601is fixedly snapped on the corresponding snapping groove204, which achieves screw-free disassembly and assembly of the plurality of fan blades60. The structure of the fan is simple and production assembly is convenient. A working principle of the embodiment of the present disclosure is as follows: During working, the center shaft10is connected to the fixing component to play a fixing role. A section of the lamp component fixed to the center shaft10is a fixed component for fixing. The center shaft10does not rotate to do work, and after the stator30is powered on to do work. The permanent magnet50drives the rotating housing20to rotate on the center shaft10. During rotation, the bearings70plays a role in separating the center shall10and the rotating housing20, ensuring normal operation of the hollow direct-current brushless motor. The plurality of fan blades60rotate along with the rotating housing20, thereby achieving a fan function. A design focus of the present disclosure is that: compared with prior art, a conventional oil bearing made by powder metallurgy is replaced with the ball hearings, which effectively prolongs service life of the hollow direct-current brushless motor and the fan. A conventional motor housing is removed and the fan blades and the rotating housing are directly integrated, the fan blades may be assembled and disassembled without screws, which saves installation steps and improves production efficiency. The central shaft is hollow, which facilitates wiring, so that the fan blades are disposed in a middle of the fan, which makes it easy to design different styles of fan with a lamp. Moreover, a connection method between the fan blades and the rotating housing is improved, and no screw is used, which solves a problem that the conventional fan blades snapping on a fan by convex circular buttons are easily fallen off when a rotating speed of a fan motor thereof is very large The technical principle of the present disclosure is described above in combination with specific embodiments. These descriptions are merely intended to explain the principle of the present disclosure and cannot be interpreted in any way as a limitation to a scope of protection of the present disclosure. Based on the explanation herein, a person skilled in the art would have been able to associate other specific embodiments of the present disclosure without involving an inventive effort, all of which fall within the scope of protection of the present disclosure. | 7,568 |
11863050 | DESCRIPTION OF EMBODIMENTS Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the embodiments, in order to facilitate understanding, structures and elements other than the main parts of the present invention will be described in a simplified or omitted manner. In addition, in the drawings, the same elements are denoted by the same reference numerals. Note that the shapes, dimensions, and the like of the respective elements illustrated in the drawings are schematically illustrated, and do not indicate actual shapes, dimensions, or the like. Note that in the following description, a direction in which a central axis J illustrated inFIG.3extends will be simply referred to as an “axial direction”, a radial direction centered on the central axis J will be simply referred to as a “radial direction”, and a circumferential direction centered on the central axis J will be simply referred to as a “circumferential direction”. In addition, in the axial direction, the right side inFIG.3will be referred to as one side, and the left side inFIG.3will be referred to as the other side. Further, in the radial direction, a side close to the central axis J will be referred to as an inner side, and a side far from the central axis J will be referred to as an outer side. Furthermore, in the drawings, an XYZ coordinate system will be illustrated as a three-dimensional orthogonal coordinate system, as needed. In the XYZ coordinate system, a Y-axis direction is a direction parallel to the central axis J, and is a left-right direction of the side cross-sectional view illustrated inFIG.2. A Z-axis direction is a direction orthogonal to a Y-axis direction, and is an up-down direction in the side view illustrated inFIG.2. An X-axis direction is a direction orthogonal to the Y-axis direction and the Z-axis direction. In any of the X-axis direction, the Y-axis direction, and the Z-axis direction, + side denotes a side on which an arrow illustrated in the drawing faces, and − side denotes an opposite side. In addition, in the following description, extending or expanding in the axial direction includes not only a case of strictly extending or expanding in the axial direction (Y-axis direction) but also a case of extending or expanding in a direction inclined within a range smaller than 45° with respect to the axial direction. Further, in the following description, extending or expanding in the radial direction includes not only a case of strictly extending or expanding in the radial direction (Y-axis direction) but also a case of extending or expanding in a direction inclined within a range smaller than 45° with respect to the radial direction. First Embodiment FIG.1is a perspective view of a hoist according to a first embodiment of the present invention. The present embodiment is an example in which the present invention is applied to a hoist for winding up a rope that connects a car of an elevator and a balance weight. The present invention is also applicable to another machine other than the hoist of the elevator. The hoist1ofFIG.1includes a sheave20around which a rope to be wound up is wound.FIG.2is a side view of the hoist1ofFIG.1, when viewed from +X side.FIG.3is a side cross-sectional view illustrating the hoist1ofFIG.2taken along a plane orthogonal to X axis and passing through a central axis J. In addition to the sheave20, the hoist1includes a motor10, which generates a driving force for causing the sheave20to rotate about the central axis J as a rotation axis, a shaft22, which transfers the driving force that has been generated by the motor10to the sheave20, a second bearing part50including a second bearing51, which axially supports the shaft22to be rotatable on the other side in the axial direction with respect to the sheave20, a first bearing part40including a first bearing41, which axially supports the shaft22to be rotatable on one side in the axial direction with respect to the sheave20, and a braking apparatus30, which brakes the rotation of the sheave20. As the first bearing41and the second bearing51, for example, self-aligning roller bearings can be used. InFIG.3, illustrations of details of the internal configurations of the first bearing41and the second bearing51are omitted. As the first bearing41and the second bearing51, other known types of bearings may be used. Note that the second bearing51may be disposed on the other side in the axial direction with respect to the motor10. The motor10is an example of the rotating machine. The motor10includes a stator11, a rotor12, a casing13, a cover member14, a cover member15, and a wind guide member16. The motor10is disposed on the other side in the axial direction with respect to the sheave20. The casing13is an example of an outer frame member. The casing13covers the stator11and the rotor12from an outer side in the radial direction over the entire circumference in the circumferential direction. The cover member14covers one side in the axial direction of the casing13. The cover member15covers the other side in the axial direction of the casing13. The wind guide member16is disposed on one side in the axial direction of the cover member15. The stator11includes coil ends11aand11b. The rotor12includes a hub part12a, which has an annular shape and is disposed on an inner side in the radial direction, a rim part12c, which has an annular shape and is disposed on an outer side in the radial direction, a spoke part12b, which extends in a radial form from the hub part12aand connects the hub part12aand the rim part12c, and a rib part12d, which extends from the spoke part12bto the other side in the axial direction and reinforces the rotor12. The rim part12cincludes a magnet at a position facing the stator11. The casing13includes leg parts102. The leg parts102are fixed to a base member200by, for example, bolts. The stator11is fixed to the casing13. The rotor12is disposed with a gap from the stator11. An end portion on the other side in the axial direction of the shaft22is fixed to the hub part12aof the rotor12by press-fitting, for example. The motor10causes the rotor12to rotate about the central axis J as a rotation axis, according to current application. The shaft22rotates about the central axis J as a rotation axis in accordance with the rotation of the rotor12. The second bearing part50includes the second bearing51. The second bearing part50includes leg parts101. The leg parts101are fixed to the base member200by, for example, bolts. The first bearing part40includes the first bearing41. The first bearing part40includes leg parts100. The leg parts100are fixed to the base member200by, for example, bolts. The base member200is a member outside the hoist1. The base member200is fixed to, for example, a floor, a side wall, or a ceiling of an elevator machine room. The sheave20is fixed to the shaft22by press-fitting, for example. The sheave20includes a rope winding part21, which has a cylindrical shape including a bore that penetrates in the axial direction. The rope winding part21includes a rope winding surface21aon its outer peripheral surface. The hoist1winds up the rope by friction between the rope and the rope winding surface21a. The braking apparatus30includes a brake disc31, and a brake clamper32, which presses a brake pad against a braking surface of the brake disc31to apply braking by friction. The brake disc31is a circular plate-shaped member having a braking surface expanding in a direction orthogonal to the axial direction. The brake disc31is fixed to the sheave20. The brake disc31rotates about the central axis J in accordance with the rotation of the sheave20. The brake clamper32is fixed to the base member200via the first bearing part40. The brake clamper32presses the brake pad against the braking surface of the brake disc31to apply braking on the rotation of the sheave20. FIG.4is a perspective view illustrating a motor10of the hoist1ofFIG.1in a partially cut-away manner. InFIG.4, except for the rotor12, illustration is given by being cut along a plane orthogonal to X axis and passing through the central axis J. As illustrated inFIG.4, in the present embodiment, the rotor12includes six spoke parts12bdisposed at equal intervals in the circumferential direction. The rotor12includes, for each of the six spoke parts12b, a rib part12d, which expands from the spoke part12bto the other side in the axial direction between the hub part12aand the rim part12c. Accordingly, in the present embodiment, the rotor12has six rib parts12d. The number of the spoke parts12band the number of rib parts12dmay be any number other than six. The number of the spoke parts12band the number of rib parts12ddo not have to be the same. The rotor12includes a bore that penetrates in the axial direction between a certain spoke part12band its adjacent spoke part12b. In the present embodiment, each of the six rib parts12dincludes surfaces parallel to the radial direction, each on a front side in a rotational direction and on a rear side in the rotational direction. The shapes of the surfaces parallel to the radial direction of the rib parts12dare substantially quadrangular. The rib part12dhas a flat plate shape expanding in the radial direction and the axial direction. When the rotor12rotates about the central axis J, the six rib parts12dalso rotate about the central axis J. The rib parts12drotate, and thus blow air. The rib part12dis an example of a blower blade. In the present embodiment, an end portion on the other side in the axial direction of the rib part12dis located on one side in the axial direction with respect to an end portion on the other side in the axial direction of the coil end11a. The coil end11ais a coil end on the other side in the axial direction, among the coil ends of the stator11. The cover member15is located on the other side in the axial direction with respect to the rib parts12d. The cover member15includes eight through holes15a, which are disposed at equal intervals in the circumferential direction. The through holes15apenetrate the cover member15in the axial direction. The wind guide member16is a cylindrical member, and its diameter gradually decreases, as approaching one side in the axial direction from the other side in the axial direction. The wind guide member16may be integrated with the cover member15, or may be a separate member. In the present embodiment, an end portion on the other side in the axial direction of the wind guide member16is fixed to the surface on one side in the axial direction of the cover member15by, for example, bolts. The through hole15areaches a position on an inner side in the radial direction from a position on an outer side in the radial direction with respect to an end portion on the other side in the axial direction of the wind guide member16. In the present embodiment, in the through hole15a, a through hole on an inner side in the radial direction with respect to an end portion on the other side in the axial direction of the wind guide member16serves as a ventilation inlet15aa(seeFIG.5). In the present embodiment, in the through hole15a, a through hole on an outer side in the radial direction with respect to an end portion on the other side in the axial direction of the wind guide member16serves as a ventilation outlet15ab(seeFIG.5). The wind guide member16expands from the surface on one side in the axial direction of the cover member15toward the rib part12dbetween the ventilation inlet15aaand the ventilation outlet15ab. FIG.5is a cross-sectional view for describing the air flow caused by the rotation of the rib parts12d. The cover member14includes a through hole14bat a position facing the coil end11bin the axial direction. The through hole14bpenetrates the cover member14in the axial direction. The coil end11bis one of the coil ends of the stator11, on one side in the axial direction. As illustrated inFIG.5, in the motor10, a flow of the wind that flows from the ventilation inlet15aaof the cover member15to the through hole14band a flow of the wind that flows from the ventilation inlet15aaof the cover member15to the ventilation outlet15abof the cover member15are present. Unless the wind guide member16is provided, the air will stagnate between the ventilation inlet15aaand the ventilation outlet15abof the cover member15, and the cooling performance of the stator11will be degraded. According to the present embodiment, the wind can be guided from the ventilation inlet15aato the ventilation outlet15abof the cover member15by the wind guide member16, the ventilation amount can be improved, and the cooling performance of the stator11can be improved. FIG.6is a view for describing an example of a position where the wind guide member16is disposed. The inventors have conducted numerical analyses on the ventilation amount of the wind that passes through the motor10to cool the stator11, and have obtained the disposed position of the wind guide member16for increasing the ventilation amount. As results of numerical analyses, it has been found that the ventilation amount can be increased by disposing the wind guide member16as illustrated inFIG.6. As illustrated inFIG.6, the position in the radial direction of an end portion of the wind guide member16on one side in the axial direction is desired to be located at a position where the ratio of the distance in the radial direction from an end portion (position A) on an inner side in the radial direction of the rib part12dto the distance in the radial direction between an end portion (position B) on an outer side in the radial direction of the rib part12dand the end portion (position A) on the inner side in the radial direction of the rib part12dfalls within 55% to 65%. Furthermore, an angle formed by the wind guide member16and the cover member15in a cross-section that passes through the central axis J and that is parallel to the axial direction (hereinafter, referred to as “wind guide angle”) is desired to fall within 35° to 45°. The graph ofFIG.8indicates results of the ventilation amounts obtained by varying the wind guide angle. As can be understood with reference toFIG.8, the wind guide angle is desired to fall within 35° to 45°. In a case where the wind guide angle is larger than 45°, exhaust is hindered by the wind that flows in from the ventilation outlet15ab, and thus the ventilation amount is reduced. In addition, in a case where the wind guide angle is smaller than 35°, the closest distance between the wind guide member16and the coil end11abecomes shorter, the cross-sectional area of the flow passage is reduced, and the ventilation amount is reduced. FIG.7is a view for describing an example of a position where the wind guide member16is disposed.FIG.7is obtained by numerical analyses similarly toFIG.6. As illustrated inFIG.7, the position in the radial direction of the end portion of the wind guide member16on one side in the axial direction is desired to be located at a position where the ratio of the distance in the radial direction from the end portion (position A) on an inner side in the radial direction of the rib part12dto the distance in the radial direction between the end portion (position B) on an outer side in the radial direction of the rib part12dand the end portion (position A) on the inner side in the radial direction of the rib part12dfalls within 55% to 65%. Furthermore, the ratio of the closest distance between the wind guide member16and the stator11(the closest distance between the wind guide member16and the coil end11ainFIG.7) to the distance in the radial direction between the end portion (position A) on an outer side in the radial direction of the rib part12dand the end portion (position B) on an inner side in the radial direction of the rib part12dis desired to fall within 20% to 30%. Even in a case where the radius of the air gap between the rotor12and the stator11(the distance from the central axis J to the air gap) is changed, the cooling performance can be improved by setting the ratio of the closest distance between the wind guide member16and the coil end11aas illustrated inFIG.7to dispose the wind guide member16. FIG.9is a view indicating results of numerical analyses of the wind pressure received by the surface orthogonal to the axial direction of the rotor12. InFIG.9, illustration of the shaft22is omitted.FIG.9indicates that the darker the color, the lower the pressure. The position in the radial direction of the end portion of the wind guide member16on one side in the axial direction is located at the position where the pressure received by the wind generated in the rib part12dis an intermediate pressure between a high pressure and a low pressure, among the positions in the radial direction of the rib part12d. This is suitable for partitioning into the ventilation inlet15aaand the ventilation outlet15abby use of the wind guide member16. This is because the intake air and the exhaust air can be separated by partitioning between the high pressure and the low pressure. Therefore, the position in the radial direction of the end portion of the wind guide member16on one side in the axial direction is desired to be located at a position where the pressure received by the wind generated in the rib part12dis an intermediate pressure between the high pressure and the low pressure, among positions in the radial direction of the rib part12d. The position at the intermediate pressure between the high pressure and the low pressure corresponds to a position where the ratio of the distance in the radial direction from the end portion (position A) on an inner side in the radial direction of the rib part12dto the distance in the radial direction between the end portion (position B) on an outer side in the radial direction and the end portion (position A) on an inner side in the radial direction of the rib part12dfalls within 55% to 65%. FIG.10is a view indicating results of numerical analyses in the example illustrated inFIG.6, and illustrates flows of the wind. The provision of the wind guide member16separates the intake air from the exhaust air with the wind guide member16as a boundary at both the ventilation inlet15aaand the ventilation outlet15ab, and thus achieves smooth ventilation. FIG.11is a perspective view illustrating an example of another cover member different from the cover member15illustrated inFIG.4. A cover member115illustrated inFIG.11covers the other side in the axial direction of the casing13. The wind guide member16is disposed on one side in the axial direction of the cover member115. The cover member115includes a band part155aat a contact portion with the wind guide member16. The present invention is also applicable to a case where the cover member115is used. However, the ventilation inlet and the ventilation outlet are made smaller by the presence of the band part155a, and air intake and air exhaust lack smoothness in some cases. FIG.12is a perspective view illustrating an example of another rotor different from the rotor12illustrated inFIG.4. The rotor112illustrated inFIG.12includes a hub part112a, which has an annular shape and is disposed on an inner side in the radial direction, a rim part112c, which has an annular shape and is disposed on an outer side in the radial direction, a spoke part112b, which extends in a radial form from the hub part112aand connects the hub part112aand the rim part112c, and a rib part112d, which expands from the spoke part112bto the other side in the axial direction and reinforces the rotor112.FIG.13is a side cross-sectional view of the rotor112ofFIG.12. The rib part112dof the rotor112inFIG.12is different in shape from the rib part12dof the rotor12inFIG.4. Other points of the rotor112are the same as those of the rotor12. The rib part112dhas a substantially triangular shape of a surface parallel to the radial direction. The rib part112dhas a flat plate shape expanding in the radial direction and the axial direction. When the rotor112rotates about the central axis J, the rib part112dalso rotates about the central axis J. The rib parts112drotate, and thus blow air. The rib part112dis an example of a blower blade. In the above embodiments, the description has been made with regard to the configuration example in which the present invention is applied to the hoist of the elevator. However, the present invention is not limited to this. The present invention is applicable to hoists for any purpose, as long as the hoist winds up a rope. In addition, the present invention is also applicable to any motors, as long as the motor causes the blower blades to rotate in accordance with the rotation of the rotor. The present invention is applicable to both an inner rotor type motor and an outer rotor type motor. The present invention is not limited to the above embodiments, and various improvements and design changes may be made without departing from the gist of the present invention. In addition, the embodiments disclosed herein are to be considered in all respects as illustrative and non-limiting ones. The scope of the present invention is indicated not by the above description but by the scope of claims, and it is intended that meanings equivalent to the claims and all modifications within the scope are included. The present application claims priority based on Japanese Patent Application No. 2020-041702 filed on Mar. 11, 2020, the entire contents of which are incorporated herein by reference. REFERENCE SIGNS LIST 1Hoist10Motor16Wind guide member20Sheave30Braking apparatus40First bearing part50Second bearing part22Shaft | 21,657 |
11863051 | Repeat use of reference characters in the present specification and drawings is intended to represent the same or analogous features or elements of the present invention. DETAILED DESCRIPTION Reference now will be made in detail to embodiments of the invention, one or more examples of which are illustrated in the drawings. Each example is provided by way of explanation of the invention, not limitation of the invention. In fact, it will be apparent to those skilled in the art that various modifications and variations can be made in the present invention without departing from the scope or spirit of the invention. For instance, features illustrated or described as part of one embodiment can be used with another embodiment to yield a still further embodiment. Thus, it is intended that the present invention covers such modifications and variations as come within the scope of the appended claims and their equivalents. As used herein, the terms “first”, “second”, and “third” may be used interchangeably to distinguish one component from another and are not intended to signify location or importance of the individual components. The terms “upstream” and “downstream” refer to the relative direction with respect to fluid flow in a fluid pathway. For example, “upstream” refers to the direction from which the fluid flows, and “downstream” refers to the direction to which the fluid flows. Embodiments of a thermal management system are provided herein that may mitigate or eliminate risks associated with shaft seals, magnetic couplings, and high-temperature motor and/or generator design. The thermal management system may include a thermal transport bus (TTB) having a first heat exchanger along a first heat exchanger circuit utilizing the same heat transfer working fluid for cooling an electric machine through a temperature-controlled bleed circuit at a second heat exchange circuit. Embodiments provided allow for a single, closed-loop and hermetically-sealed system for the TTB. Such embodiments further allow for mitigating or eliminating risks associated with contamination of the TTB with lubricant, electric machine coolant, or other undesired fluids. Embodiments of the thermal management system include a dedicated secondary circuit, such as a second heat exchange circuit described below, thermally isolated from a primary TTB circuit along a first heat exchange circuit, and configured to maintain or control the temperature of heat transfer fluid through an electric machine, a pump, a compressor, or a fluid flow device generally. The thermally isolated secondary circuit shares the same fluid and pressure as the primary TTB circuit, such as through a seal configured to allow for desired fluid communication therethrough, or through an open charging circuit. The temperature and mass flow rate of the fluid through the thermally isolated secondary circuit allows for desired cooling, thermal attenuation, or heat transfer for the electric machine substantially independent of the primary TTB flow. Referring now to the drawings,FIGS.1-4provide schematic embodiments of a thermal management system1000. The system1000includes a pump, compressor, or other fluid flow mechanism1100. The fluid flow mechanism1100has an electric machine1110configured to drive a flow device1120. In certain embodiments, a driveshaft1113operably couples the flow device1120to the electric machine1110to drive the flow device1120to generate a desired flow of heat transfer fluid through a first heat exchange circuit1101and a second heat exchange circuit1102. A conduit1115is extended through the fluid flow mechanism1100to allow for a heat transfer fluid to flow therethrough in thermal communication. A housing1130surrounds the fluid flow mechanism1100. In a particular embodiment, the conduit1115is extended through one or more pathways extended through the electric machine1110to provide thermal communication between the heat transfer fluid and components of the electric machine1110(e.g., rotors, stators, casings, etc.). In a still particular embodiment, the conduit1115is extended through the housing1130or between the housing11130and the electric machine1110. In various embodiments, the housing1130, the first heat exchange circuit1101, and the second heat exchange circuit1102are hermetically sealed to contain the heat transfer fluid therewithin. Various embodiments of the flow device1120include an impeller having blades within the housing1130configured to generate a pressurized flow of the heat transfer fluid through the heat exchange circuits1101,1102. The first heat exchange circuit1101and the second heat exchange circuit1102are substantially thermally isolated from one another. Thermal isolation includes each heat exchange circuit1101,1102configured to contain and flow the heat transfer fluid through the respective circuits. Changes in thermal load or temperature of the heat transfer fluid at the first heat exchange circuit1101in thermal communication with a first heat exchanger1140is substantially separate or isolated from changes in thermal load or temperature of the heat transfer fluid at the second heat exchange circuit1102in thermal communication with the fluid flow mechanism1100and the second heat exchanger1150. In various embodiments, the first heat exchange circuit1101is in substantially parallel flow arrangement with the second heat exchange circuit1102. The second heat exchange circuit1102is in fluid communication with the conduit1115through the fluid flow mechanism1100. In certain embodiments, the first heat exchange circuit1101is configured as a primary thermal transport bus (TTB). The primary TTB is a device for transporting thermal energy from a physical location or system to another physical location or system using a thermal transport fluid flowing through a conduit extending between these physical locations and/or systems. In particular embodiments, the system1000is in closed-loop flow arrangement from an inlet1108of the flow device1120to an outlet1104of the flow device1120. In the embodiment shown, the first heat exchange circuit1101is configured in thermal communication with one or more first heat exchangers1140. More specifically, for the exemplary embodiment shown, the one or more first heat exchangers1140includes a plurality of first heat exchangers1140. More specifically, still, for the embodiment shown, the plurality of first heat exchangers1140includes one or more thermal load heat exchangers1142and one or more thermal bus heat exchangers1144. In such a manner, it will be appreciated that the one or more first heat exchangers1140, and in particular for the embodiment shown, the one or more thermal load heat exchangers1142and thermal bus heat exchangers1144, are configured to receive or provide heat or thermal energy to the heat transfer fluid through the first heat exchange circuit1101. More specifically, in one embodiment, the one or more thermal load heat exchangers1142are configured to reject heat or thermal energy from the heat transfer fluid through the first heat exchange circuit1101, and the one or more thermal bus heat exchangers1144are configured to provide heat or thermal energy to the heat transfer fluid through the first heat exchange circuit1101. In a particular embodiment, the thermal load heat exchangers1142are positioned upstream of the thermal bus heat exchangers1144along the first heat exchange circuit1101. However, it should be appreciated that in other embodiments the thermal load heat exchangers1142and the thermal bus heat exchangers1144may be positioned in any desired or operable configuration based on desired heat retention or heat release thereto. Furthermore, various embodiments may include any desired quantity of thermal bus heat exchangers1142or thermal load heat exchangers1144as may be applicable to removing or providing thermal energy in accordance with aspects of the present disclosure. In still certain embodiments, the second heat exchange circuit1102is configured as a secondary TTB configured in thermal communication with one or more of a second heat exchanger1150. In various embodiments, the second heat exchanger1150is configured to receive heat or thermal energy from the heat transfer fluid through the second heat exchange circuit1102. In a particular embodiment, the second heat exchanger1150is an electric machine-cooler heat exchanger configured to remove heat or thermal energy from the heat transfer fluid through the second heat exchange circuit, such as to provide cooling or thermal attenuation to the fluid flow mechanism1100, or particularly the electric machine1110through which the one or more conduits1115is extended in thermal communication thereto. In still particular embodiments, such as depicted inFIG.2, the second heat exchange circuit1102is in closed-loop flow arrangement with the conduit1115through the fluid flow mechanism1100. During operation of the system1000, the flow device1120is configured to receive and induce, pressurize, or compress the flow of heat transfer fluid across the flow device1120from the suction side or inlet1108to the pressure side or outlet1104. The thermal bus heat exchangers1144may extract or receive heat or thermal energy from the heat transfer fluid. The thermal load heat exchangers1142may then provide heat or thermal energy to the heat transfer fluid as the fluid flows in a loop back to the inlet1108of the flow device1120. In various embodiments, the thermal bus heat exchanger1144at a lubrication system, a fuel system, an air system, a hydraulic system, or combinations thereof. The various systems may be any one or more systems such as may be included with a propulsion system (e.g., propulsion system10inFIG.5), an auxiliary power unit, a hybrid-electric power system, a fan propulsion system, or an aircraft in general (e.g., vehicle100inFIG.5). In still certain embodiments, the thermal bus heat exchanger1144receives heat or thermal energy from the heat transfer fluid and transfers the heat or thermal energy to a lubricant, a liquid and/or gaseous fuel, air or oxidizer generally, or a hydraulic fluid. The heat or thermal energy may be expelled, such as with a flow of air into atmospheric condition, or the thermal energy may be utilized to condition a flow of lubricant, fuel, air, or hydraulic fluid to a desired physical property (e.g., temperature, viscosity, or other desired physical property of fluid). Various embodiments of the thermal load heat exchanger1142include any one or more systems that may provide or transmit heat or thermal energy to the heat transfer fluid at the system1000. Exemplary embodiments of the thermal load heat exchanger1142are provided further below. The first heat exchange circuit1101forming a primary TTB may be configured to maintain or control the temperature of a first portion of heat transfer fluid in thermal communication with the first heat exchanger1140separately from the second heat exchange circuit1102forming a thermally isolated circuit sharing the same heat transfer fluid as the first heat exchange circuit1101. The temperature and mass flow rate of the heat transfer fluid through the second heat exchange circuit1102allows for desired cooling or thermal attenuation at the electric machine1110substantially independent of the primary TTB flow through the first heat exchange circuit1101. Referring now toFIG.1, in a particular embodiment, a first portion1105of the second heat exchange circuit1102is extended from upstream of the second heat exchanger1150to provide the heat transfer fluid from the fluid flow mechanism1100to thermal communication with the second heat exchanger1150to remove heat or thermal energy from the heat transfer fluid. In one embodiment, such as depicted inFIG.1, the first portion1105is extended from the outlet1104of the flow device1120. A second portion1106of the second heat exchange circuit1102is extended from downstream of the conduit1115, or downstream of the electric machine1110, to the inlet1108of the flow device1120. A seal1125at least partially separates the flow device1120from the electric machine1110, such as to thermally separate the first heat exchange circuit1101from the second heat exchange circuit1102. In such configurations, the second heat exchange circuit1102is in an open charging relationship with the first heat exchange circuit1101. The open charging relationship allows for fluid communication between the first heat exchange circuit1101and the second heat exchange circuit1102. The open charging relationship may allow for the first heat exchange circuit1101to form an expansion tank or reservoir for the second heat exchange circuit1102. Pressure changes at the second heat exchange circuit1102may be accommodated by the open charging relationship with the first heat exchange circuit1101. As such, the open charging relationship may mitigate over-pressurization at the system1000, or undesired changes or fluctuations in pressure of the heat transfer fluid in the system1000. A single flow device1120forming a single stage impeller may provide flow and pressure for the first portion of heat transfer fluid to flow through the first heat exchange circuit1101and for the second portion of heat transfer fluid to flow through the second heat exchange circuit1102. The seal1125may include any appropriate type of filler material between the driveshaft1113and the surrounding housing, conduits, circuits, or other walls or surfaces. Exemplary seals include rotary seals, labyrinth seals, shaft sleeves, V-seals, lip seals, or other appropriate seals for rotating-to-static interfaces. The seal1125may allow for desired levels of leakage or fluid communication thereacross. However, it should be appreciated that leakages or fluid communication across the seal1125may occur while allowing for the first heat exchange circuit1101to be substantially thermally isolated, separate, parallel flow from the second heat exchange circuit1102. Referring now toFIG.2, in another exemplary embodiment, the flow device1120at the fluid flow mechanism1100may include a first flow device1121and a second flow device1122. The first flow device1121is configured to provide a first portion of the heat transfer fluid to the first heat exchange circuit1101. The second flow device1122is configured to provide a second portion of the heat transfer fluid to the second heat exchange circuit1102. In a particular embodiment, the seal1125is positioned between the first flow device1121and the second flow device1122, such as to thermally isolate the heat exchange circuits1101,1102from one another. In such configurations, the first flow device1121and the second flow device1122provide respective flows of the first portion and the second portion of heat transfer fluid through the first heat exchange circuit1101and the second heat exchange circuit1102, respectively. In a particular embodiment, the first flow device1121is configured as a compressor, a pump, or other device to pressurize the flow of fluid. In another exemplary embodiment, the second flow device1122is configured as a pump inducer. The second flow device1122configured as a pump inducer may act to raise the pressure of the heat transfer fluid. During operation of the system1000, the first flow device1121flows the heat transfer fluid through the first heat exchange circuit1101such as described above. The second flow device1122is configured to receive and pressurize the flow of heat transfer fluid across the second flow device1122from a suction side proximate to the electric machine1110to the first portion1105of the second heat exchange circuit1102. The second heat exchanger1150, such as an electric machine-cooler heat exchanger, may extract or receive heat or thermal energy from the heat transfer fluid. The cooled heat transfer fluid then flows in a loop through the conduit1115and to the suction side of the second flow device1122. Referring now to the exemplary embodiment ofFIG.3, the embodiment provided combines the flow devices1121,1122such as is provided inFIG.2with an open charging relationship provided inFIG.1. InFIG.3, a charging port1107fluidly connects the first heat exchange circuit1101upstream of the first heat exchanger1140and downstream of the first flow device1121to the first portion1105of the second heat exchange circuit1102upstream of the second heat exchanger1150. The charging port1107allows the flow of heat transfer fluid to be re-charged at the second heat exchange circuit1102. Additionally, or alternatively, the charging port1107sets a reference pressure for the second heat exchanger circuit1102. Referring now toFIG.4, a schematic embodiment of an energy conversion system2000including embodiments of the thermal management system1000described with regard toFIGS.1-3is provided. The energy conversion system2000includes the fluid flow mechanism1100, the first heat exchange circuit1101, and the second heat exchange circuit1102such as described substantially in regard to the embodiments provided inFIGS.1-3. InFIG.4, the energy conversion system2000includes the thermal management system1000as a part of a turbomachine for a bottoming Rankine cycle. In a particular embodiment, the energy conversion system2000may include at the first heat exchange circuit1101a recuperator or heat addition system2100. The heat addition system2100may form a particular embodiment of the first heat exchanger1140described in regard toFIGS.1-3. In various embodiments, the recuperator2100is a waste heat recovery system, such as positioned at an exhaust of a combustion system, a turbine system, or a closed cycle engine system. The recuperator2100is configured to provide heat or thermal energy to the heat transfer fluid at the first heat exchange circuit1101. The fluid flow mechanism1100includes a third flow device1123operably coupled to the driveshaft1113. A first portion1101aof the first heat exchange circuit1101extends from the first flow device1121in thermal communication with the recuperator2100then to the third flow device1123. In a particular embodiment, the third flow device1123is a turbine or expander configured to allow the flow of heat transfer fluid to expand through a second portion1101bof the first heat exchange circuit1101. One or more additional first heat exchangers1140, such as a fluid heater2200, is positioned in thermal communication with the second portion1101bof the first heat exchange circuit1101downstream of the third flow device1123along the second portion1101bof the first heat exchange circuit1101. The fluid heater2200is configured to receive heat or thermal energy from the heat transfer fluid at the second portion1101bof the first heat exchange circuit1101. The energy conversion system2000further includes a third heat exchange circuit1103. The third heat exchange circuit1103includes a first portion1103aextended from the downstream or pressure side of the third flow device1123to provide the flow of heat transfer fluid to the second heat exchange circuit1102upstream of the second heat exchanger1150. The third heat exchange circuit1103further includes a second portion1103bextended from the second flow device1122to the third flow device1123and the conduit1115at the electric machine1110. InFIG.4the second flow device1122is positioned along the driveshaft1113between the first flow device1121and the third flow device1123. The electric machine1110is positioned between the first flow device1121and the second flow device1122. InFIGS.2-3, the second flow device1122is positioned between the first flow device1121and the electric machine1110. It should be appreciated that the second flow device1122may be positioned such as depicted inFIGS.2-4. Furthermore, it should be appreciated that the first flow device1121and the third flow device1123may be re-arranged, such as may be suitable based on the configuration of the electric machine1110and heat exchangers. Various embodiments depicted and described herein include positions and flow arrangements of the heat exchange circuits such as to provide one or more improvement described herein. The energy conversion system2000depicted inFIG.4may include a hermetically sealed turbomachine formed by the fluid flow mechanism1100for a bottoming Rankine Cycle. The system2000is cooled with an internal and substantially independently temperature controlled second heat exchange circuit1102charged by the first heat exchange circuit1101. The independent temperature control is allowed by the thermal isolation of the first heat exchange circuit1101and the second heat exchange circuit1102, such as described above. The internal second heat exchange circuit1102is pumped with a hermetically sealed fluid flow mechanism1100such as described herein. As such, the first, second, and third heat exchange circuits1101,1102,1103may be joined with the housing1130of the fluid flow mechanism to hermetically seal the heat transfer fluid therewithin. The driveshaft1113drives the flow devices1121,1122,1123to provide motive flow for cooling the electric machine1110. Furthermore, the driveshaft1113drives the flow devices1121,1122,1123to provide motive flow for cooling the third flow device1123forming a power turbine. The second portion1101bof the first heat exchange circuit1101may further include the fluid heater2200configured as a cooling heat exchanger (e.g., the thermal load heat exchanger1142). The fluid heater2200receives heat or thermal energy from the heat transfer fluid at the second portion1101bof the first heat exchange circuit to heat an oxidizer, such as air or oxygen (02), a liquid and/or gaseous fuel (e.g., a kerosene-based fuel, diesel fuel, natural gas, a hydrocarbon fuel generally, or other appropriate fuel, or combinations thereof), a lubricant (e.g., oil or oil-based lubricant, a synthetic lubricant, etc.), a hydraulic fluid, a refrigerant, or other desired fluid. It should be appreciated that the heat transfer fluid provided herein may be supercritical fluid (SCF) generally, such as any desired fluid within the system1000at a temperature and pressure above its critical point. In a particular embodiment, the heat transfer fluid is supercritical carbon dioxide. In other embodiments, the heat transfer fluid is an SCF that is water, methane, ethane, propane, ethylene, propylene, methanol, ethanol, acetone, or nitrous oxide, or other appropriate fluid, or combinations thereof. In still other embodiments, the heat transfer fluid is any appropriate fluid for receiving and transferring heat or thermal energy within the system1000such as described herein. Referring now toFIG.5, an exemplary embodiment of a vehicle100including a propulsion system10and the thermal management system1000according to aspects of the present disclosure is provided. In an embodiment, the vehicle100is an aircraft including an aircraft structure or airframe105. The airframe105includes a fuselage110to which wings120and an empennage130are attached. The propulsion system10is attached to one or more portions of the airframe105. In various embodiments, the thermal management system1000is a system configured to desirably distribute thermal loads, such as to add or remove heat from one or more fluids or structures, such as, but not limited to, oxidizer at the propulsion system, fuel, lubricant, hydraulic fluid, pneumatic fluid, or cooling fluid for an electric machine, electronics, computing system, environmental control system, gear assembly, or other system or structure. In certain instances, the propulsion system10is attached to an aft portion of the fuselage110. In certain other instances, the propulsion system10is attached underneath, above, or through the wing120and/or portion of the empennage130. In various embodiments, the propulsion system10is attached to the airframe105via a pylon or other mounting structure. In still other embodiments, the propulsion system10is housed within the airframe, such as may be exemplified in certain supersonic military or commercial aircraft. Various embodiments of the vehicle100include a computing system140, such as avionics or other electronics or computing devices configured to control the vehicle100or the propulsion system10. The vehicle100may further include an environmental control system (ECS)150, such as to provide thermally conditioned air to a cabin of the vehicle, the computing system140, a vehicle surface or propulsion system anti-icing system160, or other system of the vehicle100or propulsion system10. In various embodiments such as described herein, the computing system140controls the electric machine or one or more heat exchangers of the thermal management system1000or energy conversion system2000to adjust one or more flow rates, pressures, heat transfer rates, temperatures, or other actuatable properties to provide thermally conditioned heat transfer fluid to one or more of the systems described herein. In certain embodiments, one or more of the first heat exchangers1140positioned in thermal communication along the first heat exchange circuit1101include a cooling system for the computing system140, a heating system for the anti-icing system160, a cooling or heating system for the ECS150, or one or more heat exchangers for a liquid and/or gaseous fuel, compressor bleed air, lubricant system, bearing system, electric machine, or other system. The thermal load heat exchanger1142may include a cooled-cooling air system, such as a heat removal system for removing heat from compressor bleed air from the propulsion system. The thermal load heat exchangers1142may include engine controls (e.g., FADEC, digital engine controls, propeller controls, or other computing devices), electric machines, motor/generators, starters, batteries, alternators, capacitors, or other heat loads generated by the propulsion system. The thermal load heat exchangers1142may include avionics, ECS, or other heat loads generated by the vehicle100. One or more components of the propulsion system10and thermal management system1000described herein may be manufactured or formed using any suitable process, such as an additive manufacturing process, such as a 3-D printing process. The use of such a process may allow such components to be formed integrally, as a single monolithic component, or as any suitable number of sub-components, or at scales and intricacies not previously allowed or conceived in the art. In particular, the additive manufacturing process may allow such component to be integrally formed and include a variety of features not possible when using prior manufacturing methods. For example, the additive manufacturing methods described herein may allow for the manufacture of the fluid flow mechanism as a single, integral component having the conduits and circuits as provided herein. In further embodiments, the additive manufacturing methods described herein allow for the manufacture of the system1000having the fluid flow mechanism, the circuits, conduits, and housing hermetically sealing the heat transfer fluid such as described herein. Additionally, additive manufacturing allows for having unique features, configurations, thicknesses, materials, densities, fluid circuits and conduits, headers, seals, and other structures that may not have been possible or practical using prior manufacturing methods. Suitable additive manufacturing techniques in accordance with the present disclosure include, for example, Fused Deposition Modeling (FDM), Selective Laser Sintering (SLS), 3D printing such as by inkjets, laser jets, and binder jets, Stereolithography (SLA), Direct Selective Laser Sintering (DSLS), Electron Beam Sintering (EBS), Electron Beam Melting (EBM), Laser Engineered Net Shaping (LENS), Laser Net Shape Manufacturing (LNSM), Direct Metal Deposition (DMD), Digital Light Processing (DLP), Direct Selective Laser Melting (DSLM), Selective Laser Melting (SLM), Direct Metal Laser Melting (DMLM), and other known processes. Suitable powder materials for the manufacture of the structures provided herein as integral, unitary, structures, or at scales and intricacies provided herein, include metallic alloy, polymer, or ceramic powders. Exemplary metallic powder materials are stainless steel alloys, cobalt-chrome, aluminum alloys, titanium alloys, nickel based superalloys, and cobalt based superalloys. In addition, suitable alloys may include those that have been engineered to have good oxidation resistance, known as “superalloys” which have acceptable strength at the elevated temperatures of operation in a gas turbine engine, e.g. Hastelloy, Inconel alloys (e.g., IN 738, IN 792, IN 939), Rene alloys (e.g., Rene N4, Rene N5, Rene 80, Rene 142, Rene 195), Haynes alloys, Mar M, CM 247, CM 247 LC, C263, 718, X-850, ECY 768, 282, X45, PWA 1483 and CMSX (e.g. CMSX-4) single crystal alloys. The manufactured objects of the present disclosure may be formed with one or more selected crystalline microstructures, such as directionally solidified (“DS”) or single-crystal (“SX”). This written description uses examples to disclose the invention, including the best mode, and also to enable any person skilled in the art to practice the invention, including making and using any devices or systems and performing any incorporated methods. The patentable scope of the invention is defined by the claims, and may include other examples that occur to those skilled in the art. Such other examples are intended to be within the scope of the claims if they include structural elements that do not differ from the literal language of the claims, or if they include equivalent structural elements with insubstantial differences from the literal languages of the claims. Further aspects of the invention are provided by the subject matter of the following clauses: 1. A thermal management system, the system comprising a fluid flow mechanism comprising an electric machine, wherein a conduit is formed through the electric machine allowing a heat transfer fluid to flow therethrough, wherein the fluid flow mechanism comprises a flow device configured to provide a first portion of the heat transfer fluid to a first heat exchange circuit and a second portion of heat transfer fluid to a second heat exchange circuit, wherein the conduit is in fluid communication with the second heat exchange circuit. 2. The system of any one or more clauses herein, wherein the second heat exchange circuit is at least partially in parallel flow arrangement with the first heat exchange circuit. 3. The system of any one or more clauses herein, wherein the fluid flow mechanism comprises a driveshaft operably coupled to the electric machine and the flow device. 4. The system of any one or more clauses herein, wherein the fluid flow mechanism comprises a housing, and wherein the housing, the electric machine, the first heat exchange circuit, and the second heat exchange circuit hermetically seal the heat transfer fluid therewithin. 5. The system of any one or more clauses herein, wherein the flow device comprises a first flow device configured to provide the first portion of the heat transfer fluid to the first heat exchange circuit, and wherein the flow device comprises a second flow device configured to provide the second portion of the heat transfer fluid to the second heat exchange circuit. 6. The system of any one or more clauses herein, wherein the fluid flow mechanism comprises a driveshaft operably coupled to the electric machine, the first flow device, and the second flow device. 7. The system of any one or more clauses herein, the fluid flow mechanism comprising a housing surrounding the driveshaft, the first flow device, and the second flow device, and wherein a seal is operably coupled to the driveshaft between the first flow device and the second flow device. 8. The system of any one or more clauses herein, wherein the seal separates the second heat exchange circuit and the first heat exchange circuit into substantially separate, parallel flows. 9. The system of any one or more clauses herein, wherein the flow device is an impeller. 10. The system of any one or more clauses herein, the system comprising a first heat exchanger positioned in thermal communication at the first heat exchange circuit. 11. The system of any one or more clauses herein, wherein a plurality of the first heat exchanger is positioned in thermal communication at the first heat exchange circuit. 12. The system of any one or more clauses herein, wherein the plurality of the first heat exchanger comprise a thermal bus heat exchanger and a thermal load heat exchanger each in thermal communication with the heat transfer fluid at the first heat exchange circuit. 13. The system of any one or more clauses herein, the system comprising a second heat exchanger positioned in thermal communication at the second heat exchange circuit. 14. The system of any one or more clauses herein, wherein the second heat exchanger is an electric machine-cooler heat exchanger positioned upstream of the conduit at the second heat exchange circuit. 15. A vehicle, the vehicle comprising a thermal management system, the system comprising a fluid flow mechanism, wherein the fluid flow mechanism comprises an electric machine, wherein a conduit is formed through the electric machine allowing a heat transfer fluid to flow therethrough, wherein the fluid flow mechanism comprises a flow device configured to provide a first portion of the heat transfer fluid to a first heat exchange circuit and a second portion of heat transfer fluid to a second heat exchange circuit, wherein the conduit is in fluid communication with the second heat exchange circuit; a thermal load heat exchanger positioned in thermal communication with the heat transfer fluid at the first heat exchange circuit, wherein the thermal load heat exchanger is configured to provide heat to the heat transfer fluid at the first heat exchange circuit; a thermal bus heat exchanger positioned in thermal communication with the heat transfer fluid at the first heat exchange circuit, wherein the thermal bus heat exchanger is configured to receive heat from the heat transfer fluid at the first heat exchange circuit; and an electric machine-cooler heat exchanger positioned in thermal communication with the heat transfer fluid at the second heat exchange circuit, wherein the electric machine-cooler heat exchanger is configured to receive heat from the heat transfer fluid at the second heat exchange circuit. 16. The vehicle of any one or more clauses herein, wherein the second heat exchange circuit is at least partially in parallel flow arrangement with the first heat exchange circuit. 17. The vehicle of any one or more clauses herein, wherein the flow device comprises a first flow device configured to provide the first portion of the heat transfer fluid to the second heat exchange circuit, and wherein the flow device comprises a second flow device configured to provide the second portion of the heat transfer fluid to the first heat exchange circuit. 18. The vehicle of any one or more clauses herein, wherein the thermal bus heat exchanger is positioned upstream of the thermal load heat exchanger along the first heat exchange circuit. 19. The vehicle of any one or more clauses herein, the thermal management system comprising a charging port fluidly connecting the first heat exchange circuit upstream of the thermal bus heat exchanger and downstream of the flow device to a first portion of the second heat exchange circuit upstream of the electric machine-cooler heat exchanger. 20. The vehicle of any one or more clauses herein, wherein the thermal load heat exchanger is one or more of a computing system, an anti-icing system, an electric machine, an environmental control system, or a propulsion system thermal load. | 35,880 |
11863052 | DETAILED DESCRIPTION The following description will be made with a vertical direction being defined on the basis of positional relationships in the case where drive devices according to example embodiments are installed in vehicles located on a horizontal road surface. That is, it is sufficient that the relative positional relationships regarding the vertical direction described in the following example embodiments are satisfied at least in the case where the drive device is installed in the vehicle located on the horizontal road surface. In the drawings, an xyz coordinate system is illustrated appropriately as a three-dimensional orthogonal coordinate system. In the xyz coordinate system, a z-axis direction corresponds to the vertical direction. A +Z side is an upper side in the vertical direction, and a −Z side is a lower side in the vertical direction. In the following description, the upper side and the lower side in the vertical direction will be referred to simply as the “upper side” and the “lower side”, respectively. An x-axis direction corresponds to a front-rear direction of the vehicle in which the drive device is installed, i.e., a direction perpendicular to the z-axis direction. In the example embodiment described below, a +X side corresponds to a forward side in the vehicle, while a −X side corresponds to a rearward side in the vehicle. A Y-axis direction corresponds to a left-right direction of the vehicle, i.e., a width direction of the vehicle, and is a direction perpendicular to both the x-axis direction and the z-axis direction. In the following example embodiments described below, a +Y side corresponds to a left side in the vehicle, while a −Y side corresponds to a right side in the vehicle. Each of the front-rear direction and the left-right direction is a horizontal direction perpendicular to the vertical direction. Note that the definition of the forward and rearward sides in the front-rear direction is not limited to the definition of the example embodiment described below, and that the +X side and the −X side may correspond to the rearward side and the forward side, respectively, of the vehicle. In this case, the +Y side corresponds to the right side of the vehicle, while the −Y side corresponds to the left side of the vehicle. Further, in the present specification, it is assumed that the term “parallel” as used herein includes both “parallel” and “substantially parallel”, and that the term “perpendicular” as used herein includes both “perpendicular” and “substantially perpendicular”. A central axis J1illustrated in the drawing as appropriate is a virtual axis extending in a direction intersecting the vertical direction. More specifically, the central axis J1extends in the Y-axis direction perpendicular to the vertical direction, that is, in the left-right direction of the vehicle. In description below, unless otherwise particularly stated, a direction parallel to the central axis J1is simply referred to as the “axial direction”, a radial direction about the central axis J1is simply referred to as the “radial direction”, and a circumferential direction about the central axis J1, that is, a direction around the central axis J1is simply referred to as the “circumferential direction”. In the present example embodiment, the left side (+Y side) corresponds to the “one side in the axial direction”, and the right side (−Y side) corresponds to the “other side in the axial direction”. An arrow θ appropriately illustrated in the drawing indicates the circumferential direction. In the following description, a side traveling counterclockwise about the central axis J1as viewed from one side (+Y side) in the axial direction in the circumferential direction, that is, a side (+θ side) on which the arrow θ faces is referred to as “one side in the circumferential direction”, and a side traveling clockwise about the central axis J1as viewed from one side in the axial direction in the circumferential direction, that is, a side (−θ side) opposite to the side on which the arrow θ faces is referred to as “the other side in the circumferential direction”. A drive device100of the present example embodiment illustrated inFIGS.1and2is a drive device that is mounted on a vehicle and rotates an axle64. A vehicle mounted on the drive device100is a vehicle having a motor such as a hybrid vehicle (HEV), a plug-in hybrid vehicle (PHV), and an electric vehicle (EV) as a power source. As illustrated inFIGS.1and2, the drive device100includes a motor20, a transmission60, a housing10having a motor housing11accommodating the motor20therein and a transmission housing12accommodating the transmission60therein, bearings71to76, an inverter unit80, and a pump94. The motor housing11and the transmission housing12are separate bodies fixed to each other. The transmission housing12is fixed to one side in the axial direction of the motor housing11. That is, the transmission housing12is connected to one side in the axial direction of the motor housing11. Each of the bearings71to76is, for example, a ball bearing. The motor20drives the drive device100. The motor20includes a rotor30rotatable about a central axis J1extending in the axial direction, and a stator40. The rotor30includes a shaft31and a rotor body32. The shaft31is rotatable about the central axis J1. The shaft31is rotatably supported by the bearings71,72,73, and74. Thus, the bearings71,72,73, and74rotatably support the rotor30. In the present example embodiment, the shaft31is a hollow shaft. The shaft31has a columnar shape about the central axis J1and extends axially. The shaft31is provided with a hole33connecting the inside of the shaft31and the outside of the shaft31. The shaft31extends across the inside of the motor housing11and the inside of the transmission housing12. An end on one side in the axial direction of the shaft31protrudes into the transmission housing12. A speed-reduction device61is connected to an end on one side in the axial direction of the shaft31. In the present example embodiment, the shaft31is configured by connecting a first shaft member31aand a second shaft member31bin the axial direction. The first shaft member31ais accommodated in the motor housing11. The first shaft member31ais provided with the hole33. The second shaft member31bis coupled to one side in the axial direction of the first shaft member31a. The outer diameter of the second shaft member31bis smaller than the outer diameter of the first shaft member31a. The end on the other side in the axial direction of the second shaft member31bis fitted into the inside of the end on one side in the axial direction of the first shaft member31a.The second shaft member31bextends from the inside of the motor housing11to the inside of the transmission housing12. The first shaft member31aand the second shaft member31bare connected to each other by spline fitting, for example. The first shaft member31ais rotatably supported by the bearings71and72. The second shaft member31bis rotatably supported by the bearings73and74. The rotor body32is fixed to the outer peripheral surface of the shaft31. More specifically, the rotor body32is fixed to the outer peripheral surface of the first shaft member31a. Although not illustrated in the drawings, the rotor body32includes a rotor core, and a rotor magnet fixed to the rotor core. The stator40is located outward the rotor30in the radial direction. The stator40is fixed inside the motor housing11. The stator40includes a stator core41and a coil assembly42. The stator core41has an annular shape surrounding the rotor30. The coil assembly42has a plurality of coils42cattached to the stator core41along the circumferential direction. The plurality of coils42care attached to the stator core41with, for example, an insulator (not illustrated) interposed between them. Although not illustrated in the drawings, the coil assembly42may include a binding member or the like which is used to bind the coils42ctogether, and may include a passage line arranged to join the coils42cto one another. The coil assembly42includes a coil end42aprotruding from the stator core41to one side in the axial direction and a coil end42bprotruding from the stator core41to the other side in the axial direction. The transmission60is connected to the motor20. The transmission60transmits the rotation of the rotor30to the axle64of the vehicle. As illustrated inFIG.1, the transmission60of the present example embodiment includes the speed-reduction device61connected to the motor20and a differential device62connected to the speed-reduction device61. The speed-reduction device61includes a first gear61a, a second gear61b, a third gear61c, and a gear shaft61d. The first gear61ais fixed to a portion of the shaft31located inside the transmission housing12. The second gear61band the third gear61care fixed to the gear shaft61d. The second gear61bmeshes with the first gear61a. The gear shaft61dextends in the axial direction about a gear axis J2extending in parallel with the central axis J1. The gear axis J2is a virtual axis located on the lower side of the central axis J1. For example, the gear axis J2is located on the rear side (−X side) of the central axis J1. The gear shaft61dis rotatably supported by the bearings75and76. The differential device62includes a ring gear62a. The ring gear62ameshes with the third gear61c. The lower end of the ring gear62ais immersed in the oil O stored in the transmission housing12. When the ring gear62arotates, the oil O is scraped up. The scraped oil O is supplied to, for example, the speed-reduction device61and the differential device62as lubricating oil. The differential device62rotates the axle64about a differential axis J3. The differential axis J3is a virtual axis extending in parallel with the central axis J1. The motor housing11accommodates the rotor30and the stator40in the inside. The motor housing11has a first housing13and a second housing14. In the present example embodiment, the first housing13corresponds to a first housing, and the second housing14corresponds to a second housing. The first housing13is a tubular member surrounding the motor20on the radial outside of the motor20. In the present example embodiment, the inner peripheral surface of the first housing13has the cylindrical shape centered on the central axis J1. The first housing13is open to the other side in the axial direction. The first housing13is fixed to the transmission housing12. The stator core41is fitted in the first housing13. The first housing13includes a first facing wall13aexpanding in the radial direction, a peripheral wall13bextending from a radially outer peripheral edge portion of the first facing wall13ato the other side in the axial direction, and a bearing holding portion13cprovided on the first facing wall13a. The first facing wall13afaces the transmission housing12in the axial direction. The first facing wall13ais located on the other side in the axial direction of the transmission housing12. The first facing wall13ais fixed to the transmission housing12. The first facing wall13ahas a hole13daxially penetrating the first facing wall13a. The hole13dhas a circular shape centered on the central axis J1. The second shaft member31bpasses through the hole13din the axial direction. As illustrated inFIG.2, the first facing wall13ahas a through hole13epenetrating the first facing wall13ain the axial direction. The through hole13eis a through hole that connects a space S located between the first facing wall13aand a second facing wall15ato be described later in the axial direction and the inside of the motor housing11. The through hole13eis provided in a portion of the first facing wall13alocated on the lower side of the bearing holding portion13c. The lower end of the through hole13eis connected to the inner peripheral surface of the peripheral wall13b. In the present example embodiment, the bearing holding portion13cis provided on the surface on the other side in the axial direction of the first facing wall13a. The bearing holding portion13cprotrudes from the surface on the other side in the axial direction of the first facing wall13ato the other side in the axial direction. As illustrated inFIG.3, the bearing holding portion13chas a cylindrical shape centered on the central axis J1. The bearing holding portion13chas a penetration portion13fpenetrating the bearing holding portion13cin the radial direction. In the present example embodiment, the penetration portion13fpenetrates a portion of the bearing holding portion13clocated above the central axis J1and on the rear side (−X side) in the radial direction. The penetration portion13fextends rearward and obliquely upward from the inner peripheral surface of the bearing holding portion13cto the outer peripheral surface of the bearing holding portion13c. As illustrated inFIG.1, the bearing holding portion13cholds the bearing72therein. The second housing14is separate from the first housing13. The second housing14is fixed to the other side in the axial direction of the first housing13. The second housing14closes the opening on the other side in the axial direction of the first housing13. As illustrated inFIG.4, the second housing14includes a lid wall14athat expands in the radial direction, and a peripheral wall14bthat extends from a radially outer peripheral edge portion of the lid wall14ato one side in the axial direction. As illustrated inFIG.1, an end on one side in the axial direction of the peripheral wall14bis in contact with an end on the other side in the axial direction of the peripheral wall13bin the first housing13. The lid wall14ahas a recess14crecessed from the surface on one side in the axial direction of the lid wall14ato the other side in the axial direction. A portion on one side in the axial direction of the recess14cis a bearing holding portion14dthat holds the bearing71therein. In the present example embodiment, the inverter unit80is attached to the motor housing11. The inverter unit80is fixed to a rear surface of the motor housing11. Although not illustrated, the inverter unit80has an inverter circuit electrically connected to the stator40. The transmission housing12accommodates the speed-reduction device61and the differential device62therein. As illustrated inFIG.2, the transmission housing12protrudes on the lower side from the motor housing11. The bottom located on the lower side of the inner surface of the transmission housing12is located on the lower side of the bottom located on the lower side of the inner surface of the motor housing11. The transmission housing12includes a third housing15fixed to the first housing13and a fourth housing16fixed to one side in the axial direction of the third housing15. The third housing15includes a second facing wall15aexpanding in the radial direction, a peripheral wall15bextending from a radially outer peripheral edge portion of the second facing wall15ato one side in the axial direction, and bearing holding portions15cand15dprovided on the second facing wall15a. The second facing wall15afaces the first facing wall13ain the axial direction. The second facing wall15ais fixed to one side in the axial direction of the first facing wall13a. The second facing wall15ahas a hole15faxially penetrating the second facing wall15a. The hole15fhas a circular shape centered on the central axis J1. The second shaft member31bpasses through the hole15fin the axial direction. The second facing wall15ahas a recess15erecessed from the surface on the other side in the axial direction of the second facing wall15atoward the one side in the axial direction. The inner peripheral edge of the recess15ehas, for example, a circular shape centered on the central axis J1when viewed in the axial direction. The opening on the other side in the axial direction of the recess15eis closed by the first facing wall13a. The space S is provided between the first facing wall13aand the second facing wall15ain the axial direction. The space S is configured by the inside of the recess15e. As illustrated inFIG.2, the second facing wall15ahas a through hole15hpenetrating the second facing wall15ain the axial direction. The through hole15his a through hole connecting the space S located between the first facing wall13aand the second facing wall15ain the axial direction and the inside of the transmission housing12. The through hole15his provided in a portion of the second facing wall15alocated on the lower side of the bearing holding portion15c. The through hole15his provided at the lower end of the bottom of the recess15e. The bottom of the recess15eis a surface located on one side in the axial direction and facing the other side in the axial direction of the inner surface of the recess15e. The lower end of the through hole15his connected to the inner peripheral surface of the recess15e. For example, the through hole15his disposed to face one side in the axial direction of the through hole13eprovided in the first facing wall13awith a gap. In the present example embodiment, the first facing wall13aand the second facing wall15aconstitute a partition wall19that separates the inside of the motor housing11and the inside of the transmission housing12. That is, the housing10has the partition wall19. The partition wall19has a through hole19aconnecting the inside of the motor housing11and the inside of the transmission housing12. The through hole19apenetrates the partition wall19in the axial direction. In the present example embodiment, the through hole19ais configured by the through hole13eprovided in the first facing wall13a, a lower end of the recess15e, and the through hole15hprovided in the second facing wall15a. In the present example embodiment, the bearing holding portions15cand15dare provided on the surface on one side in the axial direction of the second facing wall15a. The bearing holding portions15cand15dprotrude to the one side in the axial direction from the surface on the one side in the axial direction of the second facing wall15a. As illustrated inFIG.5, the bearing holding portion15chas a cylindrical shape centered on the central axis J1. The bearing holding portion15dhas a cylindrical shape centered on the gear axis J2. As illustrated inFIG.1, the bearing holding portion15cholds the bearing73therein. The bearing holding portion15dholds the bearing75therein. The fourth housing16includes a lid wall16aexpanding in the radial direction, a peripheral wall16bextending from the radially outer peripheral edge portion of the lid wall16ato the other side in the axial direction, and bearing holding portions16cand16dprovided on the lid wall16a. The end on the other side in the axial direction of the peripheral wall16bis in contact with the end on one side in the axial direction of the peripheral wall15bof the third housing15in the axial direction. In the present example embodiment, the bearing holding portions16cand16dare provided on the surface on the other side in the axial direction of the lid wall16a. The bearing holding portions16cand16dprotrude from the surface on the other side in the axial direction of the lid wall16ato the other side in the axial direction. Although not illustrated, the bearing holding portion16chas a cylindrical shape centered on the central axis J1. The bearing holding portion16dhas a cylindrical shape centered on the gear axis J2. The bearing holding portion16cholds the bearing74therein. The bearing holding portion16dholds the bearing76therein. For example, the oil O is accommodated in the transmission housing12. The oil O is stored in a lower region in the transmission housing12. The oil O is used as a refrigerant for cooling the motor20. The oil O is also used as lubricating oil for the speed-reduction device61and the differential device62. An oil equivalent to a lubricating oil (ATF: Automatic Transmission Fluid) for an automatic transmission having a relatively low viscosity is preferably used as the oil O so that the oil O can perform functions of a lubricating oil and a cooling oil. In the present example embodiment, the oil O corresponds to the first fluid. In the present example embodiment, the pump94is attached to the transmission housing12. The pump94is attached to a lower surface of the transmission housing12. The pump94is a pump that causes the oil O to flow into a second supply flow path92described later. In the present example embodiment, the pump94is an electric pump. The pump94may be a mechanical pump rotated by the shaft31or the gear shaft61d. Although not illustrated, a space between the first housing13and the second housing14in the axial direction, a space between the first housing13and the third housing15in the axial direction, and a space between the third housing15and the fourth housing16in the axial direction are sealed by seal members. The seal member is, for example, a liquid gasket. In the present example embodiment, the first housing13, the second housing14, the third housing15, and the fourth housing16are fixed with bolts. More specifically, as illustrated inFIG.6, the first housing13and the second housing14are fixed to each other by bolts10a. The first housing13and the third housing15are fixed to each other by bolts10b. The third housing15and the fourth housing16are fixed to each other by bolts10c.A plurality of bolts10a, a plurality of bolts10b, and a plurality of bolts10care provided to surround the central axis J1. The plurality of bolts10afix a plurality of protrusions13kprovided on the outer peripheral surface of the first housing and a plurality of protrusions14kprovided on the outer peripheral surface of the second housing14, respectively. The protrusion13kis provided at an end on the other side in the axial direction of the outer peripheral surface of the first housing13. The protrusion13kprotrudes radially outward. As illustrated inFIG.7, the plurality of protrusions13kis disposed at intervals along the circumferential direction. The protrusion13khas a female screw hole13precessed from the surface on the other side in the axial direction of the protrusion13ktoward the one side in the axial direction. In the present example embodiment, the female screw hole13ppenetrates the protrusion13kin the axial direction. The female screw hole13pmay be a hole having a bottom on one side in the axial direction. The protrusion14kis provided at an end on one side in the axial direction of the outer peripheral surface of the second housing14. The protrusion14kprotrudes radially outward. The plurality of protrusions14kis disposed at intervals along the circumferential direction. The surface on one side in the axial direction of the protrusion14kis in contact with the surface on the other side in the axial direction of the protrusion13k. The protrusion14khas a fixing hole14paxially penetrating the protrusion14k. When viewed in the axial direction, the fixing hole14pand the female screw hole13poverlap each other. The bolt10apasses through the fixing hole14pfrom the other side in the axial direction and is tightened into the female screw hole13p. Thus, the first housing13and the second housing14are fixed with the bolt10a. The plurality of bolts10bfix a plurality of protrusions13mprovided on the outer peripheral surface of the first housing and a plurality of protrusions15mprovided on the outer peripheral surface of the third housing15, respectively. The protrusion13mis provided at an end on one side in the axial direction of the outer peripheral surface of the first housing13. The protrusion13mprotrudes radially outward. The plurality of protrusions13mare disposed at intervals along the circumferential direction. The circumferential position of the protrusion13mis shifted from the circumferential position of the protrusion13k. The circumferential position of the protrusion13mis, for example, a circumferential central position between the protrusions13kadjacent in the circumferential direction. The protrusion13mhas a fixing hole13qaxially penetrating the protrusion13m. The protrusion15mis provided at the end on the other side in the axial direction of the outer peripheral surface of the third housing15. The protrusion15mprotrudes radially outward. The plurality of protrusions15mare disposed at intervals along the circumferential direction. The surface on the other side in the axial direction of the protrusion15mis in contact with the surface on one side in the axial direction of the protrusion13m. The protrusion15mhas a female screw hole15qrecessed from the surface on the other side in the axial direction of the protrusion15mto the one side in the axial direction. In the present example embodiment, the female screw hole15qpenetrates the protrusion15min the axial direction. The female screw hole15qmay be a hole having a bottom on one side in the axial direction. When viewed in the axial direction, the fixing hole13qand the female screw hole15qoverlap each other. The bolt10bpasses through the fixing hole13qfrom the other side in the axial direction and is tightened into the female screw hole15q. Thus, the first housing13and the third housing15are fixed with the bolt10b. As described above, in the present example embodiment, the first housing13and the third housing15are fixed to each other by the bolt10btightened from the same side as the bolt10afor fixing the first housing13and the second housing14. That is, the bolt10bfor fixing the first housing13and the third housing15is inserted into the fixing hole13qand the female screw hole15qin the same direction as the bolt10afor fixing the first housing13and the second housing14. As illustrated inFIG.6, the bolt10cfixes a protrusion15nprovided at the end on one side in the axial direction of the outer peripheral surface of the third housing15and a protrusion16nprovided at the end on the other side in the axial direction of the outer peripheral surface of the fourth housing16. Although not illustrated, a plurality of protrusions15nand a plurality of protrusions16nare provided at intervals in the circumferential direction. The protrusion15nand the protrusion16nprotrude radially outward. The circumferential positions of the protrusion15nand the protrusion16nmay be the same as the circumferential positions of the protrusion13mand the protrusion15m, or may be positions shifted in the circumferential direction. The protrusion15nhas a female screw hole15rrecessed from the surface on one side in the axial direction of the protrusion15nto the other side in the axial direction. In the present example embodiment, the female screw hole15rpenetrates the protrusion15nin the axial direction. The female screw hole15rmay be a hole having a bottom on the other side in the axial direction. The protrusion16nhas a fixing hole16raxially penetrating the protrusion16n. The bolt10cpasses through the fixing hole16rfrom one side in the axial direction and is tightened into the female screw hole15r. Thus, the third housing15and the fourth housing16are fixed with the bolt10c. As described above, in the present example embodiment, the third housing15and the fourth housing16are fixed to each other by the bolt10afor fixing the first housing13and the second housing14and the bolt10ctightened from the side opposite to the side where the bolt10bfor fixing the first housing13and the third housing15is tightened. That is, the bolt10cfor fixing the third housing15and the fourth housing16is inserted into the fixing hole16rand the female screw hole15rin a direction different from the bolt10afor fixing the first housing13and the second housing14and the bolt10bfor fixing the first housing13and the third housing15. As described above, in the present example embodiment, the first housing13and the third housing15are fixed by the bolt10bfrom the same side as the side on which the first housing13and the second housing14are fixed by the bolt10ain the axial direction. Therefore, the work of fixing the first housing13and the second housing14and the work of fixing the first housing13and the third housing15can be performed from the same side in the axial direction, that is, from the other side in the axial direction in the present example embodiment. Accordingly, assembling workability of the housing10can be improved. Here, in the present example embodiment, the transmission housing12has a shape protruding radially outward from the motor housing11. In such a case, when an attempt is made to fix the first housing13and the third housing15by inserting the bolt from the side where the transmission housing12is located with respect to the motor housing11, that is, from one side in the axial direction, it is necessary to arrange the fixing portion of the bolt on the radially outer side in order to avoid interference with the transmission housing12itself. Therefore, the housing10tends to be enlarged. On the other hand, for example, when the first housing13, the third housing15, and the fourth housing16are fastened together by bolts inserted from one side in the axial direction, the first housing13and the third housing15can be fixed while suppressing an increase in size of the housing10. However, in this case, when the bolt is removed to separate the motor housing11and the transmission housing12, the third housing15and the fourth housing16constituting the transmission housing12are also separated. Therefore, in the state of not being fixed to the motor housing11, the transmission housing12cannot be handled in a combined state. As a result, the easy of assembly of the housing tends to deteriorate. In addition, workability tends to deteriorate when performing maintenance of the drive device100, replacing the transmission60, and the like. In addition, the axial force by the bolt necessary for suitably maintaining the sealing property may be different between the seal member provided between the first housing13and the third housing15in the axial direction and the seal member provided between the third housing15and the fourth housing16in the axial direction. Therefore, when the first housing13, the third housing15, and the fourth housing16are fastened together with the same bolt, it may be difficult to suitably apply an axial force to the seal members disposed between the respective housings. Therefore, problems such as a decrease in sealability between the housings and difficulty in adjusting the axial force of the bolt are likely to occur. The problem in the case of fastening the first housing13, the third housing15, and the fourth housing16together by the bolt inserted from one side in the axial direction is the same as the problem in the case of fastening the first housing13, the second housing14, and the third housing15together by the bolt inserted from the other side in the axial direction. In view of the above problem, according to the present example embodiment, as described above, the first housing13and the third housing15are fixed by the bolt10bfrom the same side as the side on which the first housing13and the second housing14are fixed by the bolt10ain the axial direction. Therefore, it is possible to suppress the interference of the bolt10bwith the transmission housing12even if the position of the portion fixed by the bolt10bis not changed to the more radially outer position. As a result, the first housing13and the third housing15can be fixed with the bolt10bwhile suppressing an increase in size of the housing10. Further, even if the bolt10bis removed, only the first housing13and the third housing15are separated, and the third housing15and the fourth housing16are not separated. Therefore, even in a state where the transmission housing is not fixed to the motor housing11, the transmission housing12can be handled in a combined state. As a result, it is possible to suppress deterioration in the ease of assembly of the housing10. In addition, it is possible to suppress deterioration of workability when performing maintenance of the drive device100, replacing the transmission60, and the like. Further, since the axial forces of the bolt10band the bolt10ccan be changed, different axial forces can be individually applied to the seal member located between the first housing13and the third housing15and the seal member located between the third housing15and the fourth housing16. As a result, sealability between the housings can be easily secured, and the axial forces of the bolts10band10ccan be easily adjusted. The same applies to the seal member between the first housing13and the second housing14. In addition, for example, if another housing is disposed between the motor housing11and the transmission housing12and the motor housing11and the transmission housing12are fixed to the another housing, the motor housing11and the transmission housing12can be separated in an assembled state. However, in this case, the number of components constituting the housing10increases by the provision of the other housing. On the other hand, according to the present example embodiment, as described above, it is possible to separate the motor housing11and the transmission housing12in an assembled state without providing the other members. Therefore, it is possible to suppress an increase in the number of components constituting the housing10. In addition, since it is not necessary to provide the other member, the weight of the drive device100can be reduced. As a result, even when the structure of the drive device100is a water-cooled structure in which the motor20is cooled by water W as in the present example embodiment, it is possible to suppress an increase in the weight of the entire drive device100. As indicated by a two-dot chain line inFIG.7, the protrusion13mprovided in the first housing13may extend in the axial direction. In this case, the end on the other side in the axial direction of the protrusion13mcan be brought close to the protrusion13k. As a result, when the work of fixing the first housing13and the third housing15is performed from the other side in the axial direction, the position where the jig and the tool for fastening the bolt10bare used can be brought close to the position where the jig and the tool are used when the work of fixing the first housing13and the second housing14is performed. In addition, axial dimensions of the jig and the tool can be shortened. Thus, the workability of the work of fixing the first housing13and the third housing15with the bolt10bcan be improved. In particular, the bolt10bcan be suitably tightened to suitably generate the axial force. The end on the other side in the axial direction of the protrusion13mindicated by a two-dot chain line inFIG.7is located, for example, on the other side in the axial direction with respect to the center in the axial direction of the first housing13. The end on the other side in the axial direction of the protrusion13mindicated by a two-dot chain line inFIG.7is located, for example, on the one side in the axial direction with respect to the end on the one side in the axial direction of the protrusion13k. Thus, the protrusion13mcan be prevented from interfering with the protrusion13k. As illustrated inFIGS.3and4, in the present example embodiment, the first housing13and the second housing14are also fixed by bolts10ddifferent from the plurality of bolts10adescribed above. As illustrated inFIG.3, the first housing13has a female screw hole13irecessed from the end surface on the other side in the axial direction of the peripheral wall13btoward the one side in the axial direction. The female screw hole13iis located between a groove portion93ato be described later and a second circumferential flow path portion52bof a second flow path50to be described later in the circumferential direction. The female screw hole13iis located radially inside a collection flow path body93cdescribed later. As illustrated inFIG.4, the second housing14has a fixing hole14epenetrating the second housing14in the axial direction. The fixing hole14eis located between a connection portion93bto be described later and a second circumferential flow path portion52bof the second flow path50to be described later in the circumferential direction. The fixing hole14eis located radially inside the collection flow path body93cdescribed later. The bolt10dpassed through the fixing hole14efrom the other side in the axial direction is tightened into the female screw hole13i. Thus, in the present example embodiment, the first housing13and the second housing14are fixed to each other at positions radially inside a collection flow path93to be described later and adjacent to the second flow path50in the circumferential direction. The housing10has a first gutter portion17. The first gutter portion17is located between the first facing wall13aand the second facing wall15ain the axial direction. That is, the first gutter portion17is located in the space S. As illustrated inFIG.8, the first gutter portion17has a gutter shape that opens upward and extends in the axial direction. The oil O flows into the first gutter portion17. The first gutter portion17is a reservoir capable of storing the oil O therein. In the present example embodiment, the first gutter portion17is located on the rear side (−X side) of the central axis J1. The first gutter portion17is located behind the hole13d. The first gutter portion17connects the first facing wall13aand the second facing wall15a. In the present example embodiment, the first gutter portion17has a first portion17aprotruding to one side in the axial direction from a surface on one side (+Y side) in the axial direction of the first facing wall13a, and a second portion17bprotruding to the other side in the axial direction from a surface on the other side (−Y side) in the axial direction of the second facing wall15a. The end on one side in the axial direction of the first portion17aand the end on the other side in the axial direction of the second portion17bare connected to each other. The axial dimension of the second portion17bis larger than the axial dimension of the first portion17a. The first gutter portion17has a bottom17cfacing upward, and a pair of side surfaces17dand17eprotruding upward from both sides of the bottom17cin the front-rear direction. The bottom17cand the pair of side surfaces17dand17eextend in the axial direction. The bottom17cand the pair of side surfaces17dand17econnect the first facing wall13aand the second facing wall15a. The pair of side surfaces17dand17eis disposed to face each other at an interval in the axial direction. The side surface17dis located on the front side (+X side) of the side surface17e. The bottom17cis inclined in the vertical direction with respect to the front-rear direction. The bottom17cis located on the lower side toward the front side (+X side). In the present example embodiment, the bottom17cis an inclined surface located on the lower side as approaching a first hole13gprovided in the first facing wall13a. Therefore, it is easy to guide the oil O in the first gutter portion17into the first hole13galong the bottom17cusing gravity. The first hole13gpenetrates the first facing wall13ain the axial direction. The first hole13gis, for example, a circular hole. The first hole13gopens at the front end of the inside of the first gutter portion17. The first hole13gis connected to the bottom17cand the side surface17d. As illustrated inFIG.5, the first gutter portion17is connected to a portion located on the lower side of the first hole13gin the surface on one side in the axial direction of the first facing wall13aand a portion located on the lower side of the second hole15gin the surface on the other side in the axial direction of the second facing wall15a. The second hole15gpenetrates the second facing wall15ain the axial direction. The second hole15gis, for example, a circular hole. The second hole15gopens at the end on a rear side (−X side) of the inside of the first gutter portion17and the end on a front side (+X side) of the inside of a second gutter portion18. As illustrated inFIG.2, the housing10has the second gutter portion18. The second gutter portion18is located inside the transmission housing12. As illustrated inFIGS.5and9, the second gutter portion18has a gutter shape that opens upward and extends in the axial direction. The oil O flows into the second gutter portion18. The second gutter portion18is a reservoir capable of storing the oil O therein. In the present example embodiment, the second gutter portion18is located on the rear side (−X side) of the central axis J1. The second gutter portion is located above the bearing holding portion15d. As illustrated inFIG.5, the end on the front (+X side) side of the second gutter portion18is located on one side (+Y side) in the axial direction of the rear end of the first gutter portion17. As illustrated inFIG.2, the second gutter portion18connects the first facing wall13aand the second facing wall15a. In the present example embodiment, the second gutter portion18has a first portion18aprotruding to one side in the axial direction from a surface on one side (+Y side) in the axial direction of the second facing wall15a, and a second portion18bprotruding to the other side in the axial direction from a surface on the other side (−Y side) in the axial direction of the lid wall16a. The end on one side in the axial direction of the first portion18aand the end on the other side in the axial direction of the second portion18bare connected to each other. As illustrated inFIG.9, the second gutter portion18has a bottom18cfacing upward, and a pair of side surfaces18dand18eprotruding upward from both sides of the bottom18cin the front-rear direction. The bottom18cand the pair of side surfaces18dand18eextend in the axial direction. The bottom18cand the pair of side surfaces18dand18econnect the second facing wall15aand the lid wall16a. The pair of side surfaces18dand18eare disposed to face each other at an interval in the axial direction. The side surface18dis located on the front side (+X side) of the side surface18e. The side surface18dis inclined in the front-rear direction with respect to the vertical direction. The side surface18dis located on the front side (+X side) as it goes upward. In the present example embodiment, the side surface18dis an inclined surface located on the lower side as approaching the second hole15g. Therefore, the oil O that has entered the second gutter portion18is easily guided to the inside of the second hole15galong the side surface18dusing gravity. The side surface18eis inclined in the front-rear direction with respect to the vertical direction. The side surface18dis located on the rear side (−X side) as it goes upward. The bottom18cis inclined in the vertical direction with respect to the front-rear direction. The bottom18cis located on the lower side toward the rear side (−X side). As illustrated inFIG.5, the second gutter portion18is connected to a portion located on the lower side of the second hole15gin the surface on one side in the axial direction of the second facing wall15a. The second gutter portion18is provided with supply holes18fand18g. The supply hole18fconnects the inside of the second gutter portion18and the inside of the bearing holding portion15c. Therefore, a part of the oil O entering the second gutter portion18is supplied to a bearing73in the bearing holding portion15cvia the supply hole18f. As illustrated inFIG.9, the supply hole18fopens to the side surface18d. The supply hole18fextends forward (+X side) and obliquely on the lower side from the side surface18d. The supply hole18gconnects the inside of the second gutter portion18and the inside of the bearing holding portion15d. Therefore, a part of the oil O entering the second gutter portion18is supplied to the bearing75in the bearing holding portion15dvia the supply hole18g. The supply hole18gis open to the bottom18c. The supply hole18gextends on the lower side and obliquely forward (+X side) from the bottom18c. As illustrated inFIG.2, the housing10includes a first flow path90and a second flow path50. The first flow path90is a flow path through which the oil O as the first fluid flows. The second flow path50is a flow path through which the water W as the second fluid flows. In the present specification, the “flow path” means a path through which a fluid flows. Therefore, the concept of “flow path” includes not only a “flow path”, in which a steady flow of a fluid in one direction is generated, but also a channel in which the fluid is allowed to temporarily stay, and a channel along which the fluid drips. Examples of the channel in which the fluid is allowed to temporarily stay include a reservoir or the like arranged to store the fluid. The first flow path90includes a first supply flow path91, a second supply flow path92, and a collection flow path93. The first supply flow path91and the second supply flow path92are supply flow paths for supplying the oil O in the transmission housing12to the inside of the motor housing11. The first supply flow path91includes a scraping-up channel91a, a shaft supply channel91b, an intra-shaft channel91c, and an intra-rotor channel90a. The scraping-up channel91ais a path in which the oil O in the transmission housing12is scraped up by the rotation of the ring gear62aof the differential device62and enters the second gutter portion18. The shaft supply channel91bis a path through which the oil O in the second gutter portion18flows into the bearing holding portion16cthrough a flow path (not illustrated) provided in the lid wall16aand flows into the shaft31from the bearing holding portion16c.When the oil O flows into the bearing holding portion16cin the shaft supply channel91b, the oil O is supplied to the bearing74held by the bearing holding portion16c. In the shaft supply channel91bof the present example embodiment, the oil O flows in from the end on one side in the axial direction of the shaft31. The intra-shaft channel91cis a path through which the oil O flowing into the shaft31from the end on one side in the axial direction of the shaft31flows to the other side in the axial direction in the shaft31. The intra-rotor channel90ais a path for the oil O in the shaft31to pass through the inside of the rotor body32from the hole33and to be scattered to the stator40. In this manner, the oil O is supplied to the rotor30and the stator40by the first supply flow path91. As illustrated inFIG.1, the second supply flow path92includes an introduction flow path portion92a, a connecting flow path portion92b, an intra-shaft channel92c, and the intra-rotor channel90a. The introduction flow path portion92aextends in the axial direction from the inside of the transmission housing12. More specifically, the introduction flow path portion92aextends from the inside of the transmission housing12to the other side in the axial direction, passes through the second facing wall15a, the first facing wall13a, and the peripheral wall13b, and extends to the second housing14. The oil O sucked from the inside of the transmission housing12by the pump94flows into the introduction flow path portion92a. In the introduction flow path portion92a, the oil O flows to the other side in the axial direction. As illustrated inFIG.3, a cross section of the flow path of the introduction flow path portion92ahas an oval shape elongated in the circumferential direction. The circumferential dimension of the introduction flow path portion92ais smaller than the circumferential dimension of the collection flow path body93cto be described later, the circumferential dimension of a first circumferential flow path portion52ato be described later, and the circumferential dimension of a second circumferential flow path portion52bto be described later. Therefore, the circumferential dimension of the introduction flow path portion92acan be made relatively small. As a result, the pressure loss generated in the oil O flowing in the introduction flow path portion92acan be reduced. Therefore, the oil O can be easily fed into the introduction flow path portion92aby the pump94. For example, the introduction flow path portion92ais located on the front side (+X side) and the lower side with respect to the central axis J1. At least a part of the introduction flow path portion92ais located radially outside the second flow path50. In the present example embodiment, almost the entire introduction flow path portion92aexcept for both axial ends is located radially outside the second flow path50. The introduction flow path portion92ais located below the second flow path50. As illustrated inFIG.1, the connecting flow path portion92bis provided in the lid wall14aof the second housing14. The connecting flow path portion92bextends upward from an end on the other side in the axial direction of the introduction flow path portion92a, and is connected to recess14c. As a result, the oil O flows into the recess14c. Part of the oil O flowing into the recess14cis supplied to the bearing71held by the bearing holding portion14d. The other part of the oil O flowing into the recess14cflows into the shaft31from the other side in the axial direction. The intra-shaft channel92cis a path through which the oil O flowing into the shaft31from the end on the other side in the axial direction of the shaft31flows to one side in the axial direction in the shaft31. As described above, in the present example embodiment, the oil O flows into the shaft31from both sides in the axial direction by the first supply flow path91and the second supply flow path92. Therefore, for example, as compared with a case where the oil O flows in only from one end in the shaft31, the oil O can be suitably flown to the entire shaft31in the axial direction. That is, it is possible to suppress that the oil O flowing in from one end in the shaft31does not reach the other end in the shaft31and does not flow to the entire inside of the shaft31. Therefore, it is easy to suitably supply the oil O to each of the bearings71and74supporting both axial ends of the shaft31. The oil O flowing through the intra-shaft channel92cflows through the intra-rotor channel90aand is supplied to the rotor30and the stator40, similarly to the intra-shaft channel91c. The oil O supplied to the stator40takes heat from the stator40by the first supply flow path91and the second supply flow path92. The oil O that has cooled the stator40falls on a lower side and accumulates in a lower region in the motor housing11. The oil O accumulated in the lower region in the motor housing11returns to the inside of the transmission housing12via the through hole19aof the partition wall19or the collection flow path93. As illustrated inFIG.2, the collection flow path93extends from the inside of the motor housing11to the inside of the transmission housing12. In the present example embodiment, the collection flow path93is provided across the third housing15, the first housing13, and the second housing14. The collection flow path93is located on the lower side of the motor20. The collection flow path93includes the groove portion93a, the connection portion93b, and the collection flow path body93c. The groove portion93ais provided on the inner peripheral surface of the motor housing11. In the present example embodiment, the groove portion93ais recessed on the lower side from a portion located on the lower side of the inner peripheral surface of the first housing13. The groove portion93aextends in the axial direction. The end on one side in the axial direction of the groove portion93ais closed. The end on the other side in the axial direction of the groove portion93ais open to the end surface on the other side in the axial direction of the peripheral wall13b. The end on the other side in the axial direction of the groove portion93ais connected to the connection portion93b. The bottom of the groove portion93ais located on the lower side toward the other side in the axial direction. That is, the bottom of the groove portion93ais an inclined surface located on the lower side toward the connection portion93b. Therefore, the oil O entering the groove portion93acan be easily guided to the connection portion93balong the bottom of the groove portion93ausing gravity. The bottom of the groove portion93ais a surface that is located on the radially outer side of the inner surface of the groove portion93aand faces the radially inner side. In the present example embodiment, the bottom of the groove portion93afaces upward. As illustrated inFIG.10, the circumferential dimension of the groove portion93ais smaller than the circumferential dimension of the through hole13e. The connection portion93bconnects the groove portion93aand the collection flow path body93c. The connection portion93bis connected to an end93fon the other side in the axial direction of the groove portion93a. In the present example embodiment, the connection portion93bis provided on the peripheral wall14bof the second housing14. The connection portion93bextends on the lower side from a portion located on the lower side of the inner peripheral surface of the peripheral wall14b. The connection portion93bopens upward. As illustrated inFIG.2, the lower end of the connection portion93bis connected to an end93gon the other side in the axial direction of the collection flow path body93c. As a result, the connection portion93bconnects the end93fon the other side in the axial direction of the groove portion93aand the end93gon the other side in the axial direction of the collection flow path body93c. The collection flow path body93cis located radially outside the groove portion93a. In the present example embodiment, the collection flow path body93cis located on the lower side of the groove portion93a. The collection flow path body93cextends in the axial direction and is connected to the inside of the transmission housing12. An end93pon one side in the axial direction of the collection flow path body93cis open to the inside of the transmission housing12. In the present example embodiment, the collection flow path body93cis provided across the second housing14, the first housing13, and the third housing15. That is, the collection flow path body93cincludes a first portion93hprovided in the first housing13, a second portion93iprovided in the second housing14, and a third portion93jprovided in the third housing15. An end93kon one side in the axial direction of the first portion93his connected to an end on the other side in the axial direction of the third portion93j. An end93mon the other side in the axial direction of the first portion93his connected to the end on one side in the axial direction of the second portion93i. The collection flow path body93cextends from the lower end of the connection portion93bto one side in the axial direction, penetrates the first housing13and the third housing15in the axial direction, and is open to the inside of the transmission housing12. The collection flow path body93cis located on the lower side of the through hole19aof the partition wall19. As illustrated inFIGS.3and4, the cross section of the flow path of the collection flow path body93chas a shape elongated in the circumferential direction. The circumferential dimension of the collection flow path body93cis larger than the circumferential dimension of the groove portion93aand the circumferential dimension of the connection portion93b.Therefore, the flow rate of the oil O that can flow into the collection flow path body93ccan be increased. As a result, the amount of the oil O that can be returned from the inside of the motor housing11into the transmission housing12can be increased. At least a part of the collection flow path body93cis located radially outside the second flow path50. As a result, at least a part of the collection flow path93is located radially outside the second flow path50. As illustrated inFIG.10, a part of the collection flow path body93cis located on the lower side of a pair of axial flow path portions51, which will be described later, disposed with the groove portion93ainterposed therebetween in the circumferential direction in the second flow path50, a first circumferential flow path portion52c, which will be described later, located on one side in the axial direction of the groove portion93ain the second flow path50, and a pair of second circumferential flow path portions52b, which will be described later, disposed with the connection portion93binterposed therebetween in the circumferential direction in the second flow path50. In the present example embodiment, since the circumferential dimension of the collection flow path body93cis larger than the circumferential dimension of the groove portion93aand the circumferential dimension of the connection portion93bas described above, the collection flow path body93ccan protrude in the circumferential direction from the groove portion93aand the connection portion93b. Therefore, the collection flow path body93ccan be easily disposed radially outside the second flow path50. The collection flow path body93cis disposed adjacent to one circumferential direction side (+θ side) of the introduction flow path portion92a. That is, in the present example embodiment, the introduction flow path portion92ais disposed adjacent to the collection flow path93in the circumferential direction. In the present example embodiment, the portion of the motor housing11where the collection flow path body93cand the introduction flow path portion92aare provided protrudes on the lower side from the other portion of the motor housing11. The collection flow path body93cis provided with a partition wall93dthat partitions the inside of the collection flow path body93cin the circumferential direction. The partition wall93dextends in the axial direction from the end93pon one side in the axial direction of the first portion93htoward the other side in the axial direction. In the present example embodiment, the partition wall93dextends from the end93kon one side in the axial direction of the first portion93hto the central portion in the axial direction of the first portion93h. In other words, the partition wall93dextends from the end on one side in the axial direction of the first housing13to the central portion in the axial direction of the first housing13. The partition wall93ddivides the collection flow path body93c, which is long in the circumferential direction, into substantially two equal parts in the circumferential direction. The partition wall93dcan improve the strength of the portion of the housing10where the collection flow path body93cis provided. Further, the axial force of the bolt10bcan be more suitably transmitted to the first housing13and the third housing15. The partition wall93dmay not extend to the axial center of the first portion93h, that is, the axial center of the first housing13. For example, an end93ron the other side in the axial direction of the partition wall93dmay be disposed at any position as long as it is located on the other side in the axial direction with respect to the end93kon one side in the axial direction of the first portion93hand located on one side in the axial direction with respect to the end93mon the other side in the axial direction of the first portion93h. As illustrated inFIG.3, the collection flow path body93chas a recessed portion93erecessed radially inward. The recessed portion93eis located at the circumferential central portion of the other side in the axial direction of the collection flow path body93c. An outer peripheral surface of a portion of the motor housing11where the recessed portion93eis provided is recessed radially inward. Thus, for example, the bolt10bfor fixing the first housing13and the third housing15can be prevented from interfering with the collection flow path body93c. As illustrated inFIGS.1and2, at least a part of the second flow path50is located radially outside the motor20. In the present example embodiment, substantially the entire second flow path50except for both axial ends is located radially outside the motor20. A portion of the second flow path50located on the lower side is located between the collection flow path body93cand the motor20in the radial direction. As illustrated inFIGS.10and11, in the present example embodiment, the second flow path50extends in a rectangular wave shape along the circumferential direction. The second flow path50includes a plurality of axial flow path portions51, a plurality of first circumferential flow path portions52a, and a plurality of second circumferential flow path portions52b. The plurality of axial flow path portions51extend in the axial direction. The plurality of axial flow path portions51are arranged at intervals in the circumferential direction. In the present example embodiment, the axial flow path portion51is provided in the motor housing11. More specifically, the axial flow path portion51is provided in the first housing13. As illustrated inFIG.10, the two axial flow path portions51located on the lower side among the plurality of axial flow path portions51are disposed with the groove portion93ainterposed therebetween in the circumferential direction. As illustrated inFIG.1, the plurality of axial flow path portions51include an axial flow path portion51cdivided into two in the axial direction by a partition wall51d. The axial flow path portion51cincludes an upstream flow path portion51aand a downstream flow path portion51b. In the present example embodiment, the upstream flow path portion51ais a portion of the axial flow path portion51clocated on one side in the axial direction with respect to the partition wall51d. In the present example embodiment, the downstream flow path portion51bis a portion of the axial flow path portion51clocated on the other side in the axial direction with respect to the partition wall51d. As illustrated inFIG.10, the first circumferential flow path portion52aand the second circumferential flow path portion52bextend in the circumferential direction. The plurality of first circumferential flow path portions52aare arranged at intervals in the circumferential direction. The plurality of second circumferential flow path portions52bare arranged at intervals in the circumferential direction. The first circumferential flow path portion52aconnects the ends on one side in the axial direction of the axial flow path portions51adjacent to each other in the circumferential direction. The second circumferential flow path portion52bconnects the ends on the other side in the axial direction of the axial flow path portions51adjacent to each other in the circumferential direction. The ends on both sides in the axial direction of the axial flow path portion51are alternately connected by the first circumferential flow path portion52aand the second circumferential flow path portion52b, so that the second flow path50has a rectangular wave shape. The plurality of first circumferential flow path portions52ainclude the first circumferential flow path portion52ccircumferentially across one side in the axial direction of the groove portion93a. The first circumferential flow path portion52cis the first circumferential flow path portion52alocated on the lowermost side among the plurality of first circumferential flow path portions52a. The circumferential dimension of the first circumferential flow path portion52cis larger than the circumferential dimension of the other first circumferential flow path portions52a. The through hole13eis located above the other circumferential side (−θ side) portion of the first circumferential flow path portion52c. The plurality of second circumferential flow path portions52binclude a pair of second circumferential flow path portions52bthat circumferentially sandwich the end on the other side in the axial direction of the groove portion93aand the connection portion93b. That is, in the present example embodiment, the end on the other side in the axial direction of the groove portion93aand the connection portion93bare located between the second circumferential flow path portions52badjacent to each other in the circumferential direction. As illustrated inFIG.11, in the present example embodiment, the first circumferential flow path portion52ais provided across the motor housing11and the transmission housing12. More specifically, the first circumferential flow path portion52ais provided across the first housing13and the third housing15. The first circumferential flow path portion52ais configured by axially connecting a portion provided on the end surface on one side in the axial direction of the first housing13and a groove recessed from the end surface on the other side in the axial direction of the third housing15to one side in the axial direction. In the present example embodiment, the second circumferential flow path portion52bis provided across the first housing13and the second housing14. That is, the second flow path50in the present example embodiment is provided across the first housing13and the second housing14. The second circumferential flow path portion52bis configured by axially connecting a portion provided on the end surface on the other side in the axial direction of the first housing13and a groove recessed from the end surface on one side in the axial direction of the second housing14to the other side in the axial direction. An end on one side in the axial direction of a partition wall52dthat partitions the pair of axial flow path portions51connected by the first circumferential flow path portion52ain the circumferential direction is disposed away from an end surface on one side in the axial direction of the first housing13on the other side in the axial direction. The end on the other side in the axial direction of a partition wall52ethat partitions the pair of axial flow path portions51connected by the second circumferential flow path portion52bin the circumferential direction is disposed away from the end surface on the other side in the axial direction of the first housing13on one side in the axial direction. In the axial flow path portion51, the water W flows in the axial direction. The directions in which the water W flows in the axial flow path portions51adjacent to each other in the circumferential direction are opposite to each other. In the first circumferential flow path portion52aand the second circumferential flow path portion52b, the water W flows in one circumferential direction (−θ direction). The first circumferential flow path portion52aconnects an end on one side in the axial direction of the axial flow path portion51through which the water W flows in the direction toward one side in the axial direction and an end on one side in the axial direction of the axial flow path portion51through which the water W flows toward the other side in the axial direction. The second circumferential flow path portion52bconnects the end on the other side in the axial direction of the axial flow path portion51through which the water W flows in the direction toward the other side in the axial direction and the end on the other side in the axial direction of the axial flow path portion51through which the water W flows in the direction toward the one side in the axial direction. As illustrated inFIG.1, the second flow path50includes an inflow flow path portion53aand an outflow flow path portion53b. In the present example embodiment, the inflow flow path portion53aand the outflow flow path portion53bpass through the inside of the inverter unit80. The water W flows into the inflow flow path portion53afrom the outside of drive device100. The water W flowing into the inflow flow path portion53aflows into the upstream flow path portion51a. The water W flowing into the upstream flow path portion51aflows around the motor20while flowing along a rectangular wave-shaped flow path configured by the axial flow path portion51, the first circumferential flow path portion52a, and the second circumferential flow path portion52b, and flows into the outflow flow path portion53bfrom the downstream flow path portion51b. The water W flowing into the outflow flow path portion53bflows out of the drive device100. As illustrated inFIG.2, the housing10includes an oil supply path95. The oil supply path95extends from the inside of the transmission housing12to penetrate the second facing wall15ain the axial direction. In the present example embodiment, the oil supply path95penetrates the first facing wall13ain the axial direction and extends to the inside of the motor housing11. As illustrated inFIG.5, the oil supply path95has a supply port13hfor supplying the oil O to the bearing72held by the bearing holding portion13c. In the present example embodiment, the supply port13his an opening that opens in the surface of the first hole13gon the other side in the axial direction of the first facing wall13a. The supply port13his open to the inside of the motor housing11. As illustrated inFIG.3, the supply port13his located above the central axis J1. The supply port13his open to the inside of the penetration portion13f. When viewed in the axial direction, the supply port13hoverlaps the penetration portion13f. In the present example embodiment, the oil supply path95includes the first hole13g, the second hole15g, the first gutter portion17, and the second gutter portion18. As indicated by a broken arrow inFIG.5, a part of the oil O that has been scraped up by the ring gear62aand entered the second gutter portion18passes through the second hole15gand flows into the first gutter portion17in the space S. The oil O flowing into the first gutter portion17flows in the first gutter portion17, passes through the first hole13g, and is supplied from the supply port13hinto the motor housing11. The oil O discharged from the supply port13hflows into the bearing holding portion13cvia the penetration portion13fand is supplied to the bearing72. According to the present example embodiment, at least a part of the second flow path50is located radially outside the motor20. Therefore, the motor20can be cooled by the water W flowing in the second flow path50. In the present example embodiment, the stator40can be cooled by the water W flowing in the second flow path50. At least a part of the collection flow path93is located radially outside the second flow path50. Therefore, the collection flow path93can be disposed close to the second flow path50. As a result, the oil O passing through the collection flow path93can be easily cooled by the water W flowing in the second flow path50. Therefore, the temperature of the oil O flowing into the transmission housing12from the collection flow path93can be lowered. Therefore, the temperature of the oil O supplied from the inside of the transmission housing12to the inside of the motor housing11by the first supply flow path91and the second supply flow path92can be made relatively low. As a result, the relatively low-temperature oil O can be supplied to the motor20accommodated in the motor housing11. Therefore, the motor20can be suitably cooled by the relatively low-temperature oil O. As described above, in the present example embodiment, the motor20can be suitably cooled by the water W and the oil O. Therefore, the cooling efficiency of the motor20can be improved. In addition, it is possible to easily cool the motor20without providing a cooler such as an oil cooler in order to cool the oil O. Therefore, the number of components of the drive device100can be reduced by the absence of the cooler. According to the present example embodiment, the second flow path50extends in a rectangular wave shape along the circumferential direction. Therefore, the portion of the housing10where the second flow path50is provided can be widened, and the motor20can be more suitably cooled by the water W flowing in the second flow path50. Therefore, the cooling efficiency of the motor20can be further improved. In addition, in a case where the housing10is divided into a plurality of members as in the present example embodiment, the second flow path50is easily formed by using each member constituting the housing10for constituting the second flow path50. Specifically, in the present example embodiment, the second flow path50can be easily formed by providing the hole axially penetrating the first housing13and closing both axial sides of the hole with the second housing14and the third housing15. According to the present example embodiment, the first circumferential flow path portion52ais provided across the first housing13and the third housing15. Therefore, for example, as compared with a case where the entire first circumferential flow path portion52ais provided in the third housing15, it is possible to suppress an increase in size of the third housing15in the axial direction. Thus, the drive device100can be prevented from increasing in size in the radial direction. In addition, since the second flow path50can be suitably extended to one side in the axial direction from the stator40, the range of the stator40that can be cooled by the second flow path50can be widened. As a result, the motor20can be more suitably cooled by the water W flowing in the second flow path50. According to the present example embodiment, the collection flow path93includes the groove portion93aprovided on the inner peripheral surface of the motor housing11and extending in the axial direction, the collection flow path body93clocated radially outside the groove portion93aand extending in the axial direction and connected to the inside of the transmission housing12, and the connection portion93bconnecting the groove portion93aand the collection flow path body93c. Therefore, at least a part of the oil O supplied into the motor housing11by the first supply flow path91and the second supply flow path92can flow into the collection flow path93from the groove portion93a. Further, the oil O flowing into the groove portion93acan be sent into the transmission housing12via the connection portion93band the collection flow path body93c. As a result, the oil O in the motor housing11can be easily returned into the transmission housing12by the collection flow path93. According to the present example embodiment, at least a part of the collection flow path body93cis located radially outside the second flow path50. Therefore, the oil O flowing in the collection flow path body93ccan be easily cooled by the water W flowing in the second flow path50. According to the present example embodiment, the connection portion93bconnects the end on the other side in the axial direction of the groove portion93aand the end on the other side in the axial direction of the collection flow path body93c. That is, the position where the groove portion93aand the collection flow path body93care connected by the connection portion93bcan be set to a position relatively distant from the transmission housing12in the axial direction. Therefore, it is possible to increase the distance by which the oil O flows from the connection portion93binto the collection flow path body93cand reaches the inside of the transmission housing12. As a result, it is possible to extend the time during which the oil O flowing in the collection flow path body93ccan be cooled by the water W flowing in the second flow path50. Therefore, the oil O flowing in the collection flow path body93ccan be suitably cooled by the water W flowing in the second flow path50. Therefore, the lower temperature oil O can be easily supplied to the motor20. As a result, the cooling efficiency of the motor20can be further improved. According to the present example embodiment, the plurality of first circumferential flow path portions52ainclude the first circumferential flow path portion52ccircumferentially across one side in the axial direction of the groove portion93a. The connection portion93bis located between the second circumferential flow path portions52badjacent to each other in the circumferential direction. As described above, the first circumferential flow path portion52cstraddles the groove portion93aon the side opposite to the side where the connection portion93bis provided in the axial direction, so that the connection portion93bcan be extended from the radially inner side of the second flow path50to the radially outer side of the second flow path50without interfering with the second flow path50. As a result, at least a part of the collection flow path body93ccan be disposed radially outside the second flow path50without interfering with the second flow path50. According to the present example embodiment, the second supply flow path92has the introduction flow path portion92aextending in the axial direction from the inside of the transmission housing12. At least a part of the introduction flow path portion92ais located radially outside the second flow path50. Therefore, the introduction flow path portion92acan be disposed close to the second flow path50. Thus, the oil O passing through the introduction flow path portion92acan be easily cooled by the water W flowing in the second flow path50. Therefore, the temperature of the oil O supplied to the inside of the motor housing11by the second supply flow path92can be made relatively low. Therefore, the motor20accommodated in the motor housing11can be more suitably cooled by the oil O. Therefore, the cooling efficiency of the motor20can be further improved. According to the present example embodiment, the introduction flow path portion92ais disposed adjacent to the collection flow path93in the circumferential direction. Therefore, the introduction flow path portion92aand the collection flow path93can be collectively disposed. This can suppress complication of the structure of the housing10. According to the present example embodiment, the collection flow path93and the second flow path50are provided across the first housing13and the second housing14, respectively. Therefore, the collection flow path93and the second flow path50can be enlarged in the axial direction. As a result, it is easy to increase the number of portions of the collection flow path93disposed close to the second flow path50. Therefore, the oil O flowing in the collection flow path93can be more easily cooled by the water W flowing in the second flow path50. In addition, since the second flow path50can be enlarged in the axial direction, the range of the motor20that can be cooled by the water W flowing in the second flow path50can be widened in the axial direction. As a result, the entire stator core41and the coil ends42aand42bprotruding from the stator core41to both sides in the axial direction can be easily cooled by the water W flowing in the second flow path50. As described above, the cooling efficiency of the motor20can be further improved. According to the present example embodiment, the first housing13and the second housing14are fixed to each other at positions radially inside the collection flow path93and adjacent to the second flow path50in the circumferential direction. In the present example embodiment, the first housing13and the second housing14are fixed to each other at the positions by the bolt10dtightened into the female screw hole13i. As a result, the first housing13and the second housing14can be fixed at positions close to both the collection flow path93and the second flow path50. Therefore, it is possible to prevent portions of the first housing13and the second housing14constituting the collection flow path93from being separated from each other. In addition, it is possible to prevent portions of the first housing13and the second housing14constituting the second flow paths50from being separated from each other. This can suppress leakage of the oil O from the collection flow path93and leakage of the water W from the second flow path50. Further, it is possible to prevent the oil O leaking from the collection flow path93from entering the second flow path50and mixing with the water W. In addition, it is possible to suppress the water W leaking from the second flow path50from entering the collection flow path93and mixing with the oil O. According to the present example embodiment, the partition wall19has the through hole19aconnecting the inside of the motor housing11and the inside of the transmission housing12. Therefore, the oil O supplied into the motor housing11can be returned into the transmission housing12from the through hole19ain addition to the collection flow path93. As a result, the amount of the oil O returned from the motor housing11into the transmission housing12can be increased. For example, when the housing10is configured by two separate members constituting the motor housing11and two separate members constituting the transmission housing12as in the present example embodiment, the motor housing11and the transmission housing12are provided separately. In such a case, conventionally, the motor housing11and the transmission housing12are provided with structures for lubricating the bearings separately. Therefore, there is a problem that the manufacturing cost of the drive device100increases due to a complicated structure of the housing10or the use of a relatively expensive bearing that does not require the supply of lubricating oil. The relatively expensive bearing requiring no supply of lubricating oil is, for example, a bearing provided with semi-solid grease. On the other hand, according to the present example embodiment, the housing10has the oil supply path95extending axially through the second facing wall15afrom the inside of the transmission housing12. The oil supply path95has the supply port13hfor supplying the oil O to the bearing72held by the first facing wall13aof the motor housing11. The supply port13his located above the central axis J1. Therefore, the oil O discharged from the supply port13hcan be dropped by gravity and supplied to the bearing72provided in the motor housing11among the bearings supporting the rotor30rotatable about the central axis J1. That is, a part of the oil O in the transmission housing12can be supplied to the bearing72provided in the motor housing11by the oil supply path95. In this manner, the bearing72provided in the motor housing11can be lubricated using the bearing lubrication structure provided in the transmission housing12. That is, in the drive device100, the bearing72provided in the motor housing11can be lubricated using the oil O in the transmission housing12while the motor housing11and the transmission housing12are configured to be separable. Therefore, it is possible to suppress complication of the structure of the housing10, and it is not necessary to use a bearing that does not require supply of lubricating oil as the bearing72. Therefore, it is possible to suppress an increase in manufacturing cost of the drive device100. According to the present example embodiment, the bearing holding portion13cis provided on the surface on the other side in the axial direction of the first facing wall13a. The oil supply path95penetrates the first facing wall13ain the axial direction and extends to the inside of the motor housing11. The supply port13his open to the inside of the motor housing11. Therefore, even when the bearing72held by the bearing holding portion13cis located inside the motor housing11, the oil O can be supplied to the bearing72by the oil supply path95. Further, according to the present example embodiment, the bearing holding portion13chas the penetration portion13fthat penetrates the bearing holding portion13cin the radial direction. The supply port13his open to the inside of the penetration portion13f. Therefore, the oil O discharged from the supply port13his easily supplied from the penetration portion13fto the inside of the bearing holding portion13c. As a result, the oil O can be more easily supplied to the bearing72. According to the present example embodiment, the oil supply path95includes the first hole13gaxially penetrating the first facing wall13a, the second hole15gaxially penetrating the second facing wall15a, and the first gutter portion17located between the first facing wall13aand the second facing wall15ain the axial direction and connecting the first facing wall13aand the second facing wall15a. The first gutter portion17is connected to a portion located on the lower side of the first hole13gin the surface on one side in the axial direction of the first facing wall13aand a portion located on the lower side of the second hole15gin the surface on the other side in the axial direction of the second facing wall15a. Therefore, the oil O in the transmission housing12can be supplied into the motor housing11through the second hole15g, the first gutter portion17, and the first hole13gin this order. As a result, the oil O in the transmission housing12can be more suitably supplied to the bearing72in the motor housing11. According to the present example embodiment, the oil supply path95includes the second gutter portion18located inside the transmission housing12. The second gutter portion18is connected to a portion located below the second hole15gin the surface on one side in the axial direction of the second facing wall15a. Therefore, for example, a part of the oil O scattered in the transmission housing12by being scraped up by the ring gear62acan be received by the second gutter portion18. In addition, at least a part of the oil O received by the second gutter portion18can flow into the second hole15g. As a result, the oil O in the transmission housing12can be more suitably supplied to the bearing72in the motor housing11through the second hole15g, the first gutter portion17, and the first hole13gin this order. In addition, according to the present example embodiment, the second facing wall15ahas the through hole15hconnecting the space S located between the first facing wall13aand the second facing wall15ain the axial direction and the inside of the transmission housing12. Therefore, for example, the oil O leaking from the inside of the first gutter portion17can be returned into the transmission housing12through the through hole15h. Thus, the oil O can be prevented from accumulating in the space S. According to the present example embodiment, the first facing wall13ahas the through hole13econnecting the space S located between the first facing wall13aand the second facing wall15ain the axial direction and the inside of the motor housing11. Therefore, the inside of the motor housing11and the inside of the transmission housing12can be connected by the through hole13e, the space S, and the through hole15h. As a result, the above-described through hole19ais formed, and at least a part of the oil O supplied into the motor housing11can be returned into the transmission housing12. The present disclosure is not limited to the above-described example embodiment, and other structures and other methods may be employed within the scope of the technical idea of the present disclosure. The first flow path may have any configuration as long as the second flow path includes the supply flow path and the collection flow path. In the above-described example embodiment, the first supply flow path91and the second supply flow path92are provided as the supply flow path, but the present disclosure is not limited thereto. As the supply flow path, only one of the first supply flow path91and the second supply flow path92may be provided. The collection flow path extending from the inside of the motor housing to the inside of the transmission housing may have any configuration as long as at least a part thereof is located radially outside the second flow path. When the motor housing has the first housing and the second housing, the collection flow path may be provided only in the first housing in the motor housing. The shape and size of the groove, the shape and size of the connection portion, and the shape and size of the collection flow path body are not particularly limited. The groove and the connection portion may not be provided. The second flow path may have any shape. The first circumferential flow path portion may not be provided across the first housing and the third housing. The second circumferential flow path portion may not be provided across the first housing and the second housing. For example, the second flow path may extend in a rectangular wave shape along the axial direction by connecting axial ends of a plurality of flow path portions extending in the circumferential direction and arranged at intervals in the axial direction. The second flow path may extend spirally. The type of the first fluid flowing into the first flow path and the type of the second fluid flowing into the second flow path are not particularly limited. The first fluid and the second fluid may be the same type of fluid. The first fluid may be an insulating liquid or water. When the first fluid is water, the surface of the stator may be subjected to an insulation treatment. The second fluid may be oil. The oil supply path extending from the inside of the transmission housing through the second facing wall in the axial direction may have any configuration as long as the oil supply passage has a supply port that is located above the central axis and supplies oil to the bearing. When the bearing holding portion provided on the first facing wall of the motor housing is provided on the surface on one side in the axial direction of the first facing wall, that is, the surface of the first facing wall facing the transmission housing side, the oil supply path may penetrate only the second facing wall and may not penetrate the first facing wall. In this case, for example, the supply port of the oil supply path is open to the space between the first facing wall and the second facing wall. The oil supply path may not have at least one of the first hole, the second hole, the first gutter portion, and the second gutter portion. The oil supply path may be formed of, for example, a tubular member such as a pipe. The oil supply path may not be provided. The number of housings constituting the housing is not particularly limited. The housing may be configured such that two housings are fixed to each other, three housings are fixed to each other, or five or more housings are fixed to each other. The housing constituting the housing may include a housing having a part of the motor housing and a part of the transmission housing. The application of the drive device to which the present disclosure is applied is not particularly limited. For example, the drive device may be mounted on a vehicle for a purpose other than the purpose of rotating the axle, or may be mounted on a device other than the vehicle. The posture when the drive device is used is not particularly limited. The central axis of the motor may be inclined with respect to the horizontal direction orthogonal to the vertical direction or may extend in the vertical direction. Features as described above in the present specification may be combined appropriately as long as no conflict arises. Features of the above-described example embodiments and the modifications thereof may be combined appropriately as long as no conflict arises. While example embodiments of the present disclosure have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the present disclosure. The scope of the present disclosure, therefore, is to be determined solely by the following claims. | 91,650 |
11863053 | DETAILED DESCRIPTION OF THE ENABLING EMBODIMENTS Example embodiments of a lubricant supported electric motor in accordance with the present disclosure will now be more fully described. Each of these example embodiments are provided so that this disclosure is thorough and fully conveys the scope of the inventive concepts, features and advantages to those skilled in the art. To this end, numerous specific details are set forth such as examples of specific components, devices and mechanisms associated with the lubricant supported electric motor to provide a thorough understanding of each of the embodiments associated with the present disclosure. However, as will be apparent to those skilled in the art, not all specific details described herein need to be employed, the example embodiments may be embodied in many different forms, and thus should not be construed or interpreted to limit the scope of the disclosure. FIGS.1-2illustrate a lubricant supported electric motor10in accordance with an aspect of the disclosure. As best illustrated inFIG.1, the lubricant supported electric motor10includes a stator12and a rotor14extending along an axis A and movably (i.e., rotatably) disposed within the stator12to define a gap16(also shown as “G” inFIG.1) therebetween. In an alternative arrangement, the stator12and the rotor14can be reversed, with the stator12extending along the axis A and the rotor14rotatably disposed around the stator12, without departing from the scope of the subject disclosure. A lubricant18is disposed in the gap16for supporting the rotor14within the stator12, and providing continuous contact between these components. The lubricant18may therefore act as a buffer (e.g., suspension) between the stator12and the rotor14minimizing or preventing contact therebetween. In other words, the lubricant18prevents direct contact between the stator12and rotor14and provides an electric lubricant supported motor10which is robust to shock and vibration loading due to the presence of the lubricant18. Additionally, and alternatively, a substantially incompressible lubricant18may be used in order to minimize the gap between the stator12and rotor14. As further illustratedFIG.1, the stator12defines a passageway20disposed in fluid communication with the gap16for introducing the lubricant18. However, the passageway20could be provided on any other components of the lubricant supported electric motor10without departing from the subject disclosure. According to an aspect, the lubricant18may be cycled or pumped through the passageway20and into the gap16in various ways. For example, a high pressure source (e.g., a pump)24of the lubricant18may be fluidly coupled to a low pressure source (e.g., a sump)26of the lubricant18, where the lubricant18may move from the high pressure source to the lower pressure source, through the passageway20and into the gap16. Rotation of the rotor14relative to the stator12may operate as a self-pump to drive lubricant18through the passageway20and into the gap16. As further illustrated inFIG.1, the rotor14is interconnected to a drive assembly22for coupling the lubricant supported electric motor10to one of the plurality of wheels of a vehicle. For example, in one instance, the drive assembly22may include a planetary gear system. Alternatively, the drive assembly22may include one or more parallel axis gears. The stator12and rotor14are configured to exert an electromagnetic force therebetween to convert electrical energy into mechanical energy, moving the rotor14and ultimately driving the wheel coupled to the lubricant supported electric motor10via the drive assembly22. The drive assemblies20may provide one or more reduction ratios between the lubricant supported electric motor10and the wheel in response to movement of the rotor14. As best illustrated inFIG.2, the rotor14presents an inner raceway28and the stator12presents an outer raceway30. The inner and outer raceways28,30collectively define at least one hydrostatic support chamber32which is established by a portion of the gap16and receives the lubricant18for supporting the rotor14within the stator12. For example, the hydrostatic support chamber32which is established in the gap16between the inner and outer raceways28,30determines a dynamic pressure developed when the lubricant supported electric motor10is in hydrodynamic mode. The gap16between the inner and outer raceways28,30also determines the pressure in the hydrostatic support chamber32when the lubricant supported electric motor10is in hydrostatic mode. In a preferred embodiment, the at least one hydrostatic support chamber32includes a plurality of hydrostatic support chambers32spaced circumferentially around and between the stator12and the rotor14and which each have their individualized pressure in the hydrodynamic and hydrostatic modes. For example, as illustrated inFIG.2, in a preferred arrangement, the at least one hydrostatic support chamber32can include four hydrostatic support chambers32circumferentially spaced from one another around the axis A. However, any number of hydrostatic support chambers32can be utilized without departing from the scope of the subject disclosure. As further illustrated inFIG.2, the stator12defines a plurality of passageways20each disposed in fluid communication with a respective one of the hydrostatic support chambers32for supplying lubricant thereto. As further illustrated inFIG.2, the lubricant supported electric motor10includes a monitoring port34disposed in fluid communication with each hydrostatic support chamber32. A sensor36is coupled to the monitoring port30for sensing the operating characteristic of the lubricant18disposed within the at least one hydrostatic support chamber28. For example, the sensor36can be a pressure sensor configured to sense a pressure of the lubricant18disposed within the at least one hydrostatic support chamber28. However, the sensor36could also be comprised of other sensors36, such as a temperature sensor for sensing a temperature of the lubricant18or a viscosity sensor for sensing a viscosity of the lubricant, without departing from the scope of the subject disclosure. As further illustrated inFIG.2, when the at least one hydrostatic support chamber32includes a plurality of hydrostatic support chambers32, a monitoring port34and sensor36can be disposed in communication with each hydrostatic support chamber32. In other words, in a preferred arrangement, each hydrostatic support chamber32includes its own respective monitoring port34and sensor36for providing individualized monitoring of the plurality of hydrostatic support chambers32. The utilization of the monitoring port34and the sensor36advantageously improves the performance of the lubricant supported electric motor10by providing the ability to detect operating characteristics of the lubricant18disposed within each of the hydrostatic support chambers28, which is used and analyzed to detect certain operating characteristics of the lubricant supported electric motor10such as oil supply faults, stable or instable motor operation, as well as others. In other words, the monitoring port30and the sensor32facilitates real-time diagnostics and prognostics for the lubricant supported electric motor10. As illustrated inFIG.2, each sensor36is preferably electrically connected to a controller38for sending the monitored operating characteristic of the lubricant18and/or hydrostatic support chamber32to the controller38for further evaluation to determine the operating characteristic of the lubricant supported electric motor10and provide the real-time diagnostics and prognostics. For example, the operating characteristics (e.g., pressure, temperature, viscosity) sensed by the plurality of sensors32can be used by the controller38to:verify correct oil flow into the hydrostatic support chamber28using a pressure-based flow model. For example, at a known oil flow rate a known pressure should result. If the pressure is too high this may indicate inner to outer raceway clearances that are too close. If the pressure is too low this may indicate leakage in the oil supply or an inner to outer raceway clearance that is too large;observe rotor vibration related to pressure fluctuations within the hydrostatic support chamber32(e.g. rotor motions such as translation, rocking, whirl). For example pressure fluctuations may be caused by the rotor moving away from being centered in the stator and changing the inner to outer raceway clearance;observe rotor centering related to pressure relationships of hydrostatic support chambers32diametrically opposed to each other. For example, when the rotor moves closer to a chamber on the top of the stator, the pressure in that chamber will increase due to the tighter clearance between the inner raceway and the outer raceway. At the same time, the rotor will move further away from the chamber at the bottom of the stator, which will decrease the pressure in the chamber due to the looser clearance between the inner and outer raceway. The combination of increasing pressure at the top and decreasing pressure at the bottom indicates that the rotor is moving off of center;estimate rotor position measurement to allow feedback control of rotor position. For example, the rotor centering position can be estimated using the method described above. This position estimate can be used to change the electric current supplied to the motor or the oil pressure supplied to the motor to cause the rotor to stay as close to a centered position as possible;estimate lubricant properties (e.g., viscosity) when used in conjunction with oil pump volumetric flow. For example, as the lubricant heats up it will typically become less viscous, which results in a lower chamber pressure for a given flow rate. Observing the decrease in pressure across multiple chambers may be an indicator of reduced viscosity; anddiagnose system faults in the lubricant, lubricant pump, and pressure sensing systems. For example, for a given flow rate of oil into a chamber, if the pressure is not within a known normal range, than a fault in the oil supply or raceways is indicated. In an embodiment, the controller38is also disposed in communication with a component of the lubricant supported electric motor10and can use the monitored characteristic of the lubricant18and/or hydrostatic support chamber28in conjunction with other measured parameters of the lubricant supported electric motor10(e.g., motor speed, motor temperature, central oil supply pressure, etc.) to provide further diagnostics and prognostics of the lubricant supported electric motor10. The incorporation of monitoring port34and sensor36advantageously provides for optimized performance and operating characteristics for the lubricant supported electric motor10in real-time. In other words, the monitoring port34and sensor36allows for the monitoring and diagnosing of the motor's performance in real-time using, for example, pressure measurements of the lubricant18in the hydrostatic support chamber32. This improved monitoring of the motor's performance ultimately leads to better overall performance of the lubricant supported electric motor10compared to its static and very conservatively designed counterparts. The foregoing description of the embodiments has been provided for purposes of illustration and description. It is not intended to be exhaustive or to limit the disclosure. Individual elements or features of a particular embodiment are generally not limited to that particular embodiment, but, where applicable, are interchangeable and can be used in a selected embodiment, even if not specifically shown or described. The same may also be varied in many ways. Such variations are not to be regarded as a departure from the disclosure, and all such modifications are intended to be included within the scope of the disclosure. | 11,885 |
11863054 | DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS With reference toFIG.1for a kinetic power generation unit1in accordance with an embodiment of the present invention, the kinetic power generation unit1has the functions of both electric motor1aand power generator1b. The kinetic power generation unit1is composed of the electric motor1aand the power generator1band the electric motor1and the power generator1brun individually, synchronously outputting the kinetic mechanical energy and electrical energy, the kinetic power generation unit1comprises a stator10, and a rotor20installed in the middle inside the stator10, wherein the stator10is in a hollow form and has a plurality of wire slots11arranged circularly on an inner wall of the stator10, wherein an exciting winding12aof the electric motor1aand a field winding12bof the power generator1bare installed alternately into the wire slots11. For example, the exciting winding12of the electric motor1ais installed into an odd-numbered wire slot11and the field winding12bof the power generator1bis installed into an even-numbered wire slot11and the electrical coil winding12aand the field winding12bare individually composed of a plurality of enameled wires. InFIG.2, the kinetic power generation unit1further comprises a storage battery30, and the current outputted by the storage battery30in a phase voltage (that is, split-phase power-on and power-off as shown inFIG.7, the field winding12bof the power generator1bhas not be electrified) is supplied to the exciting winding12aof the electric motor1athrough a DC brushless drive controller40in a sequential phase change and electrical conduction method. When the exciting: winding12aof the electric motor1ais electrically powered on, a magnetic field coupling will be generated inside the stator10to drive the rotor20to rotate. After the rotor20rotates, the kinetic energy is outputted and supplied. Since the rotor20is rotates in the middle surrounded by the exciting winding12aof the electric motor1aand the field winding12bof the power generator1b, and the exciting winding12aof the electric motor1aor the field winding12bof the power generator1bmove to cut the magnetic lines of force, an induction potential is generated, and the generated electricity is led out from a terminal, and the electricity can be outputted as AC power for use after the voltage of the electricity is transformed by a voltage regulator (AVR)50and a transformer60. On the other hand, the electricity can be outputted as DC power for charging the aforementioned storage battery30to increase the power of the storage battery30after the electricity is rectified by another transformer60and a rectifier70, thereby becoming a loop of an interconnected cycling cogeneration. Since the exciting winding12aof the electric motor1aand the field winding12bof the power generator1bare installed alternately in the wire slots11of the stator10, and the DC brushless drive controller40changes their phases and electrically conducts both of them to supply current to the exciting winding12aof the electric motor1aand the exciting coil winding12aof the electric motor1aand the field winding12bof the power generator1bhave a same magnetic field coupling and the rotor20is shared by the electric motor1aand the power generator1b, so that the rotational directions of the rotor20of the electric motor1aand the rotor20of the power generator1bis the same. According to Fleming left hand rule (sometimes called motor rule) and Fleming right hand rule (sometimes called generator rule), the interference effect (or negative effect) of the counter electromotive force generated by the electric motor1aafter its power-on will be the positive effect of the power generator1b; and the interference effect (or negative effect) of the counter electromotive force generated by the power generator1bwill be the positive effect of the electric motor1a. InFIG.3, there can be a plurality of kinetic power generation units1,2,3applied to the present invention, and the kinetic power generation units1,2,3are connected in series with each other and installed coaxially with the same rotor20to increase the output and electric power. In an embodiment of the present invention, the exciting coil winding12aof the electric motor1aand the field winding12bof the power generator1bof the kinetic power generation unit1are a single phase or the three-phase configuration; the three-phase configuration uses a three-phase coil winding of a delta connection as shown inFIG.4, and wherein, a connection point a, b, c between phases of the exciting coil winding of the electric motor1ais an output point of the three-phase alternating current (that is, the voltage output point of the counter electromotive force); and a center point A, B, C of each coil winding is a voltage input point and the field winding of the power generator uses the delta connection which are all the voltage output points thereby completing the configuration of the three-phase coil winding. In an embodiment of the present invention, the exciting coil winding12aof the electric motor1aand the field winding12bof the power generator1bof the kinetic power generation unit1are a single phase or the three-phase configuration; the three-phase configuration uses a three-phase coil winding of a Y connection as shown inFIG.5, and wherein an end point a, b, c on each phase while the Y connection applying to the exciting coil winding12ais a power output point; a neutral point of the coil winding is in a delta connection, and center points A, B, C of the delta connection are voltage input points, and the center point A, B, C are voltage output points while the Y connection applying to the field winding12bof the power generator1b, thereby completing the configuration of the three-phase coil winding. In an embodiment of the present invention, the exciting coil winding12aof the electric motor1aand the field winding12bof the power generator1bof the kinetic power generation unit1are the delta & Y connections configurations; the three-phase configuration uses a three-phase coil winding of the delta & Y connections as shown inFIG.6. The Y connection as shown inFIG.5is configured inside the delta connection as shown inFIG.4, and the end point in each phase of the Y connection is coupled to the connection point a, b, c between phases of the delta connection, and the connection point a, b, c between phases of the delta connection is a three-phase alternating current output point (that is, the voltage output point of the counter electromotive force), and the center points A, B, C of each coil winding of the delta connection are the phase voltage input points, and the field winding12bof the power generator1bare voltage output points, thereby achieving the joint combination effect. The DC brushless drive controller40sequentially changes phases of the exciting coil winding of the electric motor1aof the kinetic power generation unit1and sends the phase to the phase voltage. In an embodiment as shown inFIG.7, the Y connection is adopted, and A˜B, A˜C, and B˜C are electrically conducted sequentially, so as to achieve the effect of sequentially changing phases and electrically conducting the coil winding. In summation, the present invention complies with the patent application requirements, and thus is duly filed for patent application. While the invention is described in some detail hereinbelow with reference to certain illustrated embodiments, it is to be understood that there is no intent to limit it to those embodiments. On the contrary, the aim is to cover all modifications, alternatives and equivalents falling within the spirit and scope of the invention as defined by the appended claims. REFERENCE NUMBERALS IN DRAWING FIGURES Kinetic power generation units1,2,3Electric motor1aPower generator1bStator10Wire slot11Exciting winding12aField winding12bRotor20Storage battery30DC brushless drive controller40Voltage regulator (AVR)50Transformer60Rectifier70 | 8,006 |
11863055 | Like reference numbers and designations in the various drawings indicate like elements. DETAILED DESCRIPTION The present invention encompasses circuits and methods relating to a power converter that can be started and operated in a reversed unidirectional manner or in a bidirectional manner while providing sufficient voltage for an associated auxiliary circuit and start-up without the added external circuitry of a voltage booster and/or a pre-charge circuit—that is, with zero external components for some embodiments, or a reduced number of external components for other embodiments. Selective Voltage Inputs to Converter Auxiliary Circuit FIG.3Ais a block diagram300of a first embodiment of the present invention, showing a dual voltage arrangement for an auxiliary circuit302for a converter circuit304. The converter circuit304may be like the converter circuit102ofFIG.1AandFIG.2A, but the voltage labels have been changed to VAand VBrather than VINand VOUT, to reflect that the converter circuit304may be operated in a reversed unidirectional manner or in a bidirectional manner. The auxiliary circuit302may be similar to the auxiliary circuit110ofFIG.1A, with the addition of an input voltage selector306having inputs A and B respectively coupled to VAand VB. The input voltage selector306functions as an analog multiplexor, allowing selection of either VAor VBas the input voltage VINPUTto the auxiliary circuit302. Internal to the auxiliary circuit302, the voltage from VAor VBfrom the selected input A or B is used to power various subcircuits, such as a UVLO circuit124and a voltage regulator122(seeFIG.1B). The input voltage selector306may be integrated on the same integrated circuit (IC) die as part of the auxiliary circuit302, or may be a separate IC die, such as in a module containing an IC die embodiment of the auxiliary circuit302. In some embodiments, control of which input, A or B, to use as the input voltage VINPUTmay be positively controlled, for example, by a selection signal (or signals) from the controller108(not shown inFIG.3A) that selects input A or input B depending on which is coupled to the greater of the two applied voltages VA, VB. Further, input voltage selection may be controlled based on a knowledge of the existing forward or reverse configuration of the converter circuit304, rather than on a determination of the relative voltages at VAand VB. Such knowledge may be, for example, a mode setting signal from the controller108or a signal from a trim or memory component within the controller108. In other embodiments, the input voltage selector306may self-select the greater of the two applied voltages VAand VB(see examples below). Thus, for example, if the converter circuit304is used in a step-down power converter run in a forward (step-down) direction, then VAwill be the greater voltage at all times, and accordingly input A of the input voltage selector306will be selected to provide power to the auxiliary circuit302. However, if that same converter circuit304is run in a reverse (step-up) direction, then VBwill be the greater voltage until after the converter circuit304is operational, and accordingly input B of the input voltage selector306will be selected to provide power to the auxiliary circuit302during a startup phase, after which input A of the input voltage selector306may be selected (or is self-selected) to provide power to the auxiliary circuit302. After startup, during steady-state operation of the converter circuit304, it is generally more efficient to continue powering the auxiliary circuit302off the lower of VAor VBas long as that voltage meets or exceeds the specified minimum voltage VMIN. Similarly, if the converter circuit304is used in a step-up power converter run in a forward (step-up) direction, then VAwill be the greater voltage until the converter circuit304is operational, and accordingly input A of the input voltage selector306will be selected to provide power to the auxiliary circuit302during a startup phase, after which input B of the input voltage selector306may be selected (or is self-selected) to provide power to the auxiliary circuit302. However, if that same converter circuit304is run in a reverse (step-down) direction, then VBwill be the greater voltage at all times, and accordingly input B of the input voltage selector306will be selected to provide power to the auxiliary circuit302. Again, after startup, during steady-state operation of the converter circuit304, it is generally more efficient to continue powering the auxiliary circuit302off the lower of VAor VBas long as that voltage meets or exceeds the specified minimum voltage VMIN. FIG.3Ashows a single input voltage selector306for the auxiliary circuit302. In alternative embodiments, each subcircuit within the auxiliary circuit302that requires selection of VAor VBas an input voltage may have its own voltage selector circuit. For example,FIG.3Bis a block diagram320of a second embodiment of the present invention, showing a dual input voltage selector arrangement for an auxiliary circuit302for a converter circuit304. In the illustrated embodiment, the voltage regulator122has a respective first input voltage selector306ahaving inputs A and B respectively coupled to VAand VB, and the UVLO circuit124has a respective second input voltage selector306bhaving inputs A and B respectively coupled to VAand VB. One advantage of the illustrated embodiment is that the first and second input voltage selectors306a,306bcan be tailored to the circuit characteristics of the voltage regulator122and the UVLO circuit124. The first and second input voltage selectors306a,306bmay be identical circuits (but possibly with different circuit values) or may be different circuits, and there are a number of ways in which either input voltage selector306a,306bmay be implemented. For example,FIG.4Ais a schematic diagram of a first embodiment306a1of the first input voltage selector306aofFIG.3B. The voltage from VAof the converter circuit304is applied to a first diode D1coupled in series with a first resistor R1. In parallel, voltage from VBof the converter circuit304is applied to a second diode D2coupled in series with a second resistor R2. An output node, VSEL, is coupled to both resistors R1, R2(note that the order of the diodes D1, D2and corresponding resistors R1, R2may be reversed without affecting operation). The diodes D1, D2essentially isolate the voltage of either input from the other input. The output voltage at VSELis the maximum of VAand VB(minus the forward voltage of each diode), and the circuit self-selects the greater voltage to output as VSEL. To favor one input over the other input during steady-state operation for efficiency reasons, the value of the resistor in the favored path can be designed to be much lower than the value of the resistor in the disfavored path. As another example,FIG.4Bis a schematic diagram of a second embodiment306a2of the first input voltage selector306aofFIG.3B. The voltage from VAof the converter circuit304is applied to a first switch S1, which is optionally coupled in parallel with a first diode D1(which, for some switch technologies, may be an inherent parasitic diode), and voltage from VBof the converter circuit304is applied to a second switch S2, which is optionally coupled in parallel with a second diode D2(which again, for some switch technologies, may be an inherent parasitic diode). An output node, VSEL, is coupled to both switches S1, S2. The switches S1, S2may be positively controlled by selection signals, such as from the controller108. If the voltage from VAof the converter circuit304is detected to be higher than the voltage from VB, the controller108can close switch S1and open switch S2to pass through the input voltage from VAto VSEL. Conversely, if the voltage from VAis detected to be lower than the voltage from VB, the controller108can open switch S1and close switch S2to pass through the input voltage from VBto VSEL. Alternatively, the state of the switches S1, S2may be set by the controller108based on information regarding whether the converter circuit304is to operate in a forward or reverse direction. If the diodes D1, D2are optionally or inherently coupled in parallel with the respective switches S1, S2, the second embodiment306a2can operate like the first embodiment306a1ofFIG.4Aand self-select the greater of the input voltages VAand VBto output as VSELindependently of the switches S1, S2. Closing the switch coupled to the forward-conducting diode passes through more of the corresponding input voltage from either VAor VBto VSELby bypassing the diode forward voltage drop with a much smaller switch voltage drop. This allows for a more efficient solution while ensuring a greater voltage margin above the VMINneeded at VINPUTto the auxiliary circuit302. Either of the embodiments306a1,306a2shown inFIGS.4A and4Bmay be used as an input voltage selector306bfor the UVLO circuit124. As should be clear, other variants of the circuits shown inFIGS.4A and4B, as well as other analog multiplexor circuits, may be used for the input voltage selectors306,306a,306b. In other embodiments, the function of an input voltage selector may be more intimately integrated within the circuitry of the voltage regulator122and/or the UVLO circuit124. For example, positively controlled or self-selecting circuitry may enable or disable subcircuits within the voltage regulator122and/or the UVLO circuit124to effectively choose a subcircuit powered by one of VAor VBfrom the converter circuit304. For example,FIG.5Ais a schematic diagram of a first variant embodiment124aof the UVLO circuit124ofFIG.3B. The voltage from VAof the converter circuit304is applied to a first under-voltage lockout circuit UVLO A, while the voltage from VBof the converter circuit304is applied to a second under-voltage lockout circuit UVLO B. An output selector310coupled to UVLO A and UVLO B selects which under-voltage lockout circuit to use to output a CTRL signal to other circuitry in response to an input Select signal. The Select signal may be generated, for example, by the controller108based on a comparison of the voltage from VAand from VB. Alternatively, the Select signal may also be generated by the controller108based on information regarding whether the converter circuit304is to operate in a forward or reverse direction. In a variant embodiment, if the under-voltage lockout circuits UVLO A, UVLO B can be individually enabled and disabled, the output selector310may be omitted and the Select signal may be used to directly select the under-voltage lockout circuit UVLO A, UVLO B to be coupled to the CTRL output. As another example,FIG.5Bis a schematic diagram of a second variant embodiment124bof the UVLO circuit124ofFIG.3B. The voltage from VAof the converter circuit304is applied to a first resistive divider subcircuit comprising series-connected resistors R1A, R2A, the central node of which is coupled by a diode D1to a first input of a comparator322. The voltage from VBof the converter circuit304is applied to a second resistive divider subcircuit comprising series-connected resistors R1B, R2B, the central node of which is coupled by a diode D2to the first input of the comparator322. A second input of the comparator322is coupled to a reference voltage source324. The resistive divider subcircuits scale the input voltages VA, VBto match the voltage input limits of the comparator322, and the diodes D1, D2isolate the voltage of either input to the other input. The scaled voltage applied to the first input of the comparator322is the maximum of VAand VB, and thus the circuit self-selects the greater voltage to compare to the reference voltage source324and outputs a CTRL signal. Similar toFIG.4B, the diodes D1, D2at the first input of the comparator322inFIG.5Bmay be replaced with switches that are positively controlled by selection signals, such as from the controller108. As should be clear, other variants of the subcircuit selection circuitry shown inFIGS.5A and5Bmay be used for the UVLO circuit124ofFIG.3B. Similarly, subcircuit selection circuitry may be used for the voltage regulator122to effectively determine which applied voltage VA, VBwill power the voltage regulator122. A significant benefit of the circuit architectures shown inFIGS.3A and3Bis that the analog multiplexor and/or subcircuit selection circuitry can be implemented within the auxiliary circuit302, thereby eliminating the need for added external circuitry of a voltage booster circuit or a pre-charge circuit. Selective Voltage Inputs to Gate Driver Circuits The embodiments ofFIGS.3A and3Bmodify the auxiliary circuit302to select the greater of the voltages VA, VBfrom the converter circuit304as a power source for the auxiliary circuit302during startup. In cases in which a normally step-up power converter is operated in the reverse direction (i.e., in a step-down mode), the pre-charge circuit204(seeFIG.2A) may be eliminated in embodiments of the present invention. It is useful to better understand the need for the pre-charge circuit204, particularly for power converters that include switched-capacitor networks.FIG.6Ais a circuit diagram of a prior-art converter circuit600comprising a single-phase symmetric cascade multiplier having a step-up ratio of 1:5 and which may be used as an instance of the converter circuit102ofFIG.1A. The converter circuit600is configured to receive an input voltage (e.g., 5V) at node Vx and transform the input voltage into a higher output voltage at node VC5(e.g., 25V). The illustrated converter circuit600would be controlled by the controller108ofFIG.1Ain known fashion. Referring to both converter circuit600andFIG.1A, node Vx corresponds to input terminal T1of converter circuit102, node VC5corresponds to the output terminal T2of converter circuit102, and node VSScorresponds to both terminals T1′ and T2′ of converter circuit102. A cascade multiplier is a switched-capacitor network that can provide a high conversion gain. As used in this disclosure, conversion gain represents (1) a voltage gain if the switched-capacitor network produces an output voltage that is larger than the input voltage (VOUT>VIN), or (2) a current gain if the switched-capacitor network produces an output voltage that is smaller than the input voltage (VIN>VouT). Energy is transferred from the input to the output by cycling the cascade multiplier through different topological states. Charge is transferred from the input voltage to the output voltage via a charge transfer path. The number and configuration of the capacitors in each topological state sets the conversion gain. In the illustrated example, the converter circuit600includes five series-connected MOSFET switches M1-M5. Each MOSFET switch M1-M5may comprise a stack of series-connected MOSFETs having common gate connections and configured to function as a single switch. For convenience in discussing switching sequences, switches M1, M3, and M5will sometimes be referred to collectively as the “odd switches” and switches M2and M4will sometimes be referred to collectively as the “even switches.” The converter circuit600also includes first and second “low-side” MOSFET phase switches M7, M8and first and second “high-side” MOSFET phase switches M6, M9. The low-side phase switches M7, M8can connect first and second phase-nodes P1, P2to a potential VSS(usually circuit ground). The high-side phase-switches M6, M9can connect the first and second phase-nodes P1, P2to Vx. For convenience in discussing switching sequences, the high-side phase-switch M6and the low-side phase-switch M8will sometimes be referred to collectively as the “even phase-switches” and the low-side phase-switch M7and the high-side phase-switch M9will sometimes be referred collectively to as the “odd phase-switches.” A first pump capacitor C1connects a first stack-node VC1between switches M1and M2to phase-node P1. Similarly, a third pump capacitor C3connects a third stack-node VC3between switches M3and M4to phase-node P1. A second pump capacitor C2connects a second stack-node VC2between switches M2and M3to phase-node P2. Similarly, a fourth pump capacitor C4connects a fourth stack-node VC4between switches M4and M5to phase-node P2. A fifth stack-node, VC5, connects to a terminal of the converter circuit600. The illustrated converter circuit600has four stages. The first stage includes switch M1, first stack-node VC1, and first pump capacitor C1; the second stage includes switch M2, second stack-node VC2, and second pump capacitor C2; the third stage includes switch M3, third stack-node VC3, and third pump capacitor C3; and the fourth stage includes switch M4, fourth stack-node VC4, and fourth pump capacitor C4. A fifth series switch M5connects the fourth stage to the fifth stack-node, VC5. A clock source in the controller108generates non-overlapping clock waveforms φ1and φ2that are coupled to and control the ON/OFF state of the various switches M1-M9. The controller108outputs a set of control-signals112to the converter circuit600which cause the series switches M1-M5, the low-side phase-switches M7, M8, and the high-side phase-switches M6, M9to change states according to a specific sequence. As a result, the converter circuit600repeatedly transitions between first and second operating states at a selected frequency. For example, during a first operating state defined by the φ1clock waveform having a logic “1” state and the φ2clock waveform having a logic “0” state, the controller108(1) closes the odd switches M1, M3, M5, the low-side phase switch M7, and the high-side phase switch M9, and (2) opens the even switches M2, M4, the high-side phase switch M6, and the low-side phase switch M8. During a second operating state defined by the φ2clock waveform having a logic “1” state and the φ1clock waveform having a logic “0” state, the controller108(1) opens the odd switches M1, M3, M5, the low-side phase switch M7, and the high-side phase switch M9, and (2) closes the even switches M2, M4, the high-side phase switch M6, and the low-side phase switch M8. The controller108controls and sequences transitions of all the switches M1-M9in such a way as to incorporate any necessary dead-time needed when transitioning between the first and second operating states. As a consequence of alternating between the first operating state and the second operating state, charge is multiplied and conveyed from Vx to VC5in known fashion. As is known in the art, switching signals to the MOSFET switches M1-M9are applied through respective gate driver circuits G1-G9so as to provide suitable voltage levels for turning each MOSFET switch OFF (blocking) or ON (conducting) in timely fashion.FIG.6Bis a block diagram of a prior art gate driver circuit610. The gate driver circuit610includes a level shifter612that is coupled to input source VDDIand sink VSSIpotentials, and to output source VDDOand sink VSSOpotentials. The level shifter612translates an input switching signal VSWIfrom one voltage domain to another voltage domain. A level-shifted output voltage VLSOof the level shifter612is coupled to a gate-drive614which provides a low-impedance version of VLSOat VSWOto drive the gate terminal of an associated MOSFET switch Mx in a timely fashion. Referring back to the converter circuit600ofFIG.6A, the gate driver circuits G1-G9are shown in simplified block form coupled to respective switches M1-M9and connected to output source and sink potentials. Each gate driver circuit G1-G9has a corresponding input switching signal EN1-EN9(one of clock waveforms φ1or φ2) which controls the switching state of the respective switch M1-M9. All of the gate driver circuits G1-G9are coupled to common input source VDDIand sink VSSIpotentials (not shown for clarity), where the VDDvoltage generated from the auxiliary circuit110is coupled to the common input source VDDI. For both power and area efficiency, gate driver circuits G1-G6and G9are coupled to nodes VC1-VC5of the converter circuit600itself as output source VDDOand sink VSSOpotentials. In the particular embodiment of FIG.6A, MOSFET switches M1-M4, M6-M9are N-type transistors, while MOSFET switch M5is a P-type transistor. This is also reflected in the output source VDDOand sink VSSOpotentials shown for each gate driver circuit G1-G9. The gate driver circuits G7-G8are coupled to the VDDvoltage generated from the auxiliary circuit110for both their input source VDDIpotential and output source VDDOpotential. In a forward step-up operational mode, nodes VC1-VC5are initially (i.e., at startup) pumped above the voltage applied at Vx due to inherent body-diode paths in parallel with each of the switches M1-M5, and, eventually, sufficient output source VDDOand sink VSSOpotentials are reached for proper operation of the gate driver circuits G1-G9whereby the switches M1-M5can take over. However, in a reverse step-down operational mode, the input voltage is applied at node VC5and the output voltage is to be generated at node Vx. Nodes Vx, VC1-VC4may start out at or close to VSSground potential; hence, sufficient output source VDDOand sink VSSOpotentials are not yet available for proper operation of the gate driver circuits G1-G9, further worsening the circular startup problem. Accordingly, a pre-charge circuit204is typically used to provide initial and adequate voltages from the input voltage at node VC5to the nodes Vx, VC1-VC4within the converter circuit600to provide sufficient output source VDDOand sink VSSOpotentials for initial operation of the gate drivers G1-G9. After startup, the voltages at nodes Vx, VC1-VC4are then adequately supplied by the nature of the converter circuit's600steady-state operation, and accordingly the pre-charge circuit204may be disabled or disconnected. FIG.7Ais a block diagram of an improved gate driver circuit700that may be used in conjunction with the converter circuit600ofFIG.6A. The improved gate driver circuit700is similar in many regards to the gate driver circuit610ofFIG.6B. Thus, the gate driver circuit700includes a level shifter612that is coupled to input source VDDIand sink VSSIpotentials, and to output source VDDOand sink VSSOpotentials. The level shifter612translates an input switching signal VSWIfrom one voltage domain to another voltage domain. A level-shifted output voltage VLSOof the level shifter612is coupled to a gate-drive614which provides a low-impedance output at VSWOto drive the gate terminal of an associated MOSFET switch Mx in a timely fashion. In addition, the gate driver circuit700includes a selector702that functions as an analog multiplexor, allowing selection of one of two input voltages, VDDOAor VDDOB, for the output source VDDOpotential used within the gate driver circuit700. The selector702may be controlled, for example, by the controller108(e.g., based on information regarding whether the converter circuit304is to operate in a forward or reverse direction), or utilize a self-selecting embodiment similar to that ofFIG.4A. Thus, the gate driver circuit700is a dual-voltage device with respect to the output source VDDOpotential. Once an input voltage, VDDOA or VDDOB, is coupled through by the selector702, the operation of the gate driver circuit700is essentially the same as a conventional gate driver. FIG.7Bis a circuit diagram of a modified converter circuit720comprising a single-phase symmetric cascade multiplier having a step-up ratio of 1:5 and which may be used as an instance of the converter circuit102ofFIG.1A. In the illustrated embodiment, gate driver circuits G1-G3, G6, and G9are instances of the gate driver circuit700ofFIG.7A. In the illustrated embodiment, a first input (e.g., VDDOA) to each of gate drivers G1-G3, G6, and G9is the same VCXnode as in the example shown inFIG.6A(i.e., VC1-VC4). A second input (e.g., VDDOB) to each of gate drivers G1-G3, G6, and G9is the reverse-direction input voltage applied (in this 4-stage example) at VC5(a “reverse mode input node”). Conventional single-source gate driver circuits G4and G5may be instances of the gate driver circuit610ofFIG.6Band remain coupled to VC5for their output source VDDOpotential as in the example shown inFIG.6A. Conventional single-source gate driver circuits G7and G8may be instances of the gate driver circuit610ofFIG.6Band also remain coupled to the VDDvoltage generated from the auxiliary circuit110for both their input source VDDIpotential and output source VDDOpotential. In a forward step-up operational mode, the selector702would select the respective stack-nodes VC1-VC4as the output source VPPOpotential for gate driver circuits G1-G3, G6, and G9. However, in a reverse step-down operational mode during the startup phase of the converter circuit720, the selector702would select the reverse mode input node (VC5, in this example) as the output source VDDOpotential for the gate driver circuits G1-G3, G6, and G9until sufficient voltage levels develop at the stack-nodes VC1-VC4to support the gate driver circuits. Accordingly, regardless of forward or reverse operational mode, a sufficient output source VDDOpotential is available for each of the gate driver circuits G1-G9. The improved gate driver circuit700thus effectively makes the illustrated converter circuit720self-biasing at startup. A significant benefit of the improved gate driver circuit700and the circuit architecture shown inFIGS.7A and7Bis that the circuitry can be implemented without the added external circuitry of a pre-charge circuit204. While a single-phase symmetric cascade multiplier has been used in the converter circuits600and720to illustrate the problem solved by the improved gate driver circuit700, it should be noted that usage of the improved gate driver circuit700is not limited to switched-capacitor networks or charge pumps. This aspect of the present invention may also be applied to inductor-based regulators using transistor switches having one or more series-stacked switch stages. Combination Embodiments Embodiments of the example shown inFIG.3Aeliminate the need for an external voltage booster circuit202. Embodiments of the examples shown inFIGS.7A and7Beliminate the need for an external pre-charge circuit204. As should be clear, embodiments of the invention may include both inventive concepts, thereby obviating the need for an external voltage booster circuit202and an external pre-charge circuit204. A combined embodiment thus allows a power converter to be started and operated in a reversed unidirectional manner or in a bidirectional manner with zero external components. Methods Another aspect of the invention includes methods for powering an auxiliary circuit, selecting a subcircuit of an auxiliary circuit, and powering a dual voltage input gate driver. For example,FIG.8is a process flow chart800showing a method of powering an auxiliary circuit of a power converter. The process includes providing a power converter including a converter circuit having a first terminal configured to be selectably coupled to a first voltage and a second terminal configured to be selectably coupled to a second voltage (Block802); coupling an auxiliary circuit to the first terminal and the second terminal of the converter circuit (Block804); and selectively coupling the greater of the first voltage or the second voltage to provide power to the auxiliary circuit (Block806). In an alternative method, selection of the first voltage or the second voltage may be based on a knowledge of the existing forward or reverse configuration of the converter circuit. As another example,FIG.9is a process flow chart900showing a method of selecting among subcircuits of a plurality of subcircuits of an auxiliary circuit of a power converter. The power converter includes providing a power converter including a converter circuit having a first terminal configured to be selectably coupled to a first voltage and a second terminal configured to be selectably coupled to a second voltage (Block902); coupling at least a first subcircuit of an auxiliary circuit to the first terminal of the converter circuit (Block904); coupling at least a second subcircuit of the auxiliary circuit to the second terminal of the converter circuit (Block906); and selectively coupling the at least one first subcircuit or the at least one second subcircuit to the greater of the first voltage or the second voltage to generate an output for the auxiliary circuit (Block908). As still another example,FIG.10is a process flow chart1000showing a method of providing power for a gate driver circuit of a corresponding transistor switch of a converter circuit of a power converter. The method includes selectively coupling a level shifter and gate-drive to one of a first output source potential or a second output source potential, wherein the first output source potential comes from a first voltage node of the converter circuit of the power converter, and the second output source potential comes from a second, different voltage node of the converter circuit of the power converter (Block1002). The methods may be used together. For example, the method ofFIG.8may be used for a first part of an auxiliary circuit, and the method ofFIG.9may be used for a second part of the auxiliary circuit. As another example, the method ofFIG.10may be used in conjunction with the method ofFIG.8and/or the method ofFIG.9. Fabrication Technologies & Options The term “MOSFET”, as used in this disclosure, includes any field effect transistor (FET) having an insulated gate whose voltage determines the conductivity of the transistor, and encompasses insulated gates having a metal or metal-like, insulator, and/or semiconductor structure. The terms “metal” or “metal-like” include at least one electrically conductive material (such as aluminum, copper, or other metal, or highly doped polysilicon, graphene, or other electrical conductor), “insulator” includes at least one insulating material (such as silicon oxide or other dielectric material), and “semiconductor” includes at least one semiconductor material. As used in this disclosure, the term “radio frequency” (RF) refers to a rate of oscillation in the range of about 3 kHz to about 300 GHz. This term also includes the frequencies used in wireless communication systems. An RF frequency may be the frequency of an electromagnetic wave or of an alternating voltage or current in a circuit. Various embodiments of the invention can be implemented to meet a wide variety of specifications. Unless otherwise noted above, selection of suitable component values is a matter of design choice. Various embodiments of the invention may be implemented in any suitable integrated circuit (IC) technology (including but not limited to MOSFET structures), or in hybrid or discrete circuit forms. Integrated circuit embodiments may be fabricated using any suitable substrates and processes, including but not limited to standard bulk silicon, silicon-on-insulator (SOI), and silicon-on-sapphire (SOS). Unless otherwise noted above, embodiments of the invention may be implemented in other transistor technologies such as bipolar, LDMOS, BCD, GaAs HBT, GaN HEMT, GaAs pHEMT and MESFET technologies. However, embodiments of the invention may be particularly useful when fabricated using an SOI or SOS based process, or when fabricated with processes having similar characteristics. Fabrication in CMOS using SOI or SOS processes enables circuits with low power consumption, the ability to withstand high power signals during operation due to FET stacking, good linearity, and high frequency operation (i.e., radio frequencies up to and exceeding 50 GHz). Monolithic IC implementation is particularly useful since parasitic capacitances generally can be kept low (or at a minimum, kept uniform across all units, permitting them to be compensated) by careful design. Voltage levels may be adjusted, and/or voltage and/or logic signal polarities reversed, depending on a particular specification and/or implementing technology (e.g., NMOS, PMOS, or CMOS, and enhancement mode or depletion mode transistor devices). Component voltage, current, and power handling capabilities may be adapted as needed, for example, by adjusting device sizes, serially “stacking” components (particularly FETs) to withstand greater voltages, and/or using multiple components in parallel to handle greater currents. Additional circuit components may be added to enhance the capabilities of the disclosed circuits and/or to provide additional functionality without significantly altering the functionality of the disclosed circuits. Circuits and devices in accordance with the present invention may be used alone or in combination with other components, circuits, and devices. Embodiments of the present invention may be fabricated as integrated circuits (ICs), which may be encased in IC packages and/or or modules for ease of handling, manufacture, and/or improved performance. In particular, IC embodiments of this invention are often used in modules in which one or more of such ICs are combined with other circuit blocks (e.g., filters, passive components, and possibly additional ICs) into one package. The ICs and/or modules are then typically combined with other components, often on a printed circuit board, to form an end product such as a cellular telephone, laptop computer, or electronic tablet, or to form a higher level module which may be used in a wide variety of products, such as vehicles, test equipment, medical devices, etc. Through various configurations of modules and assemblies, such ICs typically enable a mode of communication, often wireless communication. CONCLUSION A number of embodiments of the invention have been described. It is to be understood that various modifications may be made without departing from the spirit and scope of the invention. For example, some of the steps described above may be order independent, and thus can be performed in an order different from that described. Further, some of the steps described above may be optional. Various activities described with respect to the methods identified above can be executed in repetitive, serial, or parallel fashion. It is to be understood that the foregoing description is intended to illustrate and not to limit the scope of the invention, which is defined by the scope of the following claims, and that other embodiments are within the scope of the claims. In particular, the scope of the invention includes any and all feasible combinations of one or more of the processes, machines, manufactures, or compositions of matter set forth in the claims below. (Note that the parenthetical labels for claim elements are for ease of referring to such elements, and do not in themselves indicate a particular required ordering or enumeration of elements; further, such labels may be reused in dependent claims as references to additional elements without being regarded as starting a conflicting labeling sequence). | 35,047 |
11863056 | DETAILED DESCRIPTION OF SOME EMBODIMENTS FIG.1is a block diagram of a switching converter10according to example embodiments. As shown inFIG.1, the switching converter10may include an inductor L1, an output capacitor C1, a first power switch PS1, a second power switch PS2, a switch driver12, a voltage feedback circuit14, and/or a load current meter100. In some example embodiments, components of the switching converter10may be included in one semiconductor package. In some example embodiments, the switching converter10may include a printed circuit board (PCB), and at least two of components of the switching converter10may be mounted as separate semiconductor packages, respectively, on the PCB. Referring toFIG.1, the switching converter10may receive an input voltage VIN and generate an output voltage VOUT. The output voltage VOUTmay be used as a supply voltage for other electronic components (e.g., a load), and the switching converter10may provide a load current ILOADto the load. The switching converter10may refer to an arbitrary electronic circuit configured to switch an apparatus on or off and generate the output voltage VOUTand be also referred to as a switching regulator. For example, the first power switch PS1and the second power switch PS2of the switching converter10may be turned on or off based on a first driving signal DRV1and a second driving signal DRV2, which are provided by the switch driver12. In some example embodiments, the first power switch PS1may include a p-type field effect transistor (PFET) as a power transistor and be turned on in response to the first driving signal DRV1that is at a low level. In some example embodiments, the second power switch PS2may include an n-type FET (NFET) as a power transistor and be turned on in response to the second driving signal DRV2that is at a high level. In some example embodiments, the second power switch PS2may be replaced by a diode having an anode to which a ground potential is applied and a cathode connected to the inductor L1. Herein, the first driving signal DRV1will be assumed to be an active-low signal, and the second driving signal DRV2will be assumed to be an active-high signal. As used herein, an on state of a switch (or a power switch) may refer to a state in which both ends of the switch are electrically connected to each other, and an off state of the switch may refer to a state in which both ends of the switch are electrically disconnected from each other. In addition, at least two components electrically connected to each other via a switch, which is in the on state, and/or a conducting wire may be referred to as being simply connected to each other, while at least two components electrically connected to each other at all times through a conducting wire may be referred to as being coupled to each other. In some example embodiments, the input voltage VIN and the output voltage VOUTmay be positive direct-current (DC) voltages, and the switching converter10may be a DC-to-DC converter. For example, the switching converter10may be a buck converter and generate an output voltage VOUTlower than an input voltage VIN. The switching converter10may be also referred to as a step-down converter. Also, the switching converter10may be a boost converter and generate an output voltage VOUThigher than an input voltage VIN. The switching converter10may be also referred to as a step-up converter. In addition, the switching converter10may be a buck-boost (or step-up/down) converter and generate an output voltage VOUTlower or higher than an input voltage VIN. Hereinafter, although the switching converter10will mainly be described with reference to a buck converter, it will be understood that example embodiments may be applied not only to other types of DC-to-DC converters but also to alternating-current (AC)-to-DC converters configured to receive AC voltages. The voltage feedback circuit14may compare the output voltage VOUTwith a target voltage and provide a feedback signal FB indicating a comparison result to the switch driver12. For example, the voltage feedback circuit14may compare the output voltage VOUTor a voltage divided from the output voltage VOUTwith at least one reference voltage and generate the feedback signal FB indicating the comparison result. The switch driver12may detect a level of the output voltage VOUTbased on the feedback signal FB. In some example embodiments, the voltage feedback circuit14may generate an activated feedback signal FB, when the output voltage VOUTor the voltage divided from the output voltage VOUTis lower than the reference voltage, and the switch driver12may activate the first driving signal DRV1in response to the activated feedback signal FB. The load current meter100may measure the load current ILOAD, which is provided by the switching converter10to the load through an output terminal. As shown inFIG.1, the load current meter100may receive the first driving signal DRV1and the second driving signal DRV2from the switch driver12and generate an output signal OUT indicating a magnitude of the load current ILOAD. The load current ILOADmay include an inductor current ILpassing through the inductor L1and/or current generated by discharging the output capacitor C1. Because the output capacitor C1is charged by the inductor current IL, the inductor current ILaccumulated for a predetermined or alternatively, desired amount of time may be equal to the load current ILOADaccumulated for the same amount of time. Accordingly, the load current meter100may measure the load current ILOADbased on the first driving signal DRV1and the second driving signal DRV2, which include information about points in time at which the inductor current ILis generated. As used herein, the load current meter100may be referred to as an apparatus configured to measure the load current ILOAD. In some example embodiments, the output signal OUT generated by the load current meter100may be used to estimate power consumption of the load. For example, as shown inFIG.1, when the output signal OUT is output, a power estimator included in the load or other components that consume the load current ILOADmay detect the magnitude of the load current ILOAD, based on the output signal OUT. The power estimator may estimate power consumption of the load based on the detected size of the load current ILOAD. The estimated power consumption may be utilized for a plurality of useful functions of, for example, estimating the remaining battery level, reducing or preventing the overheating of the load, or detecting an abnormal event. As used herein, the output signal OUT may be assumed to be provided to the power estimator. In some example embodiments, the load current ILOADmay fluctuate within a very small range. For example, the load may be set to a power saving mode to reduce power consumption, and the load current ILOADmay have a significantly reduced magnitude in the power saving mode than in a normal mode. As described below with reference to the drawings, the output signal OUT may accurately indicate a load current ILOADhaving a very small magnitude, and thus, power consumption of the load may be accurately estimated. Examples of the load current meter100will be described below with reference toFIGS.2A,2B, and8. The switch driver12may generate the first driving signal DRV1and the second driving signal DRV2such that the output voltage VOUTis maintained at approximately a target voltage. As shown inFIG.1, the switch driver12may receive the feedback signal FB from the voltage feedback circuit14, receive a peak signal PK from a peak current detector16, and generate the first driving signal DRV1and the second driving signal DRV2based on the peak signal PK and the feedback signal FB. In some example embodiments, the switch driver12may include a plurality of logic gates, perform a logic operation on the peak signal PK and the feedback signal FB, and generate the first driving signal DRV1and the second driving signal DRV2. For example, the switch driver12may generate a deactivated first driving signal DRV1and an activated second driving signal DRV2in response to an activated peak signal PK, and thus, the inductor current ILmay be reduced. In addition, the switch driver12may generate an activated first driving signal DRV1and a deactivated second driving signal DRV2in response to the activated feedback signal FB, and thus, the inductor current ILmay increase. The peak current detector16may detect a peak of the inductor current ILpassing through the inductor L and provide a peak signal PK to the switch driver12. For example, the peak current detector16may detect the inductor current IL, and generate an activated peak signal PK when a magnitude of the detected inductor current ILcorresponds to a magnitude defined based on a first reference voltage VREF1. The switch driver12may deactivate the first driving signal DRV1in response to the activated peak signal PK, and thus, the inductor current ILmay be reduced. As a result, the peak of the inductor current ILmay be limited to the magnitude defined based on the first reference voltage VREF1, and requirements of the switching converter10, for example, electromagnetic interference (EMI) requirements, may be satisfied. In some example embodiments, for example, the first reference voltage VREF1may be varied by the switch driver12, and thus, the peak of the inductor current ILmay be adjusted. In some example embodiments, unlike that shown inFIG.1, the peak current detector16may detect current passing through the first power switch PS1or a peak of the current passing through the first power switch PS1instead of the inductor current IL. FIGS.2A and2Bare block diagrams of a load current meter100according to example embodiments. The load current meter100ofFIG.1Amay receive a first driving signal DRV1and a second driving signal DRV2and generate a count value CNT indicating a magnitude of a load current ILOAD. The load current meter100may include a pulse generation circuit110, a reference current generation circuit120, a clock generation circuit130, and a counter140. The pulse generation circuit110may generate a control pulse (or a pulse signal) PLS based on the first driving signal DRV1and the second driving signal DRV2. In some example embodiments, the pulse generation circuit110may generate the control pulse PLS having a width (or activation width), which is based on the first driving signal DRV1and the second driving signal DRV2. As described above with reference toFIG.1, the first driving signal DRV1and the second driving signal DRV2may include information about points in time at which an inductor current ILis generated. Accordingly, the control pulse PLS may include information about the inductor current ILaccumulated for a predetermined or alternatively, desired amount of time. As used herein, the control pulse PLS may be assumed to be at a high level when the control pulse PLS is activated as an active-high signal, and a width of the control pulse PLS may refer to a time period during which the control pulse PLS is maintained at a high level. Examples of the pulse generation circuit110will be described below with reference toFIG.3. The reference current generation circuit120may supply a reference current RI to the clock generation circuit130based on an input voltage (supply voltage) and the control pulse PLS). The reference current generation circuit120may include a current sink and a current mirror to generate the reference current RI from the input voltage and further include a capacitor or a switch to control the reference current RI. Examples of the reference current generation circuit120will be described below with reference toFIG.7. The clock generation circuit130may generate a clock signal CLK based on the reference current RI and the control pulse PLS. The clock generation circuit130may charge a first capacitor in a section in which the control pulse PLS is activated, and float the first capacitor in a section in which the control pulse PLS is deactivated. The clock generation circuit130may charge a second capacitor in which the control pulse PLS is deactivated, and generate the clock signal CLK while discharging the second capacitor when a voltage of the second capacitor becomes equal to a voltage of the first capacitor. Examples of the clock generation circuit130will be described below with reference toFIG.5. The counter140may provide a counted number CNT of cycles of the clock signal CLK generated by the clock generation circuit130. The load current meter100may generate an output signal indicating a magnitude of a load current based on a switching period of a switching converter and the counted number CNT of cycles of the clock signal CLK. For example, the counter140may count clock signals CLK from a point in time at which the control pulse PLS is activated, and be reset by storing the counted number CNT of cycles at a point in time at which the next control pulse PLS starts to be activated. The magnitude of the load current ILOADmay be calculated based on the counted number CNT of cycles per switching period. The load current ILOADmay be calculated by using the average of counted numbers CNT of cycles. Similar to that ofFIG.2A, the load current meter100ofFIG.2Bmay include a pulse generation circuit110, a reference current generation circuit120, a clock generation circuit130, and/or a counter140and further include a logic circuit150. The logic circuit150may include a plurality of logic gates. The logic circuit150may receive the control pulse PLS and generate a control signal Ctrl for controlling the reference current generation circuit120and the clock generation circuit130. The logic circuit150may directly receive the first driving signal DRV1and the second driving signal DRV2instead of the control pulse PLS and generate the control signal Ctrl. The logic circuit150may control other components (e.g., the counter140) included in the load current meter100. FIG.3is a block diagram of a pulse generation circuit110according to example embodiments. Referring toFIG.3, the pulse generation circuit110may generate a control pulse PLS based on a first driving signal DRV1and a second driving signal DRV2. For example, a first transistor M1and a second transistor M2of the switching converter10may be turned on or turned off based on the first driving signal DRV1(or a push signal) and the second driving signal DRV2(or a pull signal), which are provided from a switch driver12. The first transistor M1may include a PFET as a power transistor and be turned on in response to a first driving signal DRV1that is at a low level. Also, the second transistor M2may include an NFET as a power transistor and be turned on in response to a second driving signal DRV2that is at a high level. As used herein, the first driving signal DRV1may be assumed to be an active-low signal, and the second driving signal DRV2may be assumed to be an active-high signal. The pulse generation circuit110may include logic operation circuits, such as an inverting circuit112and/or an OR logic gate114. For example, the pulse generation circuit110may receive the first driving signal DRV1, invert the first driving signal DRV1by using the inverting circuit112, perform an OR operation on the inverted first driving signal DRV1and the second driving signal DRV2by using the OR logic gate114, and generate the control pulse PLS. FIG.4is a timing diagram of an example of an operation of a switching converter according to example embodiments. Referring toFIGS.3and4, the switch driver12may receive an input voltage VDD and generate a first driving signal DRV1and a second driving signal DRV2to provide an output voltage VOUT. At each of a point in time t11and a point in time t14, the switch driver12may generate the first driving signal DRV1that is activated. Thus, a first transistor M1may be turned on, and the input voltage VDD may be applied by the first transistor M1to a switch node. An inductor current ILflowing through an inductor L1may start to increase. At each of points in time t12and t15, the inductor current ILmay reach an upper limit, that is, a peak current IPEAK. The switch driver12ofFIG.3may generate a deactivated first driving signal DRV1and an activated second driving signal DRV2. Accordingly, the inductor current ILmay be gradually reduced due to a turned-off first transistor M1and a turned-on second transistor M2. At each of points in time t13and t16, the inductor current ILmay be approximately 0 (zero), and the switch driver12may generate a deactivated second driving signal DRV2. Thus, a second switch SW2may be turned off, and the inductor current ILmay be maintained at approximately 0. A width of each of sections between the points in time t11to t16may vary according a state of a load, that is, a magnitude of a load current required by the load. A pulse generation circuit110may generate a control pulse PLS based on the first driving signal DRV1and the second driving signal DRV2. For example, the pulse generation circuit110may receive the first driving signal DRV1, invert the first driving signal DRV1by using an inverter operation circuit112, perform an OR operation on the inverted first driving signal DRV1and the second driving signal DRV2by using an OR operation circuit114, and generate the control pulse PLS. The pulse generation circuit110may generate an activated control pulse PLS from the point in time t11to a point in time t13, and generate a deactivated control pulse PLS from the point in time t13to the point in time t14. The pulse generation circuit110may generate an activated control pulse PLS from the point in time t14to the point in time t16, and generate a deactivated control pulse PLS from the point in time t16. FIG.5is a circuit diagram of a clock generation circuit according to example embodiments. Referring toFIG.5, the clock generation circuit130may generate a clock signal CLK based on a control pulse PLS and a reference current I1. The clock generation circuit130may include a current source136, a plurality of switches (e.g., second, third and fourth switches S2, S3, and S4), an inverter operation circuit132, a first capacitor C1, a second capacitor C2, and/or a comparator134. The comparator134may generate a clock signal CLK based on a first voltage V1and a second voltage V2. The comparator134may generate the clock signal CLK, which is activated, when the second voltage V2is higher than or equal to the first voltage V1. The clock generation circuit130may turn on or turn off the second switch S2by using the generated clock signal CLK. The current source136may generate a reference current I1from a positive supply voltage VDD. The current source136may include a current sink and a current mirror to generate the reference current I1. The third switch S3may be connected between the current source136and the first capacitor C1, and the fourth switch S4may be connected between the current source136and the second capacitor C2. Also, a fifth switch S5may be connected in parallel to the first capacitor C1, and the second switch S2may be connected in parallel to the second capacitor C2. The clock generation circuit130may turn on or turn off the third switch S3and the fourth switch S4by using the control pulse PLS. For example, the clock generation circuit130may turn on the third switch S3and turn off the fourth switch S4in a section in which the control pulse PLS is activated. The clock generation circuit130may turn off the third switch S3and turn on the fourth switch S4in a section in which the control pulse PLS is deactivated. The first capacitor C1may be connected between a second node ND2and a ground node, while the second capacitor C2may be connected between a third node ND3and the ground node. To obtain a different charging rate due to the reference current I1, capacitances of the first capacitor C1and the second capacitor C2may be variously designed. The comparator134may be connected to the third switch S3and the first capacitor C1at the second node ND2and be connected to the fourth switch S4and the second capacitor C2at the third node ND3. The comparator134may compare a first voltage V1of the second node ND2with a second voltage V2of the third node ND3and generate a clock signal CLK indicating a comparison result. As used herein, a voltage (e.g., the first voltage V1) of the second node ND2connected to the first capacitor C1may be referred to as a voltage of the first capacitor C1, and a voltage (e.g., the second voltage V2) of the third node ND3connected to the second capacitor C2may be referred to as a voltage of the second capacitor C2. When the third switch S3is turned on, the first capacitor C1may be charged by the reference current. When the third switch S3is turned off, the first voltage V1of the first capacitor C1charged by the reference current may be maintained. For example, before the third switch S3is turned on and the first capacitor C1is charged by the reference current, the clock generation circuit130may temporarily turn on the fifth switch S5to put the first capacitor C1into a discharged state. When the fourth switch S4is turned on, the second capacitor C2may be charged by the reference current. When the second voltage V2with which the second capacitor C2is charged by the reference current becomes equal to the first voltage V1, the comparator134may output an activated clock signal CLK. When the second switch S2is turned on in response to the activated clock signal CLK, the second capacitor C2may be discharged, and the comparator134may output a deactivated clock signal CLK. FIGS.6A and6Bare timing diagrams of examples of an operation of a clock generation circuit according to example embodiments. FIGS.6A and6Bshow signals of the clock generation circuit130ofFIG.5over time. It is assumed that a control pulse PLS is generated by the pulse generation circuit110ofFIG.3. Referring toFIGS.5and6A, in a section in which the control pulse PLS is activated from a point in time t21to a point in time t23, a first capacitor C1may be charged by a reference current. In a section in which the control pulse PLS is deactivated from the point in time t23to a point in time t28, the first capacitor C1may be floated and remain charged with a first voltage V1. A second capacitor C2may be charged by the reference current from the point in time t23to a point in time t24. When the first voltage V1of the first capacitor C1becomes equal to a second voltage V2of the second capacitor C2at the point in time t24, the comparator134may output an activated clock signal CLK. When the second switch S2is turned on in response to the activated clock signal CLK, the second capacitor C2may be discharged, and the second voltage V2of the second capacitor C2may be charged again by the reference current from the point in time t24to a point in time t25. Until the point in time t28at which the control pulse PLS is deactivated, the second capacitor C2may generate the clock signal CLK by repeating charging and discharging operations. A load current may be calculated based on the number of clock signals CLK generated during one period of a switching converter or one period of the control pulse PLS. For example, charges QCstored by an output terminal during one period of the switching converter or one period (TON+TOFF=T) of the control plug PLS may be as shown in Equation 1 below. QC=½*Ipeak*TON, [Equation 1] wherein TONmay be a section in which the control pulse PLS is activated. A peak current Ipeakmay be a designed value and detected by a peak current detector. Charges QDconsumed by the load current during one period of the switching converter or one period (TON+TOFF) of the control pulse PLS may be as shown in Equation 2 below. QD=IL*T=IL*(n+1)*TON, [Equation 2] wherein T may denote one period of the switching converter or one period (TON+TOFF) of the control pulse PLS, and n may denote the counted number of clock signals generated during one period. QC=QD, and thus, a load current (IL) may be 12(n+1)*Ipeak. To generate a load current having a very small magnitude, one period of the switching converter may be relatively increased, and the section in which the control pulse PLS is deactivated from the point in time t23to the point in time t28may be increased. Referring toFIG.6B, in a section (TOFF) in which the first capacitor C1is floated and remains charged with the first voltage V1, a leakage current may occur in the first capacitor C1, and thus, the first voltage V1may be gradually reduced. Referring toFIG.6B, the first capacitor C1may be charged with the first voltage V1in the same manner as inFIG.6Auntil a point in time t31. In a section in which the control pulse PLS is deactivated from the point in time t31to a point in time t37, the first capacitor C1may be floated, and a voltage level of the first voltage V1may be lowered from an initial charged state over time due to the leakage current. The second voltage V2of the second capacitor C2may be charged by the reference current from the point in time t31to a point in time t32. When the first voltage V1of the first capacitor C1becomes equal to the second voltage V2of the second capacitor C2at the point in time t32, the comparator134may output the activated clock signal CLK. However, unlike the example embodiments ofFIG.6A, because the first voltage V1of the first capacitor C1has a lower voltage level due to the leakage current, a point in time at which the first voltage V1of the first capacitor C1becomes equal to the second voltage V2of the second capacitor C2may be advanced. For example, assuming that a leakage current does not occur in the first capacitor C1, the clock signal may be generated at a point in time t33. However, when the leakage current occurs in the first capacitor C1, the clock signal CLK may be generated at the point in time t32. When the second switch S2is turned on in response to the activated clock signal CLK, the second capacitor C2may be discharged, and the second capacitor C2may be charged again with the second voltage V2by the reference current from the point in time t32to a point in time t34. When the first voltage V1of the first capacitor C1becomes equal to the second voltage V2of the second capacitor C2at the point in time t34, the comparator134may output the activated clock signal CLK. Because the voltage level of the first voltage V1of the first capacitor C1is continuously reduced due to the leakage current, a period from the point in time t32to the point in time t34in which a second clock signal2is generated may be shorter than a period from the point in time t31to the point in time t32in which a first clock signal1is generated. Until the point in time t37at which the control pulse PLS is deactivated, the second capacitor C2may generate the clock signal CLK by repeating charging and discharging operations. In some example embodiments, a period of generation of the clock signal CLK may be gradually reduced due to the leakage current caused in the first capacitor C1. The number of clock signals CLK caused during one period of the switching converter or one period of the control pulse PLS may increase more than when the leakage current does not occur, and there may be an error in the magnitude of the load current calculated based on the number of clock signals CLK. A method of compensating for the error caused by the leakage current will be described in detail with reference toFIGS.7to9. FIG.7is a circuit diagram of a reference current generation circuit120according to example embodiments. Referring toFIG.7, the reference current generation circuit120may have a structure configured to generate a reference current I1from a reference voltage V_ref. For example, the reference current generation circuit120may include a current sink and a current mirror. The reference current generation circuit120may include a first switch S1and an input compensation capacitor C3to compensate for a leakage current generated by a clock generation circuit. The reference current generation circuit120may apply the reference voltage V_ref as a third voltage V3of an OP amplifier122and provide the reference current I1based on the applied voltage. The reference current generation circuit120may directly apply the reference voltage V_ref to the third voltage V3of the OP amplifier122by turning on the first switch S1, and charge the input compensation capacitor C3. The reference current generation circuit120may apply the third voltage V3to the OP amplifier122by using the charged input compensation capacitor C3by turning off the first switch S1. In some example embodiments, when a leakage current occurs in the input compensation capacitor C3, a voltage level of the third voltage V3may be reduced, and a reference current supplied through a current source124may also be reduced. FIG.8is a circuit diagram of a load current meter500according to example embodiments. Referring toFIG.8, the load current meter500may control a first switch S1, which is included in a reference current generation circuit, and a third switch S3and a fourth switch S4, which are included in a clock generation circuit, based on a control pulse generated by a pulse generation circuit. The load current meter500may control an input compensation capacitor C3to compensate for an error caused by a leakage current of a first capacitor C1. The load current meter500may turn off the first switch S1for a period corresponding to a period for which the third switch S3is turned off by deactivating the control pulse. In some example embodiments, a second voltage applied to the comparator504may slowly increase due to a leakage current of the input compensation capacitor C3as much as a first voltage V1applied to a comparator504is reduced due to the leakage current of the first capacitor C1. Thus, an error caused during the measuring of a load current may be reduced. For example, the first switch S1and the third switch S3may be controlled in the same state based on the same signal, and the first switch S1may be turned on or off faster than the third switch S3considering time at which a reference current I1is generated based on a third voltage V3. FIG.9is a timing diagram of an example of an operation of a switching converter according to example embodiments. Referring toFIGS.8and9, the load current meter500may generate a reference current I1based on a third voltage V3by using a current mirror. In a section in which a control pulse PLS is activated from a point in time t41to a point in time t42, both a first switch S1and a third switch S3may be turned on, and a first capacitor C1may be charged by the reference current I1Because the first switch S1remains turned on, the third voltage V3may be maintained at a constant voltage level based on a reference voltage V_ref. From the point in time t42to a point in time t44in a section in which the control pulse PLS is deactivated, both the first switch S1and the third switch S3may be turned off, and a fourth switch S4may be turned on. When the fourth switch S4is turned on, a second capacitor C2may be charged by the reference current I1. Because the first switch S1remains turned off, the third voltage V3may be supplied by an input compensation capacitor C3, and a voltage level of the third voltage V3may be gradually reduced due to a leakage current, and the reference current I1may also be gradually reduced. Accordingly, because a rate at which the second capacitor C2is charged is reduced as much as a first voltage V1of the first capacitor C1is reduced, an error may be reduced during the generation of a clock signal CLK, as compared to example embodiments in which a leakage current occurs only in the first capacitor C1. When the number of clock signals CLK generated during one period of the switching converter or one period of the control pulse PLS is measured, an error caused by causing a leakage current caused by the leakage current generated in the first capacitor C1may be compensated for by using the input compensation capacitor C3. For example, by reducing the third voltage V3at a rate corresponding to a reduction in the first voltage V1due to the leakage current, an error caused during the measuring of a load current may be reduced. FIG.10is a graph showing the switching of a measurement mode of a load current meter100, according to example embodiments. For example, the graph ofFIG.10shows a hysteresis loop, which occurs during the switching of the measurement mode of the load current meter100. As described above with reference to the drawings, the load current meter100may be set to a first mode when a load current ILOADis small, and may be set to a second mode when the load current ILOADis large. For example, the load current ILOADmay be reduced as a count CNT increases, and increase as the count CNT is reduced. To reduce or prevent errors and inaccuracy due to frequent switching of the measurement mode, the load current meter100may provide hysteresis to the switching of the measurement mode. For example, as shown inFIG.10, the load current meter100may be set to the second mode when the count CNT is lower than a first threshold value THR1, and may be set to the first mode when the count CNT is higher than a second threshold value THR2. Herein the second threshold value THR2may be higher than the first threshold value THR1(THR2>THR1). For example, when the load current is very small, a switching period of a switching converter configured to supply the load current ILOADmay become relatively long, and an error caused by a leakage current may further increase. Thus, the load current meter100may operate in the first mode including an operation for compensating for the leakage current. When the load current is larger than or equal to a preset current level, the switching period of the switching converter configured to supply the load current ILOADmay become relatively short. In some example embodiments, the load current meter100may operate in the second mode in which the effect of the leakage current is not considered. FIG.11is a flowchart of a method of measuring a load current, according to example embodiments. Referring toFIGS.1,2A,2B, and11, the pulse generation circuit110may generate a control pulse, based on a power switch driving signal of the switching converter10(S110). The switching converter10may adjust a period of the power switch driving signal to control a load current or an output voltage. The control pulse may be activated in a section in which a first driving signal or a second driving signal is activated. The reference current generation circuit120may generate a reference current based on the control pulse (S120). The generation of the reference current may include adjusting the reference current to compensate for a leakage current caused during the generation of a clock signal during a switching period. The generation of the reference current may further include generating the reference current based on a reference voltage, charging an input compensation capacitor with the reference voltage in a section in which the control pulse is activated, and generating the reference voltage based on charges stored in the input compensation capacitor after the section in which the control pulse is activated. The generation of the reference current may further include turning on a first switch configured such that one end of the first switch receives an input current and the other end thereof is connected to the input compensation capacitor in the section in which the control pulse is activated and turning off the first switch such that a node at which the input compensation capacitor is connected to the first switch is floated in a section in which the control pulse is deactivated. The clock generation circuit130may generate the clock signal based on the control pulse and the reference current (S130). The generation of the clock signal may further include charging a first capacitor by the reference current in the section in which the control pulse is activated, floating the first capacitor in the section in which the control pulse is deactivated, and charging a second capacitor by the reference current. The generation of the clock signal may further include generating an activated clock signal when a second voltage of the second capacitor becomes higher than a first voltage of the first capacitor and discharging the second capacitor in response to the activated clock signal. The counter140may count the number of cycles of the clock signal during the switching period of the switching converter10(S140). The counting of the number of cycles of the clock signal may further include generating an output signal indicating a magnitude of a load current based on the switching period of the switching converter10and the counted number of cycles of the clock signal. The load current meter100may convert information about the load current into digital information based on the switching period of the switching converter10or the counted number of cycles of the clock signal and provide the digital information to a user. FIG.12is a block diagram of a system600according to example embodiments. As shown inFIG.12, the system600may include a power management integrated circuit (PMIC)610and/or a load620. The system600may refer to a system configured to provide an arbitrary function by performing an operation due to power consumption of the load620. For example, the system600may be a computing system (e.g., a personal computer, a server, a mobile phone, and a wearable device), a vehicle (e.g., a car, a ship, and an electric kickboard), or a sub-system included in the above-described systems. The PMIC610may include a switching converter612, which is as described above with reference to the drawings, and provide a positive supply voltage VDD generated by the switching converter612to the load620. In addition, the PMIC610may provide, to the load620, a status signal STA including information about power consumption of the load620. For example, as described above with reference to the drawings, the switching converter612may generate an output signal corresponding to a magnitude of a load current provided to the load620, and the PMIC610may provide the output signal or a status signal STA including information about a magnitude of a load current detected based on the output signal, to the load620. The load620may receive the positive supply voltage VDD from the PMIC610and operate based on the positive supply voltage VDD. In addition, the load620may receive the status signal STA from the PMIC610, and estimate or detect power consumption of the load620, based on the status signal STA. As described above with reference to the drawings, the load620may accurately estimate or detect power consumption due to an accurately measured load current. The load620may control the PMIC610by using a control signal CTR. For example, the load620may transmit the control signal CTR indicating a magnitude of the positive supply voltage VDD and an entry into or release from a power saving mode to the PMIC610. The PMIC610may adjust the magnitude of the positive supply voltage VDD based on the control signal CTR and/or stop or restart the generation of the positive supply voltage VDD. FIG.13is a block diagram of a system700according to example embodiments. In some example embodiments, the system700may be an integrated circuit (IC) included in one semiconductor package, such as a System-on-Chip (SoC). In some example embodiments, the system700may include a PCB and semiconductor packages mounted on the PCB. As shown inFIG.13, the system700may include at least one processor710, an input/output (I/O) interface720, a modem730, a memory740, and/or a PMIC750. The at least one processor710, the I/O interface720, the modem730, and the memory740may respectively operate based on pieces of power provided by first to fourth positive supply voltages VDD1to VDD4supplied from the PMIC750. For example, the at least one processor710may execute a series of instructions or process signals based on the first positive supply voltage VDD1. The I/O interface720may process an input signal received from the outside of the system700, based on the second positive supply voltage VDD2, and generate an output signal to be provided to the outside of the system700. The modem730may process a received signal through a communication channel or generate a signal to be transmitted through the communication channel, based on the third positive supply voltage VDD3. The memory740may store data based on the fourth supply voltage VDD4, and include a volatile memory device, such as dynamic random access memory (DRAM) and static RAM (SRAM), and/or a non-volatile memory device, such as flash memory and resistive RAM (RRAM). The PMIC750may include a plurality of switching converters752, each of which may generate one of the first to fourth positive supply voltages VDD to VDD4from an input voltage VIN. As described above with reference to the drawings, each of the plurality of switching converters752may more accurately measure even a load current having a smaller magnitude, which is provided thereto. One or more of the elements disclosed above may include or be implemented in one or more processing circuitries such as hardware including logic circuits; a hardware/software combination such as a processor executing software; or a combination thereof. For example, the processing circuitries more specifically may include, but is not limited to, a central processing unit (CPU), an arithmetic logic unit (ALU), a digital signal processor, a microcomputer, a field programmable gate array (FPGA), a System-on-Chip (SoC), a programmable logic unit, a microprocessor, application-specific integrated circuit (ASIC), etc. While the inventive concepts have been particularly shown and described with reference to example embodiments thereof, it will be understood that various changes in form and details may be made therein without departing from the spirit and scope of the following claims. | 42,232 |
11863057 | Like reference symbols in the various drawings indicate like elements. DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS To aid understanding, this document is organized as follows. First, to help introduce discussion of various embodiments, a high bandwidth variable frequency modulation system is introduced with reference toFIG.1that operates to maintain a predetermined relationship between a crossover frequency and switching frequency. Second, that introduction leads into a description with reference toFIGS.2A-3Bof exemplary high bandwidth variable frequency modulation circuits, including exemplary crossover frequency tuning engines.FIG.4depicts an exemplary digital embodiment of a crossover frequency tuning engine. Third, with reference toFIG.5, exemplary results of operation of a crossover frequency tuning engine are described in application to an exemplary variable frequency modulation circuit. Fourth, with reference toFIGS.6-7, the discussion turns to exemplary methods of maintaining and setting, respectively, a predetermined crossover frequency to switching frequency relationship. Finally, the document discusses further embodiments, exemplary applications and aspects relating to high bandwidth variable frequency modulation circuits employing crossover frequency tuning. FIG.1depicts an exemplary high bandwidth frequency modulation circuit105including an exemplary crossover frequency tuning engine125employed in an illustrative use-case scenario. In the depicted scenario100, the variable frequency modulation circuit (e.g., a regulated power supply circuit)105is shown in the context of use in a desktop computer110. The regulated power supply circuit105provides power (e.g., regulated voltage and/or regulated alternating current (AC) and/or direct current (DC)) to a load111. The load111may, for example, be one or more processors, graphics cards, other electronic components, or some combination thereof. The regulated power supply circuit105includes a feedback control module115which monitors a feedback signal corresponding to the output provided to the load111. The feedback control module115contains compensation module116and an error amplifier117. The feedback control module115provides an output signal (Cout) to a switch module120. The switch module120contains a frequency modulator121configured to provide a control signal corresponding to a switching frequency (fsw) based on a signal Ccorcorresponding to the signal Coutfrom the feedback control module115. A power stage122of the switch module120operates at least one switch according to fswto transfer power from an input (Vin) to the load111. In the depicted example, a crossover frequency tuning engine (XFTE)125is connected between the feedback control module115and the switch module120by gain stage circuit128. As depicted, the XFTE125receives the input signal Coutfrom the feedback control circuit128. The XFTE125includes a high pass filter module126and a gain module127and is configured to generate a gain in response to a transient in the switching frequency. The XFTE125identifies a transient in the switching frequency and a corresponding correction output signal via the signal Coutreceived from the feedback control module115. As depicted, during a transient in the switching frequency, the XFTE125generates a transient correction signal (Ctrans) corresponding to a non-unity gain. The gain stage circuit128applies the contribution of Ctransto Coutto generate a corrected signal Ccor. During a steady state switching frequency, Ctranscorresponds to unity gain and Cout, accordingly, passes through gain stage circuit128to switch module120with no contribution from XFTE125. In various embodiments a transient switching frequency may, by way of example and not limitation, correspond to a step up in demand of load111(e.g., calculating bitcoin, processing graphics) or a step down in demand of load111(e.g., entering a lower power state, such as a ‘sleep mode’). The XFTE125selectively contributes correction signal Ctransin response to a transient in the switching frequency as detected through the feedback control module115based on a predetermined relationship between the switching frequency and the crossover frequency (e.g., a ratio fsw:fc). Accordingly, the XFTE125is configured to advantageously maintain the predetermined relationship of fsw:fcduring a transient such as, for example, in load111. In various embodiments the predetermined relationship may, for example, include a range of ratios between switching frequency and crossover frequency such that the XFTE125advantageously maintains an approximate switching frequency to crossover frequency ratio within a predetermined range. The XFTE125generates the signal Ctransaccording to the predetermined relationship of the switching frequency to the crossover frequency. An exemplary Bode plot130illustrates an example effect of the XFTE125on frequency characteristics of the variable frequency modulation circuit105. An example unmodified relationship between frequency and magnitude is shown by curve135(e.g., obtained by sweeping a network analyzer across a range of frequencies and measuring a response, such as a voltage). The curve135crosses 0 dB (indicated by line145) at the crossover frequency (fc), indicated by point150. The XFTE125responds to a transient in the switching frequency by applying a gain146which shifts the curve135, as shown by shifted curve140. The crossover frequency is, thus, shifted, as shown by point155. Accordingly, a bandwidth of the regulated power supply circuit105may be advantageously increased by maintaining the predetermined relationship between the crossover frequency and switching frequency and, thereby, maintaining desired stability, noise, and transient response characteristics of the circuit even during transients in the switching frequency. FIG.2Adepicts an exemplary electrical schematic of high bandwidth variable frequency modulation circuit200including an exemplary crossover frequency tuning engine125configured to maintain a predetermined crossover frequency to switching frequency relationship. In the depicted example, the high bandwidth variable frequency modulation circuit200includes a feedback control module115configured to monitor a frequency-modulated power output (Vout) relative to a reference signal (Vref) and generate a control output signal (Cout). An XFTE125receives Coutand contributes a transient correction signal Ctransin response to a transient detected in Coutaffecting the switching frequency (fsw) of a switch module120to maintain a predetermined relationship between the switching frequency and crossover frequency of the circuit200. A gain stage circuit128applies Ctransas applicable to Coutto generate a corrected control signal (Ccor). The switch module120is configured to generate the frequency-modulated power output (Vout) based on the control signal Ccor. In the depicted example, the feedback control module115generates an error signal using the error amplifier117. The error amplifier117compares Voutto Vrefand passes the result to compensation module116. The compensation module116generates a compensation signal based on the error signal. Compensation may include, by way of example and not limitation, filtering, PID control, or some combination thereof. Switch module120includes a frequency modulator121configure to receive Ccorand generate a switching frequency signal to drive the power stage122had a corresponding switching frequency (fsw). The power stage122includes, for example, a pulse generator210configured to generate pulses corresponding to fsw, and switches215configure to frequency modulate a power source (Vin) according to the fsw. The power stage122provides the resulting power (Vout) to a load through an LC filter. In various embodiments base frequency characteristics of the circuit200, including, by way of example and not limitation, noise, stability, and transient response, may be determined according to predetermined values of the capacitor and inductor in the LC filter. In the depicted example, the XFTE125receives the control output signal Coutfrom the compensation module116into a high pass filter module126. The high pass filter module126may, for example, AC couple the XFTE125such that the XFTE125only contributes to the control signal Ccorin response to a transient (higher frequency component) detected in the control output signal Cout. A gain module127generates the transient control signal Ctransbased on the predetermined relationship between the switching frequency and the crossover frequency of the circuit200in response to a transient in Cout. The gain stage circuit128applies Ctransto Coutto generate Ccor. Accordingly, the switching frequency control signal output by the frequency modulator121is offset by the output of the XFTE125in response to a transient. The crossover frequency is thereby shifted, and the predetermined relationship between the crossover frequency and the switching frequency is maintained. Adjustment of the crossover frequency in response to a transient in the switching frequency advantageously allows desired characteristics of the circuit200depending on the predetermined relationship between the crossover frequency and the switching frequency such as, for example, noise, transient response, and stability, to be maintained even when the switching frequency shifts in response to a transient in the load, input voltage, other transient affecting the switching frequency, or some combination thereof. Accordingly, the effective bandwidth of the circuit200is effectively increased by advantageously maintaining the desired characteristics of the circuit200across a wider range of switching frequencies. FIG.2Bdepicts an exemplary electrical schematic of high bandwidth variable frequency modulation circuit201including the exemplary crossover frequency tuning engine125. The circuit201may, for example, be configured as a variation of the circuit200. The feedback control module115includes a compensation module116and an error amplifier117, such as is described in relation toFIG.2A. In the depicted example, the feedback control module115of circuit201is configured such that the compensation module116receives a signal corresponding to the output (Vout) and provides a compensated output error amplifier117. The error amplifier117compares the compensated signal to the reference signal (Vref) and generates the control output signal (Cout). The XFTE125is configured to contribute a transient control signal to Coutvia the gain stage circuit128in response to a transient detected in Cout. The switch module120is operated in response to the corrected control signal (Ccor) to generate Voutwhile maintaining the predetermined relationship between the crossover frequency and the switching frequency. Accordingly, the XFTE125may effectively and advantageously increase the bandwidth of the circuit201. In various embodiments the circuit200and the circuit201may, for example, be analog circuits, digital circuits, or some combination thereof. The XFTE125may, for example, be configured as an analog circuit, a digital circuit, or some combination thereof. In various embodiments the feedback control module115the switch module120, the XFTE125, or some combination thereof may be provided with an analog to digital converter (ADC), a digital to analog converter (DAC), or some combination thereof. FIG.3Adepicts an exemplary electrical schematic of a high bandwidth constant on time switching regulator circuit300including the exemplary crossover frequency tuning engine125configured to receive a control signal from a low pass filter325. In this depicted digital implementation, a feedback control module115of the switching regulator circuit300includes an analog error amplifier117. The error amplifier117compares an output voltage (Vout) of the circuit300with a reference voltage (Vref) and provides a resulting signal to a compensation module116. The compensation module116converts the analog input from the error amplifier117into a digital signal by analog to digital converter310. The output of the analog to digital converter310is filtered by a low pass filter315. The resulting signal is passed to a linear time-insensitive circuit (e.g., proportional, integral; proportional-integral-derivative), embodied in the depicted example as a digital PID (proportional, integral, derivative) circuit320. The digital PID circuit320generates a feedback control signal (CPID) which is filtered by a second low pass filter325. The resulting control output signal (Cout) is provided to an XFTE125. The low pass filter325may, for example, be configured to filter out the switching frequency of the switch module120. The XFTE125generates a transient control signal (Ctrans) in response to a transient detected in Coutaccording to a predetermined relationship between a switching frequency and crossover frequency of the circuit300. A gain stage circuit128(e.g., a multiplier, a shift register, other appropriate gain stage components, or some combination thereof) applies the contribution of Ctransto Coutto generate a corrected control signal (Ccor). A frequency modulator121of the switch module120includes a digital voltage-controlled oscillator (VCO)365. The digital VCO365generates a frequency control signal according to Ccor. A pulse generator210receives the frequency control signal and operates a plurality of switches215in a power stage122at a frequency represented by the frequency control signal. In the depicted example, the pulse generator210includes interleaving management module370and digital pulse width modulators (PWMs)375, which sequentially operate the switches215at the switching frequency in response to the frequency control signal to generate the determined Vout. Accordingly, the constant on time switching regulator circuit300may be advantageously operated across a wider bandwidth of switching frequencies by selective contribution of the XFTE125during a transient in the switching frequency to maintain a predetermined relationship between the crossover frequency and the switching frequency of the circuit300. FIG.3Bdepicts an exemplary electrical schematic of a high bandwidth constant on time switching regulator circuit301including the exemplary crossover frequency tuning engine125configured to receive a control signal from a PID circuit320and to dynamically contribute control input to an LP filter325. The error amplifier117compares an output voltage (Vout) of the circuit300with a reference voltage (Vref) and provides a resulting signal to a compensation module116. The compensation module116converts the analog input from the error amplifier117into a digital signal by analog to digital converter310. The output of the analog to digital converter310is filtered by a low pass filter315. The resulting signal is passed to a digital PID (proportional-integral-derivative) circuit320. The digital PID circuit320generates a feedback control signal (CPID), which is received as an input by the XFTE125. The XFTE125selectively generates transient control signal Ctransand a transient filter control signal CtransFin response to a transient in the switching frequency detected in the PID control output signal CPIDaccording to a predetermined relationship between the crossover frequency and the switching frequency for the circuit301. In various embodiments Ctransand CtransFmay be separate signals or the same signal. The low pass filter325is configured to dynamically adjust at least one cutoff frequency in response to CtransF. For example, the low-pass filter325and XFTE125may be configured such that a frequency range of the low-pass filter325increases when the switching frequency increases and decreases when the switching frequency decreases. The low pass filter325may, by way of example and not limitation, adjust a cutoff frequency to continue to filter out the switching frequency in response to a transient in the switching frequency. In various embodiments the low pass filter325may, by way of example and not limitation, be adjusted to mitigate effects of equivalent series resistance (ESR) of an output capacitor bank. Accordingly, the XFTE125may advantageously dynamically control the bandwidth of the circuit301in response to a transient in switching frequency. The dynamically controlled bandwidth may advantageously increase an effective bandwidth of the circuit301while maintaining desired design characteristics of the circuit301. The output CPIDof the digital PID320is filtered by the (dynamic) lowpass filter325to generate a filtered control output signal (Cout). The transient control signal Ctransgenerated by the XFTE125is applied to the filtered control output signal Coutby the gain stage circuit128to generate a corrected control signal (Ccor). Accordingly, the XFTE125selectively modifies the filtered control output signal provided to the digital VCO365to advantageously maintain the predetermined switching frequency to crossover frequency relationship of the circuit301during a transient in the switching frequency as detected in an output of the digital PID320. FIG.4depicts an exemplary block diagram of the crossover frequency tuning engine125configured for a digital variable frequency modulation circuit. In the depicted digital circuit400, the crossover frequency tuning engine (XFTE)125receives a control output signal (Cout) such as, for example, is described in relation toFIGS.2A-3B. The Coutsignal may, for example, be a digital word. The XFTE125applies a high pass filter module126to Cout. The filtered control output signal is processed by a processor405of a gain module127. The processor is operably connected to a memory module410and a storage module415. As depicted, the memory module410is random access memory (RAM) and the storage module415is nonvolatile memory (NVM). The processor405, for example, may be operably connected and configured to execute a program of instructions retrieved from the storage module415and loaded in the memory module410. The program of instructions may, for example, be a software program including, by way of example and not limitation, firmware. The gain module127may include one or more components including, by way of example and not limitation, ASICs (application-specific integrated circuits), FPGAs (field-programmable gate arrays), DSPs (digital signal processors), microcontrollers, or some combination thereof. The gain module127generates a transient control signal (Ctrans), in response to a transient in Cout, according to a predetermined relationship between switching frequency and crossover frequency of a corresponding circuit. The Ctranssignal may, for example, represent a shift to be applied to a digital word by a shift register. A gain stage circuit128includes a shift register420configured to apply the Ctranssignal to the Coutsignal by shifting the Coutword according to the Ctransoffset. Accordingly, the gain stage circuit128generates a corrected control signal (Ccor). The Ccorsignal made, for example, be a digital word shifted in response to a transient in the Coutinput received by the XFTE125. In various embodiments the digital control signal correction circuit400may be provided with one or more ADCs, DACs, or some combination thereof. The circuit400may advantageously increase an effective bandwidth of a digital and/or analog variable frequency modulation circuit by modifying a digital control output signal in response to a transient to maintain a predetermined crossover frequency to switching frequency relationship, and so maintain desired circuit characteristics such as, by way of example and not limitation, noise, stability, and transient response. FIG.5depicts exemplary results of operating of the crossover frequency tuning engine125in the exemplary switching regulator circuit300ofFIG.3A. Frequency response (e.g., Bode plots) of an exemplary circuit are depicted. The exemplary circuit may be, for example, a circuit such as is described in relation toFIG.3A. The first set of frequency response plots500correspond to an exemplary steady state value of the circuit. A magnitude plot505illustrates the magnitude of the frequency response corresponding to a range of frequencies, and phase plot510illustrates a phase of the frequency response at corresponding frequencies. The solid lines507and512depict frequency response of the entire circuit. The dashed lines506and511illustrate frequency response of a feedback control circuit such as, for example feedback control module115ofFIG.3A. The peak in the line507corresponds, for example, to a resonant frequency of the inductor and capacitor of the power stage122depicted inFIG.3A. A crossover frequency at steady state of the circuit is shown where the frequency response line507of the circuit crosses unity gain (0 dB), indicated on the magnitude plot505as fc1. The second set of frequency response plots501correspond to an exemplary crossover frequency tuned circuit such as shown inFIG.3Aduring a transient (e.g., in signal Cout) in which the XFTE125contributes a transient control signal (e.g., Ctrans) a control signal provided to a switch module120. A magnitude plot515illustrates the magnitude of the frequency response corresponding to a range of frequencies, and a phase plot520illustrates a phase of frequency response at corresponding frequencies. Solid lines517and522depict frequency response of the entire circuit. The dashed lines516and521illustrate frequency response of the feedback control circuit. In the depicted example, the XFTE125has contributed a transient control signal corresponding to a gain of approximately 20 decibels. Accordingly, the magnitude curves of the solid lines517and the magnitude curves of the dashed lines516of plot515are shifted upwards relative to the steady state curves shown in plot505. Thus, the crossover frequency of the circuit has been shifted higher, as indicated in magnitude plot515by fc2. For example, the transient response plots501may correspond to an increased switching frequency. The XFTE125may generate a transient control signal corresponding to a gain configured to adjust the crossover frequency such that a predetermined relationship between the crossover frequency in the switching frequency of the circuit is maintained. Accordingly, bandwidth of the circuit is dynamically increased according to the change in switching frequency such that, for example, desired characteristics of the circuit (e.g., noise, transient response, and stability) are maintained. FIG.6depicts an exemplary method for tuning crossover frequency in a variable frequency modulation circuit with a crossover frequency tuning engine. In the depicted method600, a control signal is received605(e.g., Cout). The control signal is compared610to a predetermined steady state value of the control signal. If the control signal is equal to the steady state value615, then no crossover frequency tuning is needed, and the method returns to step605. If the control signal is not equal to the steady state value615, then a scaling factor is determined620to restore a predetermined crossover frequency to switching frequency relationship. The scaling factor is applied625to modify the crossover frequency of the circuit to maintain the predetermined crossover frequency to switching frequency relationship (e.g., by applying a transient control signal to a control output signal to generate a corrected control signal). The method600repeats in a substantially continuous process. Accordingly, a predetermined relationship between a crossover frequency in a switching frequency of a circuit may be advantageously maintained by application of the scaling factor (e.g., a gain) to one or more control signals of a variable frequency modulation circuit. In various embodiments the steady state value, the predetermined relationship, or some combination thereof may be embodied in analog circuit components (e.g., values of resistors, inductors, capacitors, transistors, amplifiers), stored in memory modules (e.g., lookup table, NVM, RAM), or some combination thereof. Various embodiments of the method600may, by way of example and not limitation, be implemented as an analog circuit, a digital circuit, or some combination thereof. In various embodiments circuits configured to implement the method600may include, for example, circuits such as described in relation toFIGS.1-4. FIG.7depicts an exemplary method for determining a predetermined crossover frequency to switching frequency relationship in a crossover frequency tuning engine. In the depicted method700, switching frequency705of a variable frequency modulation circuit (e.g., as described in relation toFIGS.1-4) is set. The switching frequency may, by way of example and not limitation, be predetermined and/or dynamically determined within a predetermined range. The stability of a feedback system (e.g., feedback control module115) of the circuit is then evaluated710. The stability may be evaluated, for example, by characterizing the circuit using a network analyzer to sweep across a range of frequencies to determine magnitude of stimulus versus response and/or phase ratio between stimulus and response (e.g., as described in relation toFIG.5). If the stability does not meet a predetermined threshold715, then compensation (e.g., PID control coefficients, capacitor and/or resistor values, filter values) is adjusted720and the stability of the feedback system is reevaluated710. Once the stability of the feedback system meets a predetermined threshold715, then a relationship (e.g., ratio) of crossover frequency to switching frequency of the circuit is set725. A transient response of the circuit is evaluated730. For example, transient response may be measured by plotting a signal of interest (e.g., Vout, Cout, fsw, Ccor, Ctrans, or some combination thereof) in time while inducing a transient (e.g., in an input power Vin, load on Vout, or some combination thereof). If a transient response is not within a predetermined threshold735, then the relationship is adjusted740and the transient less response is reevaluated730. Once the transient response is within the predetermined threshold735(e.g., if a signal of interest recovers to a stable value within a predetermined time), then the noise of the circuit is evaluated745. If the noise exceeds a predetermined threshold750, then the relationship is adjusted740, and step730through750are repeated. Once the noise is within a predetermined threshold750, then the relationship is determined and the method700is complete. The relationship may be saved, for example, in a data store, may be implemented in circuit components, or some combination thereof. The method700may, by way of example and not limitation, be performed on a simulated circuit, on a test circuit, on an actual circuit, or some combination thereof. Accordingly, a predetermined relationship may be set between crossover frequency and a switching frequency of a circuit to achieve desired characteristics of stability, noise, and transient response. A predetermined relationship may, for example, be implemented in an XFTE125such that the XFTE125is configured to advantageously generate a transient control signal to maintain the predetermined relationship in response to a transient in the switching frequency of the circuit. Although various embodiments have been described with reference to the figures, other embodiments are possible. For example, although an exemplary system in a scenario100has been described with reference toFIG.1, other implementations may be deployed in other industrial, scientific, medical, commercial, and/or residential applications. In various embodiments, various components and/or subcircuits may, by way of example and not limitation, be implemented as analog circuits, digital circuits, or some combination thereof. For example, a filter (e.g., low pass filter325) circuit may be implemented as a digital and/or analog circuit. The low pass filter325may, by way of example and not limitation, be implemented as at least one resistor(s) with adjustable resistance values (e.g., by a transistor(s)) and an active gain component(s) (e.g., an operational amplifier), a digital filter (e.g., microprocessor, ASIC, FPGA, digital signal processor), or some combination thereof. In various embodiments, circuits may include, for example, finite impulse response (FIR) filter circuits and/or infinite impulse response (IIR) filter circuits. In various embodiments a variable frequency modulation circuit with an XFTE may be implemented in various use-case scenarios. An XFTE-equipped variable frequency modulation circuit may, for example, be implemented as a power regulator. The variable frequency modulation circuit may, by way of example and not limitation, be implemented as a constant on-time regulator, a constant off-time regulator, a constant duty-cycle regulator, or some combination thereof. In various embodiments an XFTE (e.g.,125) may be configured such that the crossover frequency is constrained below ½ of the switching frequency (for example, below the Nyquist frequency). in various embodiments the crossover frequency may be constrained between, by way of example and not limitation, ⅓ and ⅛ of the switching frequency. In various embodiments the XFTE may be configured to maintain a predetermined crossover frequency to switching frequency relationship of a circuit by generating a gain (e.g., by gain module127) using a lookup table, a predetermined algorithm, an analog circuit with predetermined and or variable component values, or some combination thereof. In various embodiments a feedback control module (e.g.,115) may include a quantum charge modulator (QCM). The QCM may, by way of example and not limitation, be configured such as is described in U.S. patent Ser. No. 10/523,102, filed by Babazadeh, et al. on Jan. 10, 2019 and issued Dec. 31, 2019, the entire contents of which are incorporated herein by reference. For example, a QCM may be configured to modulate the frequency of a switch signal to achieve a fast transient response while holding the average frequency constant over a predetermined number of N cycles. In an illustrative example, a QCM may include a compensation processor configured to compensate an error signal and generate a compensation signal by performing operations to maintain an average switching frequency over the N cycles in response to the transient. The compensation signal may be a function of a real phase deviation ΔTSWbetween a stable pulse modulated signal having a cycle period TSWbefore the transient and a measured pulse modulated signal having a cycle period TSW_Mafter the transient. A forgetting factor may be used to calculate the phase deviations. The QCM may provide a compensation free, stable, and high-performance response over power stage component changes. In various embodiments, the QCM may, by way of example and not limitation, be configured to modulate the compensation signal output from a conventional linear time-invariable (LTI) (e.g., a proportional and integrator (PI), a proportional-integral-derivative (PID)) compensator module (e.g., a digital and/or analog PID, such as digital PID of), for example, as a function of a difference between a measured period after the transient and a period during steady-state operation prior to the transient. By selectively applying a forgetting factor in certain predetermined operating conditions, the average frequency may be held constant over the predetermined N cycles of the power converter. Accordingly, various embodiments including a QCM and XFTE (e.g.,125) may advantageously dynamically control bandwidth to provide a significantly increased effective bandwidth of a variable frequency modulation circuit. In various embodiments, some bypass circuits implementations may be controlled in response to signals from analog or digital components, which may be discrete, integrated, or a combination of each. Some embodiments may include programmed, programmable devices, or some combination thereof (e.g., PLAs, PLDs, ASICs, microcontroller, microprocessor), and may include one or more data stores (e.g., cell, register, block, page) that provide single or multi-level digital data storage capability, and which may be volatile, non-volatile, or some combination thereof. Some control functions may be implemented in hardware, software, firmware, or a combination of any of them. Computer program products may contain a set of instructions that, when executed by a processor device, cause the processor to perform prescribed functions. These functions may be performed in conjunction with controlled devices in operable communication with the processor. Computer program products, which may include software, may be stored in a data store tangibly embedded on a storage medium, such as an electronic, magnetic, or rotating storage device, and may be fixed or removable (e.g., hard disk, floppy disk, thumb drive, CD, DVD). Although an example of a system, which may be in a desktop environment, has been described with reference to the above figures, other implementations may be deployed in other processing applications, such as portable and/or networked environments. Temporary auxiliary energy inputs may be received, for example, from chargeable or single use batteries, which may enable use in portable or remote applications. Some embodiments may operate with other DC voltage sources, such as 1.5V, 3V, 6V, 9V, and/or 12V (nominal) batteries, for example. Alternating current (AC) inputs, which may be provided, for example from a 50/60 Hz power port, or from a portable electric generator, may be received via a rectifier and appropriate scaling. Provision for AC (e.g., sine wave, square wave, triangular wave) inputs may include a line frequency transformer to provide voltage step-up, voltage step-down, and/or isolation. Although particular features of an architecture have been described, other features may be incorporated to improve performance. For example, caching (e.g., L1, L2, . . . ) techniques may be used. Random access memory may be included, for example, to provide scratch pad memory and or to load executable code or parameter information stored for use during runtime operations. Other hardware and software may be provided to perform operations, such as network or other communications using one or more protocols, wireless (e.g., infrared) communications, stored operational energy and power supplies (e.g., batteries), switching and/or linear power supply circuits, software maintenance (e.g., self-test, upgrades), and the like. One or more communication interfaces may be provided in support of data storage and related operations. Some systems may be implemented as a computer system that can be used with various implementations. For example, various implementations may include digital circuitry, analog circuitry, computer hardware, firmware, software, or combinations thereof. Apparatus can be implemented in a computer program product tangibly embodied in an information carrier, e.g., in a machine-readable storage device, for execution by a programmable processor; and methods can be performed by a programmable processor executing a program of instructions to perform functions of various embodiments by operating on input data and generating an output. Various embodiments can be implemented advantageously in one or more computer programs that are executable on a programmable system including at least one programmable processor coupled to receive data and instructions from, and to transmit data and instructions to, a data storage system, at least one input device, and/or at least one output device. A computer program is a set of instructions that can be used, directly or indirectly, in a computer to perform a certain activity or bring about a certain result. A computer program can be written in any form of programming language, including compiled or interpreted languages, and it can be deployed in any form, including as a stand-alone program or as a module, component, subroutine, or other unit suitable for use in a computing environment. Suitable processors for the execution of a program of instructions include, by way of example, both general and special purpose microprocessors, which may include a single processor or one of multiple processors of any kind of computer. Generally, a processor will receive instructions and data from a read-only memory or a random-access memory or both. The essential elements of a computer are a processor for executing instructions and one or more memories for storing instructions and data. Generally, a computer will also include, or be operatively coupled to communicate with, one or more mass storage devices for storing data files; such devices include magnetic disks, such as internal hard disks and removable disks; magneto-optical disks; and optical disks. Storage devices suitable for tangibly embodying computer program instructions and data include all forms of non-volatile memory, including, by way of example, semiconductor memory devices, such as EPROM, EEPROM, and flash memory devices; magnetic disks, such as internal hard disks and removable disks; magneto-optical disks; and CD-ROM and DVD-ROM disks. The processor and the memory can be supplemented by, or incorporated in, ASICs (application-specific integrated circuits). In some implementations, each system may be programmed with the same or similar information and/or initialized with substantially identical information stored in volatile and/or non-volatile memory. For example, one data interface may be configured to perform auto configuration, auto download, and/or auto update functions when coupled to an appropriate host device, such as a desktop computer or a server. In some implementations, one or more user-interface features may be custom configured to perform specific functions. Various embodiments may be implemented in a computer system that includes a graphical user interface and/or an Internet browser. To provide for interaction with a user, some implementations may be implemented on a computer having a display device, such as a CRT (cathode ray tube) or LCD (liquid crystal display) monitor for displaying information to the user, a keyboard, and a pointing device, such as a mouse or a trackball by which the user can provide input to the computer. In various implementations, the system may communicate using suitable communication methods, equipment, and techniques. For example, the system may communicate with compatible devices (e.g., devices capable of transferring data to and/or from the system) using point-to-point communication in which a message is transported directly from the source to the receiver over a dedicated physical link (e.g., fiber optic link, point-to-point wiring, daisy-chain). The components of the system may exchange information by any form or medium of analog or digital data communication, including packet-based messages on a communication network. Examples of communication networks include, e.g., a LAN (local area network), a WAN (wide area network), MAN (metropolitan area network), wireless and/or optical networks, the computers and networks forming the Internet, or some combination thereof. Other implementations may transport messages by broadcasting to all or substantially all devices that are coupled together by a communication network, for example, by using omni-directional radio frequency (RF) signals. Still other implementations may transport messages characterized by high directivity, such as RF signals transmitted using directional (i.e., narrow beam) antennas or infrared signals that may optionally be used with focusing optics. Still other implementations are possible using appropriate interfaces and protocols such as, by way of example and not intended to be limiting, USB 2.0, Firewire, ATA/IDE, RS-232, RS-422, RS-485, 802.11 a/b/g, Wi-Fi, Ethernet, IrDA, FDDI (fiber distributed data interface), token-ring networks, multiplexing techniques based on frequency, time, or code division, or some combination thereof. Some implementations may optionally incorporate features such as error checking and correction (ECC) for data integrity, or security measures, such as encryption (e.g., WEP) and password protection. In various embodiments, the computer system may include Internet of Things (IoT) devices. IoT devices may include objects embedded with electronics, software, sensors, actuators, and network connectivity which enable these objects to collect and exchange data. IoT devices may be in-use with wired or wireless devices by sending data through an interface to another device. IoT devices may collect useful data and then autonomously flow the data between other devices. Various examples of modules may be implemented using circuitry, including various electronic hardware. By way of example and not limitation, the hardware may include transistors, resistors, capacitors, switches, integrated circuits, other modules, or some combination thereof. In various examples, the modules may include analog logic, digital logic, discrete components, traces and/or memory circuits fabricated on a silicon substrate including various integrated circuits (e.g., FPGAs, ASICs), or some combination thereof. In some embodiments, the module(s) may involve execution of preprogrammed instructions, software executed by a processor, or some combination thereof. For example, various modules may involve both hardware and software. A number of implementations have been described. Nevertheless, it will be understood that various modifications may be made. For example, advantageous results may be achieved if the steps of the disclosed techniques were performed in a different sequence, or if components of the disclosed systems were combined in a different manner, or if the components were supplemented with other components. Accordingly, other implementations are contemplated within the scope of the following claims. | 42,145 |
11863058 | DETAILED DESCRIPTION OF EMBODIMENTS Computer systems may include multiple circuit blocks configured to perform specific functions. Such circuit blocks may be fabricated on a common substrate and may employ different power supply voltage levels. Power management units (commonly referred to as “PMUs”) may include multiple voltage regulator and/or power converter circuits configured to generate regulated voltage levels for various power supply signals. Such voltage regulator circuits may employ both passive circuit elements (e.g., inductors, capacitors, etc.) as well as active circuit elements (e.g., transistors, diodes, etc.). Different types of voltage regulator circuits may be employed based on power requirements of load circuits, available circuit area, and the like. One type of commonly used voltage regulator circuit is a power or buck converter circuit. Such power converter circuits include multiple switches (also referred to as “power switches”) and a switch node that is coupled to a regulated power supply node via an inductor. One switch is coupled between an input power supply node and the switch node and is referred to as the “high-side switch.” Another switch is coupled between the switch node and a ground supply node and is referred to as the “low-side switch.” When the high-side switch is closed (referred to as “on-time”), energy is applied to the inductor, resulting in an increase in the current flowing through the inductor. During this time, the inductor stores energy in the form of a magnetic field in a process referred to as “magnetizing” the inductor. When the high-side switch is opened and the low-side switch is closed, energy is no longer being applied to the inductor and the voltage across the inductor reverses, which results in the inductor functioning as a current source with the energy stored in the inductor's magnetic field supporting the current flowing into the load. The process of closing and opening the high-side and low-side switches is performed periodically to maintain a desired voltage level on the power supply node. The power switches included in buck converters may be operated in different modes. In some cases, a buck converter may employ pulse width modulation (PWM), in which the frequency with which the power converter circuit cycles is fixed, but the period of time that the high-side switch is closed is adjusted based on a comparison of an output voltage of the buck converter to a reference voltage. In other cases, a power converter circuit may employ pulse frequency modulation (PFM), in which the frequency with which the buck converter cycles (including on-time, off-time, and idle time) is adjusted based on the load current. When current flows through the inductor during each active cycle, a power converter circuit is said to be operating in continuous conduction mode (or “CCM”). Alternatively, when there is not current flowing during one or more of the active cycles, the power converter circuit is said to be operating in discontinuous conduction mode (or “DCM”). As load current changes, a power converter circuit may switch modes of operation in order to efficiently provide the desired voltage level on a regulated power supply node. In some cases, dual regulation modes may be employed. For example, PWM mode may be combined with pulse skipping mode (PSM), PFM mode, or burst mode, to accommodate varying load current ranges. During such transitions in regulation mode, a power converter may experience a loss in efficiency due to different criteria that affect the switching between regulation modes. For example, if a threshold is set to high for transitions from PFM to PWM operation, the power converter circuit can skip clock cycles, which can increase inductor current ripple. Such inductor current ripple can translate to increased ripple on the regulated power supply node, possible affecting the operation of load circuits. The embodiments illustrated in the drawings and described below may provide techniques for a power converter circuit to determine an average current delivered to the load during each cycle and using this information to adjust the switching frequency. By using the average current delivered to the load to adjust the switching frequency, efficiency of the power converter circuit may be maintained through transitions in regulation mode, thereby preventing spurious clock cycle skips and increases in inductor current ripple. A block diagram of a power converter circuit is depicted inFIG.1. As illustrated, power converter circuit100includes control circuit101, switching circuit102, and inductor104. Switching circuit102is coupled to input power supply node107and inductor104via switch node110. Inductor104is further coupled to regulated power supply node109. Switching circuit102is configured to source charge current113to regulated power supply node109via inductor104during active cycles112. As described below, during an active cycle, a high-side switch included in switching circuit102may be activated allowing charge current113to flow from input power supply node107into switch node110, through inductor104, and into regulated power supply node109. As charge current113flows into regulated power supply node109, energy is stored in the magnetic field of inductor104. When an active cycle is halted, the high-side switch is de-activated and a low-side switch included in switching circuit102is activated, coupling switch node110to a ground supply node. While switch node110is coupled to ground, inductor104continues to source current to regulated power supply node109using the energy stored in its magnetic field. In various embodiments, control circuit101is also configured to generate control signals108, which are used to initiate and halt active cycles112. Control signals108may be generated differently in various regulation modes. As described below, control circuit101may be further configured to switch regulation modes based on comparisons of the voltage level to various threshold values or other determined values. As part of switching regulation modes, control circuit101is configured to determine average inductor current105for a particular one of active cycles112. In some embodiments, control circuit101is also configured to perform a comparison of average inductor current105to threshold106and, based on a result of the comparison, deactivate switching circuit102for a different active cycle of active cycles112that is subsequent to the particular active cycle. By deactivating various ones of active cycles112using average inductor current105, control circuit101can switch between PWM and PSM modes on a cycle-by-cycle basis, allowing a rapid response to load transients. Turning toFIG.2, a block diagram of an embodiment of switch circuit102is depicted. As illustrated, switch circuit102includes devices201and202, and logic circuit203. Device201is coupled between input power supply node107and switch node110, and is controlled by signal206. In a similar fashion, device202is coupled between switch node110and ground supply node205, and is controlled by signal207. In various embodiments, device201may be implemented as a p-channel metal-oxide semiconductor field-effect transistor (MOSFET), Fin field-effect transistor (FinFET), gate-all-around field-effect transistor (GAAFET), or any other suitable transconductance device. Device202may, in other embodiments, be implemented as an n-channel MOSFET, FinFET, GAAFET, or any other suitable transconductance device. In response to an activation of signal206, device201is configured to couple input power supply node107to switch node110, allowing current to flow into switch node110and then into inductor104, thereby magnetizing inductor104. In response to an activation of signal207, device202is configured to couple switch node110to ground supply node205. With switch node110coupled to ground supply node205, energy is no longer being supplied to inductor104, causing the magnetic field of inductor104to collapse. As the magnetic field collapses, inductor104functions as a current source, providing current to regulated power supply node109. Logic circuit203is configured to generate signal206and signal207using control signals108. In various embodiments, logic circuit203may be configured, in response to an activation of control signal108, to activate signal206and deactivate signal207. Logic circuit203may be further configured, in response to a deactivation of control signals108, to deactivate signal206and activate signal207. In some embodiments, logic circuit203may include any suitable combination of logic gates, sequential logic circuit elements, MOSFETs, FinFETs, GAAFETs, or any other suitable transconductance devices. As used herein, when a signal is activated, it is set to a logic or voltage level that activates a load circuit or device. The logic level may be either a high logic level or a low logic level depending on the load circuit. For example, an active state of a signal coupled to a p-channel MOSFET is a low logic level (referred to as an “active low signal”), while an active state of a signal coupled to an n-channel MOSFET is a high logic level (referred to as an “active high signal”). Turning toFIG.3, a block diagram of an embodiment of control circuit101is depicted. As illustrated, control circuit101includes controller circuit301, a pulse width modulation circuit (denoted as “PWM circuit302”), a pulse frequency modulation circuit (denoted as “PFM circuit303”), and mode circuit304. Controller circuit301is configured to generate controls signals108using mode signal306, PWM signal307, transition signal308, and PFM signal309. In various embodiments, controller circuit301may be configured to activate control signals108in response to a determination that PFM signal309has been activated. Additionally, controller circuit301may be further configured to de-activate and re-activate control signals108based on a value of PWM signal307. In other embodiments, controller circuit301may be configured to override the effect of PWM signal307using skip signal308. For example, if PWM signal307is activated to begin a new charge cycle, controller circuit301may not initiate the new charge cycle in response to a determination that skip signal308has been activated. In various embodiments, controller circuit301may be implemented using any suitable combination of combinatorial and sequential logic circuits. PWM circuit302is configured to generate PWM signal307using inductor current111and a voltage level of regulated power supply node109. As described below, PWM circuit302is configured to compare the voltage level of regulated power supply node to PWM reference507to generate an error signal. PWM circuit302may, in various embodiments, be configured to sense inductor current111using a voltage level of switch node110, and then compare a slope-compensated version of the sensed inductor current to the error signal to generate PWM signal307. PFM/PSM circuit303is configured to generate PFM/PSM signal309using the voltage level of regulated power supply node109, PFM reference311, and PSM reference312. When power converter circuit100is operating in PFM mode, PFM/PSM circuit303is configured to compare the voltage level of regulated power supply node109to PFM reference311to generate PFM/PSM signal309. When power converter circuit100is operating in PSM mode, PFM/PSM circuit303is configured to compare the voltage of regulated power supply node109to PSM reference312. In various embodiments, PFM reference311and PSM reference312are greater than PWM reference310, to keep an error signal generated in PWM circuit302at a minim voltage level during PFM or PSM operation. If the voltage reference of PWM and PFM/PSM modes were to be the same, a voltage level of the error signal could go high at a light load condition. This may push the PWM comparator to produce wider pulses, thereby causing PWM mode to take over and cause the switching regulator to oscillate between PWM and PFM (or PWM and PSM). Mode circuit304is configured to generate mode signal306and transition signal308. The two signals are, in various embodiments, used independently of each other to control mode transitions in power converter circuit100. To generate transition signal308, mode circuit304is further configured to generate an average inductor current. In various embodiments, the average current can be used to control transitions between PWM operation and PFM operation (or PWM operation and PSM operation). It is noted that power converter circuit100may be configured to operate in PWM and PFM operation modes, or PWM and PSM operation modes. In some embodiments, when power converter circuit100is switching between PWM and PSM operation modes, transition signal308may additionally be used to selectively skip particular active cycles during PSM operation. To generate mode signal306, mode circuit304is further configured to detect durations of on-time periods as well as DCM operation. Based on the values of the detected on-time periods or the detection of DCM operation, mode circuit304may activate or de-active mode signal306which can cause controller circuit301to generate control signals108according to different ones of the operation modes. For example, in various embodiments, mode circuit304is configured, in response to a detection of DCM operation or short minimum on-times, to change the state mode signal306to cause power converter circuit100to exit PWM operation and enter PFM operation (or PSM operation). In other embodiments, mode circuit304is configured, in response to detection of long minimum on-times, change the state of mode signal306to cause power converter circuit100to exit PFM (or PSM) operation and enter PWM operation. Although mode signal306and transition signal308are depicted as single wires, in some cases, both mode signal306and transition signal308may include multiple bits of information transmitted using multiple wires. In various embodiments, mode circuit304may be implemented using a state machine or other suitable sequential logic circuit, a microcontroller, or as a general-purpose processor configured to execute software or program instructions. Turning toFIG.4, a block diagram of an embodiment of mode circuit304is depicted. As illustrated, mode circuit304includes average current circuit401, comparator402, and logic circuit403. Average current circuit401is configured to generate average signal405using the voltage level of regulated power supply node109, the on-time of switch node110, and the voltage level of input power supply node107. In various embodiments, average signal405corresponds to an average inductor current during a given active cycle of a plurality of active cycles being performed when power converter circuit100is operating in PWM regulation mode. Comparator402is configured to generate transition signal308using average signal513and skip reference404. To generate transition signal308, comparator402may be further configured to perform a comparison of average signal513and skip reference404, and determine a value for transition signal308based on a result of the comparison. In some embodiments, comparator402is configured to activate transition signal308in response to a determination that average signal513is less than transition reference404. In various embodiments, comparator402may be implemented as a Schmitt trigger circuit or any other suitable type of comparator circuit configured to generate a digital output signal based on a comparison of at least two analog voltage levels. Logic circuit402is configured to generate mode signal306using DCM detection signal406, short min-time signal407, and long min-time signal408. In various embodiments, DCM detection signal406may be activated in response to a detection of DCM operation. In some cases, DCM detection signal406may be generated by detecting zero crossings of inductor current111. In various embodiments, short min-time signal407is a threshold for a minimum on-time of an active cycle during PWM operation. In a similar fashion, long min-time signal408is a threshold for PWM comparator output (307) on-time to decide when to transition to PWM. By using different thresholds for the different operating modes, there is hysteresis between the transition between PFM and PWM operation to prevent oscillation between the operation modes (referred to as “mode chattering”). In various embodiments, logic circuit402is configured to set mode signal306to a value that causes power converter circuit100to exit PWM operation mode and enter PFM (or PSM) in response to an activation of DCM detection signal406and a determination that an active cycle satisfies the threshold of short min-time signal407. In other embodiments, logic circuit402is configured to set mode signal306to a value that causes power converter circuit100to exit PFM (or PSM) mode, and enter PWM mode in response to a determination that the error signal generated by PWM circuit302activates after the threshold specified by long min-time signal408. Logic circuit402may, in various embodiments, be implemented a state machine or other suitable sequential logic circuit, a microcontroller, or as a general-purpose processor configured to execute software or program instructions. Turning toFIG.5, a block diagram of an embodiment of PWM circuit302is depicted. As illustrated, PWM circuit302includes comparators501-502, current sensor circuit504, and slope compensation circuit505. Comparator circuit502is configured generate error signal509using a voltage level of regulated power supply node109and PWM reference507. To generate error signal509, comparator circuit502may be configured to compare the voltage level of regulated power supply node109and PWM reference507, and determine a voltage level of error signal509based on a result of the comparison. In some embodiments, a voltage level of error signal509may be proportional to a difference between the voltage level of regulated power supply node109and PWM reference507. It is noted that PWM reference507may be a different value than PFM reference402as depicted inFIG.4. In various embodiments, comparator circuit502may be implemented using a differential amplifier circuit or any suitable amplifier circuit configured to generate an output signal whose voltage level is based on the respective voltage levels of at least two input signals. Current sensor circuit504is configured to generate sensed current510using a voltage level of switch node110. As describe below, current sensor circuit504may be configured to compare the voltage level of switch node110to a voltage across a device that is a replica of switch device201. Slope compensation circuit505is configured to generate compensation current512. As described below, slope compensation circuit505may be configured to generate compensation current512such that it is a periodic current ramp. It is noted that slope compensation is used to improve the stability of power converter circuit100by increasing a frequency at which a feedback loop of power converter circuit100can operate, thereby improving a response of power converter circuit100to transients in load current demand. Sensed current510and compensation current512are combined on node514to generate sense signal511. In various embodiments, node514is coupled to a ground supply node via a resistor (not shown), and as sensed current510and compensation current512flow into the ground supply node via the resistor, the voltage drop across the resistor corresponds to sense signal511. Comparator501is configured to generate PWM signal307using error signal509and sense signal511. In various embodiments, comparator501may be configured to activate PWM signal307in response to a determination that a voltage level of sense signal511is less than a voltage level of error signal509. Comparator501may, in some embodiments, be implemented as a Schmitt trigger circuit or any other suitable type of comparator circuit configured to generate a digital output signal based on a comparison of at least two analog voltage levels. Turning toFIG.6, a block diagram of an embodiment of slope compensation circuit505is depicted. As illustrated, slope compensation circuit505includes devices602-604, amplifier circuit605, current source606, capacitor607and switch609. Switch609is coupled between node613and ground supply node205. In various embodiments, switch609is configured to couple node613to ground in order to discharge capacitor607and reset the circuit at the end of an active cycle. In various embodiments, switch609may be implemented using one or more transistors coupled between node613and ground supply node205, whose control terminals are coupled to a reset signal (not shown). Current source606is coupled between input power supply node107and node613, and is configured to generate bias current614. In various embodiments, current source606may be implemented using a variety of circuit topologies including a supply and/or temperature independent reference circuit and one or more current mirror circuits. Capacitor607is coupled between node613and ground supply node205. When switch609is open, capacitor607is charged by bias current614generating a linearly increasing voltage level on node613. In various embodiments, capacitor607may be implemented using a metal-oxide-metal (MOM) structure, a metal-insulator-metal (MIM) structure, or any other suitable capacitor structure available in a semiconductor manufacturing process. Device604is coupled between node610and node612, and is controlled by a voltage level of node611. Resistor608is coupled between node612and ground supply node205. An output of amplifier circuit605is coupled to node611, while the inputs of comparator circuit605are coupled to nodes612and613. In some embodiments, amplifier circuit605, device604, and resistor608form a voltage-to-current converter circuit configured to generate a current flowing in device604that is proportional to the voltage level of node613. In various embodiments, comparator circuit605may be implemented as a differential amplifier circuit, while device604may be implemented as an n-channel MOSFET, FinFET, GAAFET, or any other suitable transconductance device. Resistor608may, in some embodiments, be implemented using polysilicon, metal, or any other suitable material available in a semiconductor manufacturing process. Device603is coupled between input power supply node107and node610, and is controlled by a voltage level of node610. In a similar fashion, device602is coupled between input power supply node107and node615, and is controlled by the voltage level of node610. In various embodiments, devices602and603form a current mirror circuit configured to replicate a current flowing through device604, which also flows through device603, into a current flowing in device602to generate compensation current512in node615. It is noted that a magnitude of the compensation current512may be modified by changing one or more physical parameters (e.g., width) of device602relative to the physical parameters of device603. In various embodiments, devices602and603may be implemented as p-channel MOSFETs, FinFETs, GAAFETs, or any other suitable transconductance device. Turning toFIG.7, a block diagram of an embodiment of current sensor circuit504is depicted. As illustrated, current sensor circuit504includes devices701-706, and amplifier circuit715. Device701is coupled between input power supply node107and node707, and is controlled by signal206. In various embodiments, device701may be a replica, or a scaled replica, of device201included in switching circuit102. Device701may, in some embodiments, be implemented as a p-channel MOSFET, FinFET, GAAFET, or any other suitable transconductance device. Amplifier circuit715is configured to compare a voltage level of node707to a voltage level of switch node110to generate a voltage on node708. In various embodiments, comparator circuit715is configured to generate the voltage on node708such that a magnitude of the voltage on node708corresponds to an amplified difference between the voltage level of node707and switch node110. In various embodiments, comparator circuit715may be implemented as a differential amplifier circuit or any other suitable amplifier circuit configured to generate an output signal whose voltage level is based on a comparison of respective voltage levels of two or more input signals. Device702is coupled between node707and node709and is controlled by the voltage of node708. The current flowing in device702may, in various embodiments, correspond to a current flowing in inductor104during an active cycle. Device702may, in some embodiments, be implemented as a p-channel MOSFET, FinFET, GAAFET, or any other suitable transconductance device. Device703is coupled between node709and ground supply node205, and is controlled by a voltage level of node709. It is noted that the current flowing through device702also flows through device703. Device704is coupled between node710and ground supply node205, and is controlled by the voltage level of node709. In various embodiments, devices703and704form a current mirror circuit configured to generate a replica of the current flowing in device703into a current flowing in device704. It is noted that a magnitude of the current flowing in device704may be modified by changing one or more physical parameters (e.g., width) of device704relative to the physical parameters of device703. Devices703and704may, in some embodiments, be implemented as n-channel MOSFETs, FinFETS, GAAFETs, or any other suitable transconductance devices. Device705is coupled between input power supply node107and node710, and is controlled by a voltage level of node710. It is noted that the current flowing in device704also flows through device705. Device706is coupled between input power supply node107and node711, and is controlled by the voltage level of node710. In various embodiments, devices705and706form a current mirror circuit configured to generate a replica of the current flowing in device705in to a current flowing in device706to generate sensed current510. It is noted that a magnitude of sensed current510may be modified by changing one or more physical parameters (e.g., width) of device706relative to the physical parameters of device705. Devices705and706may, in some embodiments, be implemented as p-channel MOSFETs, FinFETS, GAAFETs, or any other suitable transconductance devices. Turning toFIG.8, a block diagram of an embodiment of average current circuit506is depicted. As illustrated, average current circuit506includes three voltage-to-current circuits (denoted as “V2I circuit801,” “V2I circuit802,” and “V2I circuit808”), Multiplier/Divider circuits803-805, buffer circuit806, signal converter circuit807, and peak detector circuit809. V2I circuit801is configured to generate current I_in810using a voltage level of input power supply node107. In various embodiments, a magnitude of current I_in810may be proportional to the voltage level of input power supply node107. In a similar fashion, V2I circuit802is configured to generate current I_vout811using a voltage level of regulated power supply node109. Multiplier/divider circuit803is configured to generate current I_duty812using current I_vout811, current I_in810, and current I_o817. It is noted that, in various embodiments, current I_o817may corresponding to charge current113. To generate current I_duty812, multiplier/divider circuit803may be further configured to multiply current I_o817by the quotient of currents I_vout811and I_in810. Buffer circuit806is configured to generate SWon815using an on-time of switch node110. In various embodiments, buffer circuit806may be implemented as a pair of inverter gates, or any other comparator. Signal converter circuit807is configured to generate V_ton819using SWon815. In various embodiments, signal converter circuit807may be implemented as a time-to-analog converter circuit. In such cases, a magnitude of V_ton819corresponds to the on-time of SWon815transitions. V2I circuit808is configured to generate current I_ton814using V_ton819. In various embodiments, a magnitude of current I_ton814may be proportional to the on-time of switch node110. Multiplier/divider circuit804is configured to generate current Iratio813using current I_T818, current I_ton814, and current I_o817. It is noted that, in various embodiments, current I_T818may correspond period of a particular one of active cycles112. To generate current I ratio813, multiplier/divider circuit804may be further configured to multiply current I_o817by the quotient of currents I_ton814and I_T818. Multiplier/divider circuit805is configured to generate average signal513using current I_duty812, current Iratio813, and current Ipeak816. To generate average signal513, multiplier/divider circuit805may be further configured to multiply current Ipeak816by the quotient of currents I_duty812and Iratio813. It is noted that multiplier/divider circuit805may be configured to generate a current corresponding to average signal513, and a voltage version of average signal513may be generated using the current and a resistor (not shown). Peak detector circuit809is configured to generate current Ipeak816using error signal509and compensation current512. As described below, peak detector circuit809may be further configured to sample compensation current512and use a sampled versioned of compensation current512in conjunction with a sampled version of error signal509to generate current Ipeak816. In various embodiments, peak detector circuit809may be configured to sample compensation current512and error signal509using one or more signals based on SWon815. In one embodiment, the average current may be calculated using the following equation: Iavg(out)=Ipeak2·tonT·VINVout.where Ipeakis the peak current, tonis the on time of the high side switch, T is the switching period, VINis the input voltage and VOUTis the output voltage. The calculations performed using the equation above may be valid for discontinuous current mode (DCM) operation, as mode transitions may be desirable at light loads to maintain the performance of the switching regulator. Turning toFIG.9, a block diagram of an embodiment of peak detector circuit809is depicted. As illustrated, peak detector circuit809includes devices901-910, amplifiers912and913, current source919, resistors916and917, and switches921-922. Device901is coupled between input power supply node107and node924, and is controlled by a voltage level of node924. In various embodiments, node924is coupled to slope compensation circuit505such that compensation current512flows in device901. Switch921is coupled between node924and node927. In various embodiments, switch921is configured to couple node924to node927in response to a detection of a falling edge of PWM signal307. Device902is coupled between input power supply node107and node926, and is controlled by a voltage level of node927. Device903is coupled between input power supply node107and node928, and is controlled by the voltage level of node927. When switch921is closes, devices901-903function as a current mirror, sampling compensation current512, such that replicas of compensations current512flow in devices902and903, respectively. Device906is coupled between node928and ground supply node205, and is controlled by a voltage level of node928. Device907is coupled between node929and ground supply node205, and is controlled by the voltage level of node928. Devices906and907function as a current mirror configured to generate a replica of the current flowing in device906in device907. Amplifier912is configured to generate a voltage on node934using error signal509and a voltage level of node926. In various embodiments, a magnitude of the voltage generated on node934is proportional to a difference between the voltage level of error signal509and the voltage level of node926. Current source919is coupled between input power supply node107and node929. In various embodiments, current source919is configured to source a bias current into node929. Current source919may, in some embodiments, be implemented as biased transconductance device (e.g., a p-channel MOSFET, FinFET, or GAAFET), part of a current mirror circuit, or any other suitable circuit configured to provide a constant current across a range of output voltage levels. Device905is coupled between node929and ground supply node205, and is controlled by a voltage level of node924. In various embodiments, the voltage level of node934causes a current to flow through device905that is proportional to the difference between the voltage level of error signal509and the voltage level of node926. Resistor916is coupled between node926and node926, allowing a current to flow between the two nodes. The current flowing in resistor916, the bias current generated by current source919, and the current flowing in devices905and907are all combined on node929. Switch922is coupled between node929and an input of amplifier913. Switch922is configured to couple node929to the input of comparator913in response to a detection of a rising edge of PWM signal307, sampling a voltage of node929generated by the combination of the current flowing in resistor916, the bias current generated by current source919, and the current flowing in devices905and907. Amplifier913is configured to generate a voltage on node933using the sampled voltage of node929and a voltage of node930. Device908is coupled between node931and node930, and is controlled by the voltage level of node933. Resistor917is coupled between node930and ground supply node205. In various embodiments, the combination of amplifier913, device908, and resistor917function as a voltage-to-current converter circuit configured to translate the sample voltage of node929to a current flowing in device908. Device909is coupled between input power supply node107and node932, and is controlled by a voltage level of node931. Device910is coupled between input power supply node107and node931, and is controlled by the voltage level of node931. Devices909and910function as a current mirror circuit configured to replicate the current flowing in device910in device909to generate current Ipeak816. Switches921and922may be implemented using one or more transistors or other suitable switching devices. For example, in some embodiments, switches921and922may be implemented using pass-gate or other similar circuits. Resistors916and917may be implemented using polysilicon, metal or any other suitable material available in a semiconductor manufacturing process. Amplifiers912and913may be implemented as differential amplifiers or any other suitable amplifier circuit configured to generate an output signal whose voltage level is based on the respective voltage levels of two or more input signals. Devices901-905and devices909-910may be implemented as p-channel MOSFETs, FinFETs, GAAFETs, or any other suitable transconductance devices. Devices906-908may be implemented as n-channel MOSFETs, FinFETs, GAAFETs, or any other suitable transconductance devices. Turning toFIG.10, a block diagram of an embodiment of signal converter circuit807is depicted. As illustrated, signal converter circuit807included current source1001, amplifiers1002and1003, device1004, capacitors1005and1006, resistor1007, and switches1008-1010. Current source1001is coupled between input power supply node107and node1020, and is configured to source a bias current into node1020. Current source919may, in some embodiments, be implemented as biased transconductance device (e.g., a p-channel MOSFET, FinFET, or GAAFET), part of a current mirror circuit, or any other suitable circuit configured to provide a constant current across a range of output voltage levels. Switch1008is configured to couple node1020to ground supply node205in response to an activation of switch signal1018. Switch1009is configured to couple node1020to node1012in response to an activation of switch signal1019. In a similar fashion, switch1010is configured to couple node1012to ground supply node205in response to an activation of switch signal1018. It is noted that switch signals1018and1019have opposite polarity. In various embodiments, switch signal1018and switch signal1019may be based on a signal whose transitions in time are to be converted to an analog voltage level. For example, switch signals1018and1019may be based on SWon815as depicted inFIG.8. Capacitor1005is coupled between node1012and ground supply node205. During periods of time when switch1009is closed and switch1010is open, capacitor1005is charged by the bias current generated by current source1001performing an integration function. Capacitor1005may, in various embodiments, be implemented using a MOM structure, a MIM structure, or any other suitable capacitor structure available on a semiconductor manufacturing process. Amplifier1002is coupled between nodes1012and1013, and is configured to buffer a voltage level of node1012onto node1013. Switch1011is coupled between node1013and1014and is configured to couple node1013to1014in response to activation of switch signal1019. Capacitor1006is coupled between node1014and ground supply node205. In various embodiments, the voltage level of node1013is stored on capacitor1006when switch1011is closed. Capacitor1006may, in various embodiments, be implemented using a MOM structure, a MIM structure, or any other suitable capacitor structure available on a semiconductor manufacturing process. Amplifier1003is configured to generate a voltage on node1015based on the respective voltage levels of nodes1014and1016. Device1004is coupled between node1017and node1016, and is controlled by a voltage level of node1015. Resistor1007is coupled between node1016and ground supply node205. In various embodiments, the combination of amplifier1003, device1004, and resistor1007function as a voltage-to-current converter circuit configured to translate the voltage level of node1014to a current flowing in device1004to generate output current1021whose value is based on the switching rate of switch signals1018and1019. It is noted that in some embodiments, output current1021may be passed through a resistor to generate a voltage whose value is based on the switching rate of switch signals1018and1019. Switches1008-1011may be implemented using one or more transistors or other suitable switching devices. For example, in some embodiments, switches1008-1011may be implemented using pass-gate or other similar circuits. Amplifiers1002and1003may be implemented as differential amplifiers or any other suitable comparator circuit configured to generate an output signal whose voltage level is based on the respective voltage levels of two or more input signals. Resistor1007may be implemented using polysilicon, metal or any other suitable material available in a semiconductor manufacturing process. A flow diagram depicting an embodiment of a method for operating a power converter circuit is illustrated inFIG.11A. The method, which may be applied to various power converter circuits, such as power converter circuit100, begins in block1101. The method includes sourcing, by a switching circuit using an input power supply, respective charge currents to a regulated power supply node during a plurality of active cycles. In various embodiments, the switching circuit is coupled to the regulated power supply node via an inductor (block1102). In various embodiments, sourcing a given charge current to the regulated power supply node includes coupling, by a switch device included in the switching circuit, a terminal of the inductor to the input power supply. The method further includes determining, by a control circuit, an average inductor current for a particular active cycle of the plurality of active cycles (block1103). In various embodiments, determining the average inductor current includes determining, by the control circuit, the average inductor current using a voltage level of the regulated power supply node, a peak current flowing in the inductor during the particular active cycle, and a duty cycle of a high-side switch included in the switching circuit, wherein the high-side switch is coupled between the input power supply node and the inductor. The method also includes performing, by the control circuit, a comparison of the average inductor current to a threshold value (block1104). In various embodiments, the method may further include comparing, by the control circuit, a skip voltage corresponding to the average inductor current to the threshold value, and activating a skip signal in response to determining the skip voltage is greater than the threshold value. The method further includes deactivating, by the control circuit and based on a result of the comparison of the average inductor current to the threshold value, the switching circuit for a different active cycle of the plurality of active cycles that is subsequent to the particular active cycle (block1105). In some embodiments, the method further includes deactivating the switching circuit in response to determining the skip signal has been activated. In some embodiments, the method may also include switching from a pulse width modulation (PWM) mode to a pulse frequency mode (PFM) in response to determining that a load current is at a first value, and switching from the PFM mode to the PWM mode in response to determining that the load current is at a second value greater than the first value. In other embodiment, the method may further include sensing a current flowing in the inductor and combining the current flowing in the inductor with a compensation current to generate a sense signal. The method may also include generating an error signal using a voltage level of the regulated power supply node and a reference voltage. In some embodiments, the method may further include performing a comparison of the sense signal to the error signal, and halting a given active cycle of the plurality of active cycles using a result of the comparison. The method concludes in block1106. FIG.11Bis a block diagram of another embodiment of operating a power converter circuit. The method performed inFIG.11Bmay be carried out by various embodiments of the circuitry discussed above. The method includes sourcing, by a includes sourcing, by a switching circuit using an input power supply, respective charge currents to a regulated power supply node during a plurality of active cycles. In various embodiments, the switching circuit is coupled to the regulated power supply node via an inductor (block1122). During operation, an on-time of the switching circuit and a discontinuous conduction mode (DCM) may be monitored as a basis for determining mode changes. The method thus further includes (while operating in the PWM mode) detecting DCM and a short minimum on-time, and in response thereto, causing the power converter to exit a PWM mode and enter a PFM/PSM mode (block1123). The method further includes detecting (while in the PFM/PSM mode), a long minimum on-time, and in response thereto, exiting the PFM/PSM mode and entering the PWM mode (block1124). A block diagram of a system-on-a-chip (SoC) is illustrated inFIG.12. In the illustrated embodiment, SoC1200includes power management circuit1201, processor circuit1202, input/output circuits1204, and memory circuit1203, each of which is coupled to power supply signal1205. In various embodiments, SoC1200may be configured for use in a desktop computer, server, or in a mobile computing application such as, e.g., a tablet, laptop computer, or wearable computing device. Power management circuit1201includes power converter circuit100which is configured to generate a regulated voltage level on power supply signal1205in order to provide power to processor circuit1202, input/output circuits1204, and memory circuit1203. Although power management circuit1201is depicted as including a single power converter circuit, in other embodiments, any suitable number of power converter circuits may be included in power management circuit1201, each configured to generate a regulated voltage level on a respective one of multiple internal power supply signals included in SoC1200. Processor circuit1202may, in various embodiments, be representative of a general-purpose processor that performs computational operations. For example, processor circuit1202may be a central processing unit (CPU) such as a microprocessor, a microcontroller, an application-specific integrated circuit (ASIC), or a field-programmable gate array (FPGA). Memory circuit1203may, in various embodiments, include any suitable type of memory such as a Dynamic Random-Access Memory (DRAM), a Static Random-Access Memory (SRAM), a Read-Only Memory (ROM), Electrically Erasable Programmable Read-only Memory (EEPROM), or a non-volatile memory, for example. It is noted that although a single memory circuit is illustrated inFIG.12, in other embodiments, any suitable number of memory circuits may be employed. Input/output circuits1204may be configured to coordinate data transfer between SoC1200and one or more peripheral devices. Such peripheral devices may include, without limitation, storage devices (e.g., magnetic or optical media-based storage devices including hard drives, tape drives, CD drives, DVD drives, etc.), audio processing subsystems, or any other suitable type of peripheral devices. In some embodiments, input/output circuits1204may be configured to implement a version of Universal Serial Bus (USB) protocol or IEEE 1394 (Firewire®) protocol. Input/output circuits1204may also be configured to coordinate data transfer between SoC1200and one or more devices (e.g., other computing systems or integrated circuits) coupled to SoC1200via a network. In one embodiment, input/output circuits1204may be configured to perform the data processing necessary to implement an Ethernet (IEEE 802.3) networking standard such as Gigabit Ethernet or 10-Gigabit Ethernet, for example, although it is contemplated that any suitable networking standard may be implemented. In some embodiments, input/output circuits1204may be configured to implement multiple discrete network interface ports. Turning now toFIG.13, various types of systems that may include any of the circuits, devices, or systems discussed above are illustrated. System or device1300, which may incorporate or otherwise utilize one or more of the techniques described herein, may be utilized in a wide range of areas. For example, system or device1300may be utilized as part of the hardware of systems such as a desktop computer1310, laptop computer1320, tablet computer1330, cellular or mobile phone1340, or television1350(or set-top box coupled to a television). Similarly, disclosed elements may be utilized in a wearable device1360, such as a smartwatch or a health-monitoring device. Smartwatches, in many embodiments, may implement a variety of different functions—for example, access to email, cellular service, calendar, health monitoring, etc. A wearable device may also be designed solely to perform health-monitoring functions, such as monitoring a user's vital signs, performing epidemiological functions such as contact tracing, providing communication to an emergency medical service, etc. Other types of devices are also contemplated, including devices worn on the neck, devices implantable in the human body, glasses or a helmet designed to provide computer-generated reality experiences such as those based on augmented and/or virtual reality, etc. System or device1300may also be used in various other contexts. For example, system or device1300may be utilized in the context of a server computer system, such as a dedicated server or on shared hardware that implements a cloud-based service1370. Still further, system or device1300may be implemented in a wide range of specialized everyday devices, including devices1380commonly found in the home such as refrigerators, thermostats, security cameras, etc. The interconnection of such devices is often referred to as the “Internet of Things” (IoT). Elements may also be implemented in various modes of transportation. For example, system or device1300could be employed in the control systems, guidance systems, entertainment systems, etc. of various types of vehicles1390. The applications illustrated inFIG.13are merely exemplary and are not intended to limit the potential future applications of disclosed systems or devices. Other example applications include, without limitation: portable gaming devices, music players, data storage devices, unmanned aerial vehicles, etc. FIG.14is a block diagram illustrating an example of a non-transitory computer-readable storage medium that stores circuit design information, according to some embodiments. In the illustrated embodiment, semiconductor fabrication system1420is configured to process design information1415stored on non-transitory computer-readable storage medium1410and fabricate integrated circuit1430based on design information1415. Non-transitory computer-readable storage medium1410, may comprise any of various appropriate types of memory devices or storage devices. Non-transitory computer-readable storage medium1410may be an installation medium, e.g., a CD-ROM, floppy disks, or tape device; a computer system memory or random-access memory such as DRAM, DDR RAM, SRAM, EDO RAM, Rambus RAM, etc.; a non-volatile memory such as Flash, magnetic media, e.g., a hard drive, or optical storage; registers, or other similar types of memory elements, etc. Non-transitory computer-readable storage medium1410may include other types of non-transitory memory as well or combinations thereof. Non-transitory computer-readable storage medium1410may include two or more memory mediums, which may reside in different locations, e.g., in different computer systems that are connected over a network. Design information1415may be specified using any of various appropriate computer languages, including hardware description languages such as, without limitation: VHDL, Verilog, SystemC, SystemVerilog, RHDL, M, MyHDL, etc. Design information1415may be usable by semiconductor fabrication system1420to fabricate at least a portion of integrated circuit1430. The format of design information1415may be recognized by at least one semiconductor fabrication system, such as semiconductor fabrication system1420, for example. In some embodiments, design information1415may include a netlist that specifies elements of a cell library, as well as their connectivity. One or more cell libraries used during logic synthesis of circuits included in integrated circuit1430may also be included in design information1415. Such cell libraries may include information indicative of device or transistor level netlists, mask design data, characterization data, and the like, of cells included in the cell library. Integrated circuit1430may, in various embodiments, include one or more custom macrocells, such as memories, analog or mixed-signal circuits, and the like. In such cases, design information1415may include information related to included macrocells. Such information may include, without limitation, schematics capture database, mask design data, behavioral models, and device or transistor level netlists. As used herein, mask design data may be formatted according to graphic data system (GDSII), or any other suitable format. Semiconductor fabrication system1420may include any of various appropriate elements configured to fabricate integrated circuits. This may include, for example, elements for depositing semiconductor materials (e.g., on a wafer, which may include masking), removing materials, altering the shape of deposited materials, modifying materials (e.g., by doping materials or modifying dielectric constants using ultraviolet processing), etc. Semiconductor fabrication system1420may also be configured to perform various testing of fabricated circuits for correct operation. In various embodiments, integrated circuit1430is configured to operate according to a circuit design specified by design information1415, which may include performing any of the functionality described herein. For example, integrated circuit1430may include any of various elements shown or described herein. Further, integrated circuit1430may be configured to perform various functions described herein in conjunction with other components. Further, the functionality described herein may be performed by multiple connected integrated circuits. As used herein, a phrase of the form “design information that specifies a design of a circuit configured to . . . ” does not imply that the circuit in question must be fabricated in order for the element to be met. Rather, this phrase indicates that the design information describes a circuit that, upon being fabricated, will be configured to perform the indicated actions or will include the specified components. The present disclosure includes references to “an “embodiment” or groups of “embodiments” (e.g., “some embodiments” or “various embodiments”). Embodiments are different implementations or instances of the disclosed concepts. References to “an embodiment,” “one embodiment,” “a particular embodiment,” and the like do not necessarily refer to the same embodiment. A large number of possible embodiments are contemplated, including those specifically disclosed, as well as modifications or alternatives that fall within the spirit or scope of the disclosure. This disclosure may discuss potential advantages that may arise from the disclosed embodiments. Not all implementations of these embodiments will necessarily manifest any or all of the potential advantages. Whether an advantage is realized for a particular implementation depends on many factors, some of which are outside the scope of this disclosure. In fact, there are a number of reasons why an implementation that falls within the scope of the claims might not exhibit some or all of any disclosed advantages. For example, a particular implementation might include other circuitry outside the scope of the disclosure that, in conjunction with one of the disclosed embodiments, negates or diminishes one or more the disclosed advantages. Furthermore, suboptimal design execution of a particular implementation (e.g., implementation techniques or tools) could also negate or diminish disclosed advantages. Even assuming a skilled implementation, realization of advantages may still depend upon other factors such as the environmental circumstances in which the implementation is deployed. For example, inputs supplied to a particular implementation may prevent one or more problems addressed in this disclosure from arising on a particular occasion, with the result that the benefit of its solution may not be realized. Given the existence of possible factors external to this disclosure, it is expressly intended that any potential advantages described herein are not to be construed as claim limitations that must be met to demonstrate infringement. Rather, identification of such potential advantages is intended to illustrate the type(s) of improvement available to designers having the benefit of this disclosure. That such advantages are described permissively (e.g., stating that a particular advantage “may arise”) is not intended to convey doubt about whether such advantages can in fact be realized, but rather to recognize the technical reality that realization of such advantages often depends on additional factors. Unless stated otherwise, embodiments are non-limiting. That is, the disclosed embodiments are not intended to limit the scope of claims that are drafted based on this disclosure, even where only a single example is described with respect to a particular feature. The disclosed embodiments are intended to be illustrative rather than restrictive, absent any statements in the disclosure to the contrary. The application is thus intended to permit claims covering disclosed embodiments, as well as such alternatives, modifications, and equivalents that would be apparent to a person skilled in the art having the benefit of this disclosure. For example, features in this application may be combined in any suitable manner. Accordingly, new claims may be formulated during prosecution of this application (or an application claiming priority thereto) to any such combination of features. In particular, with reference to the appended claims, features from dependent claims may be combined with those of other dependent claims where appropriate, including claims that depend from other independent claims. Similarly, features from respective independent claims may be combined where appropriate. Accordingly, while the appended dependent claims may be drafted such that each depends on a single other claim, additional dependencies are also contemplated. Any combinations of features in the dependent claims that are consistent with this disclosure are contemplated and may be claimed in this or another application. In short, combinations are not limited to those specifically enumerated in the appended claims. Where appropriate, it is also contemplated that claims drafted in one format or statutory type (e.g., apparatus) are intended to support corresponding claims of another format or statutory type (e.g., method). Because this disclosure is a legal document, various terms and phrases may be subject to administrative and judicial interpretation. Public notice is hereby given that the following paragraphs, as well as definitions provided throughout the disclosure, are to be used in determining how to interpret claims that are drafted based on this disclosure. References to a singular form of an item (i.e., a noun or noun phrase preceded by “a,” “an,” or “the”) are, unless context clearly dictates otherwise, intended to mean “one or more.” Reference to “an item” in a claim thus does not, without accompanying context, preclude additional instances of the item. A “plurality” of items refers to a set of two or more of the items. The word “may” is used herein in a permissive sense (i.e., having the potential to, being able to) and not in a mandatory sense (i.e., must). The terms “comprising” and “including,” and forms thereof, are open-ended and mean “including, but not limited to.” When the term “or” is used in this disclosure with respect to a list of options, it will generally be understood to be used in the inclusive sense unless the context provides otherwise. Thus, a recitation of “x or y” is equivalent to “x or y, or both,” and thus covers 1) x but not y, 2) y but not x, and 3) both x and y. On the other hand, a phrase such as “either x or y, but not both” makes clear that “or” is being used in the exclusive sense. A recitation of “w, x, y, or z, or any combination thereof” or “at least one of . . . w, x, y, and z” is intended to cover all possibilities involving a single element up to the total number of elements in the set. For example, given the set [w, x, y, z], these phrasings cover any single element of the set (e.g., w but not x, y, or z), any two elements (e.g., w and x, but not y or z), any three elements (e.g., w, x, and y, but not z), and all four elements. The phrase “at least one of . . . w, x, y, and z” thus refers to at least one element of the set [w, x, y, z], thereby covering all possible combinations in this list of elements. This phrase is not to be interpreted to require that there is at least one instance of w, at least one instance of x, at least one instance of y, and at least one instance of z. Various “labels” may precede nouns or noun phrases in this disclosure. Unless context provides otherwise, different labels used for a feature (e.g., “first circuit,” “second circuit,” “particular circuit,” “given circuit,” etc.) refer to different instances of the feature. Additionally, the labels “first,” “second,” and “third” when applied to a feature do not imply any type of ordering (e.g., spatial, temporal, logical, etc.), unless stated otherwise. The phrase “based on” is used to describe one or more factors that affect a determination. This term does not foreclose the possibility that additional factors may affect the determination. That is, a determination may be solely based on specified factors or based on the specified factors as well as other, unspecified factors. Consider the phrase “determine A based on B.” This phrase specifies that B is a factor that is used to determine A or that affects the determination of A. This phrase does not foreclose that the determination of A may also be based on some other factor, such as C. This phrase is also intended to cover an embodiment in which A is determined based solely on B. As used herein, the phrase “based on” is synonymous with the phrase “based at least in part on.” The phrases “in response to” and “responsive to” describe one or more factors that trigger an effect. This phrase does not foreclose the possibility that additional factors may affect or otherwise trigger the effect, either jointly with the specified factors or independent from the specified factors. That is, an effect may be solely in response to those factors, or may be in response to the specified factors as well as other, unspecified factors. Consider the phrase “perform A in response to B.” This phrase specifies that B is a factor that triggers the performance of A, or that triggers a particular result for A. This phrase does not foreclose that performing A may also be in response to some other factor, such as C. This phrase also does not foreclose that performing A may be jointly in response to B and C. This phrase is also intended to cover an embodiment in which A is performed solely in response to B. As used herein, the phrase “responsive to” is synonymous with the phrase “responsive at least in part to.” Similarly, the phrase “in response to” is synonymous with the phrase “at least in part in response to.” Within this disclosure, different entities (which may variously be referred to as “units,” “circuits,” other components, etc.) may be described or claimed as “configured” to perform one or more tasks or operations. This formulation [entity] configured to [perform one or more tasks] is used herein to refer to structure (i.e., something physical). More specifically, this formulation is used to indicate that this structure is arranged to perform the one or more tasks during operation. A structure can be said to be “configured to” perform some tasks even if the structure is not currently being operated. Thus, an entity described or recited as being “configured to” perform some tasks refers to something physical, such as a device, circuit, a system having a processor unit and a memory storing program instructions executable to implement the task, etc. This phrase is not used herein to refer to something intangible. In some cases, various units/circuits/components may be described herein as performing a set of tasks or operations. It is understood that those entities are “configured to” perform those tasks/operations, even if not specifically noted. The term “configured to” is not intended to mean “configurable to,” An unprogrammed FPGA, for example, would not be considered to be “configured to” perform a particular function. This unprogrammed FPGA may be “configurable to” perform that function, however. After appropriate programming, the FPGA may then be said to be “configured to” perform the particular function. For purposes of U.S. patent applications based on this disclosure, reciting in a claim that a structure is “configured to” perform one or more tasks is expressly intended not to invoke 35 U.S.C. § 112(f) for that claim element. Should Applicant wish to invoke Section 112(f) during prosecution of a United States patent application based on this disclosure, it will recite claim elements using the “means for” [performing a function] construct. Different “circuits” may be described in this disclosure. These circuits or “circuitry” constitute hardware that includes various types of circuit elements, such as combinatorial logic, clocked storage devices (e.g., flip-flops, registers, latches, etc.), finite state machines, memory e.g., random-access memory, embedded dynamic random-access memory), programmable logic arrays, and so on. Circuitry may be custom designed, or taken from standard libraries. In various implementations, circuitry can, as appropriate, include digital components, analog components, or a combination of both. Certain types of circuits may be commonly referred to as “units” (e.g., a decode unit, an arithmetic logic unit (ALU), functional unit, memory management unit (MMU), etc.). Such units also refer to circuits or circuitry. The disclosed circuits/units/components and other elements illustrated in the drawings and described herein thus include hardware elements such as those described in the preceding paragraph. In many instances, the internal arrangement of hardware elements within a particular circuit may be specified by describing the function of that circuit. For example, a particular “decode unit” may be described as performing the function of “processing an opcode of an instruction and routing that instruction to one or more of a plurality of functional units,” which means that the decode unit is “configured to” perform this function. This specification of function is sufficient, to those skilled in the computer arts, to connote a set of possible structures for the circuit. In various embodiments, as discussed in the preceding paragraph, circuits, units, and other elements may be defined by the functions or operations that they are configured to implement. The arrangement of such circuits/units/components with respect to each other and the manner in which they interact form a microarchitectural definition of the hardware that is ultimately manufactured in an integrated circuit or programmed into an FPGA to form a physical implementation of the microarchitectural definition. Thus, the microarchitectural definition is recognized by those of skill in the art as structure from which many physical implementations may be derived, all of which fall into the broader structure described by the microarchitectural definition. That is, a skilled artisan presented with the microarchitectural definition supplied in accordance with this disclosure may, without undue experimentation and with the application of ordinary skill, implement the structure by coding the description of the circuits/units/components in a hardware description language (HDL) such as Verilog or VHDL. The HDL description is often expressed in a fashion that may appear to be functional. But to those of skill in the art in this field, this HDL description is the manner that is used to transform the structure of a circuit, unit, or component to the next level of implementational detail. Such an HDL description may take the form of behavioral code (which is typically not synthesizable), register transfer language (RTL) code (which, in contrast to behavioral code, is typically synthesizable), or structural code (e.g., a netlist specifying logic gates and their connectivity). The HDL description may subsequently be synthesized against a library of cells designed for a given integrated circuit fabrication technology, and may be modified for timing, power, and other reasons to result in a final design database that is transmitted to a foundry to generate masks and ultimately produce the integrated circuit. Some hardware circuits or portions thereof may also be custom-designed in a schematic editor and captured into the integrated circuit design along with synthesized circuitry. The integrated circuits may include transistors and other circuit elements (e.g., passive elements such as capacitors, resistors, inductors, etc.) and interconnect between the transistors and circuit elements. Some embodiments may implement multiple integrated circuits coupled together to implement the hardware circuits, and/or discrete elements may be used in some embodiments. Alternatively, the HDL design may be synthesized to a programmable logic array such as a field programmable gate array (FPGA) and may be implemented in the FPGA. This decoupling between the design of a group of circuits and the subsequent low-level implementation of these circuits commonly results in the scenario in which the circuit or logic designer never specifies a particular set of structures for the low-level implementation beyond a description of what the circuit is configured to do, as this process is performed at a different stage of the circuit implementation process. The fact that many different low-level combinations of circuit elements may be used to implement the same specification of a circuit results in a large number of equivalent structures for that circuit. As noted, these low-level circuit implementations may vary according to changes in the fabrication technology, the foundry selected to manufacture the integrated circuit, the library of cells provided for a particular project, etc. In many cases, the choices made by different design tools or methodologies to produce these different implementations may be arbitrary. Moreover, it is common for a single implementation of a particular functional specification of a circuit to include, for a given embodiment, a large number of devices (e.g., millions of transistors). Accordingly, the sheer volume of this information makes it impractical to provide a full recitation of the low-level structure used to implement a single embodiment, let alone the vast array of equivalent possible implementations. For this reason, the present disclosure describes structure of circuits using the functional shorthand commonly employed in the industry. Numerous variations and modifications will become apparent to those skilled in the art once the above disclosure is fully appreciated. It is intended that the following claims be interpreted to embrace all such variations and modifications. | 70,237 |
11863059 | DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS In the following description, various aspects of the invention will be described. For the purposes of explanation, specific details are set forth in order to provide a thorough understanding of the invention. It will be apparent to one skilled in the art that there are other embodiments of the invention that differ in details without affecting the essential nature thereof. Therefore, the invention is not limited by that which is illustrated in the figures and described in the specification, but only as indicated in the accompanying claims, with the proper scope determined only by the broadest interpretation of said claims. In some cases, for clarity or conciseness, individual elements of the invention are discussed separately. Nonetheless, any combination of elements of the invention that is not self-contradictory is considered by the inventors to be within the scope of the invention. In addition, some embodiments of the invention are described in terms of “comprising” a set of components, that is, the embodiment includes at least the listed components. Any embodiment of the invention described herein as “comprising” a set of components is considered by the inventor to include within its scope embodiments of the invention that “consist of” the listed components, that is, embodiments that include those components and no others. As used herein, the term “battery” is used generically to refer to a device capable of generating electricity from chemical reactions. Unless specifically stated otherwise, the term as used herein includes both single-cell and multiple-cell batteries. As used herein, the abbreviation “VPSU” stands for “Versatile Power Stack Unit.” As used herein, the abbreviation “EV” stands for “Electric Vehicle.” As used herein, the abbreviation “PWM” stands for “Pulse-Width Modulation.” As used herein, within reference numbers, the italicized letter “n” is used to indicate that each of the n identical units incorporates a separate instance of the component. As a non-limiting example, in a system with three units, the notation Xn indicates that one of component X is present in each of the first unit (Xa), the second unit as (Xb), and in the third unit (Xc). As used herein, with reference to numerical quantities, the term “about” refers to a tolerance of ±20% about the nominal value. A. VPSU—General Description Disclosed herein is the Versatile Power Stack Unit (VPSU), a novel means and method for controlling the power source of an electric motor, especially a motor controller driver for the electric motor of an EV. As will be explained below, the VPSU provides high efficiency while limiting the ripple current experienced by the batteries, but avoids the difficulties of systems and methods known in the art. In commercially available EVs, the battery pack frequently comprises a large number of batteries (e.g. the battery pack of the Tesla 3 comprises 2976 cells, 31 columns of 96 batteries). The manufacturer connects occasionally a high-power DC/DC upconverter; in general, since the system operates at constant voltage, there is no DC/DC converter. The ripple capacitor residing in the motor controller is necessarily large, and the converter is also large, expensive, and uneconomical. In contrast, in the VPSU disclosed herein, a relatively small battery pack is used (20 in the typical non-limiting embodiments illustrated in the examples given below). The VPSU thus uses small voltages and currents, making it much easier to store energy and to change the voltage. Each VPSU can be treated as an independent unit, a “brick” in the construction of a larger system, and thereby provides the flexibility to construct a system of any desired size. Reference is now made toFIG.2, which is a schematic diagram of one non-limiting embodiment1000of the VPSU disclosed herein. The VPSU is connected to a battery pack33that may contain any number of battery stacks, each of which comprises a plurality of batteries connected in series. In some embodiments, the battery pack includes more than one such serially connected stack of batteries; in these embodiments, the stacks of serially connected batteries are connected in parallel one to the other. For purposes of illustration, the embodiment illustrated inFIG.2comprises one stack; this configuration is not intended to be limiting in any way. The VPSU is connected to the battery pack by a switch. The switch may be mechanical (a non-limiting example is a relay) or electrical (non-limiting examples include power transistors, such as MOSFET, SiC, GaN, or IGBT). In preferred embodiments of the invention, the switch is a Bidirectional Short Circuit Protected Multipole Switch (BDPS), described in detail below. In some non-limiting embodiments of the invention, the switch is single pole, single throw. In some other non-limiting embodiments of the invention, the switch is double pole, single throw. In some other non-limiting embodiments of the invention, the switch is triple pole, single throw. Since each VPSU is connected separately to the battery system, all of the switches other than voltage out switch can be rated to low voltage, thereby yielding low loss. In typical non-limiting embodiments of the invention, the switches are rated to 60 V; the voltage out switch is rated according to the maximum output of the serially connected VPSUs, which can be up to hundreds of volts. The VPSU output voltage is generated by a DC/DC voltage converter43, preferably a buck-boost converter, which is connected via a switch during discharge time in parallel with the battery stack. Any commercially available DC/DC converter that is appropriate for the desired application can be used. For example, in the embodiment shown inFIG.2, the nominal input voltage is 37 V, and the nominal output voltage is 37 V as well. Examples of commercially available buck-boost converters that can be used in this system include the Texas Instruments model LM5176, which can achieve efficiencies of 97.7% at 50 V, 97.5% at 25 V, and 99.1% at 37 V. Another commercially available buck-boost converter that can be used in this system is the Analog Devices model LT8228, which has similar efficiencies and a 4.5-100 V input range. In preferred embodiments, the negative input of the DC/DC voltage converter is in electrical connection with a negative terminal connector configured to accept the negative terminal of a battery. The battery pack is connected to lower column switch31and upper column switch34via charge/discharge switch35. The lower column switch is connected to ground signal or, in embodiments in which the system comprises multiple VPSU units, to the upper column switch of a second VPSU during high voltage charging or to the voltage output switch45during battery discharge (e.g. while the vehicle is being driven). In some embodiments of the invention, the VPSU additionally includes an optional battery management system (BMS)32. The BMS is configured to monitor the status of each battery cell, controls the current during high voltage charging, and maintains voltage equality among the battery cells by bypassing the current through the resistor and transistor when the batteries are being charged. When the batteries are being charged, the charger is connected to the VPSU via the upper column switch. Charge/discharge switch35is located between the battery stack and upper column switch34, and controls the current direction when the batteries are being charged in low voltage or high voltage mode. In preferred embodiments, battery current is monitored by battery current sensor36, which is typically located between the positive terminal of the battery stack and the charge/discharge switch. The battery current sensor can be any suitable device known in the art; non-limiting examples include magnetic sensors and shunt resistor measuring devices (low or high side). The negative terminal of the lowest battery in the stack is connected to the DC/DC converter negative signal path and is the common GND signal for the DC/DC. The positive terminal of the uppermost battery in the stack is connected to the DC/DC converter signal path through charge/discharge switch35and DC/DC switch38during the discharge. Data and management signals are provided to the VPSU by a data connection to the motor controller or EV computer by a galvanically isolated interface49that contains the VPSU address in the battery system. When the motor controller or EV main processor calculates that the voltage needs to be changed, then a data packet with the new VPSU arrangement or the new level of voltage output is sent to the micro-controller unit (MCU)41of the VPSU via a communication channel50. Each data packet from the motor controller or EV main processor contains information about the VPSU voltage; the MCU is described in detail below. The MCU of the VPSU follows the command and adjusts the feedback value (reference voltage) of the DC/DC converter. In such case the DC/DC output voltage is varied following the commands. In order to enable a fast response time, the MCU changes the analog feedback value via a D/A output signal. This procedure takes on the order of microseconds. Micro-controller unit (MCU)41is the main control of the VPSU. The MCU computing power must be sufficient to allow a dynamic charge current loop with a frequency of >100 kHz while maintaining low power consumption, preferably less than 100 mW. In preferred embodiments of the invention, the MCU comprises a 32-bit microprocessor and an at least 10 bit digital to analog (D/A) converter. In especially preferred embodiments of the invention, the MCU comprises an at least 12 bit D/A converter. The MCU comprises at least 8 channels of 12 bits of fast analog to digital (A/D) conversion. The MCU comprises at least 64 kb of flash memory for program and long-term nonvolatile data storage and at least 24 kb RAM for computations. The MCU is in data connection with the main processor and/or EV motor controller, and controls the voltage output of the VPSU and maintains the switch combination for the VPSU mode, as described below. In preferred embodiments of the invention, the MCU is powered by a power supply53in electrical connection with the battery stack. The MCU analog input reads analog signals in the VPSU such as the total battery current determined by battery current sensor36. As described in detail below, the MCU is programmed to receive and to send a variety of instructions and signals to various points in the motor control system. For example, the MCU can change the voltage set point of the DC/DC converter and send a “shut down” (SD) signal to the DC/DC converter as necessary. Depending on the details of the construction of the converter, other signals can be sent by the converter to the MCU. The MCU switch control48sets the VPSU switches in order to maintain the correct battery arrangement. Voltage output switch45connects the VPSU to the battery output terminal or, in cases in which more than one VPSU is used, to the lower column switch of the next VPSU. In preferred embodiments of the invention, the voltage output switch is a BDPS. A low-pass filter420is connected between the DC/DC converter input and the positive terminal of the battery stack via at least one switch. In many DC/DC applications, an input capacitor acts as a DC/DC input voltage stabilizer and an inductor is used to reduce RFI/EMI radiation arising from the high speed DC/DC current switching. In embodiments that incorporate an input capacitor, this filter minimizes PWM signals that create ripple current through the batteries. In the non-limiting embodiment shown inFIG.2, the low-pass filter comprises a charge pool capacitor (Cpool)54, a filter inductor (Lfilter)42and a filter capacitor (Cfilter)52. Cpoolis connected in parallel with the battery and acts as a charge pool that supplies current during the DC/DC PWM ON state to limit the current drawn by the battery, with Lfilterconnected in series with the positive terminal of the battery stack via the DC/DC switch38, the charge/discharge switch35, and Cfilterconnected to the voltage input of the DC/DC converter. Cfilterprovides a capacitative connection between the positive DC/DC terminal and DC/DC ground terminal in order to smooth the DC/DC input ripple current. When the PWM is in the OFF state (current is 0), Cfilteris charged by Lfilter, and Cpoolis charged by the battery. Cfilteris charged from the inductor current, which continues due to the stored magnetic field. When the PWM ON state signal is provided, Cpool, which is in parallel with the battery, is discharged by the inductor Lfilter. Cfilterdischarges into the DC/DC converter and Lfiltercreates a magnetic field. This activity acts to limit current ripple from the battery. We note that the battery internal resistance is ˜100 times higher than the impedance of the capacitors, and essentially isolates Cpoolfrom the positive side of the battery. In preferred embodiments of the invention, a positive terminal connector configured to accept a positive terminal of a battery is in electrical connection with the positive input of said DC/DC voltage converter via the converter's DC/DC input switch and the low-pass filter. It is important to understand that the VPSU is not simply a DC/DC converter; it is a system that converts the motor controller PWM frequency, typically 5-30 kHz, to a frequency of 350-500 kHz, the internal operating frequency of the DC/DC converter, which incidentally is also a PWM signal. In the VPSU, the batteries no longer experience the PWM of the motor, but rather only the order of magnitude higher PWM frequency of the DC/DC converter, which enables the use of much smaller coils and capacitors than are used in EV battery systems known in the art. Because each individual VPSU is relatively small, and uses relatively small capacitors and inductor, the voltage does not reach high values, and the coils do not saturate due the motor high current. B. Bidirectional Short-Circuit Protected Multipole Switch (BDPS) Reference is now made toFIG.3, which presents a schematic diagram of one non-limiting embodiment1100of a BDPS. The BDPS is designed to enable high-efficiency use of the batteries in the system. In addition to serving as the switch for the battery system voltage output, in preferred embodiments of the invention, the system operational mode is set by a BDPS. Each BDPS is a multipole, single-throw negative blocking current, transistor. The BDPS separates a command level and a switching level. The two levels are separated by charge pump95for a low voltage system (typically 100 V or less), shunt100, and high-side shunt amplifier101. The high-side shunt amplifier has a wide common mode signal range. N gate drivers96(N≥1) accept command level signals, and each gate driver generates switching level signals in order to drive one of N transistor gates99. In the non-limiting embodiment illustrated inFIG.3, N=3. If fully galvanic separation is needed, then the charge pump comprises a transformer and the current sense is done by an isolated amplifier or magnetic current sensor. The charge pump is operated with high-speed oscillator circuitry to generate a VGS voltage of >10 V. The current for the charge pump is low, typically about 20 μA; due to the high speed of the oscillator, the time from the issuing of the command to the VGS generation by the gate driver is less than 5 μs. When ON(n) command is asserted92, the charge pump starts to operate immediately, but a low-pass filter94delays the signal for about 5 μs. After this delay, the ON(n) signal propagates to the gate driver to open transistor n on the normally open (N.O.) side of the BDPS. When one of the gate drivers96is asserted, a diode connected as a common cathode97opens the gate of transistor98located on the common side of the BDPS. This single transistor TR1on the common side reduces the number of components and increases the reliability of the BDPS relative to semiconductor switches known in the art. The current is measured at shunt resistor100. A high-speed common mode high side amplifier101sends the current to the MCU41of the VPSU for control and management. If a short circuit were to occur, increasing the current above a preprogrammed level, then a comparator triggers a memory element flipflop, shutting down transistor n. Even if transistor n is shut down, the remaining N.O. transistors can continue to operate normally. As an example of the efficiency of the BDPS, a typical commercially available transistor (e.g. ON-Semiconductor Corporation model NTMTS0D7N06CLTXG) has a drain to source resistance of 680 μΩ. The BDPS comprises two connected transistors and a 1 mΩ shunt resistor for a total resistance of 2.36 mΩ. In a typical embodiment of the invention, the VPSU comprises two BDPSs for two stacks, each of which comprises 10 batteries. If each battery has an internal resistance of 45 mΩ, then the total connection efficiency of the BDPS is 99%(=1-2.3645(102)). C. Discharge Mode The VPSU has two operational modes, “discharge” and “charge.” The discharge mode is the normal state when the batteries are in use, e.g. while the EV is moving under battery power. The MCU dynamically sets the output voltage of the DC/DC converter according to the current needs of the output device (e.g. the motor controller of the electric motor). For example, in an exemplary non-limiting embodiment in which the battery stack comprises ten 3.7 V batteries, the output voltage from the DC/DC converter can be set at any value from 25 and 50 volts. The simultaneous use of multiple VPSUs will be described in detail below. One main advantage of the use of the MCU and DC/DC converter to set the output voltage dynamically in a linear mode is that by raising or lowering the output voltage as needed, the PWM duty cycle to the motor winding can be kept close to 50%, thereby achieving optimal DC/DC efficiency and lowering high harmonics produced by the motor driver and therefore reducing RFI/EMI interference and lowering or the eliminating ripple current. In discharge mode, the MCU (41) obtains an instruction to set the voltage from the motor controller or the EV's main processor. From this instruction, the MCU changes the voltage set point46of the DC/DC converter. The MCU can also send a “shut down” (SD) signal to the DC/DC converter as necessary. In some embodiments of the invention, other signals are sent by the MCU to the DC/DC converter, depending on the details of the construction of the converter. Concurrently with the setting of the DC/DC converter voltage set point, MCU switch control48sets the VPSU switches in order to maintain the correct total battery arrangement. MCU analog input40reads analog signals in the VPSU, for example, the battery current as determined by battery current sensor36. The current from the DC/DC converter is measured by a converter current sensor44that in typical embodiments is constructed similarly to the battery current sensor. The MCU comprises means for measuring the battery parameters such as voltage and temperature. This data is collected by the MCU and transmitted to the motor controller and/or EV main processor via the MCU—processor data connection described above in order to manage the power and keep the PWM signals close to a 50% duty cycle. In embodiments of the invention in which the VPSU comprises a BMS, the BMS system32communicates the battery status to the MCU, and in some embodiments, to the motor controller and/or EV main processor as well. D. Charge Mode When the batteries are being charged, the VPSU is placed in the charge mode. Under typical conditions of use, the VPSU will be placed in the charging state approximately daily. The system is connected to an external source of electricity in order to recharge the batteries. In principle, the batteries can be charged while remaining in serial connection. This method requires a high-voltage charger and consequently can be hazardous. One advantage of the VPSU architecture is that it permits a low-voltage charge mode, in which the battery charging is performed via the VPSU DC/DC converter. In such an arrangement, the VPSU DC/DC converter controls the charging of the local VPSU batteries. This mode thereby enables the use of an external low voltage power source which, in preferred embodiments, supplies a voltage of ≤50 V, for battery recharging. In low voltage charging mode, the charging process is controlled by the VPSU rather than by an external charger. In this mode, the voltage adjust set of the MCU sets the voltage necessary to charge the batteries connected to the VPSU, while the SD signal controls the power output bridge of the DC/DC converter to establish a dynamic current loop. This current loop is based on the current measurement, voltage adjust, battery temperature monitoring, and the DC/DC bridge output on-off sequence. The requirement that the MCU be able to perform all of these functions in tandem determines the limit, described above, that the MCU must have computing power sufficient to allow a dynamic charge current loop with a frequency of >100 kHz. The control of the charging process by the VPSU can be thought of in terms of a parameter α=ΔV/ΔI where V and I are the charge voltage and current, respectively. The VPSU topology then allows measurement of and response to the battery's “ionization factor” Δα/ΔT, where T is the battery temperature. This parameter is called the battery's ionization factor because the temperature rises after the battery ionization has reached its maximum value. This ionization factor is used as an input to a linear feedback loop in order to mitigate the temperature rise during the charging process, thereby improving the performance and extending the lifetime of the batteries; see, for example, Ma, S.; Jiang, M.; Tao, P.; Song, C.; Wu, J.; Wang, J.; Deng, T.; Shang, W. “Temperature Effect and Thermal Impact in Lithium-Ion Batteries: a Review,”Prog. Nat. Sci.-Mater.2018, 28, 653. E. Multiple VPSUs Another advantage of the VPSU is that multiple VPSUs can be combined in a single apparatus. Reference is now made toFIG.4A, which illustrates schematically a non-limiting embodiment1200of the invention in which two VPSUs (lower VPSU200A and upper VPSU200B) are connected to form a “stack.” The lower column switch31aof the lower VPSU is connected to the 0 V (power ground) signal. The charge/discharge switch35aof the lower VPSU and the DC/DC switch38aswitchably connect the battery stack to the positive (+) power input of the DC/DC converter43aof the lower VPSU. The negative (−) power input of the DC/DC converter of the lower VPSU is continuously connected to the battery stack33a. The upper column switch34aof the lower VPSU is disconnected, and the voltage output switch45aof the DC/DC converter of the lower VPSU is connected to the lower column switch31bof the upper VPSU. The charge/discharge switch35band DC/DC switch38bof the upper VPSU switchably connect the battery stack33battached to the DC/DC converter43bof the upper VPSU, while the negative power input of the DC/DC converter of the upper VPSU is continuously connected to the battery stack connected to the upper VPSU. The voltage out switch45bof the DC/DC converter of the upper VPSU is switchably connected to positive terminal70of the motor controller, while negative terminal71of the motor controller is connected to ground. While the above example describes an embodiment in which two VPSU units are connected, one of ordinary skill in the art will readily understand that any number of additional units can be connected in an analogous manner. If V0to Vmaxrepresents the output voltage range of a particular VPSU, then the voltage range for n VPSU units connected as described above will be V0to ΣnVmax. As a non-limiting example, for a VPSU comprising the DC/DC converters described above and connected to ten 3.7 V batteries connected in series, the output voltage range will be 25-50 V, a stack of four such VPSU units (220A-D), as shown in the non-limiting embodiment1210illustrated inFIG.4B, can generate linear voltage over the entire range of 25-200 V, etc. As was explained above, for electric motors, when the speed of the motor increases, the back EMF, which generates voltage opposing the motor driver voltage, increases. thereby reducing the torque. The wide linear voltage range made possible by the multiple VPSU arrangement allows the motor controller to increase the applied voltage when the motor speed is increased. In many cases, this large voltage range can eliminate any need for mechanical gear to provide the torque needs of the motor. When the torque required from the electric motor is high, a combination of multiple VPSUs can be connected in parallel in order to enable high current output. Reference is now made toFIG.4C, which illustrates a non-limiting embodiment1220in which multiple (four in this case) VPSUs (230A-D) are connected in parallel. Multiple VPSUs can be arranged in serial, in parallel, or in any combination thereof. F. Charging Modes for Multiple-VPSU Systems In contrast to systems known in the art that require 400V-600V grids and expensive devices, the VPSU arrangement allows charging over a much lower voltage range of 8-100 VDC, thereby simplifying the system and increasing its safety. As with the case of a single VPSU, there are two charging configurations for multiple-VPSU systems, high voltage and low voltage, as well. During the charging process, the VPSUs are controlled by the EV host computer, with the motor control acting as a data communication bridge. In high voltage mode, the VPSUs are connected in series with the upper column switch of one VPSU connected to the lower column switch; the charging is then done according to the battery manufacturer's specifications. Reference is now made toFIG.5, which shows schematically a typical arrangement for high-voltage charging of a non-limiting embodiment1300of a system comprising plurality of VPSUs (three in this case,240A-C). The lower column switch of the first VPSU is connected to ground, while each succeeding lower column switch is connected to the upper column switch of the preceding VPSU. The upper column switch of the final VPSU is connected to an external high voltage charger. The charge/discharge switches are set to the charge position, and the DC/DC input switches are opened. In this configuration, the charge current flows from the high voltage charger to all of the batteries in series. The charging process is controlled by the charger and, in embodiments in which they are present, by the BMS of each VPSU, which acts to equalize the charge and current on the batteries. In low voltage charge mode, the VPSUs are connected in parallel. The upper column switch of each VPSU is connected to a low voltage (≤50V) charger. In preferred embodiments of the invention, a single low voltage charger is used for the entire system. The DC/DC converter can act as a step-up converter if the input charging voltage is insufficient to charge the batteries; for typical DC/DC converters, the minimum input voltage is 8 VDC. The voltage and current output are controlled via the MCU, the DC/DC voltage being controlled by an analog output of the local microprocessor. The current is regulated by switching the DC/DC bridge with the SD signal from the MCU. In embodiments in which fast charging with a charge current above 1 C is required, the system includes means for measuring the battery temperature, the means being in data connection with the MCU, and the temperature is monitored by the MCU during the charging process; if the temperature or rate of temperature rise exceeds a predetermined limit, the charge current is reduced accordingly. As explained above for the case of charging of a single VPSU, the use of the ionization factor as input to a feedback loop enables the system to limit the temperature rise during the charging cycle, thereby improving the performance and extending the lifetime of the batteries. In addition, since each VPSU unit controls the charging of the batteries in that unit separately, in contrast to systems known in the prior art, the system disclosed herein enables separate and precise charging of small groups of batteries (i.e. each stack of batteries associated with a particular VPSU unit) rather than simultaneous and non-directed charging all of the batteries in the motor control system. In addition, the low voltage charging mode enables charging of simple, low-voltage charging of a plurality of VPSUs that comprise an entire battery pack, and can typically reach a total output voltage of 400-500 V. This capability (low-voltage charging of a high-voltage system) does not exist in battery systems known in the art. Reference is now made toFIG.6, which shows schematically a typical arrangement for low-voltage charging of a non-limiting embodiment1400of a system comprising plurality of VPSUs (three in this case,250A-C). In this charging mode, the VPSUs are connected in parallel through their upper column switches. Any number of VPSUs can be charged simultaneously as long as the charger can supply the current required. If the charger cannot supply the current needed for a fast charge, then the software in the EV computer extends the charging time, and charging takes place at the maximum current output of the charger. In preferred embodiments of the invention, each MCU stores the charging voltage vs. the charging current and battery cell temperatures at predetermined intervals (typically once per second). The charging parameters are stored in the MCU non-volatile memory, thereby enabling the system to determine and operate at the optimum charging voltage, current, and duration during each charge cycle. The manufacturer's recommended charging voltage (e.g. for a 3.7 V battery, the recommended charging voltage is typically 4.2 V) can be changed according to the running average of the voltage-current-temperature ratio. Reference is now made toFIG.7, which shows a graph of battery voltage and battery current as a function of time for a typical VPSU charging cycle. The 0.1 C point is reached soon after the charge cycle begins; normally, this is far below the minimum discharge point. The VPSU slowly increases the charging voltage to the nominal charging voltage. The charging current undergoes an abrupt jump to the 1 C charging point; the charging voltage continues to increase, however. EXAMPLES The following non-limiting examples are presented in order to assist a person of ordinary skill in the art to make and use the invention disclosed herein, and to illustrate the advantages of the instant invention over systems known in the art. In the examples that follow, the circuit diagrams presented are of emulations that accurately replicate the behavior of real systems. Unless otherwise stated, in the emulations, in order to simplify the calculations, the batteries are treated as having no internal resistance, and the internal resistance of the batteries is emulated by a single resistor having the same resistance as the total internal resistance of the batteries. The behavior of the circuits under realistic conditions is simulated by using the commercially available TINA Spice simulation ver. 11 software (DesignSoft). Example 1 As an illustration of the problems of approaches known in the art, and of battery losses under typical conditions, a simulation was run on a circuit that emulates a battery system of an EV. Reference is now made toFIG.8, which presents a circuit diagram for a circuit2000that emulates a battery system similar to that of a commercially available TESLA model 3 EV. The circuit includes voltage source121with a nominal voltage of 345.6 V and a nominal internal resistance of zero; the internal resistance of the battery is emulated by an external resistor123with resistance 108 mΩ. Meters122and124indicate the points in the circuit in which the simulated power at the battery output and after the resistor (i.e. taking into account losses due to internal resistance), are determined, respectively. The simulated battery output current and voltage are determined at the points indicated as current meter126and voltmeter125, respectively. The load is simulated with an inductor128that has an inductance of 30 pH and a serial resistance of 20 mΩ. Oscillator127generates a 16 kHz signal at 50% PWM. Four MOSFET transistors129a-d, connected in parallel, carry the current through the inductor. This emulation is expected to provide a realistic simulation of the waveforms observed in an actual EV system, since the electric motor usually behaves as an inductive load with a wiring resistance loss. Reference is now made toFIGS.9A-9D, which show graphically the calculated battery voltage (V), battery power loss (W), battery power (W), and motor current (A), respectively, as a function of time for a simulation run using circuit shown inFIG.10. The results of this simulation clearly show the power losses in the battery. The results presented inFIG.9show that for a battery voltage of 337.43 V (FIG.9A), the average power drawn from the battery (FIG.9C) at a 50% PWM duty cycle is 26.47 kW (37.92 kW RMS). The power loss in the battery (FIG.9B) averages 1.3 kW (2.04 kW RMS), i.e. under these conditions, 7.7% of the battery power expended is lost as heat; note that while the battery power from the motor controller perspective is calculated from the average value, the battery losses due to internal resistance are calculated as RMS. Example 2 Reference is now made toFIG.10, which presents a schematic diagram of a second circuit2100that emulates low voltage prior art EV battery systems, the EV battery pack drives an inductive load such as a motor coil winding of an electric vehicle. The system presented in this example is analogous to the one depicted inFIG.8, and comprises 20 batteries151divided into two stacks of 10 batteries. Within each stack the batteries are connected in series, and the two stacks are connected in parallel. Each battery in the circuit is a simulation of a Panasonic NCR 18650B lithium ion battery having a nominal voltage of 3.7 V and an internal resistance of 45 mΩ. The total internal resistance is represented by 225 mΩ resistor153. The arrangement of the batteries shown inFIG.10provides a nominal 1 C current of 6.5 A. The simulated battery power is determined at the point indicated by meter152, and the power loss at the point indicated by meter154. Signal generator155produces a 16 kHz signal that drives transistor156. The current flows through inductor157, which in the non-limiting example of the circuit shown inFIG.10has an inductance of 85 μH, and the output current and voltage are determined continuously at the points indicated by sensors158and159, respectively. Reference is now made toFIGS.11A-11D, which present graphically time-dependent battery voltage (V), battery power loss (W), battery power (W), and battery current (A), respectively, during a simulation run using the circuit shown schematically inFIG.10. The peak current load is 14.83 A. During each cycle, the total battery voltage drops from 37 V to 33.66 V. The average battery power is 174.97 W, the motor behaving as a low pass filter, and the RMS power loss is 17.69 W. Thus, the power loss in this system is 10.1%. Example 3 Reference is now made toFIG.12, which presents a schematic diagram of a third circuit2200that emulates the DC/DC portion of the VPSU disclosed herein. In this simulated circuit, the 3.7 V batteries161are arranged in two parallel stacks of 10 batteries each, with the internal resistance of the batteries represented by resistor163. The nominal voltage is 37 V and the nominal 1 C current is 6.5 A. Power meter162measures the output of the batteries, while power meter164measures the power losses due to internal resistance. A low pass filter165at the batteries and a bulk capacitor166simulate the path between the DC/DC converter and the batteries. The DC/DC converter167is controlled by the FB voltage level pin168and the current limits. The FB, CS, and CS0 pins of the MCU control the output voltage and the maximum current of transistor bridge169. The DC/DC converter is a buck-boost converter with a single inductor170; the quality of the inductor, particularly the winding resistance and bridge transistor loss from switching and RDS losses, is crucial for maximization of the total system efficiency. The output capacitors171smooth and filter the DC output voltage. The output voltage is determined at the point indicated by voltmeter172, and the current output at the point indicated by current meter174. The load, represented by 2.64Ω resistor175, has sharp rise and fall times and is quite close to actual motor loads. A signal generator173drives the current through the 2.64 resistor via power MOSFET transistor176. Reference is now made toFIGS.13A-13G, which present graphically the calculated time-dependent load voltage (V), load current (A), load power (W), battery loss (V), battery power (W), battery voltage (V), and battery current (A), respectively, during a simulation run using the circuit depicted inFIG.12. The graphs show the values of these parameters over 0.5 ms, starting 1 ms after the beginning of the simulation, after the initial transients have died down. The load current (FIG.13B) is the same as that shown inFIG.5Cfor the simulation run on an emulator of a system known in the prior art. In contrast to systems known in the art, however, the voltage output (FIG.13A) is stable at a constant 37 V. The battery voltage and current (FIGS.13F and13G, respectively) likewise are stable, with no ripple voltage or current. The battery power (FIG.13E) is 257.98 W, the battery power loss (FIG.13D) is 10.96 W, i.e. only 4.24% is lost as heat, a significant improvement over systems known in the art. Example 4 Reference is now made toFIG.14, which is a schematic diagram2300emulating a system known in the art, but without the ripple capacitor. In the simulations, the load, represented by a 2.64Ω resistor180, shows a sharp rise and fall time and is quite close to actual motor loads. A signal generator drives the current through the 2.64Ω resistor via power MOSFET transistor. Reference is now made toFIGS.15A-15G, which present graphically the time-dependent load voltage (V), load current (A), load power (W), battery loss (W), battery power (W), battery voltage (V), and battery current (A), respectively, during a simulation run using the circuit depicted inFIG.14. The graphs show the values of these parameters over 0.5 ms, starting 1 ms after the beginning of the simulation, after the initial transients have died down. the simulation run on an emulator of a system known in the prior art. the voltage output (FIG.15A) is not stable at a constant 37 V. The battery voltage and current (FIGS.15F and15G, respectively) are not stable, but rather demonstrate significant ripple voltage and current. The battery power (FIG.15E) is 239.13 W, the battery power loss (FIG.15D) 26.51 W RMS, i.e. 11% is lost as heat, a significant loss in comparison to the instant invention. | 39,343 |
11863060 | DETAILED DESCRIPTION In order that the objects, technical solutions and advantages of the present disclosure may be more clearly understood, embodiments of the present disclosure will be described in further detail below in combination with the detailed embodiments with reference to accompanying drawings. It should be noted that all the expressions using “first” and “second” in the embodiments of the present disclosure are used for distinguishing two different entities or different parameters with the same name. It can be seen that “first” and “second” are merely for the convenience of expressions and should not be understood as limiting the embodiments of the present disclosure, and the subsequent embodiments will not be described regarding this one by one. In some embodiments, the present disclosure provides a control circuit of a buck converter. As shown inFIG.1, the control circuit of a buck converter may include:a first transistor Q1, a base electrode of the first transistor Q1receiving an output voltage Voutof the buck converter;a second transistor Q2, a base electrode of the second transistor Q2being connected to an emitter electrode of the first transistor Q1;a third transistor Q3, a base electrode of the third transistor Q3being connected to a collector electrode of the second transistor Q2;a first resistor R1, one end of the first resistor R1being connected to a collector electrode of the third transistor Q3;a second resistor R2, one end of the second resistor R2being connected to the other end of the first resistor R1;a fourth transistor Q4, a collector electrode of the fourth transistor Q4being connected to the other end of the second resistor R2;a third resistor R3, one end of the third resistor R3being connected to an emitter electrode of the third transistor Q3, and the other end of the third resistor R3receiving a phase voltage of the buck converter;a fourth resistor R4, one end of the fourth resistor R4being connected to an emitter electrode of the fourth transistor Q4and the other end of the fourth resistor R4being connected to a collector electrode of the first transistor Q1;a fifth resistor R5and a sixth resistor R6, one end of the fifth resistor R5being connected to the base electrode of the third transistor Q3, the other end of the fifth resistor R5being connected to one end of the sixth resistor R6, and the other end of the sixth resistor R6being connected to a base electrode of the fourth transistor Q4;a seventh resistor R7, one end of the seventh resistor R7being connected to an emitter electrode of the second transistor Q2and the other end of the seventh resistor R7being connected to a collector electrode of the first transistor Q1; anda comparator, a positive input end of the comparator being connected to the collector electrode of the fourth transistor Q4, a negative input end of the comparator being connected to the collector electrode of the third transistor Q3, and an output end of the comparator being connected to a controller of the buck converter. In the control circuit of a buck converter provided herein, when a phase voltage of a buck converter changes, a controller in the buck converter is controlled to output a signal for turning off a lower MOS transistor (QlowerinFIG.2), so that after the signal is transmitted through the line, the lower MOS transistor (Qlower) can be controlled to be exactly turned off just when the current is reversed. Such an accurate reverse current detection function can reduce the voltage loss of the buck converter, thereby improving the efficiency of a system in standby or having a light load. In some embodiments, a resistance value of the first resistor R1is equal to a resistance value of the second resistor R2and a resistance value of the third resistor R3is equal to a resistance value of the fourth resistor R4. In some embodiments, in the control circuit shown inFIG.1, the magnitude of the current flowing through the first resistor R1may be: I1=12β1(Va-Vthn1-IaR5-I1R3-Vphase)2 wherein Vthn1is a voltage drop when the third transistor Q3is conducting, β1is a parameter of the third transistor Q3, and a difference between a left side of the fifth resistor R5(the side close to the base electrode of the third transistor Q3) and an upper side of the third resistor R3(the side close to the emitter electrode of the third transistor Q3) may be obtained by calculating the value of (Va−Vthn1−IaR5−I1R3−Vphase). The magnitude of the current flowing through the second resistor R2may be: I2=12β2(Va-Vthn2-I2R4-Vgnd)2 wherein Vthn2is a voltage drop when the fourth transistor Q4is conducting, β2is a parameter of the fourth transistor Q4, and a difference between a left side of the sixth resistor R6(the side close to the base electrode of the fourth transistor Q4) and an upper side of the fourth resistor R4(the side close to the emitter electrode of the fourth transistor Q4) can be obtained by calculating the value of (Va−Vthn2−I2R4−Vgnd). The resistance value of the first resistor R1is equal to the resistance value of the second resistor R2, the resistance value of the third resistor R3is equal to the resistance value of the fourth resistor R4, and the parameters of the third transistor Q3and the fourth transistor Q4are also the same. Therefore, when the voltage VSWCat the collector electrode of the fourth transistor Q4and the voltage Vgndcat the collector electrode of the third transistor Q3are equal, I1is equal to I2, and it can be obtained by the foregoing formula: Vphase−Vgnd−IaR5 In some embodiments, since Vgndcan be approximated to zero, Vphase−IaR5can be approximated. According to the principle of the transistor, it can be seen that a voltage at the base electrode of the second transistor Q2inFIG.1is equal to a voltage at the emitter electrode thereof, and a voltage at the base electrode of the first transistor Q1is equal to a voltage at the emitter electrode thereof. Due to the connection of the base electrode of the second transistor Q2and the emitter electrode of the first transistor Q1,a voltage at the base electrode of the first transistor Q1is equal to a voltage at the emitter electrode of the second transistor Q2, namely, Vout=IaR7. Thus, according to Vout=IaR7and Vphase=IaR5, it can be obtained that: Vphase=R5R7Vout. At this time, the voltages input by the positive input end of the comparator and the negative input end of the comparator are equal, and the comparator can output a preset signal, and the controller in the buck converter can output a signal for controlling the turning off of the lower MOS transistor Qlowerafter receiving the preset signal. Therefore, it is possible to achieve accurate turning off of the lower MOS transistor Qlowerin different output voltages by adjusting the resistance values of the fifth resistor R5and the seventh resistor R7. In some embodiments, the controller is configured to generate a signal to control turning off of the lower MOS transistor Qlowerin the buck converter in response to receiving the preset signal. Specifically, as shown inFIG.2, the output end of the comparator shown inFIG.1can be connected to the controller in the buck converter shown inFIG.2, so that the controller outputs a signal to turn off the lower MOS transistor Qlowerafter receiving a preset signal output by the comparator, so that after the signal is transmitted through the line, the lower MOS transistor (Qlower) can be controlled to be exactly turned off just when the current is reversed. Such an accurate reverse current detection function can reduce the voltage loss of the buck converter, thereby improving the efficiency of a system in standby or having a light load. Based on the same inventive concept, in some embodiments, the present disclosure also provides a server including the control circuit of a buck converter in any of the foregoing embodiments. The foregoing are exemplary embodiments disclosed herein, but it should be noted that various changes and modifications could be made therein without departing from the scope of the disclosed embodiments as defined by the appended claims. The functions, steps, and/or actions of the method claims in accordance with the disclosed embodiments described herein do not need to be performed in any particular order. Furthermore, although elements disclosed in the embodiments may be described or claimed in the singular form, they may be contemplated as in a plurality unless limitation to the singularity is explicitly stated. It will be understood that, as used herein, the singular forms “a” and “an” are intended to include the plural forms as well, unless the context clearly supports the exception. The foregoing sequence of the embodiments of the present disclosure has been presented for purposes of illustration only and is not intended to represent the advantages or disadvantages of the embodiments. Those of ordinary skill in the art will appreciate that the discussion of any embodiment above is intended to be exemplary only, and is not intended to imply that the scope of the disclosure of the embodiments of the present disclosure, including the claims, is to be limited to these examples. Combinations of the features in the foregoing embodiments or in different embodiments are also possible within the framework of the embodiments of the present disclosure, and many other variations of the different aspects of the embodiments of the present disclosure as described above are not provided in detail for the sake of brevity. Thus, it is intended that the scope of protection of the embodiments of the present disclosure cover the omissions, modifications, equivalents, and improvements falling within the spirit and principle of the embodiments of the present disclosure. | 9,795 |
11863061 | DETAILED DESCRIPTION Reference now will be made in detail to embodiments, one or more examples of which are illustrated in the drawings. Each example is provided by way of explanation of the embodiments, not limitation of the present disclosure. In fact, it will be apparent to those skilled in the art that various modifications and variations can be made to the embodiments without departing from the scope or spirit of the present disclosure. For instance, features illustrated or described as part of one embodiment can be used with another embodiment to yield a still further embodiment. Thus, it is intended that aspects of the present disclosure cover such modifications and variations. Example aspects of the present disclosure are directed to a switch that can be configured to provide an electrical output for powering one or more loads associated with a lighting fixture or other device even when the switch is in a position to disconnect AC electrical power from the lighting fixture or other device. For instance, a lighting fixture can include one or more auxiliary loads, such as a controller, wireless communication device, camera, environmental detection sensor, etc. In some applications, the auxiliary load can receive electrical power from a battery positioned onboard the auxiliary load. However, since the auxiliary load relies on the battery for electrical power, a user must regularly replace the battery to ensure proper operation of the auxiliary load. Replacing the battery can be a rather burdensome task. According to example embodiments of the present disclosure, electrical power (e.g., DC power) can be provided to a lighting fixture or other device regardless of a position of a switch used to provide AC power to the lighting fixture or other device. In this way, one or more auxiliary loads associated with the lighting fixture can remain powered without requiring a battery or other dedicated power source onboard the lighting fixture. In some embodiments, a switch can include a first terminal, a second terminal, and a third terminal. The switch can include a first output node and a second output node. The first output node can be coupled to the second terminal. The second output node can be coupled to the third terminal. The switch can include a contactor coupled to the first terminal and movable between a first position and a second position. When the contactor is in the first position, the first terminal can be coupled to the second terminal. When the contactor is in the second position, the first terminal can be coupled to the third terminal. As will be discussed below in more detail, the switch can be configured to output direct current power at the first output node, and alternating current power at the second output node. In some embodiments, the switch can include a power converter coupled between the first output node and the second terminal. The power converter can be configured to convert alternating current power to direct current power. As such, when the contactor is in the first position, the power converter can receive alternating current power from a power source and can convert the alternating current power to direct current power. In some implementations, the first output node can be coupled to an auxiliary load of the lighting system. As such, when the contactor is in the first position, the switch can deliver DC power to the auxiliary load via the first output node. When the contactor is in the second position, the switch can provide AC power at the second output node. In some implementations, the switch can be coupled to the light source and auxiliary load of the lighting system via the second output node. In this way, the switch can provide power to both the light source and the auxiliary load when the contactor is in the second position. The switch according to example embodiments of the present disclosure can provide a number of technical effects and benefits. For instance, when a user toggles the switch to deactivate (e.g., turn off) the light source, the switch can continue to provide electrical power to one or more auxiliary loads via the first output node. In this way, use of energy storage devices (e.g., batteries) to power the auxiliary load can be reduced. Example aspects of the present disclosure are discussed with reference to a switch used to power a lighting fixture for purposes of illustration and discussion. Those of ordinary skill in the art, using the disclosures provided herein, will understand that switches according to example embodiments of the present disclosure can be used with other loads without deviating from the scope of the present disclosure. Referring now to the FIGS.,FIG.1depicts a lighting fixture100disposed within a ceiling120that extends between a first surface122and a second surface124along a vertical direction V. As shown, the ceiling120can separate a first space130(e.g., positioned beneath the ceiling110) from a second space140(e.g., positioned above the ceiling110) along the vertical direction V. In some implementations, the first space130can include a room (e.g., kitchen, living room, etc.) of a residential home, and the second space140can include an attic positioned above the room. The lighting fixture100can include a light source110to provide illumination for the first space130. As shown, the light source110can be disposed within the lighting fixture100. In some implementations, the light source110can include an array of light emitting diodes (LEDs) or any other suitable light source. As discussed below, operation of the light source110can be controlled via manipulation of a switch150, such as a wall switch. Referring briefly now toFIG.2, the lighting fixture100can include a power converter112configured to receive an input power from a power source160(e.g., an AC or DC power source) and convert the input power to an output power suitable for powering the light source110. In some instances, the light source110can include an array of LED light sources, and the power converter112can be configured to provide different driving currents to each of the LED light sources. For instance, the power converter112can include one or more of a multi-channel driver circuit, a current splitter circuit, one or more current regulators, and/or other devices that can be used to independently provide a driver current to each of the LED light sources. As mentioned above, the switch150can be used to control operation of the light source110. More specifically, the switch150can be used to selectively couple the light source110to the power source160. For instance, the switch150can be a single pole single throw (SPST) switch movable between a first position152and a second position154. When the switch150is in the first position152, the light source110is not coupled to the power source160. However, when the switch150is in the second position154, the light source110is coupled to the power source160. In this way, the switch150can be used to activate (e.g., turn on) and deactivate (e.g., turn off) the light source110. Referring now toFIG.3, a lighting system200according to example embodiments of the present disclosure can include the light source110and an auxiliary load210. In some implementations, the auxiliary load210can include one or more sensor(s)220operable to sense one or more parameters associated with the first space130. For instance, the one or more parameters can include an environmental parameter, such as amount of smoke present within the first space130. In this way, the sensor(s)220can collect data that can be used to monitor the first space130. It should be appreciated, however, that the sensor(s)220can be configured to detect any parameter. For example, the sensors(s)220can be operable to detect an amount of carbon monoxide (CO) present within the first space130. In this way, the sensor(s)220can collect data that can be used to determine whether there is a CO leak within the first space130. As shown inFIGS.3and4, the auxiliary load210can additionally or alternatively include one or more control device(s)230. For instance, the control device(s)230can include at least one processor232and associated memory device234configured to perform a variety of computer-implemented functions (e.g., performing the methods, steps, calculations and the like disclosed herein). As used herein, the term “processor” refers not only to integrated circuits referred to in the art as being included in a computer, but also refers to a controller, microcontroller, a microcomputer, a programmable logic controller (PLC), an application specific integrated circuit (ASIC), a Field Programmable Gate Array (FPGA), and other programmable circuits. Examples of the memory device234can include computer-readable media including, but not limited to, non-transitory computer-readable media, such as RAM, ROM, hard drives, flash drives, or other suitable memory devices. The memory device234can store information accessible by the processor(s)232, including computer-readable instructions236that can be executed by the processor(s)232. The computer-readable instructions236can be any set of instructions that, when executed by the processor(s)232, cause the processor(s)232to perform operations. The computer-readable instructions236can be software written in any suitable programming language or can be implemented in hardware. In some implementations, the computer-readable instructions236can be executed by the processor(s)232to perform operations, such as generating a control action associated with presenting an alarm or notification based on an environmental parameter associated with the room or space in which the lighting fixture100is located. For instance, the processor(s)232can generate a control action based on data received from the sensor(s)220. The memory device234can further store data238that can be accessed by the control device(s)230. In example embodiments, the data238can include data received from the sensor(s)220. Additionally or alternatively, as shown inFIG.4, the control device(s)230can include a communication interface240. In example embodiments, the communications interface240can include associated electronic circuitry that can be used to communicatively couple the control device(s)230with other devices, such as the sensor(s)220. In some embodiments, the communication interface240can allow the control device(s)230to communicate directly with other devices. As will be discussed below, the communication interface240can provide for communication with other devices over a network170(FIG.5). Referring toFIG.5, the network170can be any suitable type of network. For instance, the network170can be a local area network (e.g., intranet), wide area network (e.g., internet), low power wireless network (e.g., Bluetooth Low Energy (BLE), Zigbee, etc.), or some combination thereof and can include any number of wired or wireless links. In general, communication over the network170can be implemented via any type of connection, using a wide variety of communication protocols, encodings or formats, and/or protection schemes. Example communication technologies used in accordance with example aspects of the present disclosure can include, for instance, Bluetooth low energy, Bluetooth mesh networking, near-field communication, Thread, TLS (Transport Layer Security), Wi-Fi (e.g., IEEE, 802.11), Wi-Fi Direct (for peer-to-peer communication), Z-Wave, Zigbee, Halow, cellular communication, LTE, low-power wide area networking, VSAT, Ethernet, MoCA (Multimedia over Coax Alliance), PLC (Power-line communication), DLT (digital line transmission), etc. Other suitable wired and/or wireless communication technologies can be used without deviating from the scope of the present disclosure. In some implementations, the control device(s)230can generate one or more control actions associated with controlling operation of the light source110. For instance, the control action(s) can include activating (e.g., turn on) or deactivating (e.g., turn off) the light source110. More specifically, the control device(s)230can command the light source110to flash (that is, activate and deactivate) at a predetermined frequency. In this way, the light source110can be used to indicate an environmental condition (e.g. fire) detected based, at least in part, on data from the sensor(s)220. In some implementations, the control device(s)230can communicate with a user device180over the network170. The user device180can be any suitable type of device, such as, for example, a personal computing device (e.g., laptop or desktop), a mobile computing device (e.g., smartphone or tablet), a wearable computing device, an embedded computing device, a remote computing device, or any other suitable type of computing device. The user device180can include one or more computing device(s)184with the same or similar components as described above with regard to the control device(s)230. For instance, the computing device184of the user device180can include one or more processors and one or more memory devices that store instructions that are executable by the processor to cause the user device180to perform operations, such as e.g., communicating one or more control signals over the network170to the control device(s)230. In this way, a user can control operation of the light source110via the user device180. In some implementations, the control device(s)230can communicate data to the user device180via the communication interface240. For instance, the control device(s)230can provide data captured by the sensor(s)220to the user device180. The information can be displayed (e.g., via a display device) or otherwise presented (e.g., via audio speakers) to the user through a suitable interface182. In this way, a user can observe data collected by the sensor(s)220. In some implementations, the control device(s)230can communicate a notification or alert indicative of a detected environmental condition (e.g., fire) to the user device180via the communication interface240. For instance, the control device(s)230can communicate an electronic message (e.g., email, short message service (SMS) text message, etc.) indicating a detected environmental condition for the first space130(FIG.1). In this way, a person using the user device180, such as a homeowner, can be apprised of the detected environmental condition. In some implementations, the lighting fixture200can include a power converter240configured to receive an input power from the power source160and convert the input power to an output power (DC power) suitable for powering the sensor(s)220and the control device(s)230. Referring now toFIGS.6and7, a system500for controlling operation of the lighting system200is provided according to example embodiments of the present disclosure. As shown, the system500can include a switch510. In some implementations, the switch510can include a first terminal520, a second terminal522, and a third terminal524. The switch510can include a contactor526coupled to the first terminal520and movable between a first position530and a second position540. When the contactor526is in the first position530, the contactor526can couple the first terminal520to the second terminal522. When the contactor526is in the second position540, the contactor526can couple the first terminal520to the third terminal524. As will be discussed below in more detail, the switch510can be coupled to the power source160and can be used to distribute electrical power to the lighting system200. A user can manipulate the position of the contactor526by toggling the switch510. In some implementations, the first terminal520can receive electrical power from the power source160, such as AC power. The AC power can be, for instance, 120V mains power or other suitable power. As mentioned above, the contactor526can be used to selectively couple the first terminal520to the second or third terminals522,524. More specifically, the second terminal522can receive the electrical power when the contactor526is in the first position530. The third terminal524can receive the electrical power when the contactor526is in the second position540. As shown, the switch510can include a first output node550coupled to the second terminal522. In this way, the switch510can output electrical power at the first output node550when the contactor526is in the first position530. In some implementations, the switch510can include a power converter560coupled between the second terminal522and the first output node550. The power converter560can be configured to convert alternating current power from the power source160to direct current power. The power converter560can include, for instance, a rectifier, one or more switching elements (e.g., transistors) and filters and/or other circuit components for converting AC power to DC power. As such, when the contactor526is in the first position530, the switch510can output DC power at the first output node550. In some implementations, the first output node550can be coupled to the auxiliary load210of the lighting system200. In this way, the switch510can provide DC power to the auxiliary load210when the contactor526is in the first position530. Still referring toFIGS.6and7, the switch510can include a second output node570coupled to the third terminal524. In this way, the switch510can output electrical power at the second output node570when the contactor526is in the second position540. More specifically, the switch510can output AC power at the second output node570. In some implementations, the second output node570can be coupled to both the light source110of the lighting system200and the auxiliary load210of the lighting system200. In this way, the switch510can provide AC power to both the light source110and the auxiliary load210when the contactor526is in the second position540. Referring now toFIGS.8and9, another system600for controlling operation of the lighting system200is provided according to example embodiments of the present disclosure. As shown, the system600can include a switch610. In some implementations, the switch610can include a first terminal620, a second terminal622, and a third terminal624. The switch610can include a contactor626coupled to the first terminal620and movable between a first position630and a second position640. When the contactor626is in the first position630, the contactor626can couple the first terminal620to the second terminal622. When the contactor626is in the second position640, the contactor626can couple the first terminal620to the third terminal624. As will be discussed below in more detail, the switch610can be coupled to the power source160and can be used to distribute electrical power to the lighting system200. A user can manipulate the position of the contactor626by toggling the switch610. In some implementations, the first terminal620can receive electrical power from the power source160, such as AC power. The AC power can be, for instance, 120V mains power or other suitable power. As mentioned above, the contactor626can be used to selectively couple the first terminal620to the second terminal622or the third terminal624. More specifically, the second terminal622can receive the electrical power when the contactor626is in the first position630. The third terminal624can receive the electrical power when the contactor626is in the second position640. As shown, the switch610can include an output node650coupled to the second terminal622and the third terminal624. In some implementations, the switch610can include a power converter660coupled between the second terminal622and the output node650. The power converter660can be configured to convert alternating current power from the power source160to direct current power. The power converter660can include, for instance, a rectifier, one or more switching elements (e.g., transistors) and filters and/or other circuit components for converting AC power to DC power. As such, when the contactor626is in the first position630, the switch610can output DC power at the output node650. When the contactor626is in the second position640, the switch610can output AC power at the output node650. The system600can include a power circuit680in electrical communication with the switch610and both the light source110and the auxiliary load210. In some implementations, the power circuit680can be onboard the lighting fixture100. In alternative implementations, the power circuit680can be external to the lighting fixture100. The power circuit680can include, for instance, one or more switching elements (e.g., transistor) and/or other circuit components for routing DC power to the auxiliary load210or AC power to both the light source110and the auxiliary load210. As will be discussed below in more detail, the power circuit680can be configured to route DC power or AC power based on a position of the contactor626. When the contactor626is in the first position630, the power circuit680receives the DC power from the switch610and routes the DC power to the auxiliary load210. When the contactor626is in the second position640, the power circuit680receives the AC power from the power source160via the switch610and routes the AC power to both the light source110and the auxiliary load210. In this way, the auxiliary load210can receive DC power when the contactor626has been moved to the first position630to deactivate (e.g., power off) the light source110. While the present subject matter has been described in detail with respect to specific example embodiments thereof, it will be appreciated that those skilled in the art, upon attaining an understanding of the foregoing may readily produce alterations to, variations of, and equivalents to such embodiments. Accordingly, the scope of the present disclosure is by way of example rather than by way of limitation, and the subject disclosure does not preclude inclusion of such modifications, variations and/or additions to the present subject matter as would be readily apparent to one of ordinary skill in the art. | 22,100 |
11863062 | Like reference symbols in the various drawings indicate like elements. DETAILED DESCRIPTION Referring now toFIG.1, a circuit10is shown having discharge circuitry11for discharging a capacitor bank12, coupled to an output of a High Voltage Power Supply13, to a discharge load14, here a resistor RDISCHARGE, here, for example, 2.5 ohms. The capacitor bank12provides a voltage, VCAP, for a system18. The High VOLTAGE POWER Supply13is coupled to a voltage source15. The voltage source15may be: an ac voltage source, in which case the High Voltage Power Supply13would, when enabled, produce a DC Voltage at output17; or, a DC voltage, in which case the High Voltage Power Supply13would be a DC-DC converter to increase the DC voltage to a higher DC voltage at output17. In either case, the DC voltage at output17is fed to the Capacitor bank12through a diode CRPSBLOCK, as shown. As will be described in more detail below, subsequent to High Voltage Power Supply being enabled and charge being stored on the capacitor bank12, the discharge circuity11discharges such stored change during a plurality of two-phase discharge cycles. More particularly, the discharge circuity11is enabled and the charge built up on the capacitor12is discharged and thereby dissipated in the discharge load14over the plurality of the two-phase discharge cycles, with current from the capacitor bank12passing through the discharge load14with increasing level over time during one portion, or phase, of each one of the two-phase discharge cycles and with current passing through the discharge load14decreasing in level over time during a subsequent, different portion, or phase of each one of the two-phase discharge cycles. In this way, the energy (charge) stored on the capacitor bank12is discharged in small packets distributed over a long period of time. More particularly, the capacitor bank12has one plate or electrode12bconnected to a reference potential, here ground, and the other plate, or electrode,12aconnected to one end of the discharge load14, as shown. The other end of the discharge load14is connected to one end of an inductor LDISCHARGE20, here, for example, 60 micro-henries, as shown; the other end of the inductor20being connected to a terminal22, as shown. The terminal22is connected to: plate12aof the capacitor bank12through a freewheeling, fly-back diode23, as shown; and to the drain D of a Field Effect Transistor (FET) QSWITCH24, as shown. The source S of the FET24is coupled to ground through a current level sensing resistor RSENSE26, as shown. The Discharge circuitry11includes a discharge controller28, as shown. In response to a power supply disable/capacitor bank discharge signal from system18, to the disable terminal of power supply13and the enable terminals of controller28, the voltage produced by the power supply13when previously enabled, is decoupled from output terminal17and the controller28changes a previous state of a signal on line30to the gate G of FET24from a signal that placed the FET24in a non-conducting condition when the power supply13was enabled to a state that places FET24in a conducting condition and thereby commences the first phase of the two-stage discharge cycle. When PET24conducts, charge on plate12aof capacitor bank12passes as discharge current through discharge load14, inductor20, FET24, current level sensing resistor26to ground, as shown inFIG.1A, thereby initiating the first portion or phase of one of a plurality of two-phase discharge cycles. During this portion or phase of the cycle, the discharge current from the capacitor bank12passes through the discharge load14, inductor20and sensing resistor26increase in level over time. The level of the discharge current from the capacitor bank12is sensed by current level sensing circuitry25, here a sensing resistor26which produces a voltage (VSENSE) on line32proportional to such sensed current level. When the voltage VSENSEreaches a predetermined voltage VREF1(such voltage VREF1being the voltage level when VSENSEcorresponds to a current level I1passing thorough R resistor26(which is the same current level passing through RDISCHARGE14), here, for example, VREF1is 1.24 volts), the controller28produces the discharge signal on line30to place FET24in a non-conducting condition. It is noted that during this first portion or first phase of the discharge cycle: (1) While a portion of the energy produced from the discharge current passing from the from the capacitor bank12through the discharge load14was dissipated in the discharge load14another portion of the energy produced from the discharge current passing from the capacitor bank12through the discharge load14was stored in the magnetic field of inductor20; (2) the voltage on the capacitor bank12was reduced; and (3), the controller28sends a sampling pulse on line35to sampler36of a timing circuit40and a capacitor C of the timing circuit40stores the sampled voltage VSENSE. When the FET24is placed in the non-conducting state, the second portion or phase of the two-phase discharge cycle commences. Three things should be noted during this second portion or phase of the two-stage discharge cycle: (1) energy stored in the inductor20during the first portion or cycle of the two-phase discharge cycle, produced from the discharge current passing from the capacitor bank12through the discharge load14, now produces a very high positive induced voltage, sometimes referred to as an inductive kick as a result of rapidly producing at open circuit when the FET24is placed in the non-conducting state. This very high positive induced voltage forward biases the diode23and the energy stored in the inductor20from the discharge current passing from the capacitor bank12now passes as current through diode23to the discharge load14, as shown inFIG.1B, for dissipation in such discharge load14; and (2) the level of the current through the discharge load14decreases over time unlike the increasing level of current passing through the discharge load14during the first portion or phase of the two-phase discharge cycle; and (3) the timing circuit40sets the time duration of the second portion or phase of the two-phase discharge cycle since the voltage VREF1stored on capacitor C, here for example, 0.012 pico-farards, discharges through a resistor R, here for example, 2 KOhms, of the timing circuit40and when such voltage on the capacitor C decreases to a predetermined level VREF2, comparator41(the voltage VREF2being selected to corresponding to a time determined a priori to be when the level of the current passing though the discharge load14reaches a current level12, here for example, VREF2is 0.62 volts) produces a voltage on line42to terminate the second portion or phase of the two-phase discharge cycle. More particularly, the controller28switches the state of the discharge signal on line30to again place the FET24in a conducting condition to commence the first portion or phase of the next two phase discharge cycle. The operation is summarized in the flowchart ofFIG.2. As noted above, the voltage on the capacitor bank12(FIG.1) decreases in level after each one of the two-phase discharge cycles and when the level of the voltage on the capacitor bank12ceases to be at a level sufficient to produce a current having a level I1, the FET24remains in the conducting condition and the capacitor bank12continues to discharge through the discharge load14until all charge is removed from the capacitor bank12, as shown by the timing diagrams inFIGS.3and4. Thus, it is noted that a constant discharge is achieved by turning the FET24“on” (in a conducting condition) when the current through the inductor14reaches the lower peak and turning the FET24“off” (in a non-conducting condition) when the current reaches the high peak. That switching of the FET24“on” and “off” creates a triangular waveform of current through the inductor and discharge resistor (FIGS.3A-3E), which results in a constant (average) power dissipation in the discharge load14. Thus, referring again toFIGS.3A-3E, to begin the high voltage capacitor discharge at time T0, a discharge command is applied from the system18to the Discharge Circuit Controller28. The discharge command signal is the inverse of the system power supply13enable signal, so that the System High Voltage Power Supply13is disabled at the same time that the discharge circuit11is enabled. This prevents an active discharge while the System High Voltage Power Supply13is still operating. The Discharge Circuit Controller28turns on the discharge FET24, QSWITCH, by applying a voltage to the gate G. The voltage on the high voltage capacitor bank12CBANK, causes a current through the discharge load14RDISCHARGE, the discharge inductor LDISCHARGE, the discharge FET24QSWITCH, and current sense resistor RSENSEto begin to increase as it passes to the ground of the circuit. When the current sense voltage VSENSEcreated by the discharge current IDISCHARGEtimes RSENSEreaches a reference voltage VREF1, then the comparator circuit34is tripped and FET24QSWITCHis turned off by the Discharge Circuit Controller28at time T1. Here, for example RSENSEis 0.08 Ohms. This occurs when the discharge current IDISCHARGEreaches I1. The current IDISCHARGEthrough RDISCHARGEand LDISCHARGEthen begins to conduct through the freewheeling, fly-back diode23CRDISCHARGE. The current IDISCHARGEdecays at a rate proportional to the time constant LDISCHARGE/RDISCHARGE. The timing circuit40, here an R-C circuit with a time constant that matches the LDISCHARGE/RDISCHARGEtime constant samples the voltage VSENSEat time T1. Here, VSENSEchanges based on the current through RSENSEand here VSENSEpeaks at, in this example, 1.4 Volts. When the voltage of the RC circuit timing circuit40decays to the trip threshold VREF2, FET24QSWITCHis turned on (time T2with IDISCHARGEat I2), and the cycle repeats with the current IDISCHARGEonce again passing through QSWITCHand RSENSE, and increasing until it reaches the current level I1again. Note that the voltage VCAPon the capacitor bank CBANKdecreases each cycle during the time that FET QSWITCHis turned on (e.g. time periods T0to T1and T2to T3). Because of this, the time it takes for the current IDISCHARGEto ramp up to the QSWITCHturn off threshold I1each cycle is slightly longer than the previous cycle by the relationship IDISCHARGE=(VCAP/RDISCHARGE)*(1−exp(−RDISCHARGE*t/LDISCHARGE)), where t is time. This means that the frequency of operation of the circuit decreases slightly as the voltage VCAPdecays. In the fashion described above an average current is created through resistor RDISCHARGEand inductor LDISCHARGE, and thereby an average power is dissipated in discharge load14RDISCHARGEby the square of IDISCHARGEtimes RDISCHARGE. It is also noted that the high current level (I1) decreases slightly each switching cycle because the voltage on the capacitor bank12has decreased slightly from the cycle before. This means that the current increases at a slightly lower rate than the previous cycle. There is a little bit of overshoot on the current each time because current is still flowing through the FET24switch as it is being turned “off”. The higher voltage on the capacitor bank12at the beginning of the discharge makes the current overshoot a little bit more than at the end of the discharge, when the voltage on the capacitor bank12is lower. The low switching current (I2) changes a little bit on the bottom side. However, since a lower peak current was reached, the decay in absolute terms of amps is slightly slower at the end of the discharge than at the beginning, so the change in the lower current level (I2) is less pronounced that the higher current level (I1). As noted above, the discharge cycles continue until the voltage on CBANK12is insufficient to cause the current through RSENSEto reach the level of I1. This means that the FET QSWITCHremains turned “on”, and the current IDISCHARGEdecays to zero at which time the voltage on capacitor CBANKalso reaches zero. This completes the discharge of the high voltage capacitor CBANK. A number of embodiments of the disclosure have been described. Nevertheless, it will be understood that various modifications may be made without departing from the spirit and scope of the disclosure. Accordingly, other embodiments are within the scope of the following claims. | 12,358 |
11863063 | DESCRIPTION OF EMBODIMENTS An embodiment of the present invention will be described with reference to the accompanying drawings. Note that in the drawings, parts that are identical or correspond to each other are provided with a same reference numeral. Overlapping descriptions of such parts will arbitrarily be simplified or omitted. Embodiment 1 FIG.1is a configuration diagram of a system to which a power conversion system according to Embodiment 1 is applied. In the upper part ofFIG.1, a direct-current power supply1is a photovoltaic facility. An alternate-current power supply2has three phases and is operated by, e.g., an electric power company. A transformer3is connected between the direct-current power supply1and the alternate-current power supply2. The power conversion system includes a power converter4, a direct-current capacitor5, direct-current switches6, an alternate-current reactor7, a first alternate-current switch8a charging resistance9, a second alternate-current switch10, an alternate-current capacitor11and a control device12. The power converter4is connected between the direct-current power supply1and the transformer3. The direct-current capacitor5is connected between the direct-current power supply1and the power converter4. The direct-current switches6are connected between the direct-current power supply1and the power converter4. The alternate-current reactor7is connected between the power converter4and the transformer3. The first alternate-current switch8is connected between the alternate-current reactor7and the transformer3. The charging resistance9is connected in parallel to the first alternate-current switch8, between the alternate-current reactor7and the transformer3. The second alternate-current switch10is connected in parallel to the first alternate-current switch8and in series to the charging resistance9, between the alternate-current reactor7and the transformer3. The alternate-current capacitor11is connected to the alternate-current reactor7side relative to the first alternate-current switch8, the charging resistance9and the second alternate-current switch10, on the output side of the power converter4. The first alternate-current switch8, the charging resistance9and the second alternate-current switch10may have any of the forms in (a), (b) and (c) in the lower part ofFIG.1. The control device12is provided so as to be capable of controlling the power converter4, the direct-current switches6, the first alternate-current switch8and the second alternate-current switch10. Next, a series of controls by the control device12will be described with reference toFIGS.2to6. FIG.2is a diagram for describing a charging start mode provided by a control device in the power conversion system according to Embodiment 1.FIG.3is a diagram for describing a boost charging mode provided by the control device in the power conversion system according to Embodiment 1.FIG.4is a diagram for describing an alternate-current voltage synchronization mode provided by the control device in the power conversion system according to Embodiment 1.FIG.5is a diagram for describing a standby mode provided by the control device in the power conversion system according to Embodiment 1.FIG.6is a diagram for describing an SVC operation mode provided by the control device in the power conversion system according to Embodiment 1. The SVC (static var compensator) operation mode refers to an operation mode in which reactive power compensation is performed. As illustrated inFIG.2, in the alternate-current voltage synchronization mode, the control device12provides a state in which the direct-current switches6are open, the first alternate-current switch8is open and the power converter4is stopped, and closes the second alternate-current switch10. At this time, an electrostatic capacitance Cdcof the direct-current capacitor5is charged via the charging resistance9. Where impedances of the transformer3and the alternate-current power supply2can be ignored, a direct-current voltage Vireaches approximately a value expressed by Expression (1) below. [Math.1]V1=2Vs1+(3ωRprcCpi)2(1) However, in Expression (1), ω is an angular speed corresponding to a frequency of the alternate-current power supply2. Rprcis a resistance value of the charging resistance9. Cpiis an electrostatic capacitance of the alternate-current capacitor11. Vsis a rated voltage of the alternate-current power supply2. Subsequently, as illustrated inFIG.3, in the boost charging mode, the control device12controls the power converter4to perform boosting, and performs charging until the direct-current voltage V1reaches a preset value that is equal to or exceeds a value according to Expression (2) below. [Math. 2] V1=√{square root over (2)}Vs(2) Subsequently, as illustrated inFIG.4, in the alternate-current voltage synchronization mode, the control device12controls the power converter4to supply reactive power to the alternate-current side, and then, when a voltage vcpiapplied to the alternate-current capacitor11has been brought into agreement with a present voltage vsof the alternate-current power supply2, the control device12closes the first alternate-current switch8. Subsequently, as illustrated inFIG.5, in the standby mode, the control device12performs control so that an output current of the power converter4becomes zero. Subsequently, as illustrated inFIG.6, in the SVC operation mode, the control device12controls the power converter4to compensate reactive power on the alternate-current side. Next, charged power of the direct-current capacitor5in the boost charging mode will be described with reference toFIG.7. FIG.7is a diagram for describing charged power of a direct-current capacitor in the power conversion system according to Embodiment 1. InFIG.7, an instantaneous active power p2is expressed by Expression (3) below. [Math.3]p2=νcpi·iR=νcpi(vs-νcpi)Rprc(3) However, in Expression (3), iRis a vector of current flowing in the charging resistance9. On dq coordinates fixed on a present voltage vector vsof the alternate-current power supply2, the present voltage vsof the alternate-current power supply is expressed by Expression (4) below. [Math.4]νs=[νsdo]=[Vs0](4) On the dq coordinates fixed on the present voltage vector vsof the alternate-current power supply2, the voltage vcpiapplied to the alternate-current capacitor11is expressed by Expression (5) below. [Math.5]νcpi=[νcpidνcpiq](5) On the dq coordinates fixed on the present voltage vector vsof the alternate-current power supply2, the instantaneous active power p2is expressed by Expression (6) below. [Math.6]p2=1Rcpi[νcpidνcpiq]·[νsd-νcpidνcpiq]=vcpid(vsd-νcpid)-νcpiq2Rprc(6) Therefore, when the present voltage vector vsof the alternate-current power supply2satisfies Expression (7) below, the instantaneous active power p2reaches a maximum value expressed by Expression (8) below. [Math.7]νcpi=[νcpidνcpiq]=[νsd20](7)[Math.8]p2max=νsd24Rprc(8) In the case of three-phase equilibrium, an instantaneous active power p1is equal to the instantaneous active power p2. Therefore, the charged power of the direct-current capacitor5can be controlled by manipulating the voltage vector Vcpiapplied to the alternate-current capacitor11. Also, power consumption of the charging resistance9can be obtained by Expression (9) below. [Math.9]pR=νs-vcpi24Rprc=(νsd-νcpid)2-vcpid2Rprc(9) The instantaneous active power p2is symmetrical with respect to vsd. When 0<vcpid<vsd/2, the instantaneous active power p1is smaller than power consumption PRof the charging resistance9. When vsd/2<vcpid<vsd, the instantaneous active power p1is larger than the power consumption pRof the charging resistance9. Therefore, for charging of the direct-current capacitor5, it is preferable that vcpidbe within a range of vsd/2<vcpid<vsd. Next, control during the boost charging mode will be described with reference toFIG.8. FIG.8is a diagram for describing control during the boost charging mode in the power conversion system according to Embodiment 1. In many cases, an impedance Lsiof the alternate-current reactor7for a frequency f of the alternate-current power supply2is sufficiently small in comparison with a complex impedance of the electrostatic capacitance Cpiof the alternate-current capacitor11and the resistance value Rpreof the charging resistance9. Therefore, if current iiflowing in the charging resistance9is not so large, the voltage vcpiapplied to the alternate-current capacitor11agrees with an output voltage viof the power converter4. At this time, the control device12sets a voltage instruction value vi* for the power converter4to a value expressed by Expression (10) below. [Math.10]vi*=(1-k2)vs(10) However, k is adjusted within a range of no less than 0 but no more than 1. Where k is 1, the direct-current capacitor5is charged with substantially theoretical maximum power. Where k is 0, the direct-current capacitor5is not charged. In normal charging, in the direct-current capacitor5, charging of the direct-current capacitor5is performed with k set to 1. If charged power is excessively large or if power consumption of the charging resistance9is excessively large, charging of the direct-current capacitor5is performed with k appropriately adjusted. Next, control during the alternate-current voltage synchronization mode will be described with reference toFIG.9. FIG.9is a diagram for describing control during the alternate-current voltage synchronization mode in the power conversion system according to Embodiment 1. In Expression (10), where k is 0, Expression (11) below holds. [Math. 11] vi*=vs≈vcpi(11) At this time, the power converter4supplies only reactive power in order to bring the voltage vcpiapplied to the alternate-current capacitor11into agreement with the present voltage vsof the alternate-current power supply2. In theory, no active power is required and thus the direct-current voltage is not lowered. If the first alternate-current switch8is closed when the voltage vcpiapplied to the alternate-current capacitor11has been brought into agreement with the present voltage vsof the alternate-current power supply2, preparation for SVC operation is completed without generation of inrush current. Next, an overview of operation of the control device12will be described with reference toFIG.10. FIG.10is a flowchart for describing an overview of operation of the control device in the power conversion system according to Embodiment 1. In step S1, the control device12closes the second alternate-current switch10. Subsequently, the control device12performs operation in step S2. In step S2, the control device12controls the power converter4so that a voltage applied to the direct-current capacitor5reaches a value that is equal to or exceeds a present value. Subsequently, the control device12performs operation in step S3. In step S3, the control device12controls the power converter4so that a voltage applied to the alternate-current capacitor11agrees with a voltage of the alternate-current power supply2. Subsequently, the control device12performs operation in step S4. In step S4, the control device12controls the power converter4so that an output current becomes zero simultaneously with closing of the first alternate-current switch8. Subsequently, the control device12performs operation in step S5. In step S5, the control device12controls the power converter4so as to compensate reactive power on the alternate-current side. According to Embodiment 1 described above, the control device12provides a state in which the direct-current switches6are open, the first alternate-current switch8is open, the second alternate-current switch10is closed and the power converter4is stopped, and controls the power converter4so that the voltage applied to the direct-current capacitor5reaches a value that is equal to or exceeds the preset voltage. Therefore, it is possible to perform initial charging of the direct-current capacitor5while suppressing inrush current without using a component having a high rating. Also, the control device12controls the power converter4so that the voltage applied to the alternate-current capacitor11agrees with the voltage of the alternate-current power supply2. Subsequently, the control device12closes the first alternate-current switch8. Therefore, it is possible to suppress inrush current flowing in the alternate-current capacitor11. Note that controlling the power converter4so that the voltage applied to the direct-current capacitor5agrees with the voltage of the direct-current power supply1and then closing the direct-current switches6enables suppressing inrush current flowing in the direct-current capacitor5when closing the direct-current switches6. Also, in a state in which the direct-current switches6are open and the first alternate-current switch8is closed, the control device12controls the power converter4so as to compensate reactive power on the alternate-current side. Therefore, reactive power on the alternate-current side can be compensated with suppression of inrush current at the time of initial charging of the direct-current capacitor5. Note that a storage battery may be used as the direct-current power supply1. In this case, the direct-current switches6may be closed after controlling the power converter4so that the voltage applied to the direct-current capacitor5agrees with a charged voltage of the storage battery, following the standby mode or the alternate-current voltage synchronization mode. In this case, also, inrush current flowing in the direct-current capacitor5when closing the direct-current switches6can be suppressed. Also, as an inrush current suppressor, a charging reactor may be used instead of the charging resistance9. In this case, also, it is possible to perform initial charging of the direct-current capacitor5while suppressing inrush current without using a component having a high rating. Also, in principle, the charging start mode is not necessary. Therefore, the mode may transition to the boost charging mode simultaneously with closure of the second alternate-current switch10. In this case, power consumption of the charging resistance9may be suppressed. Also, in the alternate-current voltage synchronization mode, if inrush current rushing in the alternate-current capacitor11can be allowed, the switch8may be turned on before the voltages of the alternate-current capacitor11and the alternate-current power supply2are brought into agreement with each other. Also, the mode may transition from the alternate-current voltage synchronization mode to the SVC operation mode without involving the standby mode. Also, the direct-current switches6may be left consistently on. Also, the direct-current power supply and the direct-current side of the power converter may directly be connected with no direct-current switches6provided. Also, the direct-current power supply and the direct-current side of the power converter may be connected via a backflow prevention semiconductor device such as a diode, with no direct-current switches6provided. Also, the present invention is applicable even if no alternate-current capacitor11is provided. Also, the alternate-current power supply2may be a single-phase power supply and the power converter4may be a single-phase power converter. In this case, in the boost charging mode, where a present voltage vsof the single-phase power supply satisfies Expression (12) below, the instantaneous active power p2reaches a maximum value expressed by Expression (13) below. [Math.12]νcpi=νs2(12)[Math.13]p2max=vs24Rprc(13) Next, an example of the control device12will be described with reference toFIG.11. FIG.11is a hardware configuration diagram of the control device in the power conversion system according to Embodiment 1. Each of functions of the control device12can be implemented by processing circuitry. For example, the processing circuitry includes at least one processor100aand at least one memory100b. For example, the processing circuitry includes at least one dedicated hardware piece200. Where the processing circuitry includes at least one processor100aand at least one memory100b, each of the functions of the control device12is implemented by software, firmware or a combination of software and firmware. At least one of the software and the firmware is described in the form of programs. At least one of the software and the firmware is stored in the at least one memory100b. The at least one processor100aimplements respective functions of the control device12by reading and executing the programs stored in the at least one memory100b. The at least one processor100ais also referred to as a central processing unit, a processing device, an arithmetic device, a microprocessor, a microcomputer or a DSP. For example, the at least one memory100bis a non-volatile or volatile semiconductor memory such as a RAM, a ROM, a flash memory, an EPROM or a EEPROM, a magnetic disk, a flexible disk, an optical disk, a compact disc, a mini-disk or a DVD. Where the processing circuitry includes the at least one dedicated hardware piece200, the processing circuitry is implemented by, for example, a single circuit, a complex circuit, a programmed processor, a parallel-programmed processor, an ASIC, an FPGA or any of combinations thereof. For example, the functions of the control device12are implemented by respective processing circuits. For example, the functions of the control device12are collectively implemented by the processing circuitry. Some of the functions of the control device12may be implemented by the dedicated hardware piece200and others of the functions may be implemented by the software or the firmware. For example, a function that controls the power converter4may be implemented by the processing circuitry serving as the dedicated hardware piece200and the functions other than the function that controls the power converter4may be implemented by the at least one processor100areading and executing programs stored in the at least one memory100b. As described above, the processing circuitry implements each of the functions of the control device12by means of the hardware piece200, the software, the firmware or any of combinations thereof. INDUSTRIAL APPLICABILITY As described above, a power conversion system according to the present invention can be used for a system that suppresses inrush current. REFERENCE SIGNS LIST 1direct-current power supply,2alternate-current power supply,3transformer,4power converter,5direct-current capacitor,6direct-current switch,7alternate-current reactor,8first alternate-current switch,9charging resistance,10second alternate-current switch,11alternate-current capacitor,12control device,100aprocessor,100bmemory,200hardware piece | 18,877 |
11863064 | DETAILED DESCRIPTION The following description provides a non-regulated power converter having a plurality of STC modules configured to be connected in parallel to a load, where each STC module is configured to deliver, over time, substantially the same output current despite differences in STC resonant output impedance, differences in tank resonant frequencies, different temperature distributions, and/or the effects of parasitic elements (e.g., resistance, inductance, or capacitance). As a result, the overall efficiency of the non-regulated power converter may be improved, and the risk of damage to the STC module(s) may be reduced or eliminated. The non-regulated power converter includes a current share (CS) interface connected to the plurality of STC modules. In some examples, each STC module is configured to communicate with the CS interface to obtain a value of the minimum output current provided by the STC modules. If a first STC module generates a first output current, a second STC module generates a second output current, and a third STC module generates a third output current, the value of the minimum output current is the minimum of the first, second, and third output currents. In some examples, the CS interface includes an analog implementation, where the CS interface is a dedicated analog pin connected to the STC modules, and the analog voltage on the dedicated pin indicates the value of the minimum output current. In some examples, the CS interface includes a digital implementation, where the CS interface is one or more digital (serial or parallel) communication lines between the STC modules, where the value of the minimum output current is digitally exchanged among the STC modules. Each STC module may include a STC circuit (e.g., including one or more resonant tanks), an output current (OC) measuring circuit, a current share (CS) circuitry, a dead time (DT) adjuster, and a control logic. The STC circuit may include one or more resonant capacitors, one or more resonant inductors, one or more filtering capacitors, and a plurality of switches (e.g., transistors). The switches may be controlled by control signals generated by the control logic. Also, the activation/deactivation of the switches may control the STC phases of an STC cycle of a respective STC module, where each STC phase generates a different resonant frequency in the resonant tank(s). In some examples, a particular STC cycle includes four STC phases, e.g., a first STC phase, a second STC phase, a third STC phase, and a fourth STC phase. In some examples, during an individual STC cycle, the first and third STC phases are activated simultaneously, which is followed by the simultaneous activation of the second and fourth STC phases. Each STC module may provide (and/or update) the value of the minimum output current on the CS interface. For example, the CS circuitry of a respective STC module may compare the value of its output current to the value of the minimum output current on the CS interface, and, if its output current is less than the minimum output current, the CS circuitry may update the value of the minimum output current with the value of the output current. In some examples, the CS circuitry includes one or more analog components. In some examples, the CS circuitry includes one or more digital components. Each STC module may obtain the value of the minimum output current from the CS interface and compare the value of the minimum output current with its respective output current. For example, the OC measuring circuit of a respective STC module may measure its average output current. If the value of its respective output current is greater than the value of the minimum output current from the CS interface, the STC module may adjust a dead time during the STC cycles. The dead time may be a period of time when the switches of the STC circuit are deactivated. The DT adjustor of a respective STC module may receive the output current from the OC measuring circuit and the minimum output current from the CS interface and adjust the dead time such that the dead time is increased at the end of an STC phase and/or the end of an STC cycle (e.g., after the deactivation of the first/third STC phases, and/or after the deactivation of the second/fourth STC phases). The control logic may receive the updated dead time, and then generate the appropriate control signals for the switches of the STC circuit such that the updated deadtime is implemented at the end of an STC phase and/or STC cycle. The increase of the dead time may cause a reduction in the output current provided by the stronger STC module so that the output currents provided by the STC modules converge towards each other. FIG.1Aillustrates a non-regulated power converter100having a plurality of switching tank converter (STC) modules102according to an aspect.FIG.1Billustrates an example of a first STC module102-1according to an aspect. The non-regulated power converter100may control the output currents provided by the STC modules102such that each STC module102is configured to deliver, over time, substantially the same current to a load104(e.g., IOUT1, IOUT2, and IOUT3being substantially the same). For example, for an STC module102that is providing more output current than other STC modules102, a current share (CS) dead time117may be added to a dead time116at the end of an STC cycle and/or end of an STC phase, where the addition of the CS dead time117may cause a reduction in the output current provided by the stronger STC module102so that the output currents provided by the STC modules102converge towards each other. The non-regulated power converter100may receive an input voltage (V-IN) and generate an output voltage (VOUT). The non-regulated power converter100may also be referred to as an STC converter. The STC converter may not include a voltage feedback loop. The non-regulated power converter100may scale down the input voltage (VIN) by a ratio N, where N is an integer number and may depend on the circuit topology of the non-regulated power converter100(e.g., the output voltage (VOUT) is VIN/N). For example, if the value of N is four, the output voltage (VOUT) of the non-regulated power converter100is VIN/4. However, the value of N may encompass any integer. As shown inFIG.1A, the STC modules102are connected in parallel with respect to each other. For example, each of the STC modules102receives the input voltage (VIN), and the outputs of the STC modules102are coupled to the load104. Each STC module102may be considered a distinct power stage that converts the input voltage (VIN) to a respective output voltage. The STC modules102may include a first STC module102-1, a second STC module102-2, and a third STC module102-3. Although three STC modules102are shown with respect toFIG.1A, the non-regulated power converter100may include any number of STC modules102such as two STC modules102or more than three STC modules102connected in parallel. The first STC module102-1may receive the input voltage (VIN) and convert the input voltage (VIN) to a first output voltage (VOUT1) with a first output current (IOUT1). The second STC module102-2may receive the input voltage (VIN) and convert the input voltage (VIN) to a second output voltage (VOUT2) with a second output current (IOUT2). The third STC module102-3may receive the input voltage (VIN) and convert the input voltage (VIN) to a third output voltage (VOUT3). As further discussed below, the non-regulated power converter100may implement an output current control mechanism that enables the output currents (e.g., IOUT1, IOUT2, IOUT3) to be substantially equal to each other. In some examples, the output current control mechanism is activated in response to a current share (CS) enable signal (ENCS). The details of the first STC module102-1are shown with respect toFIG.1B. However, it is noted that the second STC module102-2and the third STC module102-3(or any other STC module102) may be the same as the first STC module102-1and may include any of the details as discussed herein. Each STC module102may receive, via a current share (CS) interface118, a value of a minimum output current (IOUT_MIN). The value of the minimum output current (IOUT_MIN) may indicate the value of the minimum output current provided by the STC modules102(e.g., the value of the minimum of IOUT1, IOUT2, IOUT3). Each STC module102may compare the value of the minimum output current (IOUT_MIN) to its respective output current. For example, the first STC module102-1compares the value of the minimum output current (IOUT_MIN) with the value of the first output current (IOUT1). The second STC module102-2compares the value of the minimum output current (IOUT_MIN) with the value of the second output current (IOUT2). The third STC module102-3compares the value of the minimum output current (IOUT_MIN) with the value of the third output current (IOUT3). With respect to a particular STC module, if the output current is greater than the minimum output current (IOUT_MIN), the STC module102may adjust the dead time116so that currents provided by the STC modules102are more equally balanced. In some examples, the dead time116is a default dead time115, but when the value of the output current is greater than the value of the minimum output current (IOUT_MIN), the dead time116is updated to include the CS dead time117plus the default dead time115. When STC modules102are connected in parallel to boost current capability, due to the differences in STC resonant output impedance, tank resonant frequency, and/or because of parasitic elements in the non-regulated power converter100, one or more STC modules102may provide current higher than the current provided by other STC modules102. However, according to the embodiments discussed herein, if a particular STC module102is generating an output current greater than the minimum output current (IOUT_MIN), in order to decrease the amount of output current for the stronger STC module102, the STC module102may increase the dead time116(e.g., add the CS dead time117to the default dead time115). In some examples, the STC module102may increase the dead time116at the end of an STC cycle (e.g., between STC cycles) and/or at the end of each STC phase, which may cause the output current to decrease towards the minimum output current (IOUT_MIN) and/or increase the minimum output current (IOUT_MIN) such that all the STC modules102provide substantially the same current. For example, the first STC module102-1may receive the value of the minimum output current (IOUT_MIN) via the CS interface118. In some examples, the value of the minimum output current (IOUT_MIN) is analog data (e.g., within an analog format). In some examples, the value of the minimum output current (IOUT_MIN) is digital data (e.g., within a digital format). The value of the minimum output current (IOUT_MIN) indicates the value of the minimum output current provided by the STC modules102(e.g., minimum of the first output current (IOUT1), the second output current (IOUT2), and the third output current (IOUT3)). In other words, the minimum output current (IOUT_MIN) may be the value of the minimum output current supplied to the load104by the weakest STC module102. For example, if the first output current (IOUT1) is25A, the second output current (IOUT2) is24A, and the third output current (IOUT3) is23A, the minimum output current (IOUT_MIN) is23A (e.g., where the third STC module102-3is considered the weakest STC module102). The CS interface118may be one or more components (e.g., analog and/or digital) configured to share the minimum output current (IOUT_MIN) with each of the STC modules102. In some examples, the CS interface118includes one or more analog components configured to provide the minimum output current (IOUT_MIN) in an analog format to the STC modules102. In some examples, the CS interface118includes a dedicated pin (e.g., an external pin or a pin external to the STC modules102), where each STC module102is connected to the dedicated pin. The voltage level (e.g., analog voltage level) on the dedicated pin may indicate the amount of the minimum output current (IOUT_MIN). In some examples, the first STC module102-1may obtain the minimum output current (IOUT_MIN) from the dedicated pin. In some examples, the CS interface118includes one or more digital components configured to provide the minimum output current (IOUT_MIN) in a digital format to the STC modules102. In some examples, the CS interface118includes one or more digital communication lines that share the minimum output current (IOUT_MIN) among the STC modules102. In some examples, the digital communication line(s) define a digital serial interface. In some examples, the digital communication line(s) define a parallel interface. The first STC module102-1, the second STC module102-2, and the third STC module102-3may be connected to each other via the digital communication line(s). In further detail, the first STC module102-1may receive the minimum output current (IOUT_MIN) from the third STC module102-3via the digital communication line(s), and if the value of the first output current (IOUT1) is less than the value of the minimum output current (IOUT_MIN), the first STC module102-1may update the value of the minimum output current (IOUT_MIN) with the value of the first output current (IOUT1). Then, the second STC module102-2may receive the minimum output current (IOUT_MIN) from the first STC module102-1via the digital communication line(s), and, if the value of the second output current (IOUT2) is less than the value of the minimum output current (IOUT_MIN), the second STC module102-2may update the value of the minimum output current (IOUT_MIN) with the value of the second output current (IOUT2). Then, the third STC module102-3may receive the minimum output current (IOUT_MIN) from the second STC module102-2via the digital communication line(s), and, if the value of the third output current (IOUT3) is less than the value of the minimum output current (IOUT_MIN), the third STC module102-3updates the value of the minimum output current (IOUT_MIN) with the value of the third output current (IOUT3), and the process continues as so forth. As shown inFIG.1B, the first STC module102-1includes a STC circuit106that receives the input voltage (VIN) and generates the first output voltage (VOUT1) with the first output current (IOUT1). The STC circuit106may include one or more resonant tanks that absorb power at one or more frequencies (e.g., resonant frequencies). The STC circuit106may include one or more resonant capacitors103, one or more resonant inductors105, and switches107. The STC circuit106may define any number of resonant branches. The switches107may be activated (e.g., conducting) or deactivated (e.g., not conducting) based on control signals121, which controls the STC phases of the STC cycles. The first STC module102-1includes an output current (OC) measuring circuit108, a current comparator110, current share (CS) circuitry112, a dead time (DT) adjustor114, and control logic120. The OC measuring circuit108is coupled to the STC circuit106. In some examples, the OC measuring circuit108is coupled to an output node of the STC circuit106. The OC measuring circuit108may measure (or detect) an average of the first output current (IOUT1). The OC measuring circuit108may include one or more electrical components configured to obtain the level (or average) of the first output current (IOUT1). In some examples, the OC measuring circuit108includes a resistor (e.g., a shunt resistor). In some examples, the OC measuring circuit108includes a current sense circuit. For example, the OC measuring circuit108may read the drain-to-source voltage of switching transistor(s) connected to the output voltage or ground and generate a signal proportional to the STC output current. The CS circuitry112may be coupled to the OC measuring circuit108. The CS circuitry112may communicate with the CS interface118to obtain the value of the minimum output current (IOUT_MIN). In some examples, the CS circuitry112may obtain the value of the minimum output current (IOUT_MIN) from a memory device at the CS circuitry112. In some examples, the CS circuitry112may determine whether the first output current (IOUT1) is the minimum output current provided by the STC modules102, and, if so, may provide the first output current (IOUT1) as the value of the minimum output current (IOUT_MIN) on the CS interface118. In some examples, the CS circuitry112includes one or more analog components such as a comparator and one or more transistors. In some examples, the CS circuitry112includes one or more digital components such as one or more digital processors and one or more memory devices that include executable instructions that when executed by the digital processors are configured to evaluate whether the first output current (IOUT1) is greater than the minimum output current (IOUT_MIN), and, depending on the results, provide or update the minimum output current (IOUT_MIN) on the CS interface118. The DT adjustor114may receive the minimum output current (IOUT_MIN) from the CS interface118and the first output current (IOUT1) from the OC measuring circuit108. The DT adjustor114may compare the value of the first output current (IOUT1) with the value of the minimum output current (IOUT_MIN). If the results of the comparison indicate that the value of the first output current (IOUT1) is greater than the value of the minimum output current (IOUT_MIN), the DT adjustor114may adjust the dead time116to be implemented at the end of the next STC phase or end of the next STC cycle. If the results of the comparison indicate that the value of the first output current (IOUT1) is equal to or less than the value of the minimum output current (IOUT_MIN), the DT adjustor114may not adjust the dead time116. In some examples, if the value of the first output current (IOUT1) is greater than the value of the minimum output current (IOUT_MIN), the DT adjustor114may determine (or compute) the CS dead time117. The DT adjustor114may compute the CS dead time117based on the difference between the value of the first output current (IOUT1) and the value of the minimum output current (IOUT_MIN). In some examples, the DT adjustor114may compute the CS dead time117based on the following equation: CS DT=K*(IOUT1−IOUT_MIN), where the parameter K is a scaling parameter. As such, the value of the CS dead time117may be proportional to the difference between the first output current (IOUT1) and the minimum output current (IOUT_MIN) (e.g., the larger the difference, the larger the CS dead time117). The CS dead time117may be an additional period of time, which is added to the default dead time115. For example, if the value of the first output current (IOUT1) is equal to or less than the value of the minimum output current (IOUT_MIN), the dead time116includes only the default dead time115(e.g., no adjustment takes place). However, if the value of the first output current (IOUT1) is greater than the value of the minimum output current (IOUT_MIN), the dead time116includes the default dead time115plus the computed CS dead time117. In some examples, the DT adjustor114includes an analog circuit having one or more analog components such as an operational transconductance amplifier (OTA) that sinks current from a (dead time) delay ramp. In some examples, the DT adjustor114includes comparator(s), transistor(s), capacitor(s), delay(s), and/or other analog components. In some examples, the DT adjustor114includes a digital circuit having one or more digital components such as register(s), analog-to-digital converter(s) (ADCs), serial to parallel (S/P) converter(s), parallel to serial (P/S) converter(s), and/or one or more digital processors. In some examples, the DT adjustor114includes a combination of one or more analog components and one or more digital components. The control logic120may generate control signals121to enable or disable the STC phases of the STC cycles of the first STC module102-1. In some examples, the control logic120(and/or the DT adjustor114) receives the CS enable signal (ENCS) which activates the current share control mechanism discussed herein. For example, if the CS enable signal (ENCS) is in a first state (e.g., a low state), the current share control mechanism is deactivated, and the dead time116includes the fixed value of the default dead time115. However, if the CS enable signal (ENCS) is in a second state (e.g., a high state), the current share control mechanism is activated, and the dead time116may be adjusted to additionally include the CS dead time117. In some examples, the control signals121may activate/deactivate the switches107of the STC circuit106to control the activation/deactivation of the STC phases. The control logic120may be connected to the DT adjustor114. Also, in some examples, it is noted that some functionality of the DT adjustor114may be included in the control logic120(or vice versa). The control logic120receives the dead time116from the DT adjustor114and is configured to generate the control signals121to control the switches107such that the updated dead time116is inserted between STC cycles or between STC phases of an STC cycle. FIG.2illustrates an example of a STC circuit206according to an aspect. The STC circuit206may be an example of the STC circuit106ofFIG.1Band may include any of the detailed discussed herein. The STC circuit206is not a voltage regulator (e.g., a non-regulated power converter), and therefore does not include a voltage feedback loop. The STC circuit206may convert the input voltage (VIN) by a ratio N, where N is an integer number that depends on the circuit topology. In some examples, N is four (e.g., VOUT=VIN/4). However, N may be any integer. As shown inFIG.2, the STC circuit206includes a resonant capacitor244(C1), a resonant inductor246(L1), a filtering capacitor248(C2), a resonant capacitor250(C3), and a resonant inductor252(L2). Also, the STC circuit206includes switch224(Q1), switch226(Q2), switch228(Q3), switch230(Q4), switch232(Q5), switch234(Q6), switch236(Q7), switch238(Q8), switch240(Q9), and switch242(Q10). Each of the switches may be a transistor (e.g., a field-effect transistor (FET), a metal-oxide-semiconductor FET (MOSFET), a bipolar junction transistor (BJT), or other types of transistors). In some examples, the gates of the switches are configured to receive control signal (e.g. the control signals121ofFIG.1B) to activate or deactivate the switches. The STC circuit206may operate in a number of STC phases such as a first STC phase, a second STC phase, a third STC phase and a fourth STC phase, which may be controlled by the switches inFIG.2. In the first STC phase, the switch224(Q1), and the switch232(Q5) are activated. In the second STC phase, the switch226(Q2), the switch234(Q6), and the switch236(Q7) are activated. In the third STC phase, the switch228(Q3), the switch238(Q8), and the switch240(Q9) are activated. In the fourth STC phase, the switch230(Q4) and the switch242(Q10) are activated. In some examples, the first STC phase and the third STC phase are simultaneously activated. In some examples, the second STC phase and the fourth STC phase are simultaneously activated. The resonant frequencies of the STC circuit206may be Frequency (Phase 1)=1/(2*pi*sqrt(C1*L1)), Frequency (Phase 2)=1/(2*pi*sqrt((C1*C2/(C1+C2))*L1)), Frequency (Phase 3)=1/(2*pi*sqrt((C3*C2/(C3+C2))*L2)), Frequency (Phase 4)=1/(2*pi*sqrt(C3*L2)). In the example ofFIG.2, the STC circuit206has a ratio value (N) of 4, where the input voltage (VIN) is divided by 4 (e.g., the first output voltage (VOUT1) is ¼ of the input voltage (VIN)). However, the techniques discussed herein may be applied to an STC circuit206having any ratio value N. An STC cycle includes the activation of the first STC phase and the third STC phase (at the same time) for a first period of time followed by the activation of the second STC phase and the fourth STC phase (at the same time) for a second period of time. In some examples, a dead time (e.g., the dead time116ofFIG.1B) exists between the first period of time and the second period of time (e.g., a dead time between STC phases within a single STC cycle). During the dead time, all switches in the STC circuit206are deactivated. In some examples, during the dead time, one or more of the switches in the STC circuit206are deactivated (but one or more switches remain activated). For example, the dead time may be inserted to avoid cross conduction and allow current to be properly discharged to OA (to enable zero current switching and/or zero voltage switching, thereby providing high efficiency). Also, in some examples, a dead time (e.g., the dead time116ofFIG.1B) exists before the activation of a subsequent STC cycle (e.g., a dead time between consecutive STC cycles). As further discussed below, referring toFIG.1B, in response to the value of the first output voltage (VOUT1) being greater than the value of the minimum output current (IOUT_MIN), the first STC module102-1may increase the dead time116between the STC phases and/or increase the dead time116at the end of the STC cycle (and before the subsequent STC cycle). FIG.3Aillustrates a digital implementation of a current share (CS) interface318among a first STC module302-1, a second STC module302-2, and a third STC module302-3.FIG.3Billustrates an example of a current share (CS) circuitry312of the first STC module302-1according to an aspect. The CS interface318may be an example of the CS interface118ofFIG.1A, and the first STC module302-1, the second STC module302-2, and the third STC module302-3may be the first STC module102-1, the second STC module102-2, and the third STC module102-3ofFIG.1A, respectively, and may include any of the details discussed herein. The CS interface318may be a digital serial interface that enables the first STC module302-1, the second STC module302-2, and the third STC module302-3to communicate in order to share the minimum output current (IOUT_MIN). The CS interface318includes one or more digital communication lines319connected between the first STC module302-1, the second STC module302-2, and the third STC module302-3. In some examples, the digital communication lines319define a digital serial interface that is configured to serially transmit digital data among the STC modules. With respect toFIG.3B, the CS circuitry312includes a memory device323and one or more digital processors325configured to execute operations of the CS circuitry312. For example, in operation301, the CS circuitry312may determine whether the value of the first output current (IOUT1) is greater than the value of the minimum output current (IOUT_MIN). If yes, in operation303, the CS circuitry312may store the value of the minimum output current (IOUT_MIN) in the memory device323, and, in operation305, the CS circuitry312may transfer the digital value of the minimum output current (IOUT_MIN), via the digital communication line(s)319, to the second STC module302-2. If no, in operation307, the CS circuitry312may update the value of the minimum output current (IOUT_MIN) with the first output current (IOUT1), and, in operation309, the CS circuitry312may store the updated value of the minimum output current (IOUT_MIN) in the memory device323. Then, in operation311, the CS circuitry312may transfer the updated digital value of the minimum output current (IOUT_MIN), via the digital communication line(s)319, to the second STC module302-2. The second STC module302-2may include the CS circuitry312and may perform the same operations, which, as shown inFIG.3A, the second STC module302-2may transfer the digital value of the minimum output current (IOUT_MIN) to the third STC module302-3via the digital communication line(s)319. The third STC module302-3may include the CS circuitry312and may perform the same operations, which, as shown inFIG.3A, the third STC module302-3may transfer the digital value of the minimum output current (IOUT_MIN) to the first STC module302-1via the digital communication line(s)319. FIG.4illustrates an analog implementation of a CS interface418according to an aspect. As shown inFIG.4, the CS interface418may include a dedicated analog pin419, which is connected to a voltage source461(VCC) via a resistor460. The analog voltage on the dedicated analog pin419may indicate the value of the minimum output current (IOUT_MIN). Each STC module (e.g., each STC module102ofFIG.1A) is connected to the dedicated analog pin419. Each STC module is configured to determine whether the value of its respective output current is less than the value of the current minimum output current (IOUT_MIN), and, if so, provide its output current as the value of the minimum output current (IOUT_MIN) on the dedicated analog pin419. For example, a first STC module (e.g., the first STC module102-1ofFIG.1A) includes a first CS circuitry412-1, a second STC module (e.g., the second STC module102-2ofFIG.1A) includes a second CS circuitry412-2, and a third STC module (e.g., the third STC module102-3ofFIG.1A) includes a third CS circuitry412-3. Each of the first CS circuitry412-1, the second CS circuitry412-2, and the third CS circuitry412-3is connected to the dedicated analog pin419. The first CS circuitry412-1, the second CS circuitry412-2, or the third CS circuitry412-3may update the value of the minimum output current (IOUT_MIN), e.g., the voltage on the dedicated analog pin419. For instance, if the first output current (IOUT1) is less than the voltage on the dedicated analog pin419, the first CS circuitry412-1may update the voltage on the dedicated analog pin419to correspond to (e.g., to be the same with) the voltage of the first output current (IOUT1). If the second output current (IOUT2) is less than the voltage on the dedicated analog pin419, the second CS circuitry412-2may update the voltage on the dedicated analog pin419to correspond to (e.g., to be the same with) the voltage of the second output current (IOUT2). If the third output current (IOUT3) is less than the voltage on the dedicated analog pin419, the third CS circuitry412-3may update the voltage on the dedicated analog pin419to correspond to (e.g., to be the same with) the voltage of the third output current (IOUT3). The first CS circuitry412-1includes an operational transconductance amplifier (OTA)456-1and a transistor458-1. The OTA456-1receives and compares the voltage on the dedicated analog pin419with the voltage of a first output current (IOUT1). The output of the OTA456-1is connected to a gate of the transistor458-1. In some examples, the transistor458-1is a P-channel transistor. The source of the transistor458-1is connected to the dedicated analog pin419. The drain of the transistor458-1may be connected to ground. If the voltage of the first output current (IOUT1) is less than the voltage on the dedicated analog pin419, the OTA456-1may activate the transistor458-1so that the voltage of the first output current (IOUT1) is the voltage on the dedicated analog pin419. The second CS circuitry412-2includes an OTA456-2and a transistor458-2. The OTA456-2receives and compares the voltage on the dedicated analog pin419with the voltage of a second output current (IOUT2). The output of the OTA456-2is connected to a gate of the transistor458-2. In some examples, the transistor458-2is a P-channel transistor. The source of the transistor458-2is connected to the dedicated analog pin419. The drain of the transistor458-2may be connected to ground. If the voltage of the second output current (IOUT2) is less than the voltage on the dedicated analog pin419, the OTA456-2may activate the transistor458-2so that the voltage of the second output current (IOUT2) is the voltage on the dedicated analog pin419. The third CS circuitry412-3includes an OTA456-3and a transistor458-3. The OTA456-3receives and compares the voltage on the dedicated analog pin419with the voltage of a third output current (IOUT3). The output of the OTA456-3is connected to a gate of the transistor458-3. In some examples, the transistor458-3is a P-channel transistor. The source of the transistor458-3is connected to the dedicated analog pin419. The drain of the transistor458-3may be connected to the ground. If the voltage of the third output current (IOUT3) is less than the voltage on the dedicated analog pin419, the OTA456-3may activate the transistor458-3so that the voltage of the third output current (IOUT3) is the voltage on the dedicated analog pin419. FIG.5illustrates a flowchart500depicting example operations of a STC module according to an aspect. Although the operations ofFIG.5are explained with reference to the first SCT module102-1ofFIGS.1A and1B, the operations may be applicable to any of the STC modules discussed herein. In operation502, the first STC module102-1may measure the average output current (IOUT1) generated by the first STC module102-1. In operation504, the first STC module102-1may obtain the value of the minimum output current (IOUT_MIN) via the CS interface118. In operation506, the first STC module102-1may determine if the CS enable signal (ENCS) is received. If no, in operation510, the dead time116is the default dead time115. If yes, in operation508, the first STC module102-1determines whether the value of the first output current (IOUT1) is greater than the value of the minimum output current (IOUT_MIN), and the operations return to operation508to re-determine (e.g., periodically or continuously) whether the value of the first output current (IOUT1) is greater than the value of the minimum output current (IOUT_MIN). If yes, in operation512, the first STC module102-1computes the CS dead time117. In some examples, the first STC module102-1computes the CS dead time117based on the difference between the value of the first output current (IOUT1) and the value of the minimum output current (IOUT_MIN), which is multiple by a parameter K (e.g., K*(IOUT1−IOUT_MIN). In operation514, the first STC module102-1adjusts the dead time116by adding the CS dead time117to the default dead time115. FIG.6illustrates control signals621of a control logic620without the current share mechanism enabled according to an aspect. For example, the control signals621may depict the enabling of the STC phases in response to the CS control signal (ENCS) being disabled. The control logic620may be an example of the control logic120ofFIG.1B. In some examples, with respect to a particular STC module, the first STC phase and the third STC phase may be activated simultaneously, and the second STC phase and the fourth STC phase may be activated simultaneously. The control signals621are provided to the STC circuit (e.g., the STC circuit106ofFIG.1B) to control the switches (e.g., the switches107ofFIG.1B) to implement the various STC phases of the STC module. For example, with respect to the STC circuit206ofFIG.2, the control signals621may include signals to control the transistors (Q1through Q10) to implement the first through fourth STC phases. An STC cycle may include the activation of the first and third STC phases followed by the activation of the second and fourth STC phases, which is followed by another STC cycle. As shown inFIG.6, a default dead time615-1(e.g., the default dead time115ofFIG.1B) may be provided within a particular STC cycle, e.g., after the period of time in which the first and third STC phases are activated but before activation of the second and fourth STC phases. In addition, in some examples, a default dead time615-2(e.g., the default dead time115ofFIG.1B) may be provided at the end of a particular STC cycle but before the activation of a subsequent STC cycle, e.g., after the activation of the second and fourth STC phases before the activation of the first and third STC phases in the next STC phase. FIG.7Aillustrates control signals721for a control logic720with the current sharing mechanism enabled according to an aspect. The control logic720may be an example of the control logic120ofFIG.1A. The control signals721may depict the enabling of the STC phases in response to the CS control signal (ENCS) being enabled according to an aspect. With respect to a particular STC module, the first STC phase and the third STC phase may be activated simultaneously, and the second STC phase and the fourth STC phase may be activated simultaneously. The control signals721are provided to the STC circuit (e.g., the STC circuit106ofFIG.1B) to control the switches (e.g., the switches107ofFIG.1B) to implement the various STC phases of the STC module. For example, with respect to the STC circuit206ofFIG.2, the control signals721may include signals to control the transistors (Q1through Q10) to implement the first through fourth STC phases. An STC cycle may include the activation of the first and third STC phases followed by the activation of the second and fourth STC phases, which is followed by another STC cycle. As shown inFIG.7A, a current share (CS) dead time717(e.g., the CS dead time117ofFIG.1B) is added to a default dead time715(e.g., the default dead time115ofFIG.1B) at the end of an STC cycle. For example, after the disabling (or deactivation) of the second and fourth STC cycles, the updated dead time includes the CS dead time717plus the default dead time715. FIG.7Billustrates a graph700depicting current waveforms of two STC modules within its resonant tank according to an aspect. For example, the current waveforms ofFIG.7Brelate to the addition of the CS dead time at the end of the STC cycle as shown inFIG.7A. The graph700includes a first current waveform701generated by a first STC module (e.g., the first STC module102-1ofFIGS.1A and1B), and a second current waveform703generated by a second STC module (e.g., the second STC module102-2ofFIG.1A). In the example ofFIG.7B, the first STC module has added the CS dead time at the end of the STC cycle. In addition, despite the addition of the CS dead time, the first STC module may still enable zero-current switching (ZCS) as indicated by points705,707,709, and711, thereby increasing the efficiency of the power converter. FIG.8Aillustrates control signals821for a control logic820with current sharing mechanism enabled according to an aspect. The control logic820may be an example of the control logic120ofFIG.1A. The control signals821may depict the enabling of the STC phases in response to the CS control signal (ENCS) being enabled according to an aspect, where a CS dead time is added after each STC phase (e.g., as opposed to being added just at the end of the STC cycle as shown inFIG.7A). With respect to a particular STC module, the first STC phase and the third STC phase may be activated simultaneously, and the second STC phase and the fourth STC phase may be activated simultaneously. The control signals821are provided to the STC circuit (e.g., the STC circuit106ofFIG.1B) to control the switches (e.g., the switches107ofFIG.1B) to implement the various STC phases of the STC module. For example, with respect to the STC circuit206ofFIG.2, the control signals821may include signals to control the transistors (Q1through Q10) to implement the first through fourth STC phases. An STC cycle may include the activation of the first and third STC phases followed by the activation of the second and fourth STC phases, which is followed by another STC cycle. As shown inFIG.8A, a CS dead time is added to a default dead time815at the end of each STC phase. In some examples, the control logic820may divide (e.g., split) the computed CS dead time into a CS dead time817-1and a CS dead time817-2. As shown inFIG.8A, the CS dead time817-1may be added to the default dead time815after the deactivation of the first and third STC phases, and the CS dead time817-2may be added to the default dead time815after the deactivation of the second and fourth STC phases. FIG.8Billustrates a graph800depicting current waveforms of two STC modules within its resonant tank according to an aspect. For example, the current waveforms ofFIG.8Brelate to the addition of the CS dead time at the end of each STC phase as shown inFIG.8A. The graph800includes a first current waveform801generated by a first STC module (e.g., the first STC module102-1ofFIGS.1A and1B), and a second current waveform803generated by a second STC module (e.g., the second STC module102-2ofFIG.1A). In the example ofFIG.8B, the first STC module has added the CS dead time at the end of each STC cycle. In addition, despite the addition of the CS dead time, the first STC module may still enable zero-current switching (ZCS) as indicated by points805,807, and809, thereby increasing the efficiency of the power converter. FIG.9illustrates an example of an analog implementation of a DT adjustor914according to an aspect. The DT adjustor914may be an example of the DT adjustor114ofFIG.1B. Generally, the DT adjustor914may compare the value of the first output current (IOUT1) with the value of the minimum output current (IOUT_MIN), and, if the first output current (IOUT1) is greater than the minimum output current (IOUT_MIN), the DT adjustor914adjusts the dead time (DT) based on the difference of the first output current (IOUT1) and the minimum output current (IOUT_MIN) for the next STC cycle (e.g., ENPHASE_X+1). In some examples, the DT adjustor914is configured to increase the dead time at the end of each STC phase. In some examples, the DT adjustor914is a circuit that can adjust the deadtime with an OTA sinking current from dead time delay ramp. The DT adjustor914includes an OTA976, a transistor974, a current source966, a voltage source962, a transistor972, a capacitor970, and a Schmitt trigger968. The OTA976compares the voltage of the minimum output current (IOUT_MIN) (referred to as V(IOUT_MIN)) and the voltage of the first output current (IOUT1) (referred to as V(IOUT_MIN)). The output of the OTA976is coupled to the gate of the transistor974. The drain of the transistor974is connected to the current source966, and the source of the transistor974may be coupled to a ground964. In some examples, the transistor974is an N-channel transistor. The current source966is connected to the voltage source962and is configured to provide a bias current (IBIAS). If V(IOUT_MIN) is greater than V(IOUT_MIN), the OTA976activates the transistor974. The gate of the transistor972receives a phase enable signal (ENPHASE_X), the source of the transistor972is coupled to the ground964, and the drain of the transistor972is connected to the current source966. The capacitor970is connected to the input of the Schmitt trigger968and the ground964. The input of the Schmitt trigger968is connected to the current source966and the capacitor970. The output of the Schmitt trigger968is a next phase enable signal (ENPHASE_X+1), which is delayed by the CS dead time. The output of the Schmitt trigger968may transition to a logic high state in response to the voltage on the capacitor970reaching a certain Schmitt trigger threshold VTH. The time from when the voltage on the capacitor970reaches the Schmitt trigger threshold VTHmay be DT=C*VTH/ICHARGE, and ICHARGE=(IBIAS−ICURRENT_SHARING). The current ICURRENT_SHARINGis the current sinked by the transistor974, which is K*(V(IOUT1)−V(IOUT_MIN)) when V(IOUT1) is higher than V(IOUT_MIN). Therefore, DTCS(current sense deadtime)=C*VTH/(IBIAS−K*(V(IOUT1)−V(IOUT_MIN)). This DT value may increase in response to the V(IOUT1) being different from V(IOUT_MIN). Also, no changes may occur on the DT in response to the V(IOUT1) being less than V(IOUT_MIN). In this case, the DT is the default, which is DTDEFAULT=C*VTH/(IBIAS). This timing may start in response to the signal ENPHASE_Xtransitioning to a logic low state, where the capacitor970is enabled to be charged because the transistor972is deactivated (e.g., turned-off), and the DT adjustor914can generate the enable signal (ENPHASE_X+1) for the next STC phase. FIG.10illustrates an example of a digital implementation of a DT adjustor1014. The DT adjustor1014may be an example of the DT adjustor114ofFIG.1Band may include any of the details discussed herein. In some examples, the DT adjustor1014is a digital finite state machine. The DT adjustor1014includes a parallel to serial (P/S) converter1082, a serial to parallel (S/P) converter1084, a first register1088, an analog-to-digital converter (ADC)1086, a second register1090, a digital processor1092, and a phase generator1094. The S/P converter1084may receive the value of the minimum output current (IOUT_MIN) via the CS interface (e.g., the digital communication line(s)319ofFIG.3A). In some examples, the S/P converter1084is connected to the digital communication lines, which includes serial digital (e.g., N-bits) data representing the value of the minimum output current (IOUT_MIN). The S/P converter1084converts the digital value of the minimum output current (IOUT_MIN) from the serial format to the parallel format. The first register1088stores the N-bits of the digital value of the minimum output current (IOUT_MIN). The ADC1086may receive the first output current (IOUT1) which is an analog format. For example, the ADC1086may be connected to the OC measuring circuit (e.g., the OC measuring circuit108ofFIG.1B). The ADC1086converts the analog value of the first output current (IOUT1) to a digital value. The second register1090is connected to the output of ADC1086, and the ADC1086stores the digital value of the first output current (IOUT1). The digital processor1092reads the digital value of the minimum output current (IOUT_MIN) from the first register1088and the digital value of the first output current (IOUT1) from the second register1090, and, in operation901, subtracts the value of the first output current (IOUT1) from the value of the minimum output current (IOUT_MIN) to obtain a result (e.g., error value). In operation1093, the digital processor1092determines whether the result (e.g., the error value) is above a threshold (e.g., zero). If yes, in operation1095, the digital processor1092updates the value of the minimum output current (IOUT_MIN) to be the value of the first output current (IOUT1), which is provided to the P/S converter1082to change the information to the serial format to be placed on the CS interface (e.g., the digital communication line(s)). Also, if yes, in operation1097, the digital processor1092computes the dead time. For example, the digital processor1092computes the dead time (DT) as the default dead time plus the CS dead time. The CS dead time is the difference of IOUT1and IOUT_MIN, multiplied by the parameter K. The computed deadtime is provided to the phase generator1094. If no, the digital processor1092determines the dead time as the default dead time, which is provided to the phase generator1094. In some examples, the phase generator1094is included as part of the DT adjustor1014. In some examples, the phase generator1094is included as part of the control logic (e.g., the control logic120ofFIG.1B). The phase generator1094may include one or more phase counters1096, and one or more dead time (DT) phase counters1098. A phase counter1096may include an n-bit counter driven with an internal clock in order to count phase delays. A DT phase counter1098may include an n-bit counter driven with an internal clock in order to count dead time delays. If the internal clock is operating at 200 Mhz to drive the control logic, a counter (e.g., a phase counter1096or DT phase counter1098) can count delays with a tick of 5 ns. For a first phase P1with a duration of 1 us, the phase generator1094may set the end of the counter to be equal to 1000 ns/5 ns=200 Decimal. After the phase P1is counted, the phase generator1094resets the counter, and updates the end of the counter register with a new digital value related to the duration of the delay DT. For example, if a DT is set equal to 100 ns, the phase generator1094sets the end of counter equal to 100 n/5 ns=20. If current sharing is active and the DT is determined to be increase, the phase generator1094updates the end of counter from the default of20to the new value. If the new DT is 150 ns, the phase generator1094changes the end of counter from 20 to 30 (150 ns/5 ns). After the DT is set, the end of counter is set to obtain the second phase P2, and the DT is set again, and so forth. FIG.11illustrates a graph1100depicting a first output current1101(e.g., from a first STC module) and a second output current1103(e.g., from a second STC module) with respect to time according to an aspect. For example, the second STC module may be considered the weaker module (e.g., providing less output current) while the first STC module may be considered the stronger module (e.g., providing more output current). At the beginning and until a line1105, the first and second STC modules are operating independently. The line1105indicates a point in time in which the first and second STC modules are connected in parallel (but without the current sharing mechanism enabled). A line1107illustrates a point in time in which the current sharing mechanism is activated. As shown inFIG.11, after the first and second STC modules are connected in parallel (e.g., after the line1105), the first output current1101and the second output current1103begin to diverge, thereby providing unequal current. However, when the current sharing mechanism is activated (at line1105), the first output current1101and the second output current1103begin to converge toward each other, which eventually results in the first output current1101and the second output current1103having substantially the same value. FIGS.12A through12Cillustrate various signals of a non-regulated converter without the current share mechanism enabled according to an aspect. For example,FIG.12Adepicts a first average output current1201(e.g., generated by a first STC module) and a second average output current1203(e.g., generated by a second STC module connected in parallel with the first STC module) over time.FIG.12Billustrates a first output current waveform1205(e.g., generated by the first STC module) and a second output current waveform1207(e.g., generated by the second STC module).FIG.12Cillustrate control signals1209that enable the STC phases of the STC modules. As shown inFIG.12A, the first average output current1201is mismatched (e.g., not substantially equal) from the second average output current1203, which can negatively affect the performance of a power converter with STC modules. Also, as shown inFIG.12C, the default dead time provided between the phases is relatively small. FIGS.13A through13Cillustrate the effects of a STC converter with the current share mechanism enabled according to an aspect. For example,FIG.13Adepicts a first average output current1301(e.g., generated by a first STC module) and a second average output current1303(e.g., generated by a second STC module connected in parallel with the first STC module) over time.FIG.13Billustrates a first output current waveform1305(e.g., generated by the first STC module) and a second output current waveform1307(e.g., generated by the second STC module).FIG.13Cillustrate control signals1309that enable the STC phases of the STC modules. As shown inFIG.13A, the first average output current1301is matched (e.g., substantially equal) to the second average output current1303, which can improve the performance of a power converter with STC modules. Also, as shown inFIG.13C, the CS dead time1316is added at the end of the STC cycles. While certain features of the described implementations have been illustrated as described herein, many modifications, substitutions, changes and equivalents will now occur to those skilled in the art. It is, therefore, to be understood that the appended claims are intended to cover all such modifications and changes as fall within the scope of the embodiments. It should be understood that they have been presented by way of example only, not limitation, and various changes in form and details may be made. Any portion of the apparatus and/or methods described herein may be combined in any combination, except mutually exclusive combinations. The embodiments described herein can include various combinations and/or sub-combinations of the functions, components and/or features of the different embodiments described. | 52,517 |
11863065 | DETAILED EMBODIMENTS OF THE INVENTION The exemplary embodiments will now be described more fully with reference to the accompanying drawings. However, the exemplary embodiments can be implemented in various forms and shall not be understood as being limited to the embodiments set forth herein; on the contrary, these embodiments are provided so that this invention will be thorough and complete, and the conception of exemplary embodiments will be fully conveyed to those skilled in the art. In the drawings, the same reference sign denotes the same or similar structure, so their detailed description will be omitted. When factors/components/the like described and/or illustrated here are introduced, the phrases “one”, “a(an)”, “the”, “said” and “at least one” refer to one or more factors/components/the like. The terms “include”, “comprise” and “have” refer to an open and included meaning, and refer to additional factors/components/the like, in addition to the listed factors/components/the like. The embodiments may use relative phrases, such as, “upper” or “lower” to describe a relative relation of one signed component over another component. It shall be understood that if the signed device reverses to turn upside down, the described component on an “upper” side will become a component on a “lower” side. In addition, the terms “first”, “second” and the like in the claims are only used as signs, instead of numeral limitations to objects. FIG.2illustrates a circuit configuration of a power module100according to a first preferable embodiment of the invention. The power module100, for example, may be cascading applied to a medium voltage power grid system, but the invention is not limited thereto. In the embodiment ofFIG.2, the power module100may comprise N cells10in cascade connection, such as the cells Cell 1, Cell 2 . . . and Cell N shown in the figure, where N is a positive integer equal to or greater than 2. Each cell10may comprise a first end101and a second end102. The first ends101of the N cells10are cascaded and may be further connected to a power supply20. The power supply20, for example, may be, but not limited to an AC power grid, and may provide a voltage Vg. The second ends102of the N cells10, for example, may be DC output ends, and connected to a DC load, respectively, and the respective DC output ends may be at voltages Vdc1, Vdc2. . . and VdCN. Hereinafter taking the cell Cell 1 as an example, a specific circuit of the respective cells in the invention is explained in detail. In the invention, each cell10is an HPFC circuit and comprises a bidirectional switching unit11and a non-controlled rectifier bridge12, wherein the bidirectional switching unit11is connected to central points N1and N2of two bridge arms121and122of the non-controlled rectifier bridge12. In this embodiment, the bidirectional switching unit11, for example, may comprise two IGBTs having a body diode and connected in series reversely, i.e., comprising switches Q1and Q2. However, it can be understood that in other embodiments, the bidirectional switching unit11also may comprise two IGBTs without a body diode and connected in parallel, but the invention is not limited thereto. In other embodiments, each cell10may further comprise a first capacitor13and may be connected in parallel to a DC end of the non-controlled rectifier bridge12. Circuit configurations of the cells Cell 2 . . . and Cell N are the same as that of the cell Cell 1, so the details are not described here. In the invention, since the circuit in which a bidirectional switching unit11is added between the central points N1and N2of the two bridge arms121and122of the non-controlled rectifier bridge12is in an H shape and the circuit can realize Power Factor Correction (PFC), it may be referred to as an “HPFC circuit”. The invention enables the HPFC circuit to realize the Power Factor Correction (PFC) by regulating duty cycles of the switches (e.g., the switches Q1and Q2in the embodiment ofFIG.2) in the bidirectional switching unit11. In particular, each cell10can operate in one of three operating modes of a modulation mode, a bypass mode, and a non-controlled rectifying mode. When cell10is operating in the modulation mode, its bidirectional switching unit11and non-controlled rectifier bridge12operate simultaneously to realize the Power Factor Correction (PFC). When cell10is operating in the bypass mode, its bidirectional switching unit11is turned on to bypass the non-controlled rectifier bridge12. When cell10is operating in the non-controlled rectifying mode, its bidirectional switching unit11is turned off and only the non-controlled rectifier bridge12is operating to rectify. The cells in the power module of the invention adopt the HPFC circuit configuration with the bypass function and the PFC function multiplexed, thereby reducing cost as compared to the “bidirectional switch+Totem-pole” scheme in the prior art. Moreover, the non-controlled rectifier bridge does not have the problem of direct conduction, which may improve the reliability of the system. Meanwhile, the introduction of the non-controlled rectifying mode enhances the fault-tolerant control capability of the system. Further, in combination withFIG.2,FIG.3illustrates a method300for controlling a power module according to the invention, comprising: step S31, configuring N cells10in cascade connection, where N is a positive integer equal to or greater than 2, each cell10comprising a bidirectional switching unit11and a non-controlled rectifier bridge12, the bidirectional switching unit11being connected to central points N1and N2of two bridge arms121and122of the non-controlled rectifier bridge12; step S32, controlling each cell10to operate in one of three operating modes of a modulation mode, a bypass mode, and a non-controlled rectifying mode, wherein among the N cells10, m1 cells operate in the bypass mode, where 0≤m1≤M1; m2 cells operate in the non-controlled rectifying mode, where 0≤m2≤M2; m3 cells operate in the modulation mode and can realize power factor correction, where 0<m3; wherein m1+m2+m3=N, M1 is the number of cells allowing bypass in the system, and M2 is the number of cells allowing non-controlled rectification in the system. In the embodiment ofFIG.2, the M1, for example, may be determined by a ratio of a grid voltage of the medium voltage power grid system to a port withstanding voltage of a single cell. For example, when a total grid voltage is 10 kV, assuming that the rated voltage of each cell is 1 kV and the system has N=12 cells cascaded, the number allowing bypass is M1=2. The M2, for example, may be determined by an allowable current distortion degree of the system and not exceed N/2. By controlling the switches in the bidirectional switching unit11, the invention can realize power factor correction (PFC) and can multiplex bypass switches (i.e., switches in the bidirectional switching unit11may also have function of bypass). With the multiplexing of the bypass function and the PFC function, the present invention reduces cost and improves reliability as compared to the “bidirectional switch+Totem-pole” scheme in the prior art (as shown inFIG.1E). In the invention, further, the cells operating in the modulation mode can eliminate influence by a port voltage of the cells in the non-controlled rectifying mode through a current closed-loop and voltage feedforward control, thereby realizing the PFC function. For example, the current closed-loop and voltage feedforward control shown inFIG.4may be adopted. In the figure, Idrefis a current reference on a d-axis representing an active power and being set according to needs, cos θAis a cosine function of an angle of the grid voltage, Igrefis an AC reference being synchronized with the grid voltage, Igis the feedback of a grid current, P is a current proportional controller, Vbkrefis a modulation voltage reference of the k-th cell in the modulation mode, and Vffis voltage feedforward. A calculating formula of a feedforward voltage Vffis: Vff=Vg-sign(ig)∑h=1m2VdchN-m1-m2, wherein Vgis a grid voltage, igis a grid current, sign is a sign function. That is, when ig>0, sign (ig)=1, when ig<0, sign (ig)=−1, and when ig=0, sign (ig)=0. Vdchrepresents an output voltage of the h-th cell in the non-controlled rectifying mode. In the formula, subscript h represents that the h-th cell is in the non-controlled rectifying mode, and m2cells in total are in the non-controlled rectifying mode. In the formula, the numerator is the grid voltage minus a total port voltage of the m2cells in the non-controlled rectifying mode, and denominator N−m1−m2represents the number of cells in the modulation mode, i.e., m3, meaning that the cells in the modulation mode averagely distribute a total voltage corresponding to the numerator. Further, all cells operating in the modulation mode may be modulated by the phase shift of carrier waves. Specifically, a driving signal of the bidirectional switch can be generated by comparing the modulation voltage reference Vbkrefwith the carrier waves. Moreover, as for the (N−m1−m2) cells operating in the modulation mode, phases of the carrier waves between the cells may have a phase difference of 2π/(N−m1−m2) sequentially. In other words, phases of the carrier waves between the m3cells operating in the modulation mode may have a phase difference of 2π/m3sequentially. Further, the invention provides a fault-tolerant control method based on the power module of the cascaded HPFC circuit.FIG.5illustrates a flow of fault-tolerant control processing inside a control period of the k-th cell. Among the N cells, as for the k-th cell, after the control period begins, whether the k-th cell itself has a fault, such as faults of DC_Link overvoltage or cell over-temperature, is detected. If no, the k-th cell selects to enter the modulation mode. If yes, the processing is classified according to fault conditions. If it is an open circuit fault of the bidirectional switching unit, the number of cells in the non-controlled rectifying mode in the system is judged. When the number m2 of cells in the non-controlled rectifying mode is less than M2, the k-th cell may select to enter the non-controlled rectifying mode, and the number of cells in the non-controlled rectifying mode is added with 1 (i.e., m2+1), or the system stops. If it is not the open-circuit fault of the bidirectional switching unit, the number of cells in the bypass mode in the system is judged. When the number m1 of cells in the bypass mode in the N cells is less than M1, the k-th cell may select to enter the bypass mode, and the number of cells in the bypass mode is added with 1 (i.e., m1+1), or the system shall stop. FIG.6illustrates a circuit configuration of a power module100-1according to a second preferable embodiment of the invention, which may form a single-phase SST, for example. In this embodiment, each cell10of the power module100-1may further comprise a DC-DC conversion cell14connected to a DC end of the non-controlled rectifier bridge12, wherein outputs of the DC-DC conversion cells14are connected in parallel to form an output of a total port voltage Vo. For this embodiment, when the bidirectional switching unit11in cell10has an open circuit fault, a preceding stage AC-DC (e.g., the non-controlled rectifier bridge12) of cell10is in the non-controlled rectifying mode and may perform DC-Link voltage-sharing control through a post stage DC-DC (e.g., the DC-DC conversion cell14).FIGS.7and8are simulation conditions of simulation waveforms based on the embodiment of a single-phase SST in the invention. The single-phase SST comprises four cells cascaded, i.e., comprising cells Cell 1 to Cell 4, wherein simulation is provided with a fault bypass of the fourth cell Cell 4, a peak of the grid voltage is 4500V, Vdc=1580V, Vo=980V, the DC-DC conversion cell14is an LLC converter, a turns ratio of the transformer is 3:2, a capacitance Cdc of the first capacitor13is 100 μF, a capacitance Co of the output capacitor is 500 μF, a switching frequency of AC-DC (i.e., the HPFC circuit) is 5 kHZ, the carrier waves of the four cells are interlaced by 90 degrees, a resonant frequency of the LLC converter is 100 kHZ, a fixed switching frequency is controlled, and an output of the LLC converter is connected with a load of 20Ω resistor. FIG.7is simulation waveforms from the modulation mode to the bypass mode based on the second preferable embodiment of the invention. As can be seen from the waveforms, at about 100 ms, the cell Cell 4, for example, due to fault, enters the bypass mode, because a DC-Link voltage of the cell Cell 4 for exiting operation is decreased, and the DC-Link voltages of other cells (Cell 1 to Cell 3) can substantially operate uniformly and stably. Waveforms of other currents and voltages are substantially normal. It shows the feasibility of switching from the normal modulation mode to the bypass mode using the structure and method of the invention. FIG.8is simulation waveforms from the modulation mode to the non-controlled rectifying mode according to the second preferable embodiment of the invention. As can be seen from the waveforms, at about 100 ms, the cell Cell 4, for example, due to the open circuit fault of the bidirectional switch, enters the non-controlled rectifying mode, and since the cell Cell 4 in the non-controlled rectifying mode stops PWM modulation, a bridge arm voltage is changed from a high-frequency PWM wave to a low-frequency square wave. Moreover, the DC-Link voltages of all cells (Cell 1 to Cell 4) can substantially operate uniformly and stably. Waveforms of other currents and voltages are substantially normal. It shows the feasibility of switching from the normal modulation mode to the non-controlled rectifying mode using the structure and method of the invention. FIG.9illustrates a circuit configuration of a power module100-2according to a third preferable embodiment of the invention. In this embodiment, a relay15is further provided at an AC side of each cell10. When the switches (IGBTs) of the bidirectional switching unit11have an open circuit fault, it is possible to select to turn on the relay15, such that the corresponding cells enter the bypass mode. Selecting the relay to bypass has a lower loss than selecting the IGBTs of the bidirectional switching unit to bypass. Therefore, when each cell is controlled, for the k-th cell, if there is a fault and a fault type is the open circuit fault of the bidirectional switching unit, the invention may preferably turn on the relay, such that the k-th cell enters the bypass mode. FIG.10illustrates a circuit configuration of a power module100-3according to a fourth preferable embodiment of the invention. In this embodiment, each cell10further comprises a capacitor branch13′ comprised of a first capacitor131and a second capacitor132connected in series. The capacitor branch13′ is connected in parallel to a DC end of the non-controlled rectifier bridge12. The first capacitor131has a voltage Vdclpacross it, and the second capacitor132has a voltage Vdclnacross it. Moreover, the bidirectional switching unit11in each cell10may comprise a first IGBT11-11, a second IGBT11-12, a third IGBT11-21, and a fourth IGBT11-22connected in series, wherein the first IGBT11-11and the second IGBT11-12are connected in series reversely to form a first switching assembly11-1, the third IGBT11-21and the fourth IGBT11-22are connected in series reversely to form a second switching assembly11-2, and a central point N3between the first switching assembly11-1and the second switching assembly11-2are connected to a central point N4between the first capacitor131and the second capacitor132. In such a way, two IGBTs connected in series can replace one IGBT in the embodiment ofFIG.2, to form the embodiment of a three-level HPFC shown inFIG.10. This embodiment can reduce the requirement for a withstanding voltage of the single IGBT. FIG.11illustrates a structure of a three-phase power system200based on a power module of the invention. The three-phase power system200comprises three power modules100A,100B, and100C connected to three phases of a three-phase power supply30through a Y connection manner or an angle connection manner. In the embodiment ofFIG.11, the circuit configuration of the three power modules100A,100B, and100C, for example, may be the circuit configuration ofFIG.6, and the three power modules100A,100B, and100C and the three phases (VgA, VgB, and VgC) of the three-phase power supply30are connected to form a three-phase SST through the Y connection manner. It can be understood that in other embodiments, the circuit configuration of the three power modules100A,100B, and100C also can be, for example, the configuration ofFIG.2,9, or10, or other circuit configurations obtained by modification of the invention, but the invention is not limited thereto. In the invention, as for the three-phase power system in a Y connection or an angle connection, the fault-tolerant control logic of each phase is the same as that of the single-phase SST, so the details are not described here. In the invention, as for the three-phase power system in a Y connection, when one cell operates in the non-controlled rectifying mode, the current PFC control function can be realized using the method of zero-sequence voltage injection. More specifically, when the power module of one phase has cells operating in the non-controlled rectifying mode, the zero-sequence voltage can be injected into AC ports of the power modules of the other two phases. A schematic diagram of the zero-sequence voltage injection is shown inFIG.12, assuming that one cell in phase A is in the non-controlled rectifying mode, the voltage of the phase modulation wave cannot be infinitely approximate to zero near a zero-crossing point, and to ensure the current not to distort, the same voltage, i.e., the zero-sequence voltage, shall be injected into the other two phases, such that a line voltage does not include harmonic waves. The simulation effect based on the embodiment of the three-phase SST inFIG.11of the invention is shown inFIG.13. In this embodiment, simulation conditions are the three-phase SST, each phase having four cells, wherein the bidirectional switching unit of the fourth cell in phase C enters the non-controlled rectifying mode after an open circuit fault at 15 ms, and a time from 15 ms to 20 ms is a transition process of detecting the fault and switching the modes. At this time, since phase-shift angles and control policies of the carrier waves cannot be timely switched, the process has a waveform containing a short section of current distortion. Next, after considering of influence of non-controlled rectifier port voltage and control of the current closed-loop with the zero-sequence voltage injection provided by the invention, the waveform of the current gets back to normal. The simulation effect shows that the zero-sequence voltage injection into the bridge arm voltage enables THD of the waveforms of the circuit to get better, and allows the cells to operate in non-controlled rectification. To sum up, in the power module and the power system based on the cascaded HPFC circuit provided by the invention, the HPFC circuit may be formed by the bidirectional switching unit and the non-controlled rectifier bridge, wherein the switches in the bidirectional switching unit can multiplex the bypass function and the PFC function and can reduce cost as compared to the existing “bidirectional switch+Totem-pole” scheme. The non-controlled rectifier bridge of the invention does not have the problem of direct conduction and can improve reliability. The invention further provides the fault-tolerant control method suitable for the cascaded HPFC circuit, and the three-phase SST based on the cascaded HPFC circuit. When the bidirectional switching unit of one cell has the open circuit fault, it is possible to select to enter the non-controlled rectifying mode or the bypass mode. The introduction of the non-controlled rectifying mode can enhance the fault-tolerant control capability of the system. As for the three-phase power system in a Y connection, when one cell operates in the non-controlled rectifying mode, the invention also can realize the current PFC control function using the method of zero-sequence voltages injection, such that the waveform of the current is good, thereby further enhancing the fault-tolerant control capability of the system, so the invention has a further advantage. Exemplary embodiments of the invention have been shown and described in detail. It shall be understood that the invention is not limited to the disclosed embodiments. Instead, the invention intends to cover various modifications and equivalent settings included in the spirit and scope of the appended claims. | 20,919 |
11863066 | DETAILED DESCRIPTION FIG.1shows a circuit diagram of a memory100supplied by a voltage supply circuit101. The voltage supply circuit101includes a positive charge pump102, a negative charge pump104and a control stage105. The control stage105is coupled to both the positive and negative charge pumps102,104. The memory100includes a plurality of memory sectors106for storing data. Each memory sector106includes a word line108, a selection line110, a word line buffer112and a selection line buffer114. The memory sector106may include multiple transistors, capacitances, diodes, and logic gates, among others that are configured to store data and not described herein in structural detail. The positive charge pump102outputs a positive voltage (VPOS) to the word line buffer112and the negative charge pump104outputs a negative voltage (VNEG) to the selection line buffer114. The word line buffer112operates the word line108of the memory sector106using the positive voltage and the selection line buffer114operates the selection line110of the memory sector106using the negative voltage. The memory sector106is densely populated with word lines108and selection lines110and parasitic capacitances (each denoted as CWL_SELinFIG.1) emerge between various word lines108and selection lines110in the memory sector106. As the density of the memory sector106increase so does the parasitic capacitance (CWL_SEL). In addition, a parasitic capacitance (each denoted as CNW SEL inFIG.1) emerges between the n-wells of the transistors of the memory sector106and the selection line110. During the ramp-up of the positive voltage (e.g., from zero voltage or supply voltage to a desired positive voltage value) and the negative voltage (e.g., from zero voltage to a desired negative voltage value), the parasitic capacitances of the memory sector106influence the positive and negative voltages. The parasitic capacitances pull the negative voltage output by the negative charge pump104to the positive voltage output by the positive charge pump102, and vice-versa. Due to the fact that the magnitude of the positive voltage is typically greater than the magnitude of the negative voltage, the effect of the parasitic capacitances is typically more pronounced on the negative voltage. The parasitic capacitances result in the positive and negative charge pumps102,104using additional or more time to reach their respective desired output voltages or, in some cases, never reaching their respective output voltages. This may result in an operational failure of the memory100or sectors106thereof. The control stage105of the voltage supply circuit101is coupled to both the positive and negative charge pumps102,104. In various embodiments, the control stage105may have an input coupled to the negative charge pump104and an output coupled to the positive charge pump102. The control stage105receives the negative voltage supplied by the negative charge pump104and controls the positive charge pump102based on the negative voltage. For example, the control stage105may compare the negative voltage to a reference voltage for the negative voltage. The control stage105determines whether the negative voltage exceeds the reference voltage. When the negative voltage exceeds the reference voltage, the control stage105may operate to inhibit a slope, rise or increase in the positive voltage supplied by the positive charge pump102to mitigate the pull of the negative voltage towards the positive voltage due to the parasitic capacitances of the memory sector106. In various embodiments described herein, the control stage105may have an input coupled to the positive charge pump102and an output coupled to the negative charge pump104. The control stage105receives the positive voltage supplied by the positive charge pump102and controls the negative charge pump104based on the positive voltage. For example, the control stage105may compare the positive voltage to a reference voltage for the positive voltage and determine whether the positive voltage is below the reference voltage. When the positive voltage is below the reference voltage, the control stage105may operate to stem a decrease in the negative voltage supplied by the negative charge pump104to mitigate the pull of the positive voltage towards the negative voltage due to the parasitic capacitances of the memory sector106. The control stage105may have respective inputs coupled to the positive and negative charge pumps102,104and respective outputs coupled to the positive and negative charge pumps102,104. The control stage105may operate simultaneously on the positive and negative charge pumps102,104as described herein. FIG.2shows a voltage supply circuit101ain accordance with an embodiment. In the embodiment ofFIG.2, the positive voltage (VPOS) is controlled based on the negative voltage (VNEG). The voltage supply circuit101aincludes a positive charge pump102a, a negative charge pump104aand a control stage105a. The positive charge pump102aincludes a positive charge pump stage116and a feedback stage118a. The feedback stage118aincludes a voltage detector120shown as a resistive ladder comprising a plurality of resistances122. The feedback stage118aincludes a feedback comparator124, a clock stage126aand a clock buffer128. The clock stage126aincludes an AND gate129. The negative charge pump104aincludes a negative charge pump stage130and a feedback stage132a. The feedback stage132aincludes a voltage detector134shown as a resistive ladder comprising a plurality of resistances136. The feedback stage132aincludes a regulation stage138including a regulating comparator140and a transistor142. The feedback stage132aincludes a feedback comparator144and a clock stage146aincluding an AND gate148. The feedback stage132aincludes a clock buffer150. The control stage105aincludes a control comparator152. In the positive charge pump102a, the positive charge pump stage116has a first input coupled to a supply voltage node154. The supply voltage node154provides a supply voltage (Vsupply) to the positive charge pump stage116. The positive charge pump stage116has a second input coupled to a first output of the clock buffer128. The positive charge pump stage116has a third input coupled to a second output of the clock buffer128. The positive charge pump stage116receives a driving clock signal (CLKp) and an inverted driving clock signal (CLKNp), over the second and third inputs, respectively. The positive charge pump stage116has an output. The positive charge pump stage116provides the positive voltage (VPOS) over the output. The voltage detector120is coupled between the output of the positive charge pump stage116and a ground voltage node156. The voltage detector120has a tap node158for providing a divided voltage (Vpdiv) representative of the positive voltage (VPOS). The ground voltage provided by the ground voltage node156is a reference voltage for the divided voltage (Vpdiv). In alternative embodiments, the reference voltage for the divided voltage (Vpdiv) may be a voltage other than the ground voltage. The feedback comparator124has an inverting input coupled to the tap node158and a non-inverting input configured to receive a first reference voltage (Vp). The feedback comparator124has an output. The first reference voltage (Vp) may be a desired voltage for the divided voltage representative of the positive voltage (VPOS) obtained at the tap node158. It is noted that the feedback comparator124may be an error amplifier. The clock stage126a(and AND gate129thereof) has a first input coupled to the output of the feedback comparator124, a second input configured to receive a clock signal (CK), a third input coupled to an output of the control stage105aand an output coupled to an input of the clock buffer128. The clock signal (CK) is supplied by an oscillator155that may be part of the voltage supply circuit101aor outside of the voltage supply circuit101a. In the negative charge pump104a, the negative charge pump stage130has a first input coupled to the ground voltage node156. The ground voltage node156provides a ground voltage to the negative charge pump stage130. The negative charge pump stage130has a second input coupled to a first output of the clock buffer150and a third input coupled to a second output of the clock buffer150. The negative charge pump stage130receives a driving clock signal (CLKn) and an inverted driving clock signal (CLKNn), over the second and third inputs, respectively, from the clock buffer150. The negative charge pump stage130has an output. The negative charge pump stage130provides the negative voltage (VNEG) over the output. The voltage detector134is coupled between the output of the negative charge pump stage130and an output of the regulation stage138. The output of the regulation stage138is taken at a first conduction terminal of the transistor142. The transistor142has a second conduction terminal coupled to the supply voltage node154and a control terminal coupled to an output of the regulating comparator140. The regulating comparator140has an inverting input coupled to the first conduction terminal of the transistor142and a non-inverting input configured to receive a reference voltage (Vregref) for the regulation stage138. The voltage detector134has a first tap node160for providing a divided voltage (Vndiv) representative of the negative voltage (VNEG). The feedback comparator144has a non-inverting input coupled to the first tap node160and an inverting input coupled to the ground voltage node156. The feedback comparator144has an output. In an embodiment, the feedback comparator144may be an error amplifier. The clock stage146a(and AND gate148thereof) has a first input coupled to the output of the feedback comparator144, a second input configured to receive the clock signal (CK) and an output coupled to an input of the clock buffer128. The control comparator152of the control stage105ahas an inverting input coupled to the ground voltage node156. The ground voltage node156provides a ground voltage or generally a reference voltage for the control comparator152. The control comparator152has a non-inverting input coupled to a second tap node162of the voltage detector134and an output coupled to the third input of the clock stage126a(or AND gate129thereof). During operation of the negative charge pump104a, the negative charge pump stage130outputs the negative voltage (VNEG). The voltage detector134of the feedback stage132areceives the negative voltage (VNEG). The voltage detector134also receives a regulation voltage provided by the regulation stage138. The voltage detector134divides the negative voltage (VNEG) and supplies at the first tap node160the divided voltage (Vndiv) representative of the negative voltage (VNEG). The divided voltage (Vndiv) is between the negative voltage (VNEG) and the regulation voltage. The feedback comparator144compares the divided voltage (Vndiv) to the ground voltage. When the divided voltage (Vndiv) is greater than the ground voltage, the feedback comparator144asserts, activates or sets to a second logical state (logical one) a clock stop signal (Sstopn). A first logical state is described herein as being zero but may alternatively be one depending on convention. Conversely, when the divided voltage (Vndiv) is less than the ground voltage, the feedback comparator144deasserts, deactivates or sets to a first logical state (e.g., zero) the clock stop signal (Sstopn). The clock stop signal (Sstopn) ensures that, in steady state operation, the negative voltage (VNEG) does not drop below the desired negative voltage value. The AND gate148of the clock stage146areceives the clock stop signal (Sstopn) and the clock signal (CK) and shortens the on-time of the clock signal (CK) based on the clock stop signal (Sstopn). For example, if the clock stop signal (Sstopn) transitions to a logical zero while the clock signal (CK) is a logical one, the AND gate148outputs the clock signal (CK) having a logical state of zero. The clock buffer150, which may include a buffer and an inverting buffer, receives the clock signal (CK) and outputs the driving clock signal (CLKn) and the inverted driving clock signal (CLKNn) to the negative charge pump stage130. The clock buffer150may operate to sharpen the edges of the clock signal (CK) and invert the clock signal (CK) to output the inverted driving clock signal (CLKNn). The negative charge pump stage130receives the driving clock signal (CLKn) and the inverted driving clock signal (CLKNn) and generates the negative voltage (VNEG) based on the driving clock signal (CLKn) and the inverted driving clock signal (CLKNn). The on-time durations of driving clock signal (CLKn) and the inverted driving clock signal (CLKNn) dictate the voltage level of the negative voltage (VNEG). Similarly, in the positive charge pump102a, the positive charge pump stage116outputs the positive voltage (VPOS). The voltage detector120of the feedback stage118areceives the positive voltage (VPOS). The voltage detector120divides the positive voltage (VPOS) and supplies at the tap node158the divided voltage (Vpdiv) that is representative of the positive voltage (VPOS). The divided voltage (Vpdiv) is between the positive voltage (VPOS) and the ground voltage. The feedback comparator124compares the divided voltage (Vpdiv) to the first reference voltage (Vp), which is the reference voltage for the divided voltage (Vpdiv). When the divided voltage (Vpdiv) is greater than the first reference voltage (Vp), the feedback comparator124deasserts, deactivates or sets to the first logical state a clock stop signal (Sstopp). Conversely, when the divided voltage (Vpdiv) is less than the first reference voltage (Vp), the feedback comparator124asserts, activates or sets to a second logical state (e.g., one) the clock stop signal (Sstopp). The AND gate129of the clock stage126areceives the clock stop signal (Sstopp) and the clock signal (CK) and shortens the on-time of the clock signal (CK) based on the clock stop signal (Sstopp). The AND gate129ends the on-time of the clock signal (CK) and transitions the clock signal (CK) to a logical zero upon deassertion of the clock stop signal (Sstopp). For example, if the clock stop signal (Sstopp) transitions to a logical zero while the clock signal (CK) is a logical one, the AND gate129outputs the clock signal (CK) having a logical state of zero. The clock buffer128, which may include a clock buffer and an inverting clock buffer, receives the clock signal (CK) and outputs the driving clock signal (CLKp) and the inverted driving clock signal (CLKNp) to the positive charge pump stage116. The clock buffer128may sharpen the edges of the clock signal (CK) and invert the clock signal (CK) to output the inverted driving clock signal (CLKNp). The positive charge pump stage116receives the driving clock signal (CLKp) and the inverted driving clock signal (CLKNp) and generates the positive voltage (VPOS) based on the driving clock signal (CLKp) and the inverted driving clock signal (CLKNp). The on-time durations of driving clock signal (CLKp) and the inverted driving clock signal (CLKNp) dictate the voltage level of the positive voltage (VPOS). Without the control stage105a, the feedback stages118a,132acontrol respective positive and negative voltages (VPOS, VNEG) independently of each other. The feedback stage118acontrols the operation of the positive charge pump stage116based on the positive voltage (VPOS) and the feedback stage132acontrols the operation of the negative charge pump stage130based on the negative voltage (VNEG). The control stage105ainterdependently operates the positive and negative charge pumps102a,104a. The control stage105auses a feedback voltage (Vfb) representative of the negative voltage (VNEG) to control the positive charge pump102a. The control stage105areceives the feedback voltage (Vfb) representative of the negative voltage (VNEG). The feedback voltage (Vfb) may be the negative voltage (VNEG) having undergone voltage division. The control comparator152receives, over its non-inverting input, the feedback voltage (Vfb). The control comparator152receives the ground voltage over its inverting input. The control comparator152compares the feedback voltage (Vfb) to the ground voltage. When the feedback voltage (Vfb) is greater than the ground voltage, the control comparator152asserts a control signal (Scont). The control signal (Scont) is inverted and provided to the third input of the AND gate129. Upon assertion of the control signal (Scont), the AND gate129ends the on-time of the clock signal (CK) and transitions the clock signal (CK) to a logical zero thereby reducing the slope of the positive voltage (VPOS). An increase of the feedback voltage (Vfb) above the ground voltage is an indication that the negative voltage (VNEG) is pulled towards the positive voltage (VPOS). The control stage105aoperates to stem the rise of the positive voltage (VPOS) to reduce the pull induced by the positive voltage (VPOS) on the negative voltage (VNEG). The control stage105aoperates to do so by causing on-time of the clock signal (CK) to end. FIG.3shows a voltage supply circuit101bin accordance with an embodiment. Similar elements of the voltage supply circuit101bdescribed with reference toFIG.3as the voltage supply circuit101adescribed with reference toFIG.2have the same reference numerals. In the embodiment ofFIG.3, the positive voltage (VPOS) is controlled based on the negative voltage (VNEG). The voltage supply circuit101bofFIG.3differs from the voltage supply circuit101adescribed with reference toFIG.2in that the control comparator152directly reduces the positive voltage (VPOS) by sinking a control current (ICTRL) therefrom. That is in contrast to reducing the on-time duration of the driving clock signal (CLKp) of the positive charge pump stage116. The voltage supply circuit101bincludes a positive charge pump102bhaving a feedback stage118band a negative charge pump102bhaving a feedback stage132b. The voltage supply circuit101bincludes a control stage105b. The clock stage126bof the feedback stage118bincludes an AND gate164having a first input coupled to the output of the feedback comparator124, a second input configured to receive a clock signal (CK) and an output coupled to an input of the clock buffer128. In contrast to the AND gate129of the clock stage126adescribed with reference toFIG.2, the AND gate164forgoes the third input and is instead similarly configured as the AND gate148of the negative charge pump104b. The control stage105bincludes, in addition to the control comparator152, a control transistor166and a control current source168. An inverting input of the control comparator152is coupled to the ground voltage node156and a non-inverting input is configured to receive the feedback voltage (Vfb) representative of the negative voltage (VNEG) from the voltage detector134. The output of the control comparator152is coupled to a control terminal of the control transistor166. The control transistor166has a first conduction terminal coupled to the output of the positive charge pump stage116and a second conduction terminal coupled to an anode of the control current source168. The control current source168has a cathode coupled to the ground voltage node156that provides a ground voltage. The control current source168supplies the control current (ICTRL). When the feedback voltage (Vfb) rises above the ground voltage, the control comparator152asserts its output control signal (Scont) rendering the control transistor166conductive and sinking the control current (ICTRL) from the positive voltage (VPOS) thereby reducing the positive voltage (VPOS) or a rise or a slope thereof. When the feedback voltage (Vfb) drops is below the ground voltage, the control comparator152does not assert its output control signal (Scont). The control transistor166operates in the nonconductive state and the control stage105bdoes not sink the control current (ICTRL) from the positive voltage (VPOS). FIG.4shows a voltage supply circuit101cin accordance with an embodiment. Similar elements of the voltage supply circuit101cdescribed with reference toFIG.4as the voltage supply circuit101adescribed with reference toFIG.2have the same reference numerals. In the embodiment ofFIG.4, the positive voltage (VPOS) is controlled based on the negative voltage (VNEG). The voltage supply circuit101cofFIG.4differs from the voltage supply circuit101adescribed with reference toFIG.2in that the voltage supply circuit101cuses voltage-controlled oscillators to control the clock signals driving the positive and negative charge pump stages116,130. The voltage supply circuit101cincludes arrangements of a voltage-controlled oscillator and error amplifier in place of the arrangements of the feedback comparator124,144and clock stage126a,146a. The voltage supply circuit101cincludes a positive charge pump102chaving a feedback stage118c. The feedback stage118cincludes a feedback error amplifier170and a voltage-controlled oscillator172. The voltage supply circuit101cincludes a negative charge pump104chaving a feedback stage132c. The feedback stage132cincludes a feedback error amplifier174and a voltage-controlled oscillator176. The voltage supply circuit101cincludes control stage105cincluding a control error amplifier178. In the positive charge pump102c, the feedback error amplifier170has an inverting input configured to receive the first reference voltage (Vp) and a non-inverting input coupled to the voltage detector120and configured to receive the divided voltage (Vpdiv) representative of the positive voltage (VPOS). The feedback error amplifier170has an output. The voltage-controlled oscillator172has a first input coupled to the output of the feedback error amplifier170, a second input and an output coupled to the input of the clock buffer128. In the negative charge pump104c, the feedback error amplifier174has a non-inverting input coupled to the ground voltage node156and an inverting input coupled to the voltage detector134and configured to receive the divided voltage (Vndiv) representative of the negative voltage (VNEG). The feedback error amplifier170has an output. The voltage-controlled oscillator176has an input coupled to the output of the feedback error amplifier174, and an output coupled to the input of the clock buffer150. The control error amplifier178has an inverting input coupled to the ground voltage node156and a non-inverting input coupled to the voltage detector134configured to receive the feedback voltage (Vfb) representative of the negative voltage (VNEG) from the voltage detector134. The control error amplifier178has an output coupled to the second input of the voltage-controlled oscillator172. In the negative charge pump104c, the feedback error amplifier174determines a difference between the divided voltage (Vndiv) representative of the negative voltage (VNEG) and the ground voltage and outputs an error signal (Verrn) representative of the difference. The voltage-controlled oscillator176receives the error signal (Verrn) and outputs the clock signal (CK) to the clock buffer150based on the error signal (Verrn). The magnitude and sign of the difference and the error signal (Verrn) determine the frequency of the clock signal (CK) used to drive the negative charge pump stage130. As the error signal (Verrn) increases, the frequency of the clock signal (CK) generated by the voltage-controlled oscillator176also increases. Similarly, in the positive charge pump102c, the feedback error amplifier170determines a difference between the divided voltage (Vpdiv) representative of the positive voltage (VPOS) and the first reference voltage (Vp). The feedback error amplifier170outputs an error signal (Verrp) representative of the difference. The voltage-controlled oscillator172receives the error signal (Verrp) and outputs the clock signal (CK) to the clock buffer128having a frequency that is based on the error voltage error signal (Verrp). The voltage-controlled oscillator172is additionally controlled by the control error amplifier178. The control error amplifier178receives determines a difference between the ground voltage node156and the feedback voltage (Vfb) and outputs the control signal (Scont) representative of the difference to the voltage-controlled oscillator172. The control signal (Scont) and the error signal (Verrp) may both contribute (e.g., by additive combination) to setting the frequency of the clock signal (CK). When the negative voltage (VNEG) drifts in the direction of the positive voltage (VPOS), a voltage level of the control signal (Scont) decreases to reduce the frequency of the clock signal (CK). In an embodiment, the voltage supply circuit101may use supply voltage control to control the frequency of the clock signal that drives the positive and negative charge pump stages116,130. FIG.5shows a voltage supply circuit101din accordance with an embodiment. Similar elements of the voltage supply circuit101ddescribed with reference toFIG.5as the voltage supply circuit101adescribed with reference toFIG.2have the same reference numerals. In the embodiment ofFIG.5, the positive voltage (VPOS) is controlled based on the negative voltage (VNEG). The voltage supply circuit101dofFIG.5uses controls the supply voltage of an oscillator to control the frequency of the clock signals driving the positive and negative charge pump stages116,130. The positive charge pump102dincludes a positive charge pump stage116and a feedback stage118d. The feedback stage118dincludes the voltage detector120, a voltage regulator180and an oscillator and clock buffer182. The negative charge pump104dincludes the negative charge pump stage130and a feedback stage132d. The feedback stage132dincludes the voltage detector134, a voltage regulator184and an oscillator and clock buffer186. The oscillator and clock buffer182,186may be a combination of an oscillator and a clock buffer, such as the clock buffer128,150described herein. Similar to the control stage105c, the control stage105dincludes a control error amplifier178. The voltage regulator180has a first input configured to receive the first reference voltage (Vp), a second input configured to receive the divided voltage (Vpdiv) representative of the positive voltage (VPOS), a third input and an output configured to provide a first regulated voltage (VREGp). The oscillator and clock buffer182has an input coupled to the output of the voltage regulator180, a first output for providing the driving clock signal (CLKp) and a second output for providing the inverted driving clock signal (CLKNp) to the positive charge pump stage116. In the negative charge pump104d, the voltage regulator184has a first input configured to receive the ground voltage, a second input configured to receive the divided voltage (Vndiv) representative of the negative voltage (VNEG) and an output configured to provide a second regulated voltage (VREGn). The oscillator and clock buffer186has an input coupled to the output of the voltage regulator184, a first output for providing the driving clock signal (CLKn) and a second output for providing the inverted driving clock signal (CLKNn) to the negative charge pump stage130. The output of the control error amplifier178is coupled to the third input of the voltage regulator. During operation, the voltage regulator180compares the divided voltage (Vpdiv) with the first reference voltage (Vp) and outputs the first regulated voltage (VREGp), based on the comparison, to both the oscillator and clock buffer182and the positive charge pump stage116. The first regulated voltage (VREGp) is the supply voltage of the oscillator and clock buffer182and the positive charge pump stage116. When the first regulated voltage (VREGp) increases, the frequency of the driving clock signals (CLKp, CLKNp) supplied by the oscillator and clock buffer182increases. Additionally, the positive voltage (VPOS) supplied by the positive charge pump stage116also increases. The voltage regulator modulates the first regulated voltage (VREGp) so that the difference between the divided voltage (Vpdiv) and the first reference voltage (Vp) is minimized. Similarly, the voltage regulator184controls the second regulated voltage (VREGn) to minimize the difference between the ground voltage and the divided voltage (Vndiv). The second regulated voltage (VREGn) is supplied to the oscillator and clock buffer186and the negative charge pump stage130. The voltage regulator180regulates its output voltage based on the combination (for example, additive combination) of the control signal (Scont) and the difference between the divided voltage (Vpdiv) and the first reference voltage (Vp) to control the first regulated voltage (VREGp). The control signal (Scont) is positively correlated with (e.g., proportional to) to the difference between the ground voltage and proportional. The voltage regulator180receives the control signal (Scont) and controls the first regulated voltage (VREGp) based on the control signal (Scont). The voltage regulator180increases or decreases the first regulated voltage (VREGp). The control error amplifier178and voltage regulator180operate to set and control the first regulated voltage (VREGp) to minimize the difference between the ground voltage and the feedback voltage (Vfb). The voltage supply circuits101a,101b,101c,101ddescribed herein control the positive voltage (VPOS) using the negative voltage (VNEG). Corresponding embodiments are described herein in which control of the negative voltage (VNEG) is based on the positive voltage (VPOS). FIG.6shows a voltage supply circuit101ein accordance with an embodiment. The voltage supply circuit101econtrols the negative voltage (VNEG) based on the positive voltage (VPOS). Similar elements of the voltage supply circuit101edescribed with reference toFIG.5as the voltage supply circuit101adescribed with reference toFIG.2have the same reference numerals. The voltage supply circuit101eincludes a positive charge pump102ehaving a feedback stage118e, a negative charge pump104ehaving a feedback stage132eand a control stage105e. The control stage105eincludes the control comparator152. The inverting input of the control comparator152is configured to receive a feedback voltage (Vfb) from the voltage detector120. The feedback voltage (Vfb) may be tapped at a node of the voltage detector120similar to the feedback voltage (Vfb) of the negative charge pump102adescribed with reference toFIG.2. The non-inverting input of the control comparator152is configured to receive a second reference voltage (Vref), which may be a desired voltage for the feedback voltage (Vfb). The negative charge pump104eincludes a clock stage146ehaving an AND gate188similarly configured as the AND gate129of the positive charge pump102aofFIG.2. The AND gate188receives the clock stop signal (Sstopn), a clock signal from the oscillator155and the control signal (Scont) and shortens the on-time duration of the clock signal based on the clock stop signal (Sstopn) and the control signal (Scont). FIG.7shows a voltage supply circuit101fin accordance with an embodiment. The voltage supply circuit101fcontrols the negative voltage (VNEG) based on the positive voltage (VPOS). The voltage supply circuit101fuses a similar control mechanism as that of the voltage supply circuit101bdescribed with reference toFIG.3albeit with the negative voltage (VNEG) being controlled based on the positive voltage (VPOS). In particular, the voltage supply circuit101fincludes a positive charge pump102fhaving a feedback stage118f, a negative charge pump104fhaving a feedback stage132fand a control stage105f. The control stage105fincludes the control comparator152, a control transistor190and a control current source192. The control transistor190has a first conduction terminal coupled to the supply voltage node, a second conduction terminal and a control terminal coupled to the output of the control comparator152. The control comparator152may be an analog or a digital comparator. The control comparator152may also be an error amplifier. The control current source192has an anode coupled to the second conduction terminal of the control transistor190and cathode coupled to the output of the negative charge pump stage130. The inverting input of the control comparator152is configured to receive the feedback voltage (Vfb) from the voltage detector120and the non-inverting input of the control comparator152is configured to receive a second reference voltage (Vref), which may be a desired voltage for the feedback voltage (Vfb). When the feedback voltage (Vfb) goes below the second reference voltage (Vref), the control comparator152asserts its output control signal (Scont) rendering the control transistor190conductive and sourcing the control current (ICTRL) to the negative voltage (VNEG) thereby positively increasing the negative voltage (VNEG) or reducing a rise or a slope thereof. Otherwise, the control transistor190is non-conductive and the control current (ICTRL) is not added to the negative voltage (VNEG). FIG.8shows a voltage supply circuit101gin accordance with an embodiment. The voltage supply circuit101gcontrols the negative voltage (VNEG) based on the positive voltage (VPOS). The voltage supply circuit101guses a similar control mechanism as that of the voltage supply circuit101cdescribed with reference toFIG.4with the difference being that the negative voltage (VNEG) is controlled based on the positive voltage (VPOS). The voltage supply circuit101gincludes a positive charge pump102ghaving a feedback stage118g, a negative charge pump104ghaving a feedback stage132gand a control stage105g. The control stage105gincludes the control error amplifier178, which controls the voltage-controlled oscillator176. The control error amplifier178determines a difference between the second reference voltage (Vref) and the feedback voltage (Vfb) and outputs the control signal (Scont) representative of the difference to the voltage-controlled oscillator176. The control signal (Scont) and the error signal (Verrn) may both contribute (e.g., by additive combination) to setting the frequency of the clock signal (CK) generated by the voltage-controlled oscillator176. When the positive voltage (VPOS) drifts in the direction of the negative voltage (VNEG), the drift is reflected in the control signal (Scont) and the frequency of the clock signal (CK) generated by the voltage-controlled oscillator176is adjusted to counter the drift. FIG.9shows a voltage supply circuit101hin accordance with an embodiment. The voltage supply circuit101hcontrols the negative voltage (VNEG) based on the positive voltage (VPOS). The voltage supply circuit101huses a similar control mechanism as that of the voltage supply circuit101hdescribed with reference toFIG.5with the difference being that the negative voltage (VNEG) is controlled based on the positive voltage (VPOS). The voltage supply circuit101hincludes a positive charge pump102hhaving a feedback stage118h, a negative charge pump104hhaving a feedback stage132hand a control stage105h. The control stage105hincludes the control error amplifier178, which controls the voltage regulator184of the negative charge pump104h. The control error amplifier178determines a difference between the second reference voltage (Vref) and the feedback voltage (Vfb) and outputs the control signal (Scont) representative of the difference to the voltage regulator184. The control signal (Scont) and the error signal (Verrn) may both contribute (e.g., by additive combination) to setting the frequency of the clock signal (CK) generated by the voltage-controlled oscillator176. When the positive voltage (VPOS) drifts in the direction of the negative voltage (VNEG), the drift is reflected in the control signal (Scont) and the frequency of the clock signal (CK) generated by the voltage-controlled oscillator176is adjusted to counter the drift. The voltage regulator184regulates its output voltage (the second regulated voltage (VREGn)) based on the combination (for example, additive combination) of the control signal (Scont) and the difference between the divided voltage (Vndiv) and the ground voltage. The control signal (Scont) is positively correlated with (e.g., proportional to) to the difference between the second reference voltage (Vref) and the feedback voltage (Vfb). The voltage regulator184receives the control signal (Scont) and controls the second regulated voltage (VREGn) based on the control signal (Scont). The voltage regulator184increases or decreases the second regulated voltage (VREGn). The control error amplifier178and voltage regulator184operate to set and control the second regulated voltage (VREGn) to minimize the difference between the second reference voltage (Vref) and the feedback voltage (Vfb). FIG.10shows signals diagrams of the positive voltage (VPOS), negative voltage (VNEG) and control signal (Scont) of the voltage supply circuit101adescribed with reference toFIG.2. Initially, the positive voltage (VPOS) increases and the negative voltage (VNEG) decreases. The control stage105aprevents the negative voltage (VNEG) from gravitating or drifting towards the positive voltage (VPOS) (e.g., due to parasitic capacitances). The control stage105aasserts the control signal (Scont). The control signal (Scont), when asserted, ends the active period of the driving clock signal (CLKp) of the positive charge pump stage116and results in reducing the slope of the positive voltage (VPOS). It is noted that the control stages105a,105b,105c,105dmay be used to control the positive voltage (VPOS) and mitigate voltage conversion during an initial transient stage when the positive voltage (VPOS) and the negative voltage (VNEG) are actively diverging from each other. During steady state conditions, the operation of the feedback stages118a,118b,118c,118dmay be sufficient keep the positive voltage (VPOS) above a desired positive voltage value, such as 16 Volts (V). Similarly, the control stages105e,105f,105g,105hmay be used to control the negative voltage (VNEG) and mitigate voltage conversion during the initial transient stage. During steady state conditions, the operation of the feedback stages118a,118b,118c,118dmay similarly be sufficient keep the positive voltage (VPOS) above a desired negative voltage value, such as −1V. The various embodiments described above can be combined to provide further embodiments. These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure. | 39,034 |
11863067 | DETAILED DESCRIPTION OF EMBODIMENTS In order to make objects, technical solutions and advantages of the embodiments of the present disclosure clearer, the technical solutions in the embodiments of the present disclosure are described clearly and completely in conjunction with the drawings in the embodiments of the disclosure hereinafter. It is apparent that the described embodiments are only some rather than all embodiments of the present disclosure. Any other embodiments obtained by those skilled in the art based on the embodiments in the present disclosure without any creative effort shall fall within the protection scope of the present disclosure. In this specification, terms “include”, “comprise” or any other variants are intended to be non-exclusive. Therefore, a process, method, article or device including a series of elements includes not only those elements but also other elements that are not enumerated, or further includes elements inherent for the process, method, article or device. Unless expressively limited otherwise, the statement “comprising (including) a . . . ” does not exclude existence of another identical element in a process, method, article or device including the enumerated elements. According to an embodiment of the present disclosure, a bridge cascade system is provided to solve a problem that: in a system applying a bridge topology, each bridge topology includes its own bootstrap power supply and driving power supply, which causes complexity of the bridge topology and a high cost for hardware thereof. Referring toFIG.2, the bridge cascade system includes at least one phase unit and a driving unit for the phase unit. The phase unit includes N bridge topologies cascaded on alternating current AC sides, where N is an integer greater than 1. It should be noted that the bridge topology is one of a complementary half-bridge topology, a complementary full-bridge topology and a Buck-boost topology. Apparently, other topologies are not excluded and fall within the protection scope of the present disclosure, which are not described in detail herein. Description is made below referring to an example in which the bridge topology is a complementary half-bridge topology. Each bridge topology includes two switch transistors connected in series. A drain of a first switch transistor is connected to a positive electrode at an input end of the bridge topology. A source of the first switch transistor is connected to a drain of a second switch transistor. A source of the second switch transistor is connected to a negative electrode at the input end of the bridge topology. In this configuration, the first switch transistor is a high-voltage side switch transistor, and the second switch transistor is a low-voltage side switch transistor. Correspondingly, a driving circuit corresponding to the high-voltage side switch transistor is a high-voltage side driving circuit, and a driving circuit corresponding to the low-voltage side switch transistor is a low-voltage side driving circuit. FIGS.2-20each show an example in which the bridge topology is a complementary half-bridge topology. When the bridge topology is of another type, such as a complementary full-bridge topology or a Buck-boost topology, specific structure and corresponding control principle of the bridge topology are similar to those shown inFIGS.2-20, and fall within the protection scope of the present disclosure, which are not described in detail herein. It should be noted that if the bridge cascade system includes two or more phase units, the phase units may be completely the same or not, which depends on an actual situation and is no limited herein, and shall all fall within the protection scope of the present disclosure. The driving unit includes one driving power supply circuit, multiple bootstrap power supply circuits and 2N driving circuits. In the phase unit, the driving power supply circuit supplies power to the driving circuits directly or through corresponding bootstrap power supply circuits, that is, the driving power supply circuit can implement a cross-bridge topology to supply power to a corresponding driving circuit. The driving circuits are configured to provide driving signals for corresponding switch transistors in the phase unit. Specifically, as shown inFIG.2, N is equal to 2, and there is 1 phase unit in the bridge cascade system, which is only an example used for the following description. The driving power supply circuit Vcc directly supplies power to a driving circuit Dr1. The driving circuit Dr1is configured to provide a driving signal for a switch transistor S1. The driving power supply circuit Vcc supplies power to a driving circuit Dr2through a bootstrap power supply circuit including elements Dboot2and Cboot2as shown inFIG.2. The driving circuit Dr2is configured to provide a driving signal for a switch transistor S2. The driving power supply circuit Vcc supplies power to a driving circuit Dr3through a bootstrap power supply circuit including elements Dboot3and Cboot3as shown inFIG.2. The driving circuit Dr3is configured to provide a driving signal for a switch transistor S3. The driving power supply circuit Vcc supplies power to a driving circuit Dr4through a bootstrap power supply circuit including elements Dboot4and Cboot4as shown inFIG.2. The driving circuit Dr4is configured to provide a driving signal for a switch transistor S4. Apparently, N may assume another value, or other numbers of phase units may be included in the bridge cascade system for which the specific structure is similar to that shown inFIG.2; these variations shall all fall within the protection scope of the present disclosure, and are not described in detail herein. In practical applications, each of the bootstrap power supply circuits may further include a resistor connected in series with a corresponding diode. The number and resistance of the resistor may depend on its specific application and are not specifically limited herein, and shall all fall within the protection scope of the present disclosure. In this embodiment, one driving power supply in the bridge cascade system is matched with multiple bootstrap power supply circuits, so as to supply power to driving circuits corresponding to the switch transistors of all bridge topology, which reduces the difficulty in designing the driving power supply for the bridge cascade system and reduces cost for the system. In practical applications, bootstrap power supply circuits in a phase unit may be connected in various manners, such as connected in series and/or in parallel. Described below are situations where the bootstrap power supply circuits in the phase unit are connected in parallel, connected in series and connected in a combination thereof, respectively.(1) The bootstrap power supply circuits in the phase unit are connected in parallel Input terminals of the bootstrap power supply circuits are each connected to an output terminal of the driving power supply circuit. Specifically, reference is made toFIG.3, which shows an example where there are 2 bridge topologies (N=2) and 3 bootstrap power supply circuits. In the description, the bootstrap power supply circuit including Cboot2and Dboot2as shown inFIG.3is referred to as a first bootstrap power supply circuit; the bootstrap power supply circuit including Cboot3and Dboot3as shown inFIG.3is referred to as a second bootstrap power supply circuit; and the bootstrap power supply circuit including Cboot4and Dboot4as shown inFIG.3is referred to as a third bootstrap power supply circuit. An input terminal of the first bootstrap power supply circuit, an input terminal of the second bootstrap power supply circuit and an input terminal of the third bootstrap power supply circuit are each connected to the output terminal of the driving power supply circuit Vcc. An output terminal of the first bootstrap power supply circuit is connected to an input terminal of the driving circuit Dr2, an output terminal of the second bootstrap power supply circuit is connected to an input terminal of the driving circuit Dr3, and an output terminal of the third bootstrap power supply circuit is connected to an input terminal of the driving circuit Dr4. The output terminal of the driving power supply circuit Vcc is further directly connected to an input terminal of the driving circuit Dr1. In other words, the input terminal of each bootstrap power supply circuit is directly connected to the output terminal of the driving power supply circuit Vcc.(2) The bootstrap power supply circuits in the phase unit are connected in series sequentially It should be noted that the bootstrap power supply circuits are connected in series in a corresponding sequence. For example, in the phase unit, there are 3 bootstrap power supply circuits, referred to as a first bootstrap power supply circuit, a second bootstrap power supply circuit, and a third bootstrap power supply circuit. Specifically, the connection sequence may be any of the following, a sequence of the driving power supply circuit, the first bootstrap power supply circuit, the second bootstrap power supply circuit, and the third bootstrap power supply circuit; and a sequence of the driving power supply circuit, the second bootstrap power supply circuit, the first bootstrap power supply circuit, and the third bootstrap power supply circuit. Apparently, other connection sequences are possible, but the mentioned two connection sequences are preferred in consideration of the withstand voltage of the bootstrap power supply circuit. A specific connection mode of the bootstrap power supply circuits sequentially connected in series depends on an actual situation and is not specifically limited herein, and all shall fall within the protection scope of the present disclosure. In practical applications, the bootstrap power supply circuits in the phase unit are sequentially connected in series according to potentials of corresponding switch transistors from high to low. When connected in series, an input terminal of the bootstrap power supply circuit corresponding to the switch transistor with the lowest potential is connected to the output terminal of the driving power supply circuit. Referring toFIG.2, for ease of description, the bootstrap power supply circuit including Cboot2and Dboot2as shown inFIG.2is referred to as the first bootstrap power supply circuit, the bootstrap power supply circuit including Cboot3and Dboot3as shown inFIG.2is referred to as the second bootstrap power supply circuit, and the bootstrap power supply circuit including Cboot4and Dboot4as shown inFIG.2is referred to as the third bootstrap power supply circuit. An input terminal of the first bootstrap power supply circuit and an input terminal of the driving circuit Dr1are both connected to the output terminal of the driving power supply circuit Vcc. An output terminal of the first bootstrap power supply circuit is connected respectively to an input terminal of the driving circuit Dr2and an input terminal of the second bootstrap power supply circuit. An output terminal of the second bootstrap power supply circuit is connected respectively to an input terminal of the driving circuit Dr3and an input terminal of the third bootstrap power supply circuit. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of the driving circuit Dr4.(3) The bootstrap power supply circuits in the phase unit are connected by a hybrid of serial connection and parallel connection There are various connection configurations for connecting the bootstrap power supply circuits in a hybrid manner of both parallel connection and cascading connection, which are not described in detail herein, and are all within the protection scope of the present disclosure. Here, assuming that there are 2 bridge topologies (N=2) and 3 bootstrap power supply circuits in the phase unit, and the 3 bootstrap power supply circuits in the one phase unit are referred to as, according to their potentials from high to low: a first bootstrap power supply circuit (including Cboot2and Dboot2), a second bootstrap power supply circuit (including Cboot3and Dboot3), and a third bootstrap power supply circuit (including Cboot4and Dboot4), exemplary illustration is made below. When N=2 and the number of bootstrap power supply circuits in the phase unit is 3, there should be 6 hybrid connection schemes for connecting the bootstrap power supply circuits. However, for a scheme in which a parallel arrangement of the first bootstrap power supply circuit and the second bootstrap power supply circuit is connected in series with the third bootstrap power supply circuit, there is a risk of high voltage directly charging bootstrap capacitors in the first bootstrap power supply circuit and the second bootstrap power supply circuit. Therefore, such scheme should not be applied. The remaining 5 schemes are described as follows.Scheme 1: as shown inFIG.4, a series arrangement of the second bootstrap power supply circuit and the third bootstrap power supply circuit is connected in parallel with the first bootstrap power supply circuit. Specifically, an output terminal of the driving power supply circuit Vcc is connected respectively to an input terminal of the first bootstrap power supply circuit, an input terminal of the second bootstrap power supply circuit and an input terminal of a driving circuit Dr1. An output terminal of the first bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr2. An output terminal of the second bootstrap power supply circuit is connected respectively to an input terminal of a driving circuit Dr3and an input terminal of the third bootstrap power supply circuit. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr4.Scheme 2: as shown inFIG.5, a series arrangement of the first bootstrap power supply circuit and the third bootstrap power supply circuit is connected in parallel with the second bootstrap power supply circuit. Specifically, an output terminal of the driving power supply circuit Vcc is connected respectively to an input terminal of the first bootstrap power supply circuit, an input terminal of the second bootstrap power supply circuit and an input terminal of a driving circuit Dr1. An output terminal of the second bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr3. An output terminal of the first bootstrap power supply circuit is connected respectively to an input terminal of a driving circuit Dr2and an input terminal of the third bootstrap power supply circuit. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr4.Scheme 3: as shown inFIG.6, a series arrangement of the first bootstrap power supply circuit and the second bootstrap power supply circuit is connected in parallel with the third bootstrap power supply circuit. Specifically, an output terminal of the driving power supply circuit Vcc is connected respectively to an input terminal of the first bootstrap power supply circuit, an input terminal of the third bootstrap power supply circuit and an input terminal of a driving circuit Dr1. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr4. An output terminal of the first bootstrap power supply circuit is connected respectively to an input terminal of a driving circuit Dr2and an input terminal of the second bootstrap power supply circuit. An output terminal of the second bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr3.Scheme 4: as shown inFIG.7, a parallel arrangement of the second bootstrap power supply circuit and the third bootstrap power supply circuit is connected in series with the first bootstrap power supply circuit. Specifically, an output terminal of the driving power supply circuit Vcc is connected respectively to an input terminal of the first bootstrap power supply circuit and an input terminal of a driving circuit Dr1. An output terminal of the first bootstrap power supply circuit is connected respectively to an input terminal of a driving circuit Dr2, an input terminal of the second bootstrap power supply circuit and an input terminal of the third bootstrap power supply circuit. An output terminal of the second bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr3. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr4.Scheme 5: as shown inFIG.8, a parallel arrangement of the first bootstrap power supply circuit and the third bootstrap power supply circuit is connected in series with the second bootstrap power supply circuit. Specifically, an output terminal of the driving power supply circuit Vcc is connected respectively to an input terminal of the second bootstrap power supply circuit and an input terminal of a driving circuit Dr1. An output terminal of the second bootstrap power supply circuit is connected respectively to an input terminal of a driving circuit Dr3, an input terminal of the third bootstrap power supply circuit and an input terminal of the first bootstrap power supply circuit. An output terminal of the first bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr2. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr4. The above descriptions are all made based on an example where N=2. When N is greater than or equal to 3, structures are similar to the above and are not described in detail herein, and shall fall within the protection scope of the present disclosure. In a case where N=3, there are multiple connection configurations, one of which is described below. It is assumed that there are 5 bootstrap power supply circuits in the phase unit. The 5 bootstrap power supply circuits are referred to as, according to their potentials from high to low: a first bootstrap power supply circuit (including Cboot2and Dboot2as shown inFIG.9), a second bootstrap power supply circuit (including Cboot3and Dboot3as shown inFIG.9), a third power supply circuit (including Cboot4and Dboot4as shown inFIG.9), a fourth bootstrap power supply circuit (including Cboot5and Dboot5as shown inFIG.9), and a fifth bootstrap power supply circuit (including Cboot6and Dboot6as shown inFIG.9). Specifically, as shown inFIG.9, an output terminal of the driving power supply circuit Vcc is connected respectively to an input terminal of the first bootstrap power supply circuit and an input terminal of a driving circuit Dr1. An output terminal of the first bootstrap power supply circuit is connected respectively to an input terminal of a driving circuit Dr2and an input terminal of the second bootstrap power supply circuit. An output terminal of the second bootstrap power supply circuit is connected respectively to an input terminal of the third bootstrap power supply circuit, an input terminal of the fourth bootstrap power supply circuit, an input terminal of the fifth bootstrap power supply circuit and an input terminal of a driving circuit Dr3. An output terminal of the third bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr4. An output terminal of the fourth bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr5. An output terminal of the fifth bootstrap power supply circuit is connected to an input terminal of a driving circuit Dr6. It should be noted that the possibility of schemes for bootstrap power supply circuits connected in a hybrid manner of cascading connection and parallel connection increases with the increase of the number of bootstrap power supply circuits, that is, the larger N is, the more possibilities there are for connecting the bootstrap power supply circuits. Other connection configurations of the bootstrap power supply circuits when N is equal to 3 or another value are not described in detail herein, and all shall fall within the protection scope of the present disclosure. It should be noted that, as shown inFIGS.2-9, the bootstrap diodes and bootstrap capacitors are used to illustrate a principle of the bootstrap power supply circuits, which do not represent actual circuits. The specific structure of the bootstrap power supply circuits depends on an actual situation and is not specifically limited herein, and shall fall within the protection scope of the present disclosure. In this embodiment, the variety of combinations of bootstrap power supply circuits in a phase unit may facilitate device selection and design optimization. It should be noted that when N=2, a high-voltage side switch transistor of a first bridge topology and a low-voltage side switch transistor of a second bridge topology share a source electrode, and bootstrap power supply circuits corresponding thereto share a ground, or the two may share a same bootstrap power supply circuit. From this perspective, in the bridge cascade system, two driving circuits may reuse one bootstrap power supply circuit, so as to reduce the number of devices in the system, especially reduce the number of bootstrap diodes, or even bootstrap capacitors. Meanwhile, the bootstrap power supply is simplified. In this case, attention should pay to a reverse voltage stress of a corresponding bootstrap diode, with which the charge support capability of the bootstrap capacitor is adjusted accordingly. For any one of the above embodiments, arbitrary two driving circuits in a same phase unit may reuse one bootstrap power supply circuit. In other words, the number of bootstrap power supply circuits in the phase unit is less than 2N−1. For example, when N=3, the number of bootstrap power supply circuits in the phase unit is 2 or 1. It should be further noted that for the situation where every two driving circuits reuse one bootstrap power supply circuit, the number of bootstrap power supply circuits would be equal to the value of subtracting 1 from 2N−1. Specifically, in the phase unit, a bootstrap power supply circuit is reused by a low-voltage side driving circuit in one of two bridge topologies that has higher potential and a high-voltage side driving circuit in the other one of the two bridge topologies that has lower potential. Based on the structure shown inFIG.3, the bootstrap diode Dboot2of the first bootstrap power supply circuit may be reused by the bootstrap capacitor Cboot2of the first bootstrap power supply circuit and the bootstrap capacitor Cboot3of the second bootstrap power supply circuit. Therefore, The bootstrap diode Dboot3of the second bootstrap power supply circuit may be omitted. Furthermore, it is possible to use only one bootstrap capacitor, and a reuse of cascaded bootstrap power supply scheme is shown inFIG.10. Referring toFIG.10, there are 2 bridge topologies (N=2), and a first bootstrap power supply circuit (including Cboot2and Dboot2as shown inFIG.10) is reused by a driving circuit Dr3and a driving circuit Dr2; that is, an output terminal of the first bootstrap power supply circuit is directly connected to an input terminal of the driving circuit Dr3and an input terminal of the driving circuit Dr2, respectively. Based on the structure shown inFIG.4, the bootstrap diode Dboot2of the first bootstrap power supply circuit may be reused by the bootstrap capacitor Cboot2of the first bootstrap power supply circuit and the bootstrap capacitor Cboot3of the second bootstrap power supply circuit. Therefore, The bootstrap diode Dboot3of the second bootstrap power supply circuit may be omitted. Furthermore, it is possible to use only one bootstrap capacitor (not shown in the drawings). For the bridge cascade system including 2 bridge topologies (N=2), in a bootstrap power supply circuit scheme in which the bootstrap power supply circuits are connected in a hybrid manner of cascading connection and parallel connection, if one bootstrap power supply circuit is reused, there are only 2 bootstrap power supply circuits in the system, that is, there could be either the cascading connection or the parallel connection. Therefore, when N=2 and there is a reused bootstrap power supply circuit, hybrid connection configurations of the bootstrap power supply circuits is not feasible. When there is reuse of bootstrap power supply circuit(s) and there are more than two bootstrap power supply circuits, it is possible to implement a hybrid scheme of cascading connection and parallel connection. By virtue of reuse of bootstrap power supply circuit(s), the system shown inFIG.9may be simplified as that shown inFIG.11. Referring toFIG.11, there are 3 bridge topologies (N=3), and a first bootstrap power supply circuit (including Cboot2and Dboot2as shown inFIG.11) is reused by a driving circuit Dr3and a driving circuit Dr2; that is, an output terminal of the first bootstrap power supply circuit is directly connected to an input terminal of the driving circuit Dr3and an input terminal of the driving circuit Dr2, respectively. Besides, a second bootstrap power supply circuit (including Cboot4and Dboot4as shown inFIG.11) is reused by a driving circuit Dr5and a driving circuit Dr4; that is, an output terminal of the second bootstrap power supply circuit is directly connected to an input terminal of the driving circuit Dr4and an input terminal of the driving circuit Dr5, respectively. It should be noted that other implementations for reuse of bootstrap power supply circuit(s) are not described in detail herein, and all shall fall within the protection scope of the present disclosure. In the embodiments, by providing bootstrap power supply circuits to replace multiple isolated power supplies and reusing one or more of the bootstrap power supply circuits, the system may be simplified, the number of devices to be applied may be reduced, and the cost for the bridge cascade system may be reduced. In the embodiments shown inFIGS.2-11, bridge topologies in the phase unit are cascaded on AC sides by a common source electrode. In addition to the above manner of cascading by a common source electrode, another manner of cascading is also possible in practical applications, such as cascading by a common drain electrode. That is, in the phase unit, bridge topologies are cascaded on AC sides by a common source electrode or by a common drain electrode. Specifically, as shown inFIGS.2-11, the bridge topologies in the phase unit are cascaded on AC sides by a common source electrode. Taking the structure shown inFIG.2as an example, a source of a switch transistor S3in a second bridge topology is connected to a source of a switch transistor S2in a first bridge topology. As shown inFIG.12andFIG.13, bridge topologies in the phase unit are cascaded on AC sides by a common drain electrode. Taking the structure shown inFIG.13as an example, a drain of a switch transistor S2in a first bridge topology is connected to a drain of a switch transistor S3in a second bridge topology. In any of the foregoing embodiments, the bootstrap power supply circuit includes a bootstrap diode (Dboot2, Dboot3, or Dboot4as shown inFIG.2) and a bootstrap capacitor (Cboot2, Cboot3, or Cboot4as shown inFIG.2). A cathode of the bootstrap diode is connected to a terminal of the bootstrap capacitor, and a connecting point is used as an output terminal of a corresponding bootstrap power supply circuit. An anode of the bootstrap diode is used as an input terminal of the corresponding bootstrap power supply circuit. Another terminal of the bootstrap capacitor is connected to a source of a corresponding switch transistor. As shown inFIG.2, taking the first bootstrap power supply circuit as an example, a cathode of the bootstrap diode Dboot2is connected to a terminal of the bootstrap capacitor Cboot2, and a connecting point is used as an output terminal of the first bootstrap power supply circuit. An anode of the bootstrap diode Dboot2is used as an input terminal of the first bootstrap power supply circuit. Another terminal of the bootstrap capacitor Cboot2is connected to a source of the switch transistor S1. In any of the foregoing embodiments, in the phase unit, the driving circuit of the bridge topology with the lowest potential is directly connected to the driving power supply circuit. As shown inFIG.2, the input terminal of the driving circuit Dr1is directly connected to the output terminal of the driving power supply circuit Vcc. It should be noted that the main differences between a case where the bootstrap power supply circuits are connected in parallel and a case where the bootstrap power supply circuits are connected in series lie in that: timings of charge refresh for the bootstrap capacitors are different and reverse voltage stress of the bootstrap diodes is different. In the case where the bootstrap power supply circuits are connected in parallel, when a bootstrap diode is reversely biased, a reverse withstand voltage is determined by voltage across from the driving circuit where the bootstrap diode is located to the driving power supply circuit Vcc, which indicates that different bootstrap diodes have different reverse withstand voltages. Specifically, the higher the bootstrap diode located, the higher the withstand voltage of the bootstrap diode. In the case where the bootstrap power supply circuits are connected in series, when a bootstrap diode is reversely biased, the reverse withstand voltage is determined by voltage difference across from the driving circuit where the bootstrap diode is located to a previous driving circuit. Correspondingly, by comprehensively considering factors of device selection and cost, it may be determined whether to apply serial connection, parallel connection or a combination of both serial connection and parallel connection to the bootstrap power supply circuits. For the bridge cascade system shown inFIG.2in which the bootstrap power supply circuits are connected in series, a negative bus of the second bridge topology is connected to a neutral point of a bridge arm of the first bridge topology, that is, the switch transistor S2and the switch transistor S3have a common source electrode. The low-voltage side switch transistor S1of the first bridge topology is powered by the driving power supply circuit Vcc alone, and other switch transistors (S2, S3and S4as shown inFIG.2) are powered by the driving power supply circuit Vcc via corresponding bootstrap power supply circuits. The bootstrap power supply circuits are cascaded, that is, the bootstrap power supply circuits are connected in series. The bootstrap capacitors corresponding to the switch transistor S2and the switch transistor S3are charged when the neutral point potential Vs1of the bridge arm of the first bridge topology is pulled down. Generally, the neutral point potential Vs1is pulled down by turning on the switch transistor S1. The bootstrap capacitor corresponding to the switch transistor S4is charged when the neutral point potential Vs2of the bridge arm of the second bridge topology is pulled down. Generally, the neutral point potential Vs2is pulled down by turning on the switch transistor S3. During a non-charging period, the bootstrap diode is reversely biased, the bootstrap capacitors are disconnected from the driving power supply Vcc, and each of the bootstrap capacitors powers a corresponding driving circuit. For the structure of the complementary half-bridge system shown inFIG.3in which the bootstrap power supply circuits are connected in parallel, the power supply scheme of the bootstrap power supply circuits is different from the structure shown inFIG.2in that the bootstrap power supply circuits are connected in parallel. Specifically, as shown inFIG.3, the bootstrap capacitors corresponding to the switch transistor S2and the switch transistor S3are charged when the neutral point potential Vs1is pulled down. For the bootstrap capacitor Cboot4corresponding to the switch transistor S4, in order to make the bootstrap diode Dboot4forward biased, the neutral point potential Vs2is required to be pulled down to the potential of a ground terminal of the driving power supply circuit Vcc, that is, Vin−. Therefore, the bootstrap capacitor Cboot4is charged when the neutral point potential Vs1and the neutral point potential Vs2potential are both pulled down. Correspondingly, there are four working status of the second bridge topology and the first bridge topology. The charging timing of the bootstrap capacitor changes due to a change of the conditions for charging the bootstrap capacitor, which also indicates that the charge support capability of the bootstrap capacitors is adjusted accordingly. In any of the above embodiments, the complementary half-bridge cascade system further includes a controller. The controller is configured to control through each driving circuit to make each bridge topology in each phase unit work in a corresponding working mode. In practical applications, each bridge topology in a same phase unit is controlled to work in either a chopping mode or a pseudo continuous conduction mode (PCCM); and the low-voltage side switch transistor in the bridge topology working in the pseudo continuous conduction mode is intermittently turned on.FIG.20shows a refresh timing of the bootstrap capacitor when the neutral point potential of a corresponding bridge arm is switched in the pseudo continuous conduction mode. InFIG.20, D represents a duty cycle of the high-voltage side switch in a corresponding bridge topology, fss represents a state switching frequency in the pseudo continuous conduction mode; Vin represents an input voltage of the corresponding bridge topology; and Vs represents a neutral point potential of a corresponding bridge arm. Three situations are described below.(1) The controller controls the bridge topologies in the phase unit to all work in the chopping mode or to all work in the pseudo continuous conduction mode, so as to make the bootstrap power supply circuits in the phase unit be charged in one cycle. It should be noted that when there are at least two bootstrap circuits connected in parallel in the phase unit, the chopping waveforms of the respective bridge topologies do not completely overlap with each other, so as to make the bootstrap power supply circuits in the phase unit be charged in one cycle. When there are bootstrap circuits connected in series in the phase unit, the chopping waveforms of the respective bridge topologies may be completely overlapped or not completely overlapped, which depends on an actual situation and is not specifically limited herein, and all shall fall within the protection scope of the present disclosure. Specifically, the controller controls bridge topologies in the phase unit to all work in the chopping mode, and controls the chopping waveforms of the bridge topologies in the phase unit to not completely overlap with each other, so as to make the bootstrap power supply circuits in the phase unit be charged in one cycle; or otherwise, the controller controls bridge topologies in the phase unit to all work in the pseudo continuous conduction mode, and controls the chopping waveforms of the bridge topologies in the phase unit to not completely overlap with each other, so as to make the bootstrap power supply circuits in the phase unit be charged in one cycle. In more detail, the controller controls bridge topologies in the phase unit to all work in the pseudo continuous conduction mode, so as to make the bootstrap power supply circuits in the phase unit be charged in one cycle. It should be noted that in the phase unit, the duty cycle of the bridge topology with high potential may be greater or less than that of the bridge topology with low potential, which depends on an actual situation and is not specifically limited herein, and all shall fall within the protection scope of the present disclosure. When the bootstrap power supply circuits are connected in series, there are various processes for the controller to control each bridge topology in the phase unit to work in the chopping mode or the pseudo continuous conduction mode. The following description is made by referring to an example where N=2 and there are 3 bootstrap power supply circuits in the phase unit.Case 1: a first bridge topology and a second bridge topology are in the pseudo continuous conduction mode Specifically, when the high-voltage side switch transistor of the first bridge topology and the high-voltage side switch transistor of the second bridge topology both require to be conducted, the pseudo continuous conduction mode is applied in place of a conduction mode to realize a charge refresh of the bootstrap capacitors. A condition for the charge refresh of the bootstrap capacitors is the same as that for the first working state, and a timing of the charge refresh is shown as inFIG.14. During a non-charging period, the voltage of the bootstrap capacitors are required to be able to support a normal operation of respective driving circuits, and a low-level bootstrap capacitor needs to have certain charge support ability when charging a high-level bootstrap capacitor.Case 2: a first bridge topology and a second bridge topology are in the chopping mode. Specifically, as shown inFIG.2, the switch transistor S1is powered by the driving power supply circuit Vcc alone. When switch transistor S1is turned on, the driving power supply circuit Vcc charges the bootstrap capacitor Cboot2and the bootstrap capacitor Cboot3. When switch transistor S3is turned on, the driving power supply circuit Vcc also charges the bootstrap capacitor Cboot4if switch transistor S1is turned on: and the bootstrap capacitor Cboot2and the bootstrap capacitor Cboot3charge the bootstrap capacitor Cboot4if switch transistor S2is turned on.FIG.15illustrates a specific example of the refresh timing of the bootstrap capacitors when the neutral point potentials of the bridge arms of the first bridge topology and the second bridge topology are switched. The first bridge topology and the second bridge topology may be at different frequencies; however, during a non-charging period, the voltage of the bootstrap capacitors is required to be able to support the normal operation of respective driving circuits, and especially a low-level bootstrap capacitor needs to have certain charge support ability when charging a high-level bootstrap capacitor.(2) The controller controls at least one bridge topology in the phase unit to work in the chopping mode and at least one bridge topology in the phase unit to work in the pseudo continuous conduction mode, so as to make the bootstrap power supply circuits in the phase unit be charged in one cycle. When the bootstrap power supply circuits are connected in series, there are various processes for the controller to control each bridge topology in the phase unit to work in the chopping mode or the pseudo continuous conduction mode. The following description is made by referring to an example where N=2 and there are 3 bootstrap power supply circuits in the phase unit.Case 1: A first bridge topology in the chopping mode and a second bridge topology in the pseudo continuous conduction mode. Specifically, referring back toFIG.2, when the high-voltage side switch transistor of the second bridge topology requires to be conducted, the pseudo continuous conduction mode is applied in place of a conduction mode to realize a charge refresh of the bootstrap capacitors. A main purpose of the second bridge topology working in the pseudo continuous conduction mode in which the low-voltage side switch transistor is turned on instead is to charge the bootstrap capacitor Cboot4. A condition for charge refresh of the capacitors is the same as the mode described in (1) as above. A specific example of the capacitor refresh timing is shown inFIG.16. Similarly, during a non-charging period, the voltage of the bootstrap capacitors are required to be able to support a normal operation of respective driving circuits, and a low-level bootstrap capacitor needs to have certain charge support ability when charging a high-level bootstrap capacitor.Case 2: A first bridge topology in the pseudo continuous conduction mode and a second bridge topology in the chopping mode. Specifically, referring back toFIG.2, when the high-voltage side switch transistor of the first bridge topology requires to be conducted, the pseudo continuous conduction mode is applied in place of a conduction mode to realize a charge refresh of the bootstrap capacitors. A condition for the charge refresh is the same as that has described above, and a timing of the charge refresh is shown as inFIG.17. The switch transistor S1has a short conduction time. It should be noted that during a non-charging period, the voltage of the bootstrap capacitors are required to be able to support a normal operation of respective driving circuits, and a low-level bootstrap capacitor needs to have certain charge support ability when charging a high-level bootstrap capacitor. It should be noted that for a bridge cascade system with a structure as shown inFIG.3, in which the bootstrap power supply circuits are connected in parallel, if the second bridge topology and the first bridge topology are both in the chopping mode, the timing diagram is as shownFIG.18, rather than that shown inFIG.15. In other working modes, the timing may be analyzed with reference toFIG.16,FIG.17, andFIG.14, which is not described in detail herein, and all shall fall within the protection scope of the present disclosure. In the bridge cascade system, when bootstrap power supply circuits are connected in a hybrid configuration of serial connection and parallel connection, there may be situations based on the conditions for charging the bootstrap capacitors and the working modes of the first bridge topology and the second bridge topology, which are not described in detail herein, and all shall fall within the protection scope of the present disclosure. It should be noted that inFIGS.2-20, Vin, Vin1, Vin2, and Vin3each represent a input end voltage of a corresponding bridge topology, and Vin1+, Vin1−, Vin2+, Vin2− and Vin3+, Vin3− each represent a voltage of electrodes at an input end of a corresponding bridge topology; and Vs, Vs1, Vs2, and Vs3each represent a neutral point potential of a bridge arm of a corresponding bridge topology. The above description is based on the case where bridge topologies in a phase unit are cascaded by a common source electrode. Hereinafter, a power supply relationship of a case where the bridge topologies in the phase unit are cascaded by a common drain electrode is described. It should be noted that when the bridge topologies in the phase unit are cascaded by a common drain electrode, in the phase unit, the high-voltage side switch transistors of two adjacent bridge topologies should not be both turned on, in order to avoid too large voltage on a corresponding bootstrap capacitor. Specifically, as shown inFIG.13, when switch transistor S1is turned on, the driving power supply circuit Vcc charges the bootstrap capacitor Cboot2via the bootstrap diode Dboot2, and the bootstrap diode Dboot3and the bootstrap diode Dboot4are reversely biased. When switch transistor S2is turned on, the bootstrap diode Dboot2is reversely biased. At this time, if switch transistor S3is turned on, the bootstrap diode Dboot3and the bootstrap diode Dboot4are both forward biased, the bootstrap capacitor Cboot2charges the bootstrap capacitor Cboot3and the bootstrap capacitor Cboot4, and the bootstrap capacitor Cboot2has sufficient charge support capacity; or otherwise, if switch transistor S4is turned on, the potential Vs1is pulled up to a positive bus of Vin2, and the bootstrap capacitor Cboot3is applied with an input voltage of Vin2, which is apparently undesirable. Therefore, following tow solutions may be applied.Solution 1: Avoiding turning on both the switch transistor S2and the switch transistor S4, of which a possible timing waveform is shown inFIG.19.Solution 2: Arranging a dedicated bootstrap power supply circuit for bootstrap capacitor Cboot3, where the dedicated bootstrap power supply circuit automatically disconnects from bootstrap capacitor Cboot2when switch transistor S4is turned on. For the above-mentioned cascade arrangements, there are many other designs of bootstrap power supply circuits between the bridge topologies, and the above is only an example. For other cascading arrangements, even for different types of power devices such as PMOS transistors, IGBT transistors and the like, a bridge cascade system may be designed using the concept provided in the present disclosure. Specific design may be carried on by reference to the analysis with regard to the common source cascade system, which is not described in detail, and all shall fall within the protection scope of the present disclosure. The features described in embodiments of the present disclosure may be replaced or combined with each other, the same or similar parts among the embodiments can be referred to each other, and each embodiment places emphasis on the difference from another embodiment. In particular, for the system or the embodiment of the system, since they are similar to the embodiment of the method, the description of the system or the embodiment of the system is simple, and reference may be made to the relevant part of the embodiment of the method. The above system and the above system embodiment are only illustrative. The units described as separate components may be or may not be separated physically, and the components shown as units may be or may not be physical units, that is, the units may be located at the same position or may be distributed onto multiple network units. Some or all modules thereof may be selected based on an actual requirement, to implement an objective of the solution in the current embodiment. Those skilled in the art may understand and implement the present disclosure without any creative effort. It is further understood by those skilled in the art that units and algorithm steps described in combination with the disclosed embodiments may be implemented by electronic hardware, computer software or a combination thereof. In order to clearly describe interchangeability of the hardware and the software, configurations and the steps are generally described above based on functions. Determination regarding implementing the functions by the hardware or the software may depend on specific applications of the technical solutions and design constraints. For each of the specific applications, those skilled in the art may adopt a specific implementation to implement the functions described above, and the implementation should fall within the scope of the present disclosure. Based on the above description of the disclosed embodiments, those skilled in the art may implement or use the present disclosure. Many modifications to these embodiments are apparent for those skilled in the art. The general principles defined herein may be applied to other embodiments without departing from the spirit or scope of the present disclosure. Therefore, the present disclosure is not limited to the embodiments illustrated herein, but is defined by the widest scope consistent with the principles and novel features disclosed herein. | 47,768 |
11863068 | LIST OF REFERENCE SYMBOLS 10semiconductor device101reference voltage generation portion (VREG portion)102switch103reduced-voltage protection portion (UVLO portion)104temperature protection portion (TSD portion)105overvoltage protection portion (OVP portion)106input buffer107oscillator portion108PWM comparator109control logic portion110driver111overcurrent protection portion (OCP portion)112comparator113error amplifier114soft start portion115input buffer116current set portion117constant-current driver118open/short detection portion119input buffer120input buffer20semiconductor device201reference voltage generation portion (VREG portion)202reduced-voltage protection portion (UVLO portion)203temperature protection portion (TSD portion)204short protection portion (SCP portion)205overvoltage protection portion (OVP portion)206overcurrent protection portion (OCP portion)207comparator208control logic portion209input buffer210oscillator portion211slope-voltage generation portion212PWM comparator213driver control portion214driver215N-channel type field effect transistor216driver217error amplifier218soft start portion219input buffer220current set portion221constant-current driver222open/short detection portion223input buffer224input buffer BEST MODE FOR CARRYING OUT THE INVENTION FIG.1is a block diagram showing a first embodiment of a semiconductor device according to the present invention. First, an overview of a semiconductor device10according to the present embodiment is described. The semiconductor device10is a 36 V-resistant white-LED driver IC; and a voltage step-up DC/DC converter and a four-channel output constant-current driver are integrated into one chip. The semiconductor device10is able to perform light control of the white LED by using any of PWM [Pulse Width Modulation] control and VDAC control. Next, features of the semiconductor device10according to the present embodiment are described. A first feature is that the input voltage range of a power-supply voltage VCC is 4.5 to 30[V]. A second feature is that a voltage step-up DC/DC converter is incorporated. A third feature is that a four-channel constant-current driver for supplying an output current ILED to a LED (the maximum electric-current value: 150 [mA]). A fourth feature is that the semiconductor device10interacts with PWM light control (the duty ratio: 0.38 to 99.5[%]). A fifth feature is that various protection functions (UVLO [Under Voltage Lock Out], OVP [Over Voltage Protection], TSD [Thermal Shut Down], OCP [Over Current Protection]) are incorporated. A sixth feature is that a detection function for detecting a LED abnormal state (open/short) is incorporated. A seventh feature is that an HSOP-M28 package (seeFIG.2) is employed. The semiconductor device10according to the present embodiment is used for drive control of a backlight of a car navigation monitor, backlights of medium- and small-sized LCD panels and the like. The semiconductor device10having the above features according to the present embodiment, as shown inFIG.1, is composed of an integration of: a reference voltage generation portion101(hereinafter, called a VREG portion101); a switch102; a reduced-voltage protection portion103(hereinafter, called a UVLO portion103); a temperature protection portion104(hereinafter, called a TSD portion104); an overvoltage protection portion105(hereinafter, called an OVP portion105); an input buffer106; an oscillator portion107; a PWM comparator108; a control logic portion109; a driver110; an overcurrent protection portion111(hereinafter, called an OCP portion111); a comparator112; an error amplifier113; a soft start portion114; an input buffer115; a current set portion116; and a constant-current driver117; an open/short detection portion118; and input buffers119and120. Here, it is possible to roughly divide the above circuit portion of the semiconductor device10into four blocks of: a VREG block (the VREG portion101); a voltage step-up DC/DC controller block (the switch102, input buffer106, oscillator portion107, PWM comparator108, control logic portion109, driver110, OCP portion111, comparator112, error amplifier113and soft start portion114); a current driver block (the input buffer115, current set portion116, constant-current driver117, open/short detection portion118and input buffers119and120); a protection block (the UVLO portion103, TSD portion104, OVP portion105). Besides, the semiconductor device10according to the present embodiment includes28external terminals (1st to 28th pins) as means for securing electric connections with outside. FIG.2is a pin arrangement diagram of the semiconductor device10andFIG.3is a table which shows pin numbers, terminal names and functions of the external terminals. InFIG.2, wide terminals disposed at both sides of central portions of the semiconductor device10are FIN terminals that are linked to subground and improve heat radiation. Next, detailed description of external connections of the semiconductor device10is performed. FIG.4is a diagram for describing the external connections of the semiconductor device10. As for external elements shown inFIG.4, it is desirable that decoupling capacitors CVCC, CREG are connected as close to IC pins as possible. Because a large current is likely to flow in a CS terminal (22nd pin), a GND terminal (7th pin) and a PGND (21st pin), it is desirable to separately wire them and lower the impedances. It is necessary to make sure that noise does not appear on a VDAC terminal (8th pin), an ISET terminal (9th pin), a RT terminal (26th pin) and a COMP terminal (28th pin). It is necessary to make sure that a PWM terminal (5th pin), a SYNC terminal (6th pin), a LED1terminal (12th pin), a LED2terminal (14th pin), a LED3terminal (15th pin) and a LED4terminal (17th pin) do not influence patterns around them because they are switched. It is desirable that thick-line portions inFIG.4are designed with wide patterns and a layout as short as possible. Here, in the semiconductor10according to the present embodiment, because a power transistor Q2is externally connected, it becomes possible to raise heat radiation. FIG.5is a setting table showing examples of constants of external elements. Here, the constants shown as examples in this figure are constants whose operations are confirmed at the power-supply voltage VCC=12[V], LED5in series and 4 in parallel, the output current ILED=50 [mA]. Accordingly, because the optimum values are different depending on use conditions and the like, it is desirable to decide on the constants after a sufficient evaluation. FIG.6is an input/output equivalent circuit diagram of the external terminals. As shown inFIG.6, electrostatic protection diodes are connected with all the external terminals of the semiconductor device10. Besides, as for the PWM terminal (5th pin), SYNC terminal (6th pin), VDAC terminal (8th pin), ISET terminal (9th pin), LEDEN1terminal (10th pin), LEDEN2terminal (11th pin), CS terminal (22nd pin), SWOUT terminal (23rd pin), EN terminal (24th pin), OVP terminal (25th pin), RT terminal (26th pin), SS terminal (27th pin), COMP terminal (28th pin) that are control related terminals, a structure is employed, in which the cathodes of electrostatic diodes on the upper sides (on a route side where electric charges are pulled out from a signal line to a power-supply line) are not connected with application terminals of the reference voltage VREG and the power-supply voltage VCC but with an application terminal of an intermediate voltage CL10V (e.g, 10[V]; see the most right below ofFIG.6). According to such structure, in a case where the power-supply voltage VCC is not applied, or in a case where the reference voltage VREG is not generated by an enable signal EN, even if a positive voltage is applied to an external terminal, an overcurrent does not flow in the reference voltage line and the power-supply voltage line via an electrostatic protection diode, accordingly, it becomes possible to protect breakdown and erroneous operation of the circuit. FIG.7is a table showing electric characteristics of the semiconductor device10that has the above structure. Here, the electric characteristics shown inFIG.7represent numerical values at the power-supply voltage VCC=12 [V], ambient temperature Ta=25[° C.] unless otherwise specified. Next, detailed description of the VREG block (VREG portion101) of the semiconductor device10is performed with reference to the aboveFIG.1and the like. The VREG portion101is a means that generates the reference voltage VREG (5[V] (Typ.)) from the power-supply voltage VCC (12[V]) input into the VCC terminal (1st pin) when the enable signal EN input into the EN terminal (24th pin) is in a high level. This reference voltage VREG is used as a power supply for an internal circuit and also used to fix a terminal at a high-level voltage outside the IC. Besides, the VREG portion101includes a UVLO function, begins operation at 2.9[V] (Typ.) or higher, and stops the operation at 2.8[V] (Typ.) or lower. Here, the VREG terminal (4th pin) is an external terminal to connect the capacitance CREG (10 μF (Typ.)) for phase compensation. By connecting such capacitance CREG for phase compensation, it becomes possible to stabilize circuit operation of the VREG portion101. Next, detailed description of a self-diagnosis function of the semiconductor device10is performed. The semiconductor device10according to the present embodiment, to represent an operation state of a protection circuit incorporated in itself, includes a function to output the FAIL1signal and the FAIL2signal in an open-drain fashion from the FAIL1terminal (3rd pin) and the FAIL2terminal (20th pin), respectively. If any of the UVLO portion103, TSD portion104, OVP portion105and OCP portion111detects an abnormal state and the output signal is brought to a low level, the control logic portion109brings the FAIL1signal to the low level via an output stage shown inFIG.8and fixes the SWOUT terminal (23rd pin) at the low level, thereby stopping the voltage step-up operation. However, because the OCP portion111is of a pulse-by-pulse type, after the SWOUT terminal is fixed at the low level for only one period decided on by the oscillation frequency FOSC of the voltage step-up DC/DC converter, the voltage step-up operation is resumed. According to such pulse-by-pulse type, because it is possible to limit a current without completely stopping the circuit operation, automatic resumption is performed with no delay even if the circuit is stopped by erroneous operation, so that it is easy for the user to operate. Besides, if at least one of the UVLO portion103, TSD portion104and OVP portion105detects an abnormal state, all the LED1terminal, LED2terminal, LED3terminal and LED4terminal (12th pin, 14th pin, 15th pin, 17th pin) are opened (high impedance). Besides, the FAIL1signal output from the FAIL1terminal (3rd pin) and the LOADSW signal output from the LOAD SW terminal (2nd pin) are signals inverted from each other; if the FAIL1signal is brought to the low level, the LOADSW signal is brought to the high level by means of the switch102. Accordingly, if any of the UVLO portion103, TSD portion104, OVP portion105and OCP portion111detects an abnormal state, a load switch (the P-channel type field effect transistor Q1inFIG.4) that is externally connected with the LOAD SW terminal (2nd pin) is turned off. Accordingly, in an abnormal time of the semiconductor device10, the voltage step-up operation is stopped, so that it becomes possible to prevent the IC from being broken, emitting smoke, or burning. On the other hand, the FAIL2signal output from the FAIL2terminal (20th pin) is brought to the low-level output via an output stage shown inFIG.9if the open/short detection portion118detects an abnormal state (an open state or a short state). Here, the FAIL2signal output from the open/short detection portion118is of a latch type, and release of the latch is performed based on on/off (and on/off of the UVLO signal) of the enable signal EN. As shown inFIG.10, if LED terminal voltages V1to V4(the respective terminal voltages of the LED1terminal to LED4terminal) that are to be maintained at a predetermined LED control voltage VLED (0.8 [V] (Typ.)) become 0.15 [V] (Typ.) or lower, the open/short detection portion118determines that the LED terminal is opened; further, if a terminal voltage VP (a divided voltage of an output voltage VOUT) of the OVP terminal (25th pin) reaches 1.7 [V] (Typ.), the open/short detection portion118transmits an instruction to the constant-current drive117so as to turn off a current output for the LED terminal that is judged to be opened and shifts the FAIL2signal to the low level. Here, in the example inFIG.10, a case where the LED1terminal is opened is shown as an example. As described above, by performing the open detection of the LED terminal and the off control of the electric-current output in a two-step fashion, it becomes possible to avoid an unnecessary shutdown. Here, as for the above open detection, it is possible to substitute overvoltage detection by the OVP portion105. Specifically, in the OVP portion105, it is detected that the terminal voltage VP of the OVP terminal reaches a predetermined overvoltage detection voltage VDOVP (2.0 [V] (Typ.)); the voltage step-up operation of the DC/DC converter is stopped; and the electric-current outputs of all the channels are turned off; accordingly, without performing the open detection, it is possible to turn off the electric-current outputs of all the channels by performing the overvoltage detection that doubles as the open detection. Here, to turn off only a channel that is opened, as described above, it is sufficient to identify the LED terminal that is opened and turn off only the channel by monitoring the LED terminal voltages V1to V4. Especially, as for adaptation to an application (a backlight drive device for a car navigation monitor and the like) in which trouble occurs in the use if the electric-current outputs of all the channels are turned off, the structure according to the present embodiment that is able to separately turn off the electric-current outputs of the respective channels is desirable. Besides, the open/short detection portion118determines that a short occurs if the LED terminal voltages V1to V4become 4.5 [V] (Typ.) or higher. In other words, if a difference between the LED terminal voltages in a normal time and an abnormal time becomes 3.7 [V] (=4.5 [V]−0.8 [V]) (Typ.) or higher, the short is detected. Here, because a forward-direction drop voltage VF of the white LED is about 3.4 [V], in the above setting example, a short is not detected even if only one LED shorts; but a short is detected if two or more LEDs short. According to the setting of such a threshold level, it becomes possible to avoid an unnecessary shutdown within an extent where a LED short occurs but serious trouble is not caused in the use. As described above, the short detection in the open/short detection portion118means a detection operation in which for example, one LED that constitutes any one of LED trains that are connected separately and externally with the LED1terminal to LED4terminal goes to a short state (a short-circuit state between the anode and the cathode); as a result of this, a forward-direction drop voltage of the entire LED train becomes low by the forward-direction drop voltage VF of the LED that goes to the short state, so that a state in which one LED terminal voltage becomes higher than the other LED terminal voltages by the forward-direction drop voltage VF of the LED is detected. Accordingly, as shown inFIG.1, the open/short detection portion118and the OVP portion105are formed as protection blocks separate from each other. Here, in the semiconductor device10according to the present embodiment, if an open/short is detected, thereafter, the short detection signal is masked. Describing with reference to the example inFIG.10, after an open of the LED1terminal is detected, the short detection signals of the other LED2terminal to LED4terminal are masked. According to such mask control, as a result of the fact that the LED1terminal is opened, even in a case where the LED terminal voltage V1drops almost to the GND, in response to this, the output voltage VOUT rises; by following this, the LED terminal voltages V2to V4rise higher than usual, this is not erroneously detected as a short. Here, the open detection signal is not masked even after the open/short detection. Besides, the above short detection signal is also masked in an off time of the output current ILED by the PWM drive. According to such mask control, even if the LED terminal voltages V1to V4leap in an off time of the output current ILED, this is not erroneously detected as a short. As for the above mask control, considering that a delay between the logic-shift timing of the PWM signal and the on/off timing of the output current ILED occurs, it is sufficient to mask from the timing of starting to flow the output current ILED (the timing the output transistor of the constant-current driver117is turned on) to the timing the PWM signal falls to the low level (seeFIG.13later described). Besides, if additional capacitance is connected with the LED1terminal to LED4terminal, the LED terminal voltages V1to V4become slow to drop and the short detection can malfunction; accordingly, it is necessary to take care. Besides, because both FAIL1signal and FAIL2signal are of the open-drain type, the FAIL1terminal and the FAIL2terminal are pulled up to the application terminal of the reference voltage VREG via resistors (resistors RFL1, RFL2inFIG.4). Next, detailed description of the current driver block (the input buffer115, current set portion116, constant-current driver117, open/short detection portion118and input buffers119and120) of the semiconductor device10is performed. Of the LED output terminals LED1to LED4, if there is an output terminal (and a train of LEDs that do not go on) that does not use the output current ILED from the constant-current driver117, it is possible to separately turn off the electric-current outputs for he LED output terminals LED1to LED4by using the LEDEN1terminal (10th pin) and the LEDEN2terminal (11th pin). FIG.11is a truth table showing a correlation between input logics of LED enable signals LEDEN1, LEDEN2and on/off states of the LED output terminals LED1to LED4. Here, if a LED terminal that is not used is opened without using the LED enable signals LEDEN1, LEDEN2, the open detection malfunctions in the open/short detection portion118. Besides, even if the electric-current output for the LED terminal is turned off by using the LED enable signals LEDEN1, LEDEN2, the input stage of the error amplifier113operates; accordingly, it is desirable that the LED1terminal to LED4terminal are not fixed to the GND but opened or connected with the application terminal of the constant voltage VREG. Besides, when the output current ILED is PWM-driven, it is desirable not to switch the LED enable signals LEDEN1, LEDEN2. Next, a method for setting the output current ILED is described in detail with reference toFIG.12. FIG.12is a circuit diagram showing structural examples of the current set portion116and the constant-current driver117. As shown inFIG.12, the current set portion116includes: an operational amplifier A1; a direct-current voltage source A2; an npn-type bipolar transistor A3; resistors A4, A5; pnp-type bipolar transistors A6to A9; and a resistor A10(the resistance value R). A first non-inverting input terminal (+) of the operational amplifier A1is connected with the VDAC terminal (8th pin). A second non-inverting input terminal (+) of the operational amplifier A2is connected with a positive-polar terminal of the direct-current voltage source A2, and a predetermined constant voltage VISET (=2.0 [V]) is applied. A negative-polar terminal of the direct-current voltage source A2is connected with a ground terminal. An inverting input terminal (−) of the operational amplifier A2is connected with the ISET terminal (9th pin). A base of the transistor A3is connected with an output terminal of the operational amplifier A1. An emitter of the transistor A3is connected with the ISET terminal. One terminal of each of the resistors A4, A5is connected with the application terminal of the reference voltage VREG. The other terminal of the resistor A4is connected with an emitter of the transistor A6. The other terminal of the resistor R5is connected with an emitter of the transistor A7. Bases of the transistors A6, A7are connected with each other and the connection node is connected with a collector of the transistor A7. A collector of the transistor A6is connected with an emitter of the transistor A8. The collector of the transistor A7is connected with an emitter of the transistor A9. Bases of the transistors A8, A9are connected with each other and the connection node is connected with a collector of the transistor A8. The collector of the transistor A8is connected with the collector of the transistor A3. A collector of the transistor A9is connected with the ground terminal via the resistor A10. On the other hand, as shown inFIG.12, the constant-current driver117includes 4 channels of output stages Ch1, Ch2, Ch3and Ch4that supply the output current ILED to the LED1terminal to LED4terminal, respectively. Here, the output stage Ch1includes: an operational amplifier B1; an N-channel type field effect transistor B2; a resistor B3(the resistance value5R); a current mirror circuit B4(the mirror ratio 1:1); a resistor B5(the resistance value5R); an operational amplifier B6; an N-channel type field effect transistor B7; an resistor B8(the resistance value5R); a current mirror circuit B9(the mirror ratio 1:10); an operational amplifier B10; a direct-current voltage source B11; N-channel type field effect transistors B12to B14; an operational amplifier B15; a direct-current voltage source B16; an N-channel type field effect transistor B17; and a resistor B18. A non-inverting input terminal (+) of the operational amplifier B1is connected the connection node of the transistor A9and the resistor A10. An inverting input terminal (−) of the operational amplifier B1is connected with one terminal of the resistor B3. The other terminal of the resistor B3is connected with the ground terminal. A drain of the transistor B2is connected with an input terminal of the current mirror circuit B4. A source of the transistor B2is connected with one terminal of the resistor B3. A gate of the transistor B2is connected with an output terminal of the operational amplifier B1. A power-supply input terminal of the current mirror circuit B4is connected with the application terminal of the reference voltage VREG. A non-inverting input terminal (+) of the operational amplifier B6is connected with an output terminal of the current mirror circuit B4and with one terminal of the resistor B5. An inverting input terminal of the operational amplifier B6is connected with one terminal of the resistor B8. Both of the other terminals of the resistors B5, B8are connected with the ground terminal. A drain of the transistor B7is connected with an input terminal of the current mirror circuit B9. A source of the transistor B7is connected with one terminal of the resistor B8. A gate of the transistor B7is connected with an output terminal of the operational amplifier B6. A power-supply input terminal of the current mirror circuit B9is connected with the application terminal of the reference voltage VREG. A non-inverting input terminal (+) of the operational amplifier B10is connected with a positive-polar terminal of the direct-current voltage source B11. A negative-polar terminal of the direct-current voltage source B11is connected with the ground terminal. A drain of the transistor B12is connected with an output terminal of the current mirror circuit B9. A source of the transistor B12is connected with an inverting input terminal (−) of the operational amplifier B10. A gate of the transistor B12is connected with an output terminal of the operational amplifier B10. A drain of the transistor B13is connected with the source of the transistor B12. Gates of the transistors B13, B14are connected with each other and the connection node is connected with the drain of the transistor B12and also connected with the ground terminal via the resistor B18. Both sources of the transistors B13, B14are connected with the ground terminal. A non-inverting input terminal (+) of the operational amplifier B15is connected with a positive-polar terminal of the direct-current voltage source B16. A negative-polar terminal of the direct-current voltage source B16is connected with the ground terminal. A drain of the transistor B17is connected with the LED1terminal. A source of the transistor B17is connected with an inverting input terminal (−) of the operational amplifier B15and also connected with a drain of the transistor B14. A gate of the transistor B17is connected with an output terminal of the operational amplifier B15. Here, because the other output stages Ch2to Ch4that constitute the constant-current driver117include the same structure as that of the above output stage Ch1, detailed description of them is skipped. In the current set portion116and the constant-current driver117that have the above structures, the output current ILED is set based on the following formula (1). ILED[mA]=min{VDAC, 2.0[V]}/RSET[kΩ]×3300 (1) In the above formula (1), a parameter min {VDAC, 2.0 [V]} is a voltage value that is the lower of the control voltage VDAC input into the VDAC terminal (8th pin) and the constant voltage VISET (=2.0 [V]) predetermined in the current set portion116. Besides, a parameter RSET is a resistance value of the resistor RSET that is externally connected with the ISET terminal (9th pin); and a parameter 3300 (Typ.) is a constant that is decided on in the constant-current driver117. Specifically, the resistor RSET is pulldowm-connected with the ISET terminal (9th pin), so that an electric current predetermined-gain times (e.g., 3300 times) larger than the reference current ISET flowing in this is set as the maximum value (e.g., 50 [mA]) of the output current ILED. Describing with reference to the example inFIG.12, in the constant-current driver117, first, by using the operational amplifier B1, transistor B2and resistor B3(the resistance value5R), a terminal voltage Va (=ISET×R) of the resistor A10is voltage/current-converted to generate an intermediate current Ia (=1/5ISET) that is ⅕ the reference current ISET. Next, by using the current mirror circuit B4, the intermediate current Ia is mirrored at 1:1 to generate an intermediate current Ib (=1/5ISET). Next, by using the resistor B5(the resistance value5R), the intermediate current Ib is current/voltage-converted to generate a terminal voltage Vb (=ISET×R). Next, by using the operational amplifier B6, transistor B7and resistor B8(the resistance value S5), the terminal voltage Vb of the resistor B5is voltage/current-converted to generate an intermediate current Ic (=1/5ISET). Next, by using the current mirror circuit B9, the intermediate current Ic is mirrored at 1:10 to generate an intermediate current Id (=2ISET) that is two times as large as the reference current ISET. And, finally, by using the current mirror circuit that includes the transistors B13, B14, the intermediate current Id is mirrored at 1:1650 to generate the output current ILED (=3300ISET) that is 3300 times as large as the reference current ISET. Here, to raise accuracy of the output current ILED, in the last-stage current mirror circuit, by using the operational amplifiers B10, B15, drain-source voltages of the transistors B13, B14are made identical to each other (e.g., 0.3 [V]). Besides, the constant-current driver117is so structured as to generate the desired output current ILED by repeating the voltage/current conversion and the current/voltage conversion based on the input reference current ISET. Accordingly, the number of resistor elements (the resistors B3, B5and B8in the example inFIG.12) used for the above conversion processes increases and trimming chances increase. As described above, according to the structure including many resistors that are able to be trimmed, by finely adjusting the resistance values, it is possible to achieve a relative uneven width of ±4% and an absolute uneven width of ±6%, which is able to contribute to reduction in the brightness unevenness and to longevity of the LED. FIG.13is a graph showing a correlation between the resistor RSET and the output current ILED. Here, it is desirable to use a resistor having 300 [kΩ] or smaller as the resistor RSET. Besides, in a case where variable control (light control of the LED) of the output current ILED is performed by using the above control voltage VDAC, it is sufficient to set the input range at a range of 0.1 to 2.0 [V]. By applying such control voltage VDAC, it becomes possible to decrease the output current ILED from the maximum value. On the other hand, in a case where 2.0 [V] or higher is input as the control voltage VDAC, as given by the above formula (1), the voltage value of the constant voltage VISET is selected; accordingly, the light control function by the control voltage VDAC is not used. Here, in a case where the light control by the control voltage VDAC is not used, from the viewpoint of avoidance of malfunction, it is sufficient not to open the VDAC terminal but connect it with the application terminal of the reference voltage VREG (5 [V]). In addition, in the semiconductor device10according the present embodiment, besides the light control of the LED that uses the above control voltage VDAC, by using the PWM signal input into the PWM terminal (5th pin), the on/off control of the reference current ISET is performed, so that it is also possible to perform the light control of the LED. Specifically, based on the PWM signal, if a pulse current is generated as the reference current ISET that serves as the reference for the output current ILED, the duty ratio of the PWM signal becomes the duty ratio of the output current ILED; accordingly, it becomes possible to seemingly decrease the output current ILED from the maximum value (or a current value decided on by the control voltage VDAC). Here, it is sufficient to dispose an on/off control means (a pulse current generation means) for the reference current ISET based on the PWM signal in the output stage (the previous stage of the constant-current driver117) of the current set portion116. Besides, in the semiconductor device10according to the present embodiment, to raise response of the output current ILED to the PWM signal, a pull-down resistor B18(500 [kΩ]) is inserted between the gate and the source of each of the transistors B13, B14in the last-stage current mirror circuit. According to such insertion of the pull-down resistor B18, because it is possible to speed up the rising of the transistors B13, B14, it becomes possible to achieve increase (at the minimum duty ratio of 0.38[%] (150 [Hz])) in the PWM light control capability. On the other hand, in a case where the light control by the PWM signal is not used (the duty ratio is 100%), it is sufficient to fix the PWM terminal at the high level (e.g., the constant voltage VREG). Here, it is desirable to insert a low pass filter (a cut-off frequency of 30 [kHz]) into the PWM terminal. FIGS.14A to14Care timing charts showing examples of the PWM light control, and each show a correlation between the PWM signal and the output current ILED. Here,FIG.14Ashows a case where the PWM signal has a frequency of 150 [Hz] and a duty ratio of 0.38[%];FIG.14Bshows a case where the PWM signal has a frequency of 150 [Hz] and a duty ratio of 50[%]. And,FIG.14Cshows a case where the PWM signal has a frequency of 20 [kHz] and a duty ratio of 50[%]. Here, although all the horizontal axes ofFIGS.14A,14B and14care each a time axis, the frequencies of the PWM signal are extremely different from each other; accordingly, the representing ranges are different from each other. Usually, the frequency of the PWM signal is set fixedly at about 100 to 200 [Hz]. Next, a voltage step-up DC/DC controller block of the semiconductor device10(a circuit block that includes: the input buffer106; oscillator portion107; PWM comparator108; control logic portion109; driver110; OCP portion111; comparator112; error amplifier113and soft start portion114) is described in detail. First, basic operation (voltage step-up operation) of the voltage step-up DC/DC controller block is described in detail with reference to the aboveFIGS.1and4. The transistor Q2is an N-channel field effect type output power transistor that is on/off-controlled depending on an output from the SWOUT terminal (23rd pin). If the transistor Q2is brought into the on state, a switch current flows in the coil L1to the ground terminal via the transistor Q2and its electric energy is stored. Here, in a case where electric charges are already accumulated in the output capacitor CVOUT in an on time of the transistor Q2, a current from the output capacitor CVOUT flows into a train of light emitting diodes that are loads (a train of LEDs connected between a lead-out terminal of the output voltage VOUT and the LED1terminal to LED4terminal, which is not shown inFIG.4, though). Besides, here, because an anode potential of the diode D1drops to almost the ground potential via the transistor Q2, the diode D1goes into a backward bias state and a current does not flow from the output capacitor CVOUT to the transistor Q2. On the other hand, if the transistor Q2is brought into the off state, the electric energy accumulated there is discharged by a backward voltage generated in the coil L1. Here, because the diode D1goes to a forward bias state, the current flowing via the diode D1flows into the LED train that is the load and into the ground terminal as well via the output capacitor CVOUT, thereby charging the output capacitor CVOUT. The above operation is repeated, so that the LED train, that is, the load, is stepped up in voltage by the output capacitor CVOUT and the smoothed output voltage VOUT is supplied. As described above, the semiconductor device10according to the present embodiment functions as a constituent component of a chopper-type voltage step-up circuit that drives the coil L1that is an energy storage element by the on/off control of the transistor Q2, thereby stepping up the power-supply voltage VCC to generate the output voltage VOUT. Next, output feedback control of the voltage step-up DC/DC controller block is described in detail. The error amplifier113amplifies a difference between the smallest value of the LED terminal voltages V1to V4applied respectively to the first to fourth inverting input terminals (−) and the predetermined LED control voltage VLED input into the non-inverting input terminal (+), thereby generating an error voltage Verr. In other words, the voltage value of the error voltage Verr goes to a higher level as the output voltage VOUT becomes lower than its target set value. On the other hand, the PWM comparator108compares the lower of the error voltage Verr and an upper limit voltage Vlmt respectively applied to the first and second non-inverting input terminals (+) with a triangular-wave voltage (lamp-wave voltage) Vosc applied to the inverting input terminal (−), thereby generating a comparison signal (PWM drive waveform) having a duty depending on the comparison result. In other words, the logic of the comparison signal goes to the high level if the error voltage Verr (or the upper limit voltage Vlmt) is higher than the triangular-wave voltage Vosc and goes to the low level if the error voltage Verr (or the upper limit voltage Vlmt) is lower than the triangular-wave voltage Vosc. Accordingly, the on duty (the ratio of an on time of the transistor Q2per unit time) of the comparison signal in a steady operation time changes depending on a relative height difference between the error voltage Verr and the triangular-wave voltage Vosc. During a time the above comparison signal is maintained at the high level, the control logic portion109holds the terminal voltage (i.e., the gate voltage of the transistor Q2) of the SWOUT terminal at the high level via the driver110. Accordingly, the transistor Q2is brought into the on state. On the other hand, during a time the comparison signal is maintained at the low level, the terminal voltage of the SWOUT terminal is held at the low level. Accordingly, the transistor Q2is brought into the off state. As described above, the voltage step-up DC/DC controller block is so structured as to perform the drive control of the transistor Q2based on monitoring results of the LED terminal voltages V1to V4(and the output voltage VOUT). Accordingly, it becomes possible to maintain the output voltage VOUT at a predetermined value. Next, the series number of the LED train that is the load is described. As described above, the voltage step-up DC/DC controller block of the semiconductor device10detects the cathode voltage (i.e., the LED terminal voltages V1to V4) of the LED train and controls the output voltage VOUT applied to the anode of the LED train so as to allow the cathode voltage to become the LED control voltage VLED (=0.8 [V] (Typ.)). The above voltage step-up operation is performed only when the PWM signal is in the high level and the output current ILED is flown into the LED train. Beside, when a plurality of LED trains are driven, the LED terminal voltage (in other words, the smallest value of the LED terminal voltages) of the LED train that has the largest forward-direction drop voltage VF of the LED is so controlled as to become identical to the LED control voltage VLED. Accordingly, the LED terminal voltages of the other trains of LED terminal voltages become a voltage that is higher by a difference in the forward-direction drop voltage VF. Here, a difference allowable voltage Vper (=3.7 [V] (Typ.)) of the forward-direction drop voltage VF is set by the following formula (2) based on a short detection voltage VDSHT (=4.5 [V] (Typ.)) and the LED control voltage VLED (=0.8 [V] (Typ.)). Vper=VDSHT−VLED (2) Besides, in detecting an open by the open/short detection portion118, 85% of an overvoltage detection reference voltage VDOVP (=2.0 [V] (Typ.)) is set as a trigger voltage (open detection voltage VD OP2(=1.7 [V] (Typ.)) (seeFIGS.7and10). When this is converted into the output voltage VOUT, the maximum value of the output voltage VOUT in the usual operation time becomes 30.6 [V]=36 [V]×0.85. Accordingly, the series number N of the LED is limited so as to make the series number N smaller than a value that is obtained by dividing the maximum value 30.6 [V] of the output voltage VOUT by the forward-direction drop voltage VF of one LED. Next, the OVP portion105is described. A divided voltage VP obtained by resistance-dividing the output voltage VOUT is input into the OVP terminal (25th pin). As described above, based on the series number N of the LED train and on the difference allowable voltage Vper of the forward-direction drop voltage VF, it is sufficient to suitably decide on the overvoltage detection reference voltage VD OVP of the OVP portion105that is compared with this. Besides, in deciding on the overvoltage detection reference voltage VDOVP, it should be decided on considering the open detection voltage VD OP2(=VDOVP×0.85) as well. Here, after the OVP portion105starts once the protection operation, the protection operation is released when the output voltage VOUT decreases to 77.5% of the overvoltage detection reference voltage VDOVP. For example, in a case where the resistance values of the resistance division circuit are ROVP1(on the voltage step-up side), ROVP2(on the GND side), when the output VOUT meets the following formula (3), the protection operation of the OVP portion105starts. VOUT≥(ROVP1+ROVP2)ROVP2×VDOVP(3) Here, when ROVP1=330 [kΩ], ROVP2=22 [kΩ], and VDOVP=2.0 [V], the protection operation of the OVP portion105starts at VOUT=32 [V] or higher Next, the oscillation frequency FOSC of the voltage step-up DC/DC converter is described. By externally connecting a pull-down resistor RT with the RT terminal (26th pin), charge and discharge currents for an internal capacitor of the oscillator portion107are decided on and it is possible to set the oscillation frequency FOSC of the triangular-wave voltage Vosc. The resistance value of the pull-down resistor RT externally connected with the RT terminal may be set in view of the following formula (4) andFIG.15, and a range of 62.6 to 523 [kΩ] is desirable. FOSC[kHz]=30×106RT[Ω]×α(4) Here, in the above formula (4), 30×106[V/A/S] is a constant (±16.6%) that is decided on inside the circuit, and α is a correction coefficient (RT: α=50 [kΩ]: 0.98, 60 [kΩ]: 0.985, 70 [kΩ]: 0.99, 80 [kΩ]: 0.994, 90 [kΩ]: 0.996, 100 [kΩ]: 1.0, 150 [kΩ]: 1.01, 200 [kΩ]: 1.02, 300 [kΩ]: 1.03, 400 [kΩ]: 1.04, 500 [kΩ]: 1.045). Besides, in the setting outside the frequency range inFIG.15, it is necessary to take care because the switching is likely to stop. Next, an external synchronization oscillation frequency FSYNC is described. When a clock for external synchronization is input into the SYNC terminal (6th pin) of the voltage step-up DC/DC converter, it is better not to perform operations such as switching to internal oscillation and the like during the manipulation. After the input logic of the SYNC terminal is switched from the high level to the low level, it takes a delay time of about 30 [μsec] (Typ.) until an internal oscillation circuit starts to operate. The clock input into the SYNC terminal is valid at only the rising edge. Besides, in a case where the external input frequency is lower than the internal oscillation frequency, because after the above delay time, the internal oscillation circuit starts to operate, such an input should be avoided. As described above, in the semiconductor device10according to the present embodiment, by using the RT terminal or the SYNC terminal, it is possible to perform variable control of the oscillation frequency FOSC of the voltage step-up DCDC converter block arbitrarily and with great accuracy. For example, in a case where the semiconductor device10according to the present embodiment is used as a backlight control means for a car navigation monitor, it is possible to avoid the oscillation frequency FOSC of the voltage step-up DC/DC converter overlapping the frequency band of radio noise by suitably setting the external synchronization oscillation frequency FSYNC from the SYNC terminal matching the switching control of a radio reception frequency; accordingly, it becomes possible to perform the backlight control of the car navigation monitor without deteriorating the reception quality of radio. Next, the OCP portion111is described with reference toFIG.16. FIG.16is a diagram for describing a connection relationship among external elements related to operation of the OCP portion111. As shown inFIG.16, a detection resistor RCS is inserted between the source of the power transistor Q2(N-channel field effect transistor) for the voltage step-up DC/DC converter and the GND, and the connection node is connected with the CS terminal (22nd pin). Besides, to reduce switching noise (spike noise), a low pass filter (a resistor RLPF and a capacitor CLPF) having a cut-off frequency of 1 to 2 [MHz] is inserted between the CS terminal and the detection resistor RCS. Here, if the time constant of the low pass filter LPF is too large, the rising of the CS terminal voltage becomes slow, and the detection operation of the OCP portion111becomes slow; accordingly, for example, it is appropriate that RLPF=100 [Ω], CLPF=1000 [pF] when the oscillation frequency FOSC=300 [kHz]. Besides, a detection current IOCP in the OCP portion111is decided on by the following formula (5) based on the overcurrent protection operation voltage VDCS (a constant voltage applied to the non-inverting input terminal (+) of the comparator112) and the detection resistor RCS. IOCP[A]=VDCS(=0.4[V]/RCS[Ω] (5) Besides, because the OCP portion111is of the pulse-by-pulse type, the SWOUT terminal is fixed at the low level for one period decided on by the oscillation frequency FOSC of the voltage step-up DC/DC converter; thereafter, the voltage step-up operation is resumed. Besides, because a large-current line is formed between the detection resistor RCS and the GND, separate wiring to the GND should be performed in the board designing. Next, the soft start portion114is described. In the semiconductor device10according to the present embodiment, the SS terminal (27th pin) is unused and opened. Besides, the open/short detection function of the open/short detection portion118is masked until the SS terminal voltage reaches a clamp voltage of 2.5 [V] (Typ.). Next, detailed description of selection of external components is performed. First, selection of the coil L1is described in detail with reference toFIGS.17A and17B. FIGS.17A,17Bare each a diagram for describing selection of the coil L1. Here, inFIG.17A, a ripple component ΔIL of the coil current IL is shown, and inFIG.17B, a circuit that constitutes input and output stages of the DC/DC converter is shown. The inductor value of the coil L1extremely influences the ripple component ΔIL (a difference between the maximum value ILMAX and the minimum value ILMIN of the coil current IL) of the coil current IL. Specifically, as shown by the following formula (6), the larger the inductor value of the coil L1is, and the higher the oscillation frequency FOSC is, the smaller the ripple component ΔIL becomes. ΔIL[A]=(VOUT-VCC)×VCCL1×VOUT×FOSC(6) Besides, when the efficiency η is represented as shown by the following formula (7a), the maximum value ILMAX of the coil current IL becomes as shown by the formula (7b). η=VOUT×IOUTVCC×ICC(7a)ILMAX[A]=ICC+ΔIL2=VOUT×IOUTVCC×η+ΔIL2(7b) If the coil current IL exceeding the rated current value of the coil L1is flown in the coil L1, the coil L1reaches magnetic saturation and the efficiency η decreases. Accordingly, the coil L1should be selected with a sufficient margin so as not to allow the maximum value ILMAX of the coil current IL to exceed the rated current value of the coil L1. Besides, to reduce loss in the coil L1and improve the efficiency η, as the coil L1, a coil that has a small resistance component (a direct-current reactor DCR, an alternating-current ACR) should be selected. Next, detailed description of selection of the output capacitor CVOUT is performed with reference toFIG.18. FIG.18is a diagram for describing selection of the output capacitor CVOUT, and a circuit that constitutes the input and output stages of the DC/DC converter is shown. As for the selection of the output capacitor CVOUT, in light of a stable domain of the output voltage VOUT and further considering an equivalent series resistance ESR necessary to smooth a ripple component ΔVOUT of the output voltage, it is sufficient to suitably decide on the output capacitor CVOUT. The ripple component ΔVOUT of the output voltage VOUT is decided on as shown by the following formula (8). ΔVOUT[V]=ILMAX×RESR+1CVOUT×IOUTη×1FOSC(8) Here, in the above formula (8), ΔIL represents the ripple component of the output current IL; RESR represents the resistance value of the equivalent series resistance ESR of the output capacitor CVOUT; and η represents the efficiency. Here, it is desirable to select the rating of the output capacitor CVOUT with a sufficient margin with respect to the output voltage VOUT. Next, selection of the input capacitor CVCC is described in detail with reference toFIG.19. FIG.19is a diagram for describing selection of the input capacitor CVCC, and a circuit that constitutes the input and output stages of the DC/DC converter is shown. As for the selection of the input capacitor CVCC, it is desirable to use an input capacitor having a low ESR that has a capacitance value capable of sufficiently interacting with a large ripple current IRMS so as to prevent a large transient voltage. The above ripple current IRMS is given by the following formula (9). IRMS[A]=IOUT×(VOUT-VCC)×VOUTVOUT(9) Besides, because the ripple current IRMS extremely depends on the characteristics of the power supply used for the input, the wiring pattern of the board and the gate-drain capacitances of the transistors Q1, Q2, it is desirable to sufficiently confirm the temperature, the load range, and the conditions of the transistors Q1, Q2in a time of use. Next, selection of the load-switch transistor Q1and its soft start are described in detail with reference toFIGS.20and21. FIG.20is a diagram for describing selection of the load-switch transistor Q1and its soft start, and a circuit that constitutes the input and output stages of the DC/DC converter is shown. Besides,FIG.21is a graph showing a correlation between the gate-source capacitance of the transistor Q1and the soft start time. In a case of a usual voltage step-up application, there is no switch on the route that extends from the application terminal of the power-supply voltage VCC to the lead-out terminal of the output voltage VOUT; accordingly, if an output short-circuit occurs, an overcurrent flows in the route, so that the coil L1and the rectifying diode D1can be broken. To avoid this, in the semiconductor device10according to the present embodiment, the P-channel type field effect transistor Q1for the load-switch is inserted between the application terminal of the power-supply voltage VCC and the coil L1. Here, as the transistor Q1, it is sufficient to select a transistor whose gate-source breakdown voltage and whose drain-source breakdown voltage are both higher than the power-supply voltage VCC. Besides, to perform the soft start of the load switch, it is sufficient to insert capacitance between the gate and source of the transistor Q1. According to this, as shown inFIG.21, it is possible arbitrarily to decide on the soft start time depending on the inserted capacitance value. However, the soft start time changes depending on the gate capacitance of the transistor Q1as well. Next, selection of the switching transistor Q2is described. There is no problem whatever MOSFET is used if the MOSFET has the absolute maximum rated current that is larger than the rated current of the coil L1and the absolute maximum rated voltage that is higher than the breakdown voltage of the output capacitor CVOUT+the forward-direction drop voltage VF of the rectifying diode D1; however, to achieve high-speed switching, a MOSFET that has a small gate capacitance (the amount of injected electric charges) should be selected, and it is desirable to use a MOSFET that has gate capacitance lager than the overcurrent protection set value. Besides, if a MOSFET that has a small on resistance is selected, it becomes possible to obtain high efficiency. Next, selection of the rectifying diode D1is described. Whatever diode may be used if the diode is a Schottky barrier diode that has an electric-current capability equal to or larger than the rated current of the coil L1and a backward breakdown voltage equal to or higher than the breakdown voltage of the output capacitor CVOUT; especially, it is sufficient select a diode whose forward-direction drop voltage VF is low. Next, detailed description of a phase-compensation setting method is performed. First, stable conditions of an application are described. As for conditions under which a feedback system with negative feedback returned becomes stable, it is necessary that the phase delay is 150° or smaller (i.e., the phase margin is 30° or larger) at a time the gain is 1 (0 [dB]). Besides, because the DC/DC converter application is sampled by the oscillation frequency FOSC, it is necessary to set the GBW (Gain-Band Width) (a product of the gain and the band width) of the entire system at a value that is 1/10 the oscillation frequency FOSC or lower. Summing up the above description, the characteristics the application aims at are the phase delay that is 150° or smaller (i.e., the phase margin is 30° or larger) at the time the gain is 1 (0 [dB]); and it is sufficient if the GBW (i.e., the frequency at the gain of 0 [dB]) at that time is 1/10 the oscillation frequency FOSC or lower. Accordingly, to improve the response according to a limit to the GBW, it is necessary to raise the oscillation frequency FOSC. To secure the stability by phase compensation, it is sufficient to cancel a secondary phase delay (−180°) due to an LC resonance by a secondary phase advance (i.e., two phase advances are used). Here, as a means for giving a phase advance, the ESR component (seeFIG.23) of the output capacitor CVOUT and the CR component (seeFIG.24) connected with the COMP terminal (28th pin) are possible. In the DC/DC converter application, as shown inFIG.22, there is invariably an LC resonance circuit. Accordingly, the phase delay at that portion becomes −180°. As shown inFIG.23, in a case where the output capacitor CVOUT is an element such as an aluminum electrolytic capacitor or the like that has a large ESR (a few ohms [Ω]), a phase advance of +90° occurs and the phase delay becomes −90°. On the other hand, in a case where the output capacitor CVOUT such as a ceramic capacitor or the like that has a low ESR is used, it is necessary to insert resistance equal to the ESR component. Here, because of a change in the phase characteristic due to the ESR, the phase advance to be inserted becomes one. Besides, as for setting of the frequency into which a phase advance is inserted, for the purpose of canceling the LC resonance, it is ideally desirable to set the frequency near the LC resonance frequency. Next, an operation sequence of the semiconductor device10is described with reference toFIG.25. FIG.25is a timing chart showing an operation sequence of the semiconductor device10. When the enable signal EN rises to high level after the power-supply voltage VCC is turned on, generation of the reference voltage VREG is started in the VREG portion101. Here, as for the enable signal EN, after the power-supply voltage VCC sufficiently rises, for example, after the power-supply voltage VCC becomes 4.5 [V] or higher, it is sufficient to turn on the enable signal EN. When the reference voltage VREG reaches 2.9 [V], it is recognized by the UVLO portion103that it is not a reduced-voltage state, and the UVLO signal rises to the high level. According to this, the internal circuit of the semiconductor device10starts to operate. Here, during a time the UVLO signal is maintained in the low level, the switch102is maintained in the off state and the terminal voltage of the LOADSW terminal (2nd pin) is maintained in the high level. Accordingly, because the load-switch transistor Q1is turned off, the voltage step-up operation of the DC/DC converter is maintained in a stop state. On the other hand, when the UVLO signal rises to the high level, the switch102is turned on, and the terminal voltage of the LOADSW terminal falls to the low level. As a result of this, the load-switch transistor Q1is turned on, and the voltage step-up operation becomes possible. For stable operation, there are predetermined input sequences for the VDAC signal, SYNC signal and PWM signal that are external signals. Specifically, it is desirable to input the VDAC signal and SYNC signal after a first predetermined time TINON elapses from the input timing of the enable signal EN; and it is desirable to input the PWM signal after a second predetermined time TPWMON elapses from the input timing of the EN signal. Here, the second predetermined time TPWMON>the first predetermined time TINON and the second predetermined time TPWMON>500 [V/A·s]×CREG [sec]. Besides, it is desirable to block the inputs of the VDAC signal and SYNC signal earlier than the EN signal by a third predetermined time TINOFF, while it is desirable to block the input of the PWM signal earlier than the EN signal by a fourth predetermined time TPWMOFF. Here, the fourth predetermined time TPWMOFF>the third predetermined time TINOFF. Besides, although not shown in this figure, it is desirable fix the logics of the LEDEN1signal and LEDEN2signal before the EN signal shifts to the high level. In the OVP portion105, when the terminal voltage of the OVP terminal (25th pin) reaches 2 [V], it is recognized as an overvoltage state, and the voltage step-up operation is stopped. Thereafter, in the OVP portion105, when the terminal voltage of the OVP terminal drops to 1.6 [V], it is recognized that the overvoltage state is released, and the voltage step-up operation is resumed. In the OCP portion111, when the terminal voltage of the CS terminal (22nd pin) reaches 0.4 [V], it is recognized as an overcurrent state, and thereafter, the voltage step-up operation of the DC/DC converter is intermittently turned on/off by the pulse-by-pulse fashion. In the TSD portion104, when the temperature of the semiconductor device10reaches 175[° C.], it is recognized as an abnormal heating state, and the voltage step-up operation of the DC/DC converter is stopped. Thereafter, in the TSD portion104, when the temperature of the semiconductor device10drops to 150[° C.], it is recognized that the abnormal heating state is released, and the voltage step-up operation is resumed. Here, if the EN signal is made fall to the low level, the generation of the reference voltage VREG is stopped. In the UVLO portion103, when this reference voltage VREG drops to 2.8 [V], it is recognized as a reduced-voltage state, and the UVLO signal falls to the low level. In this way, the internal circuit of the semiconductor device10stops the operation. Next, detailed description of a second embodiment of the semiconductor device according to the present invention is performed. FIG.26is a block diagram showing a second embodiment of the semiconductor device according to the present invention. First, an overview of a semiconductor device20according to the present embodiment is described. The semiconductor device20is a 36 V-resistant white-LED driver IC; and a voltage step-up and -down DC/DC converter of a current mode and a four-channel output constant-current driver are integrated into one chip. The semiconductor device20is able to perform light control of the white LED by using any of PWM [Pulse Width Modulation] control and VDAC control. Next, of features of the semiconductor device20according to the present embodiment, especially, points different from the first embodiment is described. A first feature is that to interact with the power-supply voltage VCC directly supplied from a battery, a voltage step-up/-down DC/DC controller block is incorporated instead of the voltage step-up DC/DC controller block. A second feature is that to use a low-ESR ceramic capacitor as the output capacitor CVOUT, the control mode of the DC/DC converter is changed from a voltage mode to an electric-current mode. A third feature is that to raise the PWM light control capability of LED luminous brightness, a duty ratio (with no overshoot) of 0.38[%] is achieved. A fourth feature is that a relative uneven width ±3% of the output current ILED and an absolute uneven width ±5% of the output current ILED are achieved. A fifth feature is that a protection function portion (SCP [Short Circuit Protection]) which detects a short in the anode and cathode of the LED and performs an appropriate protection operation is incorporated. Here, the semiconductor device20according to the present embodiment is used for drive control of a backlight of a car navigation monitor, backlights of medium- and small-sized LED panels and the like. The semiconductor device20according to the present embodiment having the above features, as shown inFIG.26, is composed of an integration of: a reference voltage generation portion201(hereinafter, called a VREG portion201); a reduced-voltage protection portion202(hereinafter, called a UVLO portion202); a temperature protection portion203(hereinafter, called a TSD portion203); a short protection portion204(hereinafter, called a SCP portion204); an overvoltage protection portion205(hereinafter, called an OVP portion205); an overcurrent protection portion206(hereinafter, called an OCP portion206); a comparator207; a control logic portion208; an input buffer209; an oscillator portion210; a slope-voltage generation portion211; a PWM comparator212; a driver control portion213; a driver214; an N-channel type field effect transistor215; a driver216; an error amplifier217; a soft start portion218; an input buffer219; a current set portion220; a constant-current driver221; an open/short detection portion222; and input buffers223and224. Here, it is possible to roughly divide the above circuit portion of the semiconductor device20into four blocks of: a VREG block (the VREG portion201); a voltage step-up and -down DC/DC controller block (the OCP portion206, comparator207, control logic portion208, input buffer209, oscillator portion210, slope-voltage generation portion211, PWM comparator212, driver control portion213, driver214, transistor215, driver216, error amplifier217and soft start portion218); a current driver block (the input buffer219, current set portion220, constant-current driver221, open/short detection portion222and input buffers223and224); and a protection block (the UVLO portion202, TSD portion203, SCP portion204and OVP portion205). Besides, the semiconductor device20according to the present embodiment includes28external terminals (1st to 28th pins) as means for securing electric connections with outside. FIG.27is a pin arrangement diagram of the semiconductor device20andFIG.28is a table which shows pin numbers, terminal names and functions of the external terminals. InFIG.27, wide terminals disposed at both sides of central portions of the semiconductor device20are FIN terminals that are linked to subground and improve heat radiation. FIG.29is a table showing electric characteristics of the semiconductor device20that has the above structure. Here, the electric characteristics shown inFIG.29represent numerical values at the power-supply voltage VCC=12 [V], ambient temperature Ta=25[° C.] unless otherwise specified. Next, detailed description of operation of each portion of the semiconductor device20is performed centering on points different from the first embodiment. First, detailed description of the current driver block of the semiconductor device20(the input buffer219, current set portion220, constant-current driver221, open/short detection portion222and input buffers223and224) is performed. FIG.30is a circuit diagram showing structural examples of the current set portion220and the constant-current driver221. As shown inFIG.30, the current set portion220includes: the operational amplifier A1; the direct-current voltage source A2; the npn-type bipolar transistor A3; the resistors A4, A5; the pnp-type bipolar transistors A6to A9; and the resistor A10(the resistance value R). The first non-inverting input terminal (+) of the operational amplifier A1is connected with the VDAC terminal (8th pin). The second non-inverting input terminal (+) of the operational amplifier A2is connected with the positive-polar terminal of the direct-current voltage source A2, and the predetermined constant voltage VISET (=2.0 [V]) is applied. The negative-polar terminal of the direct-current voltage source A2is connected with the ground terminal. The inverting input terminal (−) of the operational amplifier A2is connected with the ISET terminal (9th pin). The base of the transistor A3is connected with the output terminal of the operational amplifier A1. The emitter of the transistor A3is connected with the ISET terminal. One terminal of each of the resistors A4, A5is connected with the application terminal of the reference voltage VREG. The other terminal of the resistor A4is connected with the emitter of the transistor A6. The other terminal of the resistor A5is connected with the emitter of the transistor A7. The bases of the transistors A6, A7are connected with each other and the connection node is connected with the collector of the transistor A7. The collector of the transistor A6is connected with the emitter of the transistor A8. The collector of the transistor A7is connected with the emitter of the transistor A9. The bases of the transistors A8, A9are connected with each other and the connection node is connected with the collector of the transistor A8. The collector of the transistor A8is connected with the collector of the transistor A3. The collector of the transistor A9is connected with the ground terminal via the resistor A10. On the other hand, as shown inFIG.30, the constant-current driver221includes 4 channels of output stages Ch1, Ch2, Ch3and Ch4that supply the output current ILED to the LED1terminal to LED4terminal, respectively. Here, the output stage Ch1includes: the operational amplifier B1; the N-channel type field effect transistor B2; the resistor B3(the resistance value4R); the current mirror circuit B4(the mirror ratio 1:1); the resistor B5(the resistance value4R); the operational amplifier B6; the N-channel type field effect transistor B7; the resistor B8(the resistance value (4/12)×R); the current mirror circuit B9(the mirror ratio 1:10); the operational amplifier B10; the direct-current voltage source B11; the N-channel type field effect transistors B12to B14; the operational amplifier B15; the direct-current voltage source B16; the N-channel type field effect transistor B17; an N-channel type field effect transistor B19; a P-channel type field effect transistor B20; resistors B21, B22; an N-channel type field effect transistor B23; and an inverter B24. The non-inverting input terminal (+) of the operational amplifier B1is connected the connection node of the transistor A9and the resistor A10. The inverting input terminal (−) of the operational amplifier B1is connected with one terminal of the resistor B3. The other terminal of the resistor B3is connected with the ground terminal. The drain of the transistor B2is connected with the input terminal of the current mirror circuit B4. The source of the transistor B2is connected with one terminal of the resistor B3. The gate of the transistor B2is connected with the output terminal of the operational amplifier B1. The power-supply input terminal of the current mirror circuit B4is connected with the application terminal of the reference voltage VREG. The non-inverting input terminal (+) of the operational amplifier B6is connected with the output terminal of the current mirror circuit B4and with one terminal of the resistor B5. The inverting input terminal of the operational amplifier B6is connected with one terminal of the resistor B8. Both of the other terminals of the resistors B5, B8are connected with the ground terminal. The drain of the transistor B7is connected with the input terminal of the current mirror circuit B9. The source of the transistor B7is connected with one terminal of the resistor B8. The gate of the transistor B7is connected with the output terminal of the operational amplifier B6. The power-supply input terminal of the current mirror circuit B9is connected with the application terminal of the reference voltage VREG. The non-inverting input terminal (+) of the operational amplifier B10is connected with the positive-polar terminal of the direct-current voltage source B11. The negative-polar terminal of the direct-current voltage source B11is connected with the ground terminal. The drain of the transistor B12is connected with the output terminal of the current mirror circuit B9. The source of the transistor B12is connected with the inverting input terminal (−) of the operational amplifier B10. The gate of the transistor B12is connected with the output terminal of the operational amplifier B10. The drain of the transistor B13is connected with the source of the transistor B12. The gates of the transistors B13, B14are connected with each other and the connection node is connected with the drain of the transistor B12and also connected with the drain of the transistor B19. All the sources of the transistors B13, B14and B19are connected with the ground terminal. The gate of the transistor B19is connected with the PWM terminal (8th pin) via the input buffer219(not shown in this figure). The non-inverting input terminal (+) of the operational amplifier B15is connected with the positive-polar terminal of the direct-current voltage source B16. The negative-polar terminal of the direct-current voltage source B16is connected with the ground terminal. The drain of the transistor B17is connected with the LED1terminal. The source of the transistor B12is connected with the inverting input terminal (−) of the operational amplifier B15and also connected with the drain of the transistor B14. The gate of the transistor B17is connected with the output terminal of the operational amplifier B15. A source of the transistor B20is connected with the application terminal of the reference voltage VREG. A drain of the transistor B20is connected with the input terminal of the current mirror circuit B9. One terminal of the resistor B21is connected with the application terminal of the reference voltage VREG. The other terminal of the resistor B21is connected with a gate of the transistor B20. One terminal of the resistor B22is connected with the gate of the transistor B20. The other terminal of the resistor B22is connected with a drain of the transistor B23. A source of the transistor B23is connected with the ground terminal. A gate of the transistor B23is connected with an output terminal of the inverter B24. An input terminal of the inverter B24is connected with the PWM terminal via the input buffer219(not shown in this figure). Here, because the other output stages Ch2to Ch4that constitute the constant-current driver221include the same structure as that of the above output stage Ch1, detailed description of them is skipped. In the current set portion220and the constant-current driver221that have the above structures, the output current ILED is set based on the following formula (10). ILED[mA]=min{VDAC, 2.0[V]}/RSET[kΩ]×3000 (10) In the above formula (10), the parameter min {VDAC, 2.0 [V]} is a voltage value that is the lower of the control voltage VDAC input into the VDAC terminal (18th pin) and the constant voltage VISET (=2.0 [V]) predetermined in the current set portion220. Besides, the parameter RSET is a resistance value of the resistor RSET that is externally connected with the ISET terminal (19th pin); and a parameter 3000 (Typ.) is a constant that is decided on in the constant-current driver221. Specifically, the resistor RSET is pulldown-connected with the ISET terminal (19th pin), so that an electric current predetermined-gain times (e.g., 3000 times) higher than the reference current ISET flowing in this is set as the maximum value (e.g., 50 [mA]) of the output current ILED. Describing specifically with reference to the example inFIG.30, in the constant-current driver221, first, by using the operational amplifier B1, transistor B2and resistor B3(the resistance value4R), the terminal voltage Va (=ISET×R) of the resistor A10is voltage/current-converted to generate the intermediate current Ia (=1/4ISET) that is ¼ the reference current ISET. Next, by using the current mirror circuit B4, the intermediate current Ia is mirrored at 1:1 to generate the intermediate current Ib (=1/4ISET). Next, by using the resistor B5(the resistance value4R), the intermediate Ib is current/voltage-converted to generate the terminal voltage Vb (=ISET×R). Next, by using the operational amplifier B6, transistor B7and resistor B8(the resistance value (4/12)×R)), the terminal voltage Vb of the resistor B5is voltage/current-converted to generate the intermediate current Ic (=3ISET) that is times as large as the reference current ISET. Next, by using the current mirror circuit B9, the intermediate current Ic is mirrored at 1:10 to generate the intermediate current Id (=30ISET) that is 30 times as large as the reference current ISET. And, finally, by using the current mirror circuit that includes the transistors B13, B14, the intermediate current Id is mirrored at 1:100 to generate the output current ILED (=3000ISET) that is 3000 times as large as the reference current ISET. Here, to raise the accuracy of the output current ILED, in the last-stage current mirror circuit, by using the operational amplifiers B10, B15, the drain-source voltages of the transistors B13, B14are made identical to each other (e.g., 0.3 [V]). Besides, the constant-current driver221is so structured as to generate the desired output current ILED by repeating the voltage/current conversion and the current/voltage conversion based on the input reference current ISET. Accordingly, resistor elements (the resistors B3, B5and B8in the example inFIG.30) used for the above conversion processes increase and the trimming chances increase. As described above, according to the structure including many resistors that are able to be trimmed, by finely adjusting the resistance values, it becomes possible to reduce the relative uneven width and the absolute uneven width of the output current ILED. Unlike the fist embodiment that amplitudes the reference current ISET without stopping by the last-stage current mirror circuit (ISET→1/5ISET→2ISET→3300ISET), the semiconductor device20according to the second embodiment is so structured as to dispersedly perform current amplification in the generation process of the output current ILED in the constant-current driver221(ISET→1/4ISET→3ISET→30ISET→3000ISET). According to such structure, as shown by a comparison ofFIG.31AandFIG.31B, it is possible to reduce the unevenness in the production of elements and the influence of stress by decreasing a difference between the transistor sizes of the transistors B13, B14that constitute the last-stage current mirror circuit. Specifically, if the difference between the transistor sizes is large, a state in which a large stress acts on only one transistor while almost no stress acts on the other transistor occurs, that is, the transistors are likely to be subjected to the influence of stress that is generated during a packaging time of the semiconductor device20; however, according to the structure in the present embodiment, because the difference between the transistor sizes becomes small, an equal stress is likely to act on both elements, so that it becomes possible to reduce the influence of stress. Of course, in designing the transistor elements, it is desirable to suitably design W/L of each element to allow the transistors to operate in a saturation domain in an actual use range (5 [mA] or more is expected) of the output current ILED. Besides, in the semiconductor device20according to the present embodiment, as shown inFIG.32, pairs of resistors R1, R2(e.g., the resistors B3and B5or the resistors B5and B8) are arranged in a zigzag layout. According to the employment of such arrangement layout, in packaging the semiconductor device20, an equal stress is likely to act on the pair of resistors R1, R2, so that it becomes possible to reduce the influence of stress. As described above, in the semiconductor device20according to the present embodiment, the accuracy of the output current ILED is improved from many sides, that is, the increase in the resistor trimming chance, dispersion of the current amplification capability, improvement in the pairing easiness of resistors and the like. According to such structure, it is possible to achieve the relative uneven width of ±3% and the absolute uneven width of ±5% of the output current ILED after the packaging and it becomes possible to contribute to reduction in the brightness unevenness and longevity of the LED. Here, as for the resistance value of the resistor RSET, as described with reference to the aboveFIG.11, it is desirable to use a resistor that has a resistance of 300 [kΩ] or lower. Here, in a case where the variable control (light control of the LED) of the output current ILED is performed by using the above control voltage VDAC, it is sufficient to set the input range at a range of 0.1 to 2.0 [V]. By applying such control voltage VDAC, it becomes possible to decrease the output current ILED from the maximum value. On the other hand, in a case where 2.0 [V] or higher is input as the control voltage VDAC, as given by the above formula (10), the voltage value of the constant voltage VISET is selected; accordingly, a non-use state in which the light control function by the control voltage VDAC is not used occurs. In a case where the light control by the control voltage VDAC is not used, from the viewpoint of avoidance of malfunction, it is sufficient not to open the VDAC terminal but connect it with the application terminal of the reference voltage VREG (5 [V]). In addition, in the semiconductor device20according the present embodiment, besides the light control of the LED that uses the above control voltage VDAC, by using the PWM signal input into the PWM terminal (8th pin), the on/off (in the example inFIG.30, the on/off of the transistors B13, B14that constitute the last-stage current mirror circuit and the on/off of the current mirror circuit B9) of the constant-current driver221is controlled, so that it is possible to perform the light control of the LED as well. Specifically, in the semiconductor device20according to the present embodiment, the on/off of the transistors B13, B14that constitute the last-stage current mirror circuit and the on/off of the current mirror circuit B9are controlled based on the PWM signal, so that the duty ratio of the PWM signal becomes the duty ratio of the output current ILED; accordingly, it becomes possible to seemingly decrease the output current ILED from the maximum value (or a current value decided on by the control voltage VDAC). Here, in the first embodiment that converts the reference current ISET into a pulse current based on the PWM signal, ringing occurs in the current mirror circuits B4, B9, which leads to an overshoot of the output current ILED; however, in the structure according to the present embodiment, such problem does not occur. Besides, in the semiconductor20according to the present embodiment, as a further measure against an overshoot, an operational amplifier that has a slow slew rate (e.g., 0.5 [V/μs]) is used as the output-stage operational amplifier B15; and fluctuation in the gate-source voltage VGS of the transistor B17is limited, so that the rising of the output current ILED is slowed down to prevent an overshoot from occurring. Further, in the semiconductor device20according to the present embodiment, as described above, because the intermediate current Id that flows into the last-stage current mirror circuit is increased from 2ISET in the first embodiment to 30ISET, the response of the transistors B13, B14rises, which allows removal of the pull-down resistor B18; accordingly, it becomes possible to further curb occurrence of an overshoot. As described above, in the semiconductor device20according to the present embodiment, from many sides such as the PWM control of the last-stage current mirror circuit, use of the operational amplifier B15having a slow slew rate, removal of the pull-down resistor B18and the like, the improvement in response of the output current ILED and the reduction in overshoot are performed. According to such structure, without causing an overshoot, it becomes possible to achieve improvement (the minimum duty ratio: 0.38% (at 150 [Hz])) in the PWM light control capability and improve the light control accuracy at a low duty. On the other hand, in a case where the light control function by the PWM signal is not used, it is sufficient to fix the PWM terminal at the high level (e.g., the constant voltage VREG). Here, it is desirable to inset a low pass filter (a cut-off frequency of 30 [kHz]) into the PWM terminal. An example of the PWM light control is already described with reference to the aboveFIGS.14A to14C. Next, the voltage step-up and -down DC/DC controller block of the semiconductor device20(the OCP portion206, comparator207, control logic portion208, input buffer209, oscillator portion210, slope-voltage generation portion211, PWM comparator212, driver control portion213, driver214, transistor215, driver216, error amplifier217and soft start portion218) is described in detail with reference to the aboveFIG.26. First, detailed description of external connections of the semiconductor device20, especially circuit elements (N-channel type field effect transistors N1, N2, diodes D2, D3, coil L2, resistors RCS, RLPF, capacitors CBS, CLPF) related to the voltage step-up and -down DC/DC converter is performed. As shown inFIG.26, a gate of the transistor N1is connected with an output terminal of the driver214via the OUTH terminal (25th pin). A drain of the transistor N1is connected with the application terminal of the power-supply voltage VCC via the resistor RCS and also connected with the CS terminal (27th pin) via the resistor RLPF. A source of the transistor N1is connected with a second power-supply terminal (low-potential terminal) of the driver214and a drain of the transistor215via the SW terminal (24th pin). One terminal of the coil L2is connected with the SW terminal. The other terminal of the coil L2is connected with an anode of the diode D3. As a lead-out terminal of the output voltage VOUT, a cathode of the diode D3is connected with the anode of the LED train that is the load. A gate of the transistor N1is connected with an output terminal of the driver216via the OUTL terminal (22nd pin). A drain of the transistor N2is connected with a connection node of the other terminal of the coil L2and the anode of the diode D3. A source of the transistor N3is connected with the ground terminal. A cathode of the diode D2is connected with SW terminal. An anode of the diode D2is connected with the ground terminal. One terminal of the capacitor CBS for bootstrap is connected with a first power-supply terminal (high-potential terminal) of the driver214via the BOOT terminal (9th pin). The other terminal of the capacitor CBS is connected with the SW terminal. One terminal of the capacitor CLPF is connected with the application terminal of the power-supply voltage VCC. The other terminal of the capacitor CLPF is connected with the CS terminal. Here, in the semiconductor device20according to the present embodiment, because the transistors N1, N2are externally connected, it becomes possible to raise heat radiation. Next, detailed description of basic operation of the voltage step-up and -down DC/DC controller block is performed. If the transistors N1, N2are brought into an on state, a current flows into the coil L2via a route X and its electric energy is stored. Besides, in a case where electric charges are accumulated in the capacitor CVOUT in an on time of the transistors N1, N2, a current from the capacitor CVOUT flows into the lead-out terminal of the output voltage VOUT. Here, because the other-terminal potential of the coil L2drops to almost the ground potential via the transistor N2, the diode D3goes into a backward bias state and a current does not flow from the capacitor CVOUT to the transistor N2. Next, if the transistors N1, N2are brought into an off state, the electric energy accumulated there is discharged by a backward voltage generated in the coil L1via a route Y, flows from the lead-out terminal of the output voltage VOUT into the LED train that is the load, and also flows into the ground terminal via the capacitor CVOUT to charge the capacitor CVOUT. Accordingly, in the semiconductor device20according to the present embodiment, by suitably controlling the duty ratios of the transistors N1, N2by means of the driver control portion213, specifically, in a time of voltage step-down operation, the duty ratios of the transistors N1, N2are decreased to a value smaller than 50%, and to the contrary, in a time of voltage step-up operation, the duty ratios of the transistors N1, N2are increased to a value larger than 50%, so that it becomes possible to easily and suitably change the voltage step-up and -down operations even with a simple structure. In other words, in the semiconductor device20according to the present embodiment, regardless of whether the power-supply voltage VCC is is higher or lower than the desired output voltage VOUT, it becomes possible to always obtain the desired output voltage VOUT. Accordingly, for example, even in a case where the power-supply voltage VCC changes in a range of 6 to 18 [V] when the desired value of the output voltage is 16 [V], it becomes possible to obtain the desired output voltage VOUT. Such structure is suitable for an application, for example, (e.g., a backlight-control LED driver IC for a car navigation monitor) that needs to interact with the power-supply voltage VCC which is directly supplied from a battery. Note that even in a structure in which the gate of the transistor N2is connected with the SW terminal, the above voltage step-un and -down operations are possible; however, the semiconductor device20according to the present embodiment achieves the breakdown voltage of 36 V, and such a high voltage is likely to be applied to the SW terminal as well. On the other hand, the gate breakdown voltage of the external transistor N2is not invariably high. Accordingly, in a structure in which the gate of the transistor N2is connected with the SW terminal, the transistor N2can be broken. Because of this, in the semiconductor device20according to the present embodiment, as a means to perform the gate control of the transistor N2, the separate driver216(which operates on the reference voltage VREG) is prepared, and by using this, the on/off control of the transistor is performed. According to such structure, there is no worry over that the transistor N2is broken even if the power-supply voltage is a high voltage. Besides, in the semiconductor device20according to the present embodiment, as a ringing protection means in a time of a light load or no load, the N-channel type field effect transistor215is integrated. A drain of the transistor215is connected with the SW terminal. A source of the transistor215is connected with the ground terminal. A gate of the transistor215is connected with a control signal output terminal of the driver control portion213. Here, it is desirable that not to cause increase in an unnecessary chip area nor decrease in conversion efficiency, the electric-current capability of the transistor215is designed to be the smallest possible capability of removing a small current of ringing noise. The driver control portion213performs the on/off control of the transistors N1, N2, while it performs the on/off control of the transistor215in a complementary fashion to this. According to such structure, even in a case where the output current drops and the coil current is entirely lowered in a light-load or no-load time, and is trapped into a state (so-called discontinuous mode) in which ringing, that is, deformation in a waveform occurs, it is possible to enable the ringing noise to escape to the ground terminal via the transistor215, so that it becomes possible to raise the stability of the voltage step-up and -down operations. Here, the language “complementary” used in the above description covers not only a case where the on/off of the transistor N1(and the transistor N2) and the on/off of the transistor215are completely opposite to each other but also a case where from viewpoints of prevention of a through current and the like, a predetermined delay is given to the on/off transition timing of the transistor N1and the transistor215. Next, output feedback control of the voltage step-up DC/DC controller block is described in detail. The error amplifier217amplifies a difference between the smallest value of the LED terminal voltages V1to V4applied respectively to the first to fourth inverting input terminals (−) and the predetermined LED control voltage VLED input into the non-inverting input terminal (+), thereby generating an error voltage Verr. In other words, the voltage value of the error voltage Verr goes to a higher level as the output voltage VOUT becomes lower than its target set value. On the other hand, the PWM comparator212compares the error voltage Verr applied to the non-inverting input terminal (+) with a slope voltage Vslp (a sum voltage of the triangular-wave voltage Vosc of the oscillator OSC and the terminal voltage (an electric-current detection signal generated at the resistor RCS) of the CS terminal (27th pin)), thereby generating a comparison signal having a duty ratio depending on the comparison result. In other words, the logic of the comparison signal goes to a high level if the error voltage Verr is higher than the slope voltage Vslp and goes to a low level if the error voltage Verr is lower than the slope voltage Vslp. Here, the on duty (the ratio of on times of the transistors N1, N2per unit time) of the comparison signal in a steady operation time changes depending on a relative height difference between the error voltage Verr and the slope voltage Vslp. During a time the above comparison signal is maintained at the low level, the driver control portion213holds the OUTH terminal and the OUTL terminal (i.e., the gate voltages of the transistors N1, N2) at the high level via the driver214and the driver216. Accordingly, the transistors N1, N2are brought into the on state. On the other hand, during a time the comparison signal is maintained at the low level, the terminal voltages of the OUTH terminal and the OUTL terminal are held at the low level. Accordingly, the transistors N1, N2are brought into the off state. As described above, in the voltage step-up and -down DC/DC controller block, the drive control of the transistors N1, N2is performed based on not only monitoring results of the LED terminal voltages V1to V4(and the output voltage VOUT) but also monitoring results of the switch current that flows in the transistor N1. Accordingly, in the semiconductor device20according to the present embodiment, even if the error voltage Verr is not able to follow a sharp change in the load, it is possible to directly perform drive control of the transistors N1, N2based on the monitoring results of the switch current that flows in the transistors N1, N2; accordingly, it becomes possible to effectively curb a change in the output voltage VOUT. In other words, in the semiconductor device according to the present embodiment, because it is not necessary to enlarge the capacity of the output capacitor CVOUT and it is possible to use a low-ESR ceramic capacitor, it becomes possible to avoid an unnecessary cost increase and a size increase in the output capacitor CVOUT. Next, improved points of the protection circuit in the semiconductor device20are described. First, in the semiconductor device20according to the present embodiment, the type of detecting a short in the open/short detection portion222is changed to a delay counter type. Specifically, a structure is employed, in which the circuit operation is not immediately made off-latch at a time any of the LED1terminal to the LED4terminal reaches 4.5 [V] but made off-latch at a time it is confirmed that any of the LED1terminal to the LED4terminal continuously exceeds 4.5 [V] for a predetermined time. By employing such type, it becomes possible to effectively prevent erroneous detection. Second, in the semiconductor device20according to the present embodiment, the protection function portion that detects shorts (chiefly a ground short) in the anode and cathode of the LED and performs suitable protection operation is incorporated. Specifically, the SCP portion204has a structure in which when the SCP204portion confirms that the terminal voltage VP of the OVP terminal is equal to or lower than a predetermined voltage for a predetermined time, the SCP204recognizes that the anode terminal of the LED train ground-shorts (or shorts to a low potential comparable to this), and off-latches the circuit operation. Besides, the open/short detection portion222has a structure in which when the open/short detection portion222confirms by using an existing open detection function portion that any of the LED terminal voltages V1to V4is equal to or lower than a predetermined voltage for a predetermined time, the open/short detection portion222recognizes that the cathode terminal of the LED train ground-shorts, and off-latches the circuit operation. By incorporating such protection function portion, it becomes possible to further raise the safety of the semiconductor device20. Third, unlike the first embodiment that turns off the load switch Q1at a stop time of the circuit operation, the semiconductor device20according to the present embodiment has a structure in which the OCP signal and the OVP signal are input into the soft start portion218, and a soft start voltage (a charging voltage for the capacitor CSS) is pulled down at a time of occurrence of trouble. According to such structure, because the soft start is performed again at a time of resumption of the circuit operation, it becomes possible to prevent a rush current at the resumption time. Finally, an energy-saving function of the semiconductor device20is described. The semiconductor device20according to the present embodiment has a structure in which the control logic portion208is equipped with a timer latch function; when it is confirmed that the PWM signal is maintained at the low level for a predetermined time, the control logic portion208shifts to an energy-saving mode (sleep mode) for lowering the consumed energy of the semiconductor device20. According to such structure, it becomes possible to achieve the energy saving of the semiconductor device20. Besides, in the above energy-saving mode, it is desirable not to block the supply route of the power-supply voltage VCC but to turn off operation of a drive current generation portion (not shown) that generates the drive current ICC for each circuit portion. Here, in the above embodiments, as application targets of the present invention, semiconductor devices that perform the drive control of a backlight of a car navigation monitor, backlights of medium- and small-sized LCD panels are described as the examples; however, the application target of the present invention is not limited to these, and the present invention is widely applicable to other load drive devices. Besides, it is possible to make various modifications to the structure of the present invention without departing from the spirit of the present invention. INDUSTRIAL APPLICABILITY The present invention is a preferred technology for a drive device that performs drive control of a load (a LED backlight of medium- and small-sized LCD panels and the like). | 95,597 |
11863069 | DETAILED DESCRIPTION Some example embodiments include a switching converter controller configured to selectively adjust its switching frequency to avoid one or more predetermined stopbands. As used herein, a “stopband” is a band of frequencies to be avoided. The same reference numbers (or other reference designators) are used in the drawings to designate the same or similar (structurally and/or functionally) features.FIG.1is a block diagram of a system100in accordance with an example embodiment. The system100represents any electrical device with a load164, a power supply102(e.g., a battery or other direct-current (DC) power source), and power management circuitry including a power stage154and a switching converter controller104. As shown, the power stage154includes: a power stage input156; a first drive signal input158; a second drive signal input160; an inductor166; switch(es)168having respective control terminals coupled to the first drive signal input158or the second drive signal input160; and a power stage output162. In different example embodiments, the topology (e.g., the arrangement of the inductor166and switch(es)168) of the power stage154may vary. Example topologies for the power stage154include a boost converter topology, a buck converter topology, or a buck-boost converter topology. In a buck converter topology, VOUT at the power stage output162is less than the input voltage (VIN) provided to the power stage input156by the power supply102. In a boost converter topology, VOUT is greater than VIN. In a buck-boost converter topology, VOUT may be greater than or less than VIN. Relative to switchless power management options, the switching converter controller104and power stage154can more efficiently provide power from the power supply102to the load164. However, some undesirable switching noise may be introduced to the load164and/or other components of the system100. To reduce or avoid switching noise at one or more predetermined stopbands, the switching converter controller104includes a control loop106having a stopband controller126. More specifically, in the example ofFIG.1, the control loop106includes: a feedback error circuit108; a pulse-frequency modulation (PFM) controller116coupled to the feedback error circuit108; a peak current (Ipeak) controller134coupled to the PFM controller116; and a stopband controller126coupled to the PFM controller116. In the example ofFIG.1, the stopband controller126includes a first stopband controller input128configured to receive an enable signal. The enable signal is optional. In some example embodiments, the enable signal is de-asserted when the control loop106performs PWM operations and/or other control options instead of PFM operations. The stopband controller126is configured to determine when the switching frequency of the switching converter controller104is within a predetermined stopband. For example, the switching frequency of the switching converter controller104may be determined by analyzing the frequency of a clock signal (“CLK”) provided by the control loop106to a driver circuit142and a second stopband controller input130of the stopband controller126. Other techniques for determining the frequency of the switching converter controller104are possible (e.g., analysis of the frequency of a high-side control signal “HS_CS” at a first driver circuit output150, analysis of the frequency of a low-side control signal “LS_CS” at a second driver circuit output152, or resulting switch activity). Once the switching frequency is obtained, the stopband controller126is configured to compare the switching frequency with one or more predetermined stopbands stored by the stopband controller126. If the switching frequency of the switching converter controller104is within a predetermined stopband, the control loop106may adjust the frequency of CLK up or down to avoid the stopband based on stopband information (SB information) output from a stopband controller output132of the stopband controller126. In some example embodiments, stopband detection operations of the stopband controller126involve detecting that the switching frequency of the switching converter controller104is within a predetermined stopband for a number of stopband detection cycles. In different example embodiments, the stopband controller126may also: delay stopband detection operations for a time interval after a stopband is detected (to reduce the number of changes to the frequency of CLK due to stopband detection); adjust the direction of change to the frequency of CLK; and/or adjust the amount of change to the frequency of CLK. In some example embodiments, the control loop106may also account for a maximum peak current (related to the inductor166of the power stage154), where the maximum peak current is based on a target efficiency for the switching converter controller104and/or a target VOUT ripple. As desired, the stopband controller126may be selectively enabled or disabled. When enabled, the operations of the stopband controller126are used with the PFM controller116. In some example embodiments, the PFM controller116only adjusts the frequency of CLK when feedback error is equal to or less than a threshold. In such example embodiments, if the feedback error is greater than the threshold, the frequency of CLK is maintained by the PFM controller116while the peak current controller134performs peak current modulation and provides a related control signal (CS1) to the driver circuit142. In one example, CS1is asserted responsive to a peak current being reached. With CS1asserted, a high-side switch of the power stage154may be turned off (e.g., by de-asserting HS_CS). In different example embodiments, the switching converter controller104or control loop106may provide other control signals (e.g., PWM control signals, multi-phase control signals, zero crossing detection signals, and/or other control signals) to the driver circuit142via additional driver circuit inputs148. In operation, the driver circuit is configured to provide drive signals (e.g., for a high-side switch and/or a low-side switch corresponding to the switch(es)168) at the first driver circuit output150and the second driver circuit output152. In the example ofFIG.1, the operations of the switching converter controller104are based at least in part on VOUT and VIN. Accordingly, the switching converter controller104may include: a first input170configured to receive VOUT from the power stage output162; and a second input172configured to receive VIN from the power supply102or the power stage input156. The control loop106includes a feedback error circuit108having: a first feedback error circuit input110configured to receive VOUT (or a scaled version of VOUT); a second feedback error circuit input112configured to receive a reference voltage (VREF); and a feedback error circuit output114. In operation, the feedback error circuit108is configured to provide a feedback error at the feedback error circuit output114responsive to VREF and VOUT. In some example embodiments, the feedback error is a current (Ierror). As an option, Ierroris adjusted based on a feedforward signal, which may be a function of VOUT and/or VIN. In the example ofFIG.1, Ierroris one of the inputs to the PFM controller116. More specifically, the PFM controller116has a first PFM controller input118, a second PFM controller input120, a first PFM controller output122, and a second PFM controller output124. The first PFM controller input118is coupled to a stopband controller output132of the stopband controller126. The second PFM controller input120is coupled to the feedback error circuit output114and is configured to receive Ierror. The first PFM controller output122is coupled to a first driver circuit input144of the driver circuit142and is configured to provide CLK to the first driver circuit input144. The second PFM controller output124is coupled to a first peak current controller input136of the peak current controller134. In some example embodiments, the first peak current controller input136receives a minimum peak current (Ipeak_min) from the PFM controller116. A second peak current controller input138of the peak current controller134is configured to receive a slope compensation signal or ramp. The peak current controller output140of the peak current controller134is configured to provide CS1to a second driver circuit input146of the driver circuit142. In some example embodiments, CS1results from peak current modulation operations of the peak current controller. For example, peak current modulation may be performed by the peak current controller134based on Ipeak_minand the slope compensation signal when feedback error (e.g., Ierror) is greater than a threshold. When feedback error (e.g., Ierror) is equal to or less than a threshold, the PFM controller116is configured to adjust the frequency of CLK based on the stopband information provided by the stopband controller126. FIG.2is a diagram of a switching converter controller104A (an example of the switching converter controller104inFIG.1) in accordance with an example embodiment. InFIG.2, the switching converter controller104A includes analog circuitry201and a stopband controller126A (an example of the stopband controller126inFIG.1) with digital circuitry. As shown, the stopband controller126A includes an oscillator258, a frequency counter262, and stopband detection logic270. More specifically, the oscillator258has an oscillator output260, which outputs a reference clock signal (CLKREF). The stopband controller126A also includes a frequency counter262having: a first frequency counter input264, a second frequency counter input266, and a frequency counter output268. The first frequency counter input264is coupled to the oscillator output260and is configured to receive CLKREF. The second frequency counter input266is coupled to a second stopband controller input130and is configured to receive CLK from the first PFM controller output122. In some example embodiments, CLKREFhas a higher frequency than CLK. In operation, the frequency counter262is configured to count the number of periods of CLKREF(the period of CLKREFbeing known) that fit into one period of CLK to determine a frequency value for CLK. The frequency value is output from the frequency counter output268to a stopband detection logic input272of the stopband detection logic270. In some example embodiments, the stopband detection logic270includes stopband detection logic output274coupled to the stopband controller output132. In operation, the stopband detection logic270is configured to provide stopband information at the stopband detection logic output274responsive to a comparison of the frequency value (received from frequency counter262) with a predetermined stopband. The stopband information indicates when the frequency value is within the predetermined stopband. In some example embodiments, the stopband controller126A is configured to provide the stopband information to the stopband controller output132responsive to determining that a switching frequency of the switching converter controller is within a predetermined stopband for multiple stopband detection cycles. In some example embodiments, the stopband controller126A is configured to: determine whether a switching frequency of the switching converter controller is within a predetermined stopband; and responsive to determining that the switching frequency of the switching converter controller is within the predetermined stopband, provide a stopband detected signal (e.g., the stopband information includes the stopband detected signal) to the stopband controller output132and delay stopband detection operations for a time interval. In some example embodiments, the stopband controller126A is configured to: store a programmable number of predetermined stopbands; compare a frequency value determined from the reference signal to the programmable number of predetermined stopbands; and output a stopband detected signal to the stopband controller output132responsive to the frequency value being within one of the programmable number of predetermined stopbands. In some example embodiments, the stopband controller126A is configured to: store a programmable stopband size for each of the programmable number of predetermined stopbands; and output a stopband detected signal and a respective stopband size (e.g., the stopband information includes the stopband detected signal and the respective stopband size) to the stopband controller output132responsive to the frequency value being within one of the programmable number of predetermined stopbands. In the example ofFIG.2, the analog circuitry201includes: a feedback error circuit108A (an example of the feedback error circuit108inFIG.1); a PFM controller116A (an example of the PFM controller116inFIG.1); a peak current controller134A (an example of the peak current controller134inFIG.1); and a driver circuit142A (an example of the driver circuit142inFIG.1. The feedback error circuit108A includes the first feedback error circuit input110, the second feedback error circuit input112, and the feedback error circuit output114. The first feedback error circuit input110is configured to receive VOUT (e.g., from a power stage output such as the power stage output162). The second feedback error circuit input112is configured to receive VREF. In operation, the feedback error circuit108A is configured to output a feedback error signal (e.g., Ierroror part of Ierror) to the feedback error circuit output114responsive to VOUT and VREF. In some example embodiments, the feedback error circuit108A includes an error amplifier106having a non-inverting (+) input, an inverting (−) input, and an output. In the example ofFIG.2, the non-inverting input of the error amplifier206is coupled to the second feedback error circuit input112. The inverting input of the error amplifier106is coupled the first feedback error circuit input110via a voltage divider that includes resistors R1and R2. With R1and R2, the inverting input of the error amplifier208receives a scaled version of VOUT. In the example ofFIG.2, the output of the error amplifier206is coupled to loop compensation circuitry208and a voltage-to-current converter210. The loop compensation circuitry208is coupled between the output of the error amplifier206and ground. In the example ofFIG.2, the loop compensation circuitry208includes a third resistor (R3) and a first capacitor (C1) in series. In other example embodiments, the loop compensation circuitry208varies. As shown, the voltage-to-current converter210includes a transistor (M1) having a control terminal coupled to the output of the error amplifier206. A first current terminal of M1is coupled to the feedback error circuit output114via a current mirror. A second current terminal of M1is coupled to a first side of a current source212. The second side of the current source212is coupled to ground. In some example embodiments, a feedforward signal (not shown) may be applied to the feedback error at the feedback error circuit output114. When used, the feedforward signal helps account for fast changes to VIN and/or VOUT. In the example ofFIG.2, the PFM controller116A includes the first PFM controller input118, the second PFM controller input120, the first PFM controller output122, and the second PFM controller output124. The first PFM controller input118is coupled to the stopband controller output132and is configured to receive stopband information (e.g., a stopband detected signal, stopband size, stopband response instructions) as appropriate. The second PFM controller input120is coupled to the feedback error circuit output114. The first PFM controller output122is coupled to: the first driver input144of the driver circuit142; and a clamp controller222. The second PFM controller output124is coupled to the peak current controller input136. In some example embodiments, the PFM controller116A is configured to provide a minimum peak current (Ipeak_min) to the second PFM controller output124. In some example embodiments, the PFM controller116A includes a PFM modulator242. The PFM modulator242includes a first PFM modulator input244, a second PFM modulator input246, and a PFM modulator output248. In the example ofFIG.2, the first PFM modulator input244is coupled to a peak current clamp230. The second PFM modulator input246is coupled to the feedback error circuit output114via a current mirror240. In other words, the first PFM modulator input244receives a reference signal, while the second PFM modulator input246receives a feedback error signal (Ierroror a scaled version of Ierror). In some example embodiments, the reference signal is a maximum peak current (Ipeak_max) provided by the peak current clamp230. In some example embodiments, the PFM modulator242is configured to adjust a frequency of CLK at the PFM modulator output248responsive to the reference signal and feedback error, where the reference signal may vary to account for stopband detection. If the feedback error is greater than a threshold, the PFM modulator242may be configured to maintain the frequency of CLK regardless of stopband detection. In such case, the peak current controller134A is configured to perform peak current modulation while the frequency of CLK is clamped to a fixed value. In the example ofFIG.2, the peak current controller134A includes a comparator218having: an inverting (−) input coupled to the first peak current controller input136; and a non-inverting (+) input to the second peak current controller input138. The inverting input of the comparator218is also coupled to a transistor (M2). In some example embodiments, M2is a replica of a high-side power transistor of a power stage (i.e., M2is a replica of one of the switch(es)168the power stage154inFIG.1). M2may be scaled relative to the high-side power transistor based on a replica ratio. The non-inverting input of the comparator218is coupled to the switch node of the high-side power transistor. Thus, the voltage at the second peak current controller input138is the drop across the high-side power transistor, which is directly proportional to the current in the inductor. Current flowing through M2via the first peak current controller input136is defined by a control loop (e.g., the control loop106inFIG.1), which acts as a reference for the inductor current. When the inductor current is equal to the current through M2times a replica factor, the voltage at the inverting and non-inverting inputs of the comparator218are the same. During switching converter operations, the logic250of the driver circuit142A turns on the high-side power transistor at the rising edge of CLK. When the high-side power transistor turns on, the inductor current rises and as a result, the voltage at the second peak current controller input138rises. When voltage at the second peak current controller input138crosses the voltage at the first peak current controller input136, the comparator218asserts CS1at the peak current controller output140. Responsive to CS1being asserted, the logic250turns off the high-side power transistor and turns on the low-side power transistor. In operation, the peak current controller134A is configured to perform peak current modulation response to Ipeak_minreceived at the first peak current controller input136, a slope compensation signal received at the second peak current controller input138, and the operations of M2. The peak current modulation operations result in CS1being selectively asserted at the peak current controller output140. As shown, the peak current controller output140is coupled to the second driver circuit input146of the driver circuit142A. In the example ofFIG.2, Ipeak_minis controlled by the peak current clamp230, which includes a first peak current clamp input232, a second peak current clamp input234, a first peak current clamp output236, and a second peak current clamp output238. The first peak current clamp input232is coupled to a clamp controller output228of a clamp controller222. As shown, the clamp controller222also includes: a first clamp controller input224coupled to the PFM modulator output248; and a second clamp controller input226coupled to the first PFM controller input118(to receive stopband information). In operation, the clamp controller222is configured to provide a clamp control signal at the clamp control output228responsive to the stopband information and/or CLK. As shown, the second peak current clamp input234of the peak current clamp230is coupled to the feedback error circuit output114via the current mirror240. The first peak current clamp output is coupled to the first PFM modulator input244. The second peak current clamp output238is coupled to the second PFM controller output124. In some example embodiments, the peak current clamp230is configured to provide Ipeak_maxto the first peak current clamp input236and Ipeak_minto the second peak current clamp output238responsive to a clamp control signal received at the peak current clamp input232and a feedback error received at the second peak current clamp input234. In the example ofFIG.2, the driver circuit142A includes the first driver circuit input144, the second driver circuit input146, the first driver circuit output150, and the second driver circuit output152. In some example embodiments, the driver circuit142A includes logic250coupled to the first driver circuit input144, the second driver circuit input146, and possibly the additional driver circuit inputs148. The logic250is coupled to: a first drive circuit252having an output coupled to the first driver circuit output250; and a second drive circuit252having an output coupled to the second driver circuit output252. In operation, the driver circuit142A is configured to provide drive signals (e.g., HS_CS and/or LS_CS) at the first driver circuit output150and the second driver circuit output152responsive to CLK received by the first driver circuit input144, CS1received by the second driver circuit input146, and/or other control signals received by additional driver circuit inputs148. The other control signals are provided by other control options256, which may be included with the analog circuitry201. Examples of the other control options256include PWM control, multi-phase control, zero crossing detection, and/or other control options. In different example embodiments, the topology of a switching converter controller such as the switching converter controller104may vary. Regardless of the particular topology, a switching converter controller topology may include a stopband controller such as the stopband controller126A to determine a switching frequency of the switching converter controller. If the switching frequency of the switching converter controller is within a predetermined stopband, the frequency of CLK provided to the driver circuit142A may be selectively adjusted to avoid the stopband. In some example embodiments, adjusting the frequency of CLK involves adjusting a reference signal (e.g., a maximum peak current) used by a PFM modulator as described herein. In some example embodiments, a first switching frequency (Fsw1) of a switching converter controller (e.g., the switching converter controller104inFIG.1, or the switching converter controller104A inFIG.2) is given as: Fsw1=2×Iload×VOUTIpeak12·L,Equation(1) where Iloadis the output current to the load, VOUT is the output voltage to the load, Ipeak1is a first peak current for the power stage inductor (e.g., inductor166), and L is the value of the power stage inductor. Meanwhile, a second switching frequency (Fsw2) of a switching converter controller (e.g., the switching converter controller104inFIG.1, or the switching converter controller104A inFIG.2) is given as: Fsw2=2×Iload×VOUTIpeak22·L,Equation(2) where Ipeak2is a second peak current for the power stage inductor (e.g., inductor166). In some example embodiments, avoiding stopbands is based on changing Ipeakas indicated in Equations 1 and 2, it can be seen that changing Ipeak. As another option, the amount of change to Ipeakmay be strategically controlled depending on stopband size, a stopband detection pattern, and/or other criteria. FIGS.3-8are graphs showing switching converter parameters as a function of time in accordance with example embodiments. InFIG.3, graph300shows Ipeak, Iload, and switching frequency (Fsw) as a function of time. Fsw is the switching frequency of the switching converter controller104inFIG.1, the switching converter controller104A inFIG.2, or related power stage switches. Stopband detection cycles or samples are also represented in graph300. As shown in graph300, Fsw decreases as Iloaddecreases, resulting in Fsw eventually being detected within a predetermined stopband302at time t1. After a second detection of Fsw within the predetermined stopband302at time t2, IpeakS decreased (e.g., the value of the reference signal used by the PFM modulator242is decreased), which results in an increase in Fsw. By adjusting Ipeakresponsive to Fsw being detected within the predetermined stopband302, Fsw can be moved away from the predetermined stopband302during scenarios in which Iloaddecreases and settles. In some example embodiments, stopband detection operations are paused or delayed after Fsw is detected as being within the predetermined stopband302. Also, in different example embodiments, the number of stopband detection samples needed before a change in Ipeakis initiated may vary. With the decrease in Ipeak, Fsw settles outside of the predetermined stopband302. In graph300, an interval304that includes t1and t2is shown, where the duration of the interval304is less than 1 ms. InFIG.4, graph400shows Ipeak, Iload, and Fsw as a function of time. Stopband detection cycles or samples are also represented in graph400. As shown in graph400, Fsw decreases as Iloaddecreases, resulting in Fsw eventually being detected within a predetermined stopband402at time t1. After a second detection of Fsw within the predetermined stopband402at time t2, Ipeakis decreased (e.g., the value of the reference signal used by the PFM modulator242is decreased), which results in an increase in Fsw. In some example embodiments, stopband detection operations are paused or delayed after Fsw is detected as being within the predetermined stopband402. Also, in different example embodiments, the number of stopband detection samples needed before a change in Ipeakis initiated may vary. After the increase in Fsw due to the decrease in Ipeak, Iloadand Fsw continue to decrease. At times t3and t4, Fsw is again detected as being within the predetermined stopband402. In response, Ipeakis increased, which causes Fsw to decrease below the predetermined stopband402. By adjusting Ipeakup or down responsive to Fsw being detected within the predetermined stopband402, Fsw can be moved away from the predetermined stopband402during scenarios in which Iloadis decreasing. InFIG.5, graph500shows Ipeak, Iload, and Fsw as a function of time. As shown in graph500, Fsw decreases as Iloaddecreases, resulting in Fsw eventually being detected within a predetermined stopband504at time t1. After a second detection of Fsw within the predetermined stopband504at time t2, Ipeakis decreased (e.g., the value of the reference signal used by the PFM modulator242is decreased), which results in an increase in Fsw. In graph500, the adjustment to Ipeakstarting at time t2results in Fsw being increased and detected within a predetermined stopband502at times t3and t4. Responsive to the stopband detection at time t4, Ipeakis increased, which results in Fsw being decreased. At times t5and t6, Fsw is detected as being within the predetermined stopband504. Responsive to the stopband detection at time t6, Ipeakis decreased, which results in Fsw being increased. At times t7and t8, Fsw is detected as being within the predetermined stopband502. Responsive to the stopband detection at time t8, Ipeakis decreased again, which results in Fsw being increased and settling above the predetermined stopbands502and504. In some example embodiments, stopband detection operations are paused or delayed for a time after a change in Ipeakdue to stopband detection (e.g., at times t2, t4, t6, and t8inFIG.5). Also, in different example embodiments, the number of stopband detection samples needed before a change in Ipeakis initiated may vary. As another option, the number of predetermined stopbands may vary. With graph500, Ipeakis adjusted up or down as needed responsive to stopband detections to move Fsw out of the predetermined stopbands502and504during scenarios in which Iloaddecreases and settles. InFIG.6, graph600shows Ipeak, Iload, and Fsw as a function of time. As shown in graph600, Fsw decreases as Iloaddecreases, resulting in Fsw eventually being detected within a predetermined stopband604at times t1and t2. Responsive to the stopband detection at times t1and t2, Ipeakis decreased from a maximum peak current (Ipeak_max), which results in Fsw being increased. At times t3and t4, Fsw is detected as being within a predetermined stopband602. Responsive to the stopband detection at times t3and t4, Ipeakis increased to Ipeak_max, which results in Fsw being decreased. At times t5and t6, Fsw is detected as being within a predetermined stopband604. Responsive to the stopband detection at times t5and t6, Ipeakis decreased from Ipeak_maxagain, which results in Fsw being increased. At times t7and t8, Fsw is detected as being within the predetermined stopband602. Responsive to the stopband detection at times t7and t8, Ipeakis decreased again, which results in Fsw being increased above of the predetermined stopband602. However, due to Iloaddecreasing, Fsw decreases and is detected as being within the predetermined stopband602at times t9and t10. Responsive to the stopband detection at times t9and t10, Ipeakis increased, which results in Fsw being decreased. At times t11and t12, Fsw is detected as being within the predetermined stopband604. Responsive to the stopband detection at times t11and t12, Ipeakis increased to Ipeak_max, which results in Fsw being decreased out of the predetermined stopband604. In some example embodiments, Ipeak_maxis selected based on a target efficiency and target VOUT ripple for a switching converter controller. In the example ofFIG.6, Ipeakadjustments do not go above Ipeak_max. In some example embodiments, stopband detection operations are paused or delayed for a time after a change in Ipeakdue to stopband detection (e.g., at times t2, t4, t6, t8, t10, and t12inFIG.6). Also, in different example embodiments, the number of stopband detection samples needed before a change in Ipeakis initiated may vary. As another option, the number of predetermined stopbands may vary. With graph600, Ipeakis adjusted up or down as needed responsive to stopband detections to move Fsw out of the predetermined stopbands602and604during scenarios in which Iloadis decreasing at different rates. Also, Ipeak_maxis accounted for (i.e., Ipeakdoes not go above Ipeak_max). InFIG.7, graph700shows Ipeak, Iloadand Fsw as a function of time. As shown in graph700, Fsw increases as Iloadincreases, resulting in Fsw eventually being detected within a predetermined stopband702at times t1and t2. Responsive to the stopband detection at times t1and t2, Ipeakis decreased, which results in Fsw being increased and settling above the predetermined stopband702. With graph700, Ipeakis adjusted up or down as needed responsive to stopband detections to move Fsw out of the predetermined stopband702during scenarios in which Iloadis increasing. In some example embodiments, stopband detection operations are paused or delayed for a time after a change in Ipeakdue to stopband detection (e.g., at times t1and t2inFIG.7). Also, in different example embodiments, the number of stopband detection samples needed before a change in Ipeakis initiated may vary. With graph700, Ipeakis adjusted up or down as needed responsive to stopband detections to move Fsw out of the predetermined stopband702during scenarios in which Iloadis increasing. InFIG.8, graph800shows Ipeak, Iload, and Fsw as a function of time. As shown in graph800, Fsw increases as Iloadincreases, resulting in Fsw eventually being detected within a predetermined stopband804at times t1and t2. Responsive to the stopband detection at times t1and t2, Ipeakis decreased, which results in Fsw being increased. At times t3and t4, Fsw is detected as being within a predetermined stopband802. Responsive to the stopband detection at times t3and t4, Ipeakis increased, which results in Fsw being decreased. At times t5and t6, Fsw is detected as being within the predetermined stopband804. Responsive to the stopband detection at times t5and t6, Ipeakis decreased again, which results in Fsw being increased. At times t7and t8, Fsw is detected as being within the predetermined stopband802. Responsive to the stopband detection at times t7and t8, Ipeakis decreased again, which results in Fsw being increased and settling above the predetermined stopband802. In some example embodiments, stopband detection operations are paused or delayed for a time after a change in Ipeakdue to stopband detection (e.g., at times t2, t4, t6, and t8inFIG.8). Also, in different example embodiments, the number of stopband detection samples needed before a change in Ipeakis initiated may vary. As another option, the number of predetermined stopbands may vary. With graph800, Ipeakis adjusted up or down as needed responsive to stopband detections to move Fsw out of the predetermined stopbands802and804during scenarios in which Iloadincreases and settles. FIG.9is a flowchart showing a switching converter controller method900in accordance with an example embodiment. The method900is performed, for example, by the switching converter controller104inFIG.1, or the switching converter controller104A inFIG.2. As shown, the method900includes monitoring Fsw of a switching converter controller or power stage switch at block902. If Fsw is not detected to be within a predetermined stopband (determination block904), the method900returns to block902. If Fsw is detected to be within a predetermined stopband (determination block904), Ipeakin the control loop (e.g., the control loop106inFIG.1) is adjusted to change Fsw at block906. For example, an increase to Ipeakwill decrease Fsw while a decrease to Ipeakwill increase Fsw. In some example embodiments, Ipeak_maxis accounted for by the switching converter controller, where Ipeak_maxis based on a target efficiency and/or target VOUT ripple for the switching converter controller. Other options include enabling/disabling stopband detection, selectively combining stopband avoidance with other control loop options (e.g., PWM control, peak/valley current control, voltage mode control, hysteretic control, constant ON/OFF time control, multi-phase control, zero crossing detection, and/or other control options), use of multiple programmable stopbands, use of programmable stopband sizes, changing the amount of change to Ipeakbased on stopband size or stopband pattern detection, determining an appropriate frequency step size to avoid stopbands or move away from stopbands, and/or changing Fsw by the frequency step size as needed to avoid repeated stopband violations. In another example embodiments, the inductor peak current is adjusted up or down without clamping the maximum or minimum inductor peak currents and also no clamping for the maximum switching frequency. FIGS.10A and10Bare flowcharts showing switching converter controller methods1000A and1000B in accordance with example embodiments. The methods1000A and1000B are performed, for example, by a state machine or digital logic of a stopband controller (e.g., the stopband controller126inFIG.1, or the stopband controller126A inFIG.2). When enabled, the method1000A ofFIG.10Aincludes measuring the frequency of CLK at block1002. If a threshold number (fixed or programmable) of CLK frequency measurements is not reached (determination block1004), the method1000A returns to block1002. If the threshold number of CLK frequency measurements is reached (determination block1004), a determination is made regarding whether a stopband is violated in all of the CLK frequency samples (determination block1006). If not, the method1000A returns to block1002. If a stopband is violated in all of the CLK frequency samples (determination block1006), a determination is made regarding whether any stopband bit or flag was previously set (determination block1008). If not, a stopband bit or flag is set at block1010, and the method1000A waits at block1012for a time interval (e.g., fixed or programmable) before returning to block1002. If a stopband bit or flag was previously set (determination block1008), an active stopband bit or flag is reset at block1014. The method1000A then waits at block1012for a time interval before returning to block1002. When enabled, the method1000B ofFIG.10Bincludes the various blocks of the method1000A ofFIG.10Aas well as blocks1018and1020to avoid oscillating between adjacent stopbands. In method1000B, if any stopband bit or flag was previously set (determination block1008), a determination is made regarding whether a stopband violation is repeated (determination block1018). If not, the method1000B proceeds to blocks1014and1012. If a stopband violation is repeated (determination block1018), a next stopband bit or flag is set at block1020. The method1000B then waits at block1012for a time interval before returning to block1002. In some example embodiments, a switching converter controller is configured to selectively adjust its switching frequency to avoid one or more predetermined stopbands. In different example embodiments, the size and number of stopbands accounted for by the switching converter controller varies. In some example embodiments, the switching converter controller includes a control loop configured to selectively adjust a clock signal responsive to stopband information and feedback error, where the clock signal determines the switching frequency of the switching converter controller. For example, the clock signal may be provided to a driver circuit configured to generate drive signals for one or more power stage switches based on the clock signal. By controlling the switching frequency of the switching converter controller to avoid the one or more predetermined stopbands, switching noise at these predetermined stopbands is avoided or reduced. As desired, the described options for selectively adjusting the switching frequency of the switching converter controller to account for one or more predetermined stopbands can be combined with other switching converter controller options (e.g., pulse-width modulation (PWM) control, peak current control, multi-phase control, zero crossing detection, and/or other control options). In this description, the term “couple” may cover connections, communications, or signal paths that enable a functional relationship consistent with this description. For example, if device A generates a signal to control device B to perform an action: (a) in a first example, device A is coupled to device B by direct connection; or (b) in a second example, device A is coupled to device B through intervening component C if intervening component C does not alter the functional relationship between device A and device B, such that device B is controlled by device A via the control signal generated by device A. As used herein, the terms “electrode”, “node”, “interconnection”, “pin”, “contact”, and “connection” are used interchangeably. Unless specifically stated to the contrary, these terms are generally used to mean an interconnection between or a terminus of a device element, a circuit element, an integrated circuit, a device or other electronics or semiconductor component. The example embodiments above may utilize switches in the form of n-type metal-oxide semiconductor field-effect transistors (nMOSFET or just “nMOS”) or pMOS transistors. Other example embodiments may utilize NPN bipolar junction transistors (BJTs), PNP BJTs, or any other type of transistor. Hence, when referring to a current electrode, such electrode may be an emitter, collector, source or drain. Also, the control electrode may be a base or a gate. A device that is “configured to” perform a task or function may be configured (e.g., programmed and/or hardwired) at a time of manufacturing by a manufacturer to perform the function and/or may be configurable (or re-configurable) by a user after manufacturing to perform the function and/or other additional or alternative functions. The configuring may be through firmware and/or software programming of the device, through a construction and/or layout of hardware components and interconnections of the device, or a combination thereof. A circuit or device that is described herein as including certain components may instead be adapted to be coupled to those components to form the described circuitry or device. For example, a structure described as including one or more semiconductor elements (such as transistors), one or more passive elements (such as resistors, capacitors, and/or inductors), and/or one or more sources (such as voltage and/or current sources) may instead include only the semiconductor elements within a single physical device (e.g., a semiconductor die and/or integrated circuit (IC) package) and may be adapted to be coupled to at least some of the passive elements and/or the sources to form the described structure either at a time of manufacture or after a time of manufacture, for example, by an end-user and/or a third-party. Circuits described herein are reconfigurable to include the replaced components to provide functionality at least partially similar to functionality available prior to the component replacement. Components shown as resistors, unless otherwise stated, are generally representative of any one or more elements coupled in series and/or parallel to provide an amount of impedance represented by the shown resistor. For example, a resistor or capacitor shown and described herein as a single component may instead be multiple resistors or capacitors, respectively, coupled in parallel between the same nodes. For example, a resistor or capacitor shown and described herein as a single component may instead be multiple resistors or capacitors, respectively, coupled in series between the same two nodes as the single resistor or capacitor. Uses of the phrase “ground” in this description include a chassis ground, an Earth ground, a floating ground, a virtual ground, a digital ground, a common ground, and/or any other form of ground connection applicable to, or suitable for, the teachings of this description. Unless otherwise stated, “about,” “approximately,” or “substantially” preceding a value means+/−10 percent of the stated value. Modifications are possible in the described embodiments, and other embodiments are possible, within the scope of the claims. | 43,180 |
11863070 | DETAILED DESCRIPTION OF THE DRAWINGS FIG.1shows a circuit diagram of a DC/DC converter1. The DC/DC converter1has a first connection pole U1+ and a second connection pole U1−, wherein a first DC voltage U1is present between the first connection pole U1+ and the second connection pole U1−. The DC/DC converter1furthermore has a third connection pole U2+ and a fourth connection pole U2−, wherein a second DC voltage U2is present between the third connection pole U2+ and the fourth connection pole U2−. The DC/DC converter1furthermore has a first switching cell2that has a capacitor3, a diode4, a semiconductor switching device5and a coil16. An optional diode18is connected in parallel with the semiconductor switching device5and wired in the blocking direction between the first connection pole U1+ and the third connection pole U2+. The diode18may thus be what is known as a body diode if the semiconductor switching device5is embodied as a field-effect transistor. The semiconductor switching device5and the coil16are wired in series between the first connection pole U1+ and the third connection pole U2+. The diode4is electrically connected, by way of its cathode, to the semiconductor switching device5, the anode of the diode18and the coil16and is electrically connected, by way of its anode, to the capacitor3and a precharging circuit11. The DC/DC converter1furthermore has a second switching cell6that has a capacitor7, a diode8, a semiconductor switching device9and a coil17. An optional diode19is connected in parallel with the semiconductor switching device9and wired in the conducting direction between the second connection pole U1− and the fourth connection pole U2−. The diode19may thus be what is known as a body diode if the semiconductor switching device9is embodied as a field-effect transistor. The semiconductor switching device9and the coil17are wired in series between the second connection pole U1− and the fourth connection pole U2−. The diode8is electrically connected, by way of its anode, to the semiconductor switching device9, the cathode of the diode19and the coil17and is electrically connected, by way of its cathode, to the capacitor7and the precharging circuit11. The DC/DC converter1has the precharging circuit11for precharging the capacitors3and7, wherein the precharging circuit11has a precharging resistor12, a driveable switching device in the form of a relay13and an optional capacitor15, wherein the precharging resistor12, the driveable switching device13and the capacitor15are connected in parallel. The capacitor3of the first switching cell2, the precharging circuit11, that is to say the parallel circuit consisting of precharging resistor12, driveable switching device13and capacitor15, and the capacitor7of the second switching cell6are wired in series between the first connection pole U1+ and the second connection pole U1−. The DC/DC converter1has a control unit14, for example in the form of a microprocessor, which is designed to drive the driveable switching device13of the precharging circuit11such that the driveable switching device13is opened during a precharging phase of the capacitors3and7in order to charge the capacitors3and7up to a desired precharging voltage. The DC/DC converter1may be designed as a step-down converter, that is to say U1>U2holds true. The capacitance of the capacitor15is typically selected to be smaller than the capacitance of the capacitors3and7. FIG.2shows a circuit diagram of a grid feedback unit100having two parallel-connected DC/DC converters1, wherein a respective DC/DC converter1corresponds to the DC/DC converter1shown inFIG.1. The grid feedback unit100is designed to feed electrical energy from a voltage intermediate circuit101into a three-phase grid102. The grid feedback unit100is based on the grid feedback unit disclosed in WO 2017/072297 A1, and expands on this, in particular with regard to the precharging circuit11. With regard to the basic functions of the grid feedback unit100, reference is therefore also made to the disclosure in WO 2017/072297 A1. The grid feedback unit100has two DC/DC converters1as shown inFIG.1and an inverter10that is electrically coupled at input to the respective third connection poles U2+ and the respective fourth connection poles U2− of the DC/DC converters1and that is electrically coupled at output to the three-phase grid102. It goes without saying that, rather than the two DC/DC converters1, just a single DC/DC converter may also be provided, or more than two DC/DC converters may be connected in parallel. Between the respective third connection poles U2+ and the respective fourth connection poles U2− of the DC/DC converters1, a filter capacitor24is wired in series together with a precharging circuit having a resistor25and a driveable switching device26. In order to precharge the filter capacitor24, the switching device26is opened. The switching device26may be designed as a relay. The inverter10has conventional semiconductor switching device22in the form of a bridge circuit. Line chokes23are provided on the grid side. The inverter10may be operated for example at a switching frequency of >60 kHz or clocked at grid frequency. The precharging circuit11is arranged in a switching region that is used exclusively for low-frequency compensation currents. These compensation currents or ripple currents that occur during operation are considerably smaller than the currents to be transferred from the DC/DC converter, meaning that only a small portion of the switching current flows through the respective switching device13. As a result, no additional impedance is introduced into the switching path and oscillations or overvoltages at the semiconductor switching device5and9are able to be avoided. A single switching device or precharging relay13is sufficient for each DC/DC converter1. This makes it possible for the precharging circuit11not to have to be designed for the entire power output of the unit. Due to the resulting low power losses, the precharging circuit11and its peripherals are able to be produced on a circuit board without additional cooling. Diodes20and21prevent reverse charging of the intermediate circuit101from the three-phase grid102. The following advantages are able to be achieved by way of the invention: saving on space; lower costs; low power losses, and thus production on an uncooled circuit board; and layout in the region of the intermediate circuit capacitors and thus at a location where the required structural height of the relay is present anyway. | 6,560 |
11863071 | DETAILED DESCRIPTION OF THE EXEMPLARY EMBODIMENTS The present disclosure is more particularly described in the following examples that are intended as illustrative only since numerous modifications and variations therein will be apparent to those skilled in the art. Like numbers in the drawings indicate like components throughout the views. As used in the description herein and throughout the claims that follow, unless the context clearly dictates otherwise, the meaning of “a”, “an”, and “the” includes plural reference, and the meaning of “in” includes “in” and “on”. Titles or subtitles can be used herein for the convenience of a reader, which shall have no influence on the scope of the present disclosure. The terms used herein generally have their ordinary meanings in the art. In the case of conflict, the present document, including any definitions given herein, will prevail. The same thing can be expressed in more than one way. Alternative language and synonyms can be used for any term(s) discussed herein, and no special significance is to be placed upon whether a term is elaborated or discussed herein. A recital of one or more synonyms does not exclude the use of other synonyms. The use of examples anywhere in this specification including examples of any terms is illustrative only, and in no way limits the scope and meaning of the present disclosure or of any exemplified term. Likewise, the present disclosure is not limited to various embodiments given herein. Numbering terms such as “first”, “second” or “third” can be used to describe various components, signals or the like, which are for distinguishing one component/signal from another one only, and are not intended to, nor should be construed to impose any substantive limitations on the components, signals or the like. Reference is made toFIGS.1and3, in whichFIG.1is a block diagram of a power converter having a smooth transition control mechanism according to an embodiment of the present disclosure, andFIG.3is a flowchart diagram of steps of the power converter having the smooth transition control mechanism according to the embodiment of the present disclosure. The power convertor of the embodiment of the present disclosure may include an oscillator circuit100, a control circuit200, a driver circuit300, a high-side switch UG and a low-side switch LG as shown inFIG.1. The control circuit200may be connected to the oscillator circuit100and the driver circuit300. The driver circuit300may be connected to a control terminal of the high-side switch UG, a control terminal of the low-side switch LG and the oscillator circuit100. A first terminal of the high-side switch UG is coupled to a common voltage VCC. A first terminal of the low-side switch LG may be connected to a second terminal of the high-side switch UG. A second terminal of the low-side switch LG may be grounded. A node LX between the first terminal of the low-side switch LG and the second terminal of the high-side switch UG may be connected to a first terminal of an inductor L. A second terminal of the inductor L may be connected to a first terminal of an output capacitor Cout. A second terminal of the output capacitor Cout may be grounded. A node between the second terminal of the inductor L and the first terminal of the output capacitor Cout may be an output terminal of the power convertor. The output terminal of the power convertor supplies an output voltage VOUT. Steps101to S111are shown inFIG.3and applicable to performed on the power converter of the embodiment of the present disclosure as shown inFIG.1, which is specifically described in the following. In step S101, the power convertor is powered on. In step S103, the oscillator circuit100outputs a clock signal CLK to the control circuit200. In step S105, the control circuit200outputs a control signal to the driver circuit300based on the clock signal CLK. The driver circuit300, according to the control signal, outputs a high-side conduction signal UGS at a first level such as a high level to the high-side switch UG to turn on the high-side switch UG. The driver circuit300, according to the control signal, outputs a low-side conduction signal LGS at a second level such as a low level to the low-side switch LG to turn off the low-side switch LG. In step S107, the control circuit200of the power convertor determines whether or not the output voltage VOUT supplied from the power convertor to an electronic device connected to the power convertor is enough for operation of the electronic device. If the power convertor does not supply enough energy to the electronic device, step S105is continually performed. In step S105, the high-side switch UG is turned on and the low-side switch LG is turned off. Conversely, if the power convertor supplies enough energy to the electronic device, step S109is performed. In step S109, the driver circuit300, according to the control signal, outputs the high-side conduction signal UGS at the second level such as the low level to the high-side switch UG to turn off the high-side switch UG. The driver circuit300, according to the control signal, outputs the low-side conduction signal LGS at the first level such as the high level to the low-side switch LG to turn on the low-side switch LG. It is worth noting that, the oscillator circuit100may receive the high-side conduction signal UGS from the driver circuit300. Then, the oscillator circuit100may, according to the high-side conduction signal UGS, determine whether or not the clock signal CLK outputted to the control circuit200needs to be adjusted to change an on-time of the high-side switch UG and an on-time of the low-side switch LG. For example, the oscillator circuit100may adjust a frequency of the clock signal CLK or time points at which one or more of pulse waves of the clock signal CLK appear. For example, in step S111, the oscillator circuit100may determine whether or not a time point Tosc at which the high-side conduction signal UGS transits from the high level to the low level is later than a transition time point Tclk of one of the pulse waves of the clock signal CLK. In the embodiment, the transition time point Tclk of the one of the pulse waves of the clock signal CLK may be a time point of a falling edge of the one of the pulse waves of the clock signal CLK, but the present disclosure is not limited thereto. In practice, the transition time point Tclk of the one of the pulse waves of the clock signal CLK may be a time point of a rising edge of the one of the pulse waves of the clock signal CLK. If the oscillator circuit100determines that the time point Tosc at which the high-side conduction signal UGS transits from the high level to the low level is not later than the transition time point Tclk of the one of the pulse waves of the clock signal CLK, the oscillator circuit100does not adjust the frequency of the clock signal CLK. Conversely, when the oscillator circuit100determines that the time point Tosc at which the high-side conduction signal UGS transits from the high level to the low level is later than the transition time point Tclk of the one of the pulse waves of the clock signal CLK, the oscillator circuit100may adjust the frequency of the clock signal CLK. For example, the oscillator circuit100may reduce the frequency of the clock signal CLK. When the oscillator circuit100determines that the time point Tosc at which the high-side conduction signal UGS transits from the high level to the low level is later than the transition time point Tclk of the one of the pulse waves of the clock signal CLK, the oscillator circuit100may delay the one of the pulse waves of the clock signal CLK such that a time point at which the one of the pulse waves of the clock signal CLK appears is later than the time point Tosc at which the high-side conduction signal UGS transits from the high level to the low level. Reference is made toFIG.2, which is a block diagram of the power converter having the smooth transition control mechanism according to the embodiment of the present disclosure. In the embodiment, the oscillator circuit100of the power converter may include a comparator CMP, a current source SC and an input capacitor Cin. A first input terminal such as a non-inverting input terminal of the comparator CMP may be connected to a first terminal of the input capacitor Cin. The first terminal of the input capacitor Cin may be connected to the current source SC. A second terminal of the input capacitor Cin may be grounded. A second input terminal such as an inverting input terminal of the comparator CMP may be coupled to a reference voltage. If necessary, the oscillator circuit100may further include a voltage divider circuit DVR. An input terminal of the voltage divider circuit DVR is coupled to an input voltage VIN. An output terminal of the voltage divider circuit DVR may be connected to the second input terminal of the comparator CMP. The voltage divider circuit DVR may output the reference voltage as described above to the second input terminal such as the inverting input terminal of the comparator CMP. For example, the voltage divider circuit DVR may include a first resistor R1and a second resistor R2. A first terminal of the first resistor R1is coupled to the input voltage VIN. A second terminal of the first resistor R1may be connected to a first terminal of the second resistor R2. A second terminal of the second resistor R2may be grounded. A node between the second terminal of the first resistor R1and the first terminal of the second resistor R2may be connected to the second input terminal of the comparator CMP. A voltage of the node between the second terminal of the first resistor R1and the first terminal of the second resistor R2may be the reference voltage coupled to the second input terminal such as the inverting input terminal of the comparator CMP as described above. It is worth noting that, a third input terminal of the comparator CMP may be connected to an output terminal of the driver circuit300and receive the high-side conduction signal UGS. An output terminal of the comparator CMP may be connected to an input terminal of the control circuit200. The comparator CMP may output the clock signal CLK signal to the control circuit200, according to a voltage of the first input terminal of the comparator CMP, a voltage of the second input terminal of the comparator CMP and the high-side conduction signal UGS. The power convertor of the embodiment of the present disclosure may include a feedback circuit. The feedback circuit is configured to feedback the output voltage Vout of the power convertor (or other related data such as a voltage divided from the output voltage Vout of the power convertor) to the control circuit200. For example, the feedback circuit may include an error amplifier ER. A first input terminal (such as a non-inverting input terminal) of the error amplifier ER may be connected to the node between the second terminal of the inductor L and the first terminal of the output capacitor Cout. A second input terminal (such as an inverting input terminal) of the error amplifier ER may be coupled to a reference voltage Vref. An output terminal of the error amplifier ER may be connected to an input terminal of the control circuit200. The error amplifier ER may amplify a difference between a voltage of the first input terminal of the error amplifier ER and a voltage of the second input terminal of the error amplifier ER to output an error amplified signal to the control circuit200. The control circuit200may output the control signal to the driver circuit300according to the error amplified signal from the error amplifier ER and the clock signal CLK from the comparator CMP of the oscillator circuit100. Reference is made toFIGS.1,4and5, in whichFIG.4is a waveform diagram of signals of the power converter having the smooth transition control mechanism according to the embodiment of the present disclosure, andFIG.5is a waveform diagram of signals of a conventional power converter. As shown inFIG.5, VIN represents the input voltage of the conventional power converter, VOUT0represents an output voltage of the conventional power converter, and IL0represents a current flowing through an inductor of the conventional power converter. In addition, LXS0represents a node signal at a node between of a second terminal of a high-side switch of the conventional power converter and a first terminal of a low-side switch of the conventional power converter, and CLK0represents a clock signal of the conventional power converter. In addition, ES0represents an error amplified signal of an error amplifier of the conventional power converter, and LFS0represents an energy signal at the node between of the second terminal of the high-side switch of the conventional power converter and the first terminal of the low-side switch of the conventional power converter. As shown inFIG.5, a frequency of the clock signal CLK0of the conventional power converter is a constant value such that an off time of the energy signal LFS0of the conventional power converter changes randomly with a change in energy. As a result, a very large ripple wave appears in the output voltage VOUT0of the conventional power converter. As shown inFIG.4, VIN represents the input voltage of the power converter of the present disclosure, VOUT represents the output voltage of the power converter of the present disclosure, and IL represents a current flowing through the inductor L of the power converter of the present disclosure. In addition, LXS represents a node signal at the node LX between of the second terminal of the high-side switch UG of the power converter of the present disclosure and the first terminal of the low-side switch LG of the power converter of the present disclosure, and CLK represents the clock signal that is received by the control circuit200from the oscillator circuit100in the power converter of the present disclosure. In addition, ES represents the error amplified signal of the error amplifier of the power converter ER of the present disclosure, and LFS represents an energy signal at the node LX between of the second terminal of the high-side switch UG of the power converter of the present disclosure and the first terminal of the low-side switch LG of the power converter of the present disclosure. As shown inFIG.4, a time required for the power converter of the present disclosure to supply the node signal LXS having enough energy (that is, a time during which the high-side switch UG needs to be turned on) may be larger than a time between two ones of the pulse waves of the clock signal CLK. Under this condition, the oscillator circuit100reduces the frequency of the clock signal CLK supplied to the control circuit200such that the frequency of the clock signal CLK is not a constant value while an off time of the energy signal LFS is a constant value. As a result, the output voltage VOUT supplied by the power converter of the present disclosure is a stable value, and only small a ripple wave appears in the output voltage VOUT of the power converter of the present disclosure. In summary, the present disclosure provides the power converter having the smooth transition control mechanism. The frequency of the clock signal can be adjusted appropriately according to the energy that the power convertor needs to supply. The control circuit controls the driver circuit to turn on or off the high-side switch and the low-side switch based on the adjusted frequency of the clock signal in real time, thereby efficiently preventing a large ripple wave from appearing in the output voltage of the power converter. Therefore, the output voltage of the power converter can be maintained at a constant value and cannot drop to a very low value. The foregoing description of the exemplary embodiments of the disclosure has been presented only for the purposes of illustration and description and is not intended to be exhaustive or to limit the disclosure to the precise forms disclosed. Many modifications and variations are possible in light of the above teaching. The embodiments were chosen and described in order to explain the principles of the disclosure and their practical application so as to enable others skilled in the art to utilize the disclosure and various embodiments and with various modifications as are suited to the particular use contemplated. Alternative embodiments will become apparent to those skilled in the art to which the present disclosure pertains without departing from its spirit and scope. | 16,642 |
11863072 | In the drawings, reference numbers may be reused to identify similar and/or identical elements. DETAILED DESCRIPTION The present disclosure relates to DC-DC converters including bypass switches to bypass the power switches in some of the operating modes of the power control system. While the foregoing description relates to a DC-DC converter in a vehicle, the present disclosure relates to DC-DC converters used in other applications. Referring now toFIG.1, a vehicle10includes a battery system12with a battery management system14. In some examples, the battery system12is a high voltage battery system operating at voltage levels above 100V, such as 300V, 400V, 800V, etc., although lower or higher voltage levels can be used. The battery system12is connected to a DC-DC converter18that changes a ratio between an input voltage Vinand an output voltage Vout. For example when the DC-DC converter operates in a buck mode, Vin>Vout. When the DC-DC converter operates in a boost mode, Vin<Vout. The DC-DC converter18is connected to a DC-AC inverter22that converts single phase DC power to three-phase AC power. One or more electric motors26are connected to the DC-AC inverter22. The one or more motors26drive one or more wheels32directly (e.g. wheel hub motors or direct drive) or indirectly via a gearbox30. While a single motor is shown driving two front or rear wheels, additional motors can be provided to drive rear or front wheels, respectively, and/or to drive each wheel independently. The battery system12is further connected to one or more DC-DC converters34. In some examples, the DC-DC converter34has an output voltage that is lower than an output voltage of the DC-DC converter18. The DC-DC converter34converts battery voltage to a level (e.g. 12V, 24V, 36V, 48V, . . . ) used by auxiliary loads38. Referring now toFIG.2, an example of the DC-DC converter18includes a single bypass switch SW0, power switches T1, T2, T3, and T4, and an inductor L1. The power switches T1, T2, T3, and T4include a control terminal and first and second terminals. The control terminal is used to turn the power switches T1, T2, T3, and T4on and off. In some examples, each of the power switches receives a pulse width modulated signal having a duty cycle between 0 and 1 inclusive. Switched On corresponds to a duty cycle 1, switched Off corresponds to a duty cycle of 0, and switching states correspond to duty cycles between 0 and 1. Each of the power switches T1to T4can include one or more power switch that are connected in parallel, series, or a combination thereof depending upon the application. A first terminal of a power switch T1is connected to a first terminal of a first side of the DC-DC converter18and to a first terminal of the bypass switch SW0. A second terminal of the power switch T1is connected to a first terminal of an inductor L1and a first terminal of the power switch T2. A first terminal of a power switch T3is connected to a first terminal of a second side of the DC-DC converter18and to a second terminal of the bypass switch SW0. A second terminal of the power switch T3is connected to a second terminal of an inductor L1and a first terminal of a power switch T4. Second terminals of the power switches T2and T4are connected together and to second terminals of the first side and second side of the DC-DC converter18, respectively. In use, the bypass switch SW0may be closed during certain operating modes when the input and output voltages of the DC-DC converter18are about the same. Referring now toFIG.3, an example of the DC-DC converter118according to the present disclosure is shown. The DC-DC converter118includes power switches T1, T2, T3, and T4, an inductor L1, and bypass switches SW1and SW3. The power switches T1, T2, T3, and T4may include voltage-controlled bipolar switching devices in the form of insulated gate bipolar transistors (IGBTs), metal-oxide semiconductor field effect transistors (MOSFETs), wide bandgap GaN or SiC devices (WBG), or other suitable power switches. The power switches T1, T2, T3, and T4include a control terminal and first and second terminals. The control terminal is used to turn the power switch on and off. Each of the power switches T1to T4can include one or more power switch that are connected in parallel, series, or a combination thereof depending upon the application. A first terminal of a power switch T1is connected to a first terminal of a first side of the DC-DC converter118and to a first terminal of the bypass switch SW1. A second terminal of the power switch T1is connected to a first terminal of an inductor L1, a first terminal of a power switch T2and a second terminal of the bypass switch SW1. A first terminal of a power switch T3is connected to a first terminal of a second side of the DC-DC converter118and to a first terminal of a second bypass switch SW3. A second terminal of the power switch T3is connected to a second terminal of an inductor L1, a first terminal of a power switch T4and a second terminal of the second bypass switch SW3. Second terminals of the power switches T2and T4are connected together and to second terminals of the first side and the second side of the DC-DC converter118, respectively. The power control system according to the present disclosure can operate in different modes. The DC-DC converter operates in a buck mode when an input side of the DC-DC converter is higher than an output side. The DC-DC converter operates in a boost mode when an input side of the DC-DC converter is lower than an output side. During a charge mode of the battery packs, power is output by the DC-DC converter to the battery system to charge the battery system. During a discharge mode of the battery packs, power is output by the DC-DC converter to the vehicle loads to discharge the battery system. During the bypass mode, the bypass switches are closed to reduce conduction and/or switching losses. The power control system according to the present disclosure transitions from the various boost/buck and charge/discharge modes to the bypass mode and from the bypass mode to the various boost/buck and charge/discharge modes (as shown inFIGS.7-10). Referring now toFIGS.4A-4B, operation of the DC-DC converters are shown in their respective bypass modes. InFIG.4A, the DC-DC converter18inFIG.2is shown with the single bypass switch SW0closed. InFIG.4B, the DC-DC converter118inFIG.3is shown with the bypass switches SW1and SW3closed. The bypass switches SW1and SW3eliminate high conduction losses from the power switches T1and T3while ensuring smoother current transition during mode changes to prevent components from breakdown. InFIG.4C, current is shown as a function of time during a mode transition for the DC-DC converter118ofFIG.3(shown at190) with the bypass switches SW1and SW3and a DC-DC converter18with a single bypass switch SW0and without the bypass switches SW1and SW3(shown at190). When the bypass SW0is closed or opened, differences in voltages and currents between the input and output may cause current to spike, which may cause damage to the components of the DC/DC converter. For example, it may cause a power switch to break down. Solid line192shows the current spike in the single bypass switch SW0of the DC-DC converter18inFIG.2when the single bypass switch SW0is closed. With the inductor L1in the bypass branch of the DC-DC converter118inFIG.3, the current cannot instantly change, which provides smoother transitions during mode changes. When the input and output levels are nearly the same, the input side of the DC/DC converter can directly supply the desired voltage and current to the output. But when doing so, power switches T1and T3will be always, or nearly always, conducting. This causes high and unnecessary conduction loss, which adversely affects the efficiency of the DC-DC converter. The bypass switches SW1and SW3can be realized using mechanical relays, contactors, and/or solid state relays. The bypass switches SW1and SW3are connected across the power switches T1and T3, respectively, to eliminate the high conduction losses produced by the power switches T1and T3when closed for longer periods with high duty cycles. Some modes may also eliminate switching losses of the power switches as will be described further below. When an output power level is higher than the voltage or current levels that the power switches can safely handle, the two bypass switches SW1and SW3can also be applied to directly link the input to the output. Both bypass switches SW1and SW3are closed in several situations. The bypass switches SW1and SW3are closed when the controller42determines that input and output voltage levels are similar and within rated voltage, current, and power of DC/DC converter (e.g., <+/−5%). The bypass switches SW1and SW3are also closed when the controller42determines that the required output level to drive the motor (or reverse in regenerative braking) exceeds the rated (or safe) power, voltage, current of the DC-DC converter. Examples when this condition may occur include driving up an incline, climbing a step (high curb, etc.), full accelerator, and/or braking hard. In some examples, only one bypass switch SW1is closed to bypass the power switch T1when the controller42stays in discharge-boost mode or charge-buck mode for a period longer than a first predetermined period. For example, the first predetermined period can be set to greater than or equal to 15 s, 30 s, 60 s, 90 s or 120 s, although shorter or longer periods can be used. In some examples, the bypass switch SW3is engaged to bypass the power switch T3when the controller42stays in discharge-buck mode or charge-boost mode for a period longer than a second predetermined period. For example, the second predetermined period can be set to greater than or equal to 15 s 30 s, 60 s, 90 s or 120 s, although shorter or longer periods can be used. When in bypass mode, the bypass switches SW1and SW3are closed and all of the power switches T1, T2, T3, and T4can be turned off. Referring now toFIG.5, states of the power switches T1, T2, T3, and T4and the bypass switches SW1and SW3are shown for various operating modes. InFIG.5, Option 1 (Opt 1) refers to operation without closing the bypass switches SW1and SW3and Option 2 (Opt 2) refers to operation with one of the bypass switches SW1and SW3closed. “SW” refers to switching on and off the power switches T1, T2, T3, and/or T4during operation (e.g. based on a pulse width modulated (PWM) signal). “ON” refers to conducting and “OFF” refers to not conducting. The power switches T1, T2, T3, and T4are controlled ON and OFF based on the voltage applied to the control terminal or gate of the corresponding power switch. In some examples, a switching signal such as a PWM signal is output to the control terminals of one or more of the power switches to allow operation in various modes such as buck or boost modes while other ones of the power switches are either continuously ON or OFF. In some examples, the bypass switches SW1and SW3are operated at relatively low frequency. In some examples, ON and OFF states last on the order of seconds or minutes to realize the desired circuit for each mode. The bypass switches SW1and SW3bypass the power switches T1and T3, respectively, while connecting the inductor L1that is already in the main circuit of DC-DC converter118into the bypass branch of the circuit. This operation ensures smoother current and voltage transitions during mode changes to prevent components from breakdown. Examples of mode changes include buck mode or boost mode to bypass mode or bypass mode to buck mode or boost mode. For example only, a battery system operates at ˜660V and supplies 75 A in a discharge-buck mode. Switching of the power switches T1and T2is being performed at 75 kHz. When operating in Opt 1, the power switch T1has about 260 W in conduction loss and the power switch T3has about 110 W of conduction loss. In bypass mode, however, the bypass switches SW1and SW3have about 6 W of conduction loss. Referring now toFIGS.6A to6F, operation of the DC-DC converter is shown when transitioning between discharge-buck mode and bypass mode or charge-boost mode and bypass mode. InFIG.6A, an initial mode is shown with power switches T1and T2switching, the power switch T3is ON and the power switch T4is OFF. The bypass switches SW1and SW3are OFF. InFIG.6B, the bypass switch SW3is turned ON across the power switch T3. InFIG.6C, the power switch T3is turned off and conduction loss in the power switch T3is eliminated. InFIG.6D, the power switch T2is turned off and switching loss in the power switch T2is eliminated. InFIG.6E, the bypass switch SW1is closed across the power switch T1. InFIG.6F, the power switch T1is turned off and conducting and switching loss in the power switch T1is eliminated. Referring now toFIGS.7to10, tables illustrating switching logic when transitioning between various modes are shown. InFIG.7, switching logic when transitioning between discharge-buck mode and bypass mode or charge-boost mode and bypass mode is shown. InFIG.8, switching logic when transitioning between bypass mode and discharge-buck mode or bypass mode and charge-boost mode is shown. InFIG.9, switching logic when transitioning between discharge-boost mode and bypass mode or charge-buck mode and bypass mode is shown. InFIG.10, switching logic when transitioning between bypass mode and discharge-boost mode or bypass and charge-buck mode is shown. The foregoing description is merely illustrative in nature and is in no way intended to limit the disclosure, its application, or uses. The broad teachings of the disclosure can be implemented in a variety of forms. Therefore, while this disclosure includes particular examples, the true scope of the disclosure should not be so limited since other modifications will become apparent upon a study of the drawings, the specification, and the following claims. It should be understood that one or more steps within a method may be executed in different order (or concurrently) without altering the principles of the present disclosure. Further, although each of the embodiments is described above as having certain features, any one or more of those features described with respect to any embodiment of the disclosure can be implemented in and/or combined with features of any of the other embodiments, even if that combination is not explicitly described. In other words, the described embodiments are not mutually exclusive, and permutations of one or more embodiments with one another remain within the scope of this disclosure. Spatial and functional relationships between elements (for example, between modules, circuit elements, semiconductor layers, etc.) are described using various terms, including “connected,” “engaged,” “coupled,” “adjacent,” “next to,” “on top of,” “above,” “below,” and “disposed.” Unless explicitly described as being “direct,” when a relationship between first and second elements is described in the above disclosure, that relationship can be a direct relationship where no other intervening elements are present between the first and second elements, but can also be an indirect relationship where one or more intervening elements are present (either spatially or functionally) between the first and second elements. As used herein, the phrase at least one of A, B, and C should be construed to mean a logical (A OR B OR C), using a non-exclusive logical OR, and should not be construed to mean “at least one of A, at least one of B, and at least one of C.” In the figures, the direction of an arrow, as indicated by the arrowhead, generally demonstrates the flow of information (such as data or instructions) that is of interest to the illustration. For example, when element A and element B exchange a variety of information but information transmitted from element A to element B is relevant to the illustration, the arrow may point from element A to element B. This unidirectional arrow does not imply that no other information is transmitted from element B to element A. Further, for information sent from element A to element B, element B may send requests for, or receipt acknowledgements of, the information to element A. In this application, including the definitions below, the term “module” or the term “controller” may be replaced with the term “circuit.” The term “module” may refer to, be part of, or include: an Application Specific Integrated Circuit (ASIC); a digital, analog, or mixed analog/digital discrete circuit; a digital, analog, or mixed analog/digital integrated circuit; a combinational logic circuit; a field programmable gate array (FPGA); a processor circuit (shared, dedicated, or group) that executes code; a memory circuit (shared, dedicated, or group) that stores code executed by the processor circuit; other suitable hardware components that provide the described functionality; or a combination of some or all of the above, such as in a system-on-chip. The module may include one or more interface circuits. In some examples, the interface circuits may include wired or wireless interfaces that are connected to a local area network (LAN), the Internet, a wide area network (WAN), or combinations thereof. The functionality of any given module of the present disclosure may be distributed among multiple modules that are connected via interface circuits. For example, multiple modules may allow load balancing. In a further example, a server (also known as remote, or cloud) module may accomplish some functionality on behalf of a client module. The term code, as used above, may include software, firmware, and/or microcode, and may refer to programs, routines, functions, classes, data structures, and/or objects. The term shared processor circuit encompasses a single processor circuit that executes some or all code from multiple modules. The term group processor circuit encompasses a processor circuit that, in combination with additional processor circuits, executes some or all code from one or more modules. References to multiple processor circuits encompass multiple processor circuits on discrete dies, multiple processor circuits on a single die, multiple cores of a single processor circuit, multiple threads of a single processor circuit, or a combination of the above. The term shared memory circuit encompasses a single memory circuit that stores some or all code from multiple modules. The term group memory circuit encompasses a memory circuit that, in combination with additional memories, stores some or all code from one or more modules. The term memory circuit is a subset of the term computer-readable medium. The term computer-readable medium, as used herein, does not encompass transitory electrical or electromagnetic signals propagating through a medium (such as on a carrier wave); the term computer-readable medium may therefore be considered tangible and non-transitory. Non-limiting examples of a non-transitory, tangible computer-readable medium are nonvolatile memory circuits (such as a flash memory circuit, an erasable programmable read-only memory circuit, or a mask read-only memory circuit), volatile memory circuits (such as a static random access memory circuit or a dynamic random access memory circuit), magnetic storage media (such as an analog or digital magnetic tape or a hard disk drive), and optical storage media (such as a CD, a DVD, or a Blu-ray Disc). The apparatuses and methods described in this application may be partially or fully implemented by a special purpose computer created by configuring a general purpose computer to execute one or more particular functions embodied in computer programs. The functional blocks, flowchart components, and other elements described above serve as software specifications, which can be translated into the computer programs by the routine work of a skilled technician or programmer. The computer programs include processor-executable instructions that are stored on at least one non-transitory, tangible computer-readable medium. The computer programs may also include or rely on stored data. The computer programs may encompass a basic input/output system (BIOS) that interacts with hardware of the special purpose computer, device drivers that interact with particular devices of the special purpose computer, one or more operating systems, user applications, background services, background applications, etc. The computer programs may include: (i) descriptive text to be parsed, such as HTML (hypertext markup language), XML (extensible markup language), or JSON (JavaScript Object Notation) (ii) assembly code, (iii) object code generated from source code by a compiler, (iv) source code for execution by an interpreter, (v) source code for compilation and execution by a just-in-time compiler, etc. As examples only, source code may be written using syntax from languages including C, C++, C#, Objective-C, Swift, Haskell, Go, SQL, R, Lisp, Java®, Fortran, Perl, Pascal, Curl, OCaml, Javascript®, HTML5 (Hypertext Markup Language 5th revision), Ada, ASP (Active Server Pages), PHP (PHP: Hypertext Preprocessor), Scala, Eiffel, Smalltalk, Erlang, Ruby, Flash®, Visual Basic®, Lua, MATLAB, SIMULINK, and Python®. | 21,295 |
11863073 | DETAILED DESCRIPTION Compensation circuits are employed to stabilize the control loop in DC-DC converters and other feedback loop-controlled circuits. The type (e.g., type-1, type-2, or type-3) of compensation circuit employed is selected based on various parameters (e.g., output filter component type and size, switching frequency, bandwidth, etc.) of the circuit being controlled. Type-2 compensation is widely used in DC-DC converters. For example, type-2 compensation may be used in applications where the frequency of the zero caused by the circuit output capacitor and its equivalent series resistance is smaller than the closed loop bandwidth of the control loop. DC-DC converters that use voltage-controlled oscillator (VCO) based pulse frequency mode (PFM) in light load conditions suffer from phase margin degradation due to the switching frequency pole. Pulse width modulation (PWM) based DC-DC converters that implement pulse skipped mode (PSM) operation exhibit similar phase margin degradation. When the phase margin degradation results in negative phase margins, instability in the form of pulse grouping results. FIG.1shows signals in a DC-DC converter exhibiting pulse grouping in pulse frequency mode (PFM) operation. In this example, the DC-DC converter is an inverting buck-boost converter with type-2 compensation. The load applied to the DC-DC converter decreases over time. As the DC-DC converter's switching frequency decreases with the decreasing load, pulse grouping or bursting occurs in the switching (see region102). Some DC-DC converters attempt to avoid this behavior by using a dummy load to limit the switching frequency of the DC-DC converter, which limits the bandwidth of the converter, and reduces efficiency with light loads. The DC-DC converters described herein track the zero of the type-2 compensation circuit with converter switching frequency to maintain phase margin across the entire switching frequency range of the DC-DC converter. By maintaining phase margin, the DC-DC converters avoid pulse grouping with light loads. FIG.2is a block diagram for an example DC-DC converter200that includes compensation zero-tracking based on switching frequency. The DC-DC converter200includes a high-side transistor202, a low-side transistor204, an inductor206, an output capacitor208, a voltage divider210, and a controller212. The DC-DC converter200is configured as an inverting buck-boost converter. Some examples of the DC-DC converter200may be configured as a buck converter, a boost converter, a buck-boost converter, or other type of DC-DC converter. The controller212controls switching of the high-side transistor202and the low-side transistor204to provide a selected output voltage (VOUT) at the output218. The voltage divider210is coupled to the controller212for provision of output voltage feedback216to the controller212. The controller212controls switching of the high-side transistor202and the low-side transistor204based on the output voltage feedback216received from the voltage divider210. The controller212includes a compensation circuit214to stabilize control of VOUT generation. The compensation circuit214tracks the zero of the compensation circuit214with converter switching frequency to maintain phase margin across the entire switching frequency range of the DC-DC converter200. FIG.3is a schematic level diagram for an example compensation circuit300that includes compensation zero-tracking based on switching frequency. The compensation circuit300is an example of the compensation circuit214. The compensation circuit300includes an error amplifier302, a capacitor304, a capacitor306, a resistor308, a resistor310, a switch312, a switch314, and a switch control circuit301. The error amplifier302includes an input coupled to the voltage divider210for receipt of the output voltage feedback216, and an input coupled to a voltage reference circuit303. The error amplifier302generates an error signal representative of the difference between the output voltage feedback216and the reference voltage received from the voltage reference circuit303. The capacitor304is coupled to the output of the error amplifier302. The resistor308includes a first terminal coupled to the output of the error amplifier302, and a second terminal coupled to a first terminal of the switch312. A second terminal of the switch312is coupled to a first terminal of the capacitor306. A second terminal of the capacitor306is coupled to ground. The resistor310includes a first terminal coupled to the output of the error amplifier302, and a second terminal coupled to a first terminal of the switch314. A second terminal of the switch314is coupled to a first terminal of the capacitor306. The switch312and the switch314may be implemented using field effect transistors (FETs) in some implementations of the compensation circuit300. The resistance of the resistor310may greater (e.g., 10 time greater) than the resistance of the resistor308. The switch control circuit301controls switching of the switch312and the switch314to vary the resistance coupling the output of the error amplifier302to the capacitor306, and vary the location of the zero corresponding to the resistance. The switch control circuit301include a phase output301B that is coupled to a control terminal of the switch312and a phase output301C that is coupled to a control terminal of the switch314. A signal ϕ1(a switch control signal) generated by the switch control circuit301controls switching of the switch312, and a signal ϕ2(a switch control signal) generated by the switch control circuit301controls switching of the switch314. The signals ϕ1and ϕ2may be complementary. That is, the signal ϕ2may be inverted relative to (an inverted version of) the signal ϕ1. Accordingly, the switch312is closed when the switch314is open, and the switch312is open when the switch314is closed, and the switch control circuit301complementarily couples the resistors308and310to the capacitor306. The switch control circuit301includes an input terminal301A that is coupled to a control terminal of the high-side transistor202or a control terminal of the low-side transistor204. The switch control circuit301includes a logic gate328, an inverter330, an inverter332, and a delay circuit305. A first input of the logic gate328is coupled to the input terminal301A, and a second input of the logic gate328is coupled to the output of the delay circuit305. The logic gate328combines the signal at the input terminal301A and the delayed signal output by the delay circuit305to produce a control signal for controlling the switch312and the switch314. The output of the logic gate328is coupled to an input of the inverter330, and an output of the inverter330(the phase output301B) is coupled to an input of the inverter332. The output of the inverter332is coupled to the phase output301C. The delay circuit305includes an inverter316, a switch318, a resistor320, a capacitor322, and a Schmitt trigger326. A first terminal of the resistor320is coupled to the input terminal301A and the input of the inverter316. A second terminal of the resistor320is coupled to an input of the Schmitt trigger326, a first terminal of the switch318, and a first terminal of the capacitor322. The output of the inverter316is coupled to a control terminal of the switch318. A second terminal of the switch318and a second terminal of the capacitor322are coupled to ground. The switch318discharges the capacitor322when the signal at the input terminal301A is low. The switch318may be implemented using a FET in some implementations of the delay circuit305. The duty cycles of the signals ϕ1and ϕ2generated by the switch control circuit301change with the frequency of the switching control signal received at the input terminal301A. The resistance between the output of the error amplifier302and the capacitor306changes with the duty cycle of the signals ϕ1and ϕ2. The resistance decreases as switching frequency increases, and increases as switching frequency decreases. FIG.4is a bode diagram for an example of the compensation circuit300. The curves402and404represent magnitude and phase when the signal ϕ2is always on (lower switching frequency), and show a phase boost in a lower frequency region (e.g., 2-100 kilohertz). The curves406and408represent magnitude and phase when the signal ϕ1is always on (higher switching frequency), and show a phase boost in a higher frequency region (20 kilohertz to one megahertz). By modulating the duty cycle of the signals ϕ1and ϕ2with switching frequency, the effective resistance applied in the compensation circuit300varies such that the effective phase curve lies between the curves404and408. FIG.5is a graph of signals in an implementation of the DC-DC converter200that includes the compensation circuit300. InFIG.5, the load applied to the DC-DC converter200is linearly reduced over time. Output voltage, current in the inductor206, and output of the error amplifier302are shown inFIG.5. The reduction in load causes the switching frequency of the DC-DC converter200to decrease. As the switching frequency of the DC-DC converter200decreases, the compensation circuit300varies the compensation resistance to avoid the pulse grouping shown inFIG.1. FIG.6is a graph showing change in duty cycle applied in the compensation circuit300with decreasing load. As the load decreases, the inductor current and switching frequency of the DC-DC converter200also decreases, and the duty cycle of the signal ϕ2smoothly increases (interval602). No undesirable perturbations in the output voltage are present. FIG.7is a schematic diagram for an example compensation circuit700that includes compensation zero-tracking based on switching frequency. The compensation circuit700is similar to the compensation circuit300, and adds capacitance multiplication circuitry. The compensation circuit700includes the error amplifier302, the capacitor306, the resistor308, the resistor310, the switch312, the switch314, and the switch control circuit301(not shown) as described with respect to the compensation circuit300. The compensation circuit700further includes an amplifier702and a switch704. The amplifier702is a buffer amplifier (e.g., a unity gain buffer). A terminal of the amplifier702is coupled to the first terminal of the capacitor306. The output of the amplifier702is coupled to a second input of the amplifier702, and to a first terminal of the switch704. A second terminal of the switch704is coupled to the first terminal of the switch312. A control terminal of the switch704is coupled to the control terminal of the switch312. When the switches switch314and switch704are closed, a capacitance multiplier circuit is formed using the amplifier702. The switch704may implemented using a FET. FIG.8is a bode diagram for an example of the compensation circuit700. The curves802and804represent magnitude and phase when the signal ϕ2is always on (lower switching frequency), and show a phase boost in a lower frequency region (e.g., 20 kilohertz to 1 megahertz). The curves806and808represent magnitude and phase when the signal ϕ1is always on (higher switching frequency), and show a phase boost in a higher frequency region (20 kilohertz to 1 megahertz). By modulating the duty cycle of the signals ϕ1and ϕ2with switching frequency, the effective capacitance of the compensation circuit is controlled, and the region of phase boost810is extended with switching frequency. Gain in the region of phase boost810remains relatively constant. FIG.9is a graph of signals in a DC-DC converter that includes an example of the compensation circuit700. InFIG.9, load current increases from about 100 microamperes (at time A) to about 4 milliamperes (at time C) over time. The change in load has little impact on the output voltage, and the inductor current and output voltage show that no pulse grouping is present. FIGS.10A-10Care graphs of magnified signals from the graph ofFIG.9.FIG.10Azooms in on the signals of the graph ofFIG.9at time A (about 100 microamperes of load current). The DC-DC converter switching frequency is about 2 kilohertz. The signal ϕ2is activated after each switching operation with delay. Due to the low switching frequency, the signal ϕ2is high, and the capacitance multiplier of the compensation circuit700, is active most of the time to provide a phase-boost. FIG.10Bzooms in on the signals of the graph ofFIG.9at time B (about 1.8 milliamperes of load current). The DC-DC converter switching frequency is about 33 kilohertz. The signal ϕ2is activated after each switching operation with delay. The signal ϕ2is high and the capacitance multiplier of the compensation circuit700is active about half of the time. FIG.10Czooms in on the signals of the graph ofFIG.9at time C (about 4 milliamperes of load current). The DC-DC converter switching frequency is about 77 kilohertz. The signal ϕ2is not activated (the signal ϕ1is always active), and the capacitance multiplier of the compensation circuit700is off. Because of the higher switching frequency, no phase boost in the lower frequency range is needed. In this description, the term “couple” may cover connections, communications or signal paths that enable a functional relationship consistent with this description. For example, if device A generates a signal to control device B to perform an action, then: (a) in a first example, device A is directly coupled to device B; or (b) in a second example, device A is indirectly coupled to device B through intervening component C if intervening component C does not substantially alter the functional relationship between device A and device B, so device B is controlled by device A via the control signal generated by device A. Modifications are possible in the described embodiments, and other embodiments are possible, within the scope of the claims. | 13,878 |
11863074 | DETAILED DESCRIPTION Examples of the methods and systems discussed herein are not limited in application to the details of construction and the arrangement of components set forth in the following description or illustrated in the accompanying drawings. The methods and systems are capable of implementation in other embodiments and of being practiced or of being carried out in various ways. Examples of specific implementations are provided herein for illustrative purposes only and are not intended to be limiting. In particular, acts, components, elements and features discussed in connection with any one or more examples are not intended to be excluded from a similar role in any other examples. Also, the phraseology and terminology used herein is for the purpose of description and should not be regarded as limiting. Any references to examples, embodiments, components, elements or acts of the systems and methods herein referred to in the singular may also embrace embodiments including a plurality, and any references in plural to any embodiment, component, element or act herein may also embrace embodiments including only a singularity. References in the singular or plural form are not intended to limit the presently disclosed systems or methods, their components, acts, or elements. The use herein of “including,” “comprising,” “having,” “containing,” “involving,” and variations thereof is meant to encompass the items listed thereafter and equivalents thereof as well as additional items. References to “or” may be construed as inclusive so that any terms described using “or” may indicate any of a single, more than one, and all of the described terms. In addition, in the event of inconsistent usages of terms between this document and documents incorporated herein by reference, the term usage in the incorporated features is supplementary to that of this document; for irreconcilable differences, the term usage in this document controls. Power devices are configured to provide output power to a load. For example, an uninterruptible power supply (UPS) is a power device configured to provide regulated, uninterrupted power to a load. Single-phase UPSs provide single-phase power, and multi-phase UPSs provide multi-phase power. For example, a three-phase UPS provides three-phase power to a load. Power devices may include one or more power converters. Power converters include devices to convert power from one state to another, such as rectifiers, inverters, power-factor-correction circuits (PFCs), DC/DC converters, and so forth. Power converters may include multiple converter legs, each configured to convert a respective portion of input power. In some power converters, the legs may be coupled in parallel and interleaved together. Each interleaved leg converts a respective portion of the power provided by the power converter. Interleaving may advantageously reduce a ripple current in the power converter. Interleaved multi-leg converters may include common components. Common components are common amongst each of the converter legs, such that power provided by each of the converter legs is provided to the common components. Conversely, individual components are components within or coupled to a single leg of the multi-leg converters. In some examples, individual components only receive power from the converter leg in which the individual component is disposed rather than receiving power from each leg in a converter. In some interleaved multi-leg converters, common components may include at least one common filter capacitor, filter inductor, relay, and/or fuse. At least because the common components receive current from each of the multiple legs, the common components may have high current ratings in order to sustain the high aggregate currents provided to the common components. Such high current ratings may introduce practical manufacturing challenges. For example, it may be difficult or impossible to acquire PCB-mounted components having sufficiently high current ratings. Moreover, regulatory requirements may be difficult or impossible to attain. For example, Underwriters Laboratories (UL) requirements may not accept hard paralleling of common components such as backfeed relays and fuses, as failure of one component may cause a voltage spike or a high current in the non-failing components. Further still, a common fuse may have a significant clearing energy as a result of the high common current conducted by the common fuse. Such significant clearing energy may disadvantageously damage adjacent components, making troubleshooting difficult and complicating efforts to live-swap components. Examples of the disclosure include a multi-leg converter having multiple parallel and/or interleaved legs. Each converter leg may include one or more individual components. The one or more individual components may include components that are implemented as common components in other topologies. The one or more individual components may include, for example, one or more fuses, relays, filters, current sensors, and so forth. In some examples, each converter leg includes a current sensor configured to provide a current-sense signal indicative of a current in the respective converter leg to at least one controller. The at least one controller may determine a current difference between the paralleled converter legs and decrease the current difference to balance the converter legs. For example, the at least one controller may control one or more power converters of the respective converter legs to modulate a current in one or more of the converter legs. Accordingly, examples of the disclosure include multi-leg power converters having an increased power density with a reduction in size of converter components, such as passive components. Current power-conversion systems, such as power converters in UPSs, may include interleaved converter legs with one or more common components coupled to all of the interleaved converter legs. Such power-conversion systems may operate inefficiently, because the common components conduct an entire converter current, which may require very large components that are difficult to acquire commercially, reduce power density, have significant clearing energy, and complicate efforts to obtain UL certification. This is a technical problem. An exemplary embodiment of a power-conversion system includes an uninterruptible power supply having a first input configured to be coupled to a primary power source, a second input configured to be coupled to a backup power source, an output configured to be coupled to a load, a first converter leg configured to provide a first voltage signal to the output, wherein the first converter leg includes at least one of a first relay or a first fuse, a second converter leg configured to provide a second voltage signal to the output and configured to be coupled in parallel with the first converter leg, wherein the second converter leg provides the second voltage signal out of phase with the first converter leg providing the first voltage signal, and wherein the second converter leg includes at least one of a second relay or a second fuse, a first current sensor coupled to the first converter leg and being configured to provide a first current-sense signal indicative of a first current in the first converter leg, a second current sensor coupled to the second converter leg and being configured to provide a second current-sense signal indicative of a second current in the second converter leg, and at least one controller configured to receive the first current-sense signal and the second current-sense signal, determine a current difference between the first converter leg and the second converter leg based on the first current-sense signal and the second current-sense signal, and decrease the current difference. At least this foregoing combination of features comprises a power-conversion system that serves as a technical solution to the foregoing technical problem. This technical solution is not routine and is unconventional. This technical solution is a practical application of the power-conversion system design that solves the foregoing technical problem and constitutes an improvement in the technical field of power converters at least by reducing a current conducted by converter components. Example power converters may be implemented in any of many types of power devices. For purposes of explanation, examples are given in which power converters are implemented in a UPS. Example power converters may be implemented in any of many types of UPSs, such as offline UPSs, online UPSs, line-interactive UPSs, single-phase UPSs, multi-phase UPSs, and so forth. Accordingly, it is to be appreciated that example power converters are not limited to implementation in UPSs, nor are example power converters limited to specific UPS topologies. FIG.1is a block diagram of a UPS100according to an example. The UPS100includes an input102, an AC/DC converter104, one or more DC busses106, a DC/DC converter108, an energy-storage-device interface110, at least one controller112(“controller112”), a DC/AC inverter114, an output116, a memory and/or storage118, and one or more communication interfaces120(“communication interfaces120”), which may be communicatively coupled to one or more external systems122(“external systems122”). The input102is coupled to the AC/DC converter104and to an AC power source (not pictured), such as an AC mains power supply. The AC/DC converter104is coupled to the input102and to the one or more DC busses106, and is communicatively coupled to the controller112. The one or more DC busses106are coupled to the AC/DC converter104, the DC/DC converter108, and to the DC/AC inverter114, and are communicatively coupled to the controller112. The DC/DC converter108is coupled to the one or more DC busses106and to the energy-storage-device interface110, and is communicatively coupled to the controller112. The energy-storage-device interface110is coupled to the DC/DC converter108, and is configured to be coupled to at least one battery124and/or another energy-storage device. In some examples, the UPS100may include one or more energy-storage devices, such as the battery124. The DC/AC inverter114is coupled to the one or more DC busses106and to the output116, and is communicatively coupled to the controller112. The output116is coupled to the DC/AC inverter114, and to an external load (not pictured). The controller112is communicatively coupled to the AC/DC converter104, the one or more DC busses106, the DC/DC converter108, the energy-storage-device interface110, the DC/AC inverter114, the memory and/or storage118, and the communication interfaces120. The input102is configured to be coupled to an AC mains power source and to receive input AC power having an input voltage level. The UPS100is configured to operate in different modes of operation based on the input voltage of the AC power provided to the input102. The controller112may determine a mode of operation in which to operate the UPS100based on whether the input voltage of the AC power is acceptable. The controller112may include or be coupled to one or more sensors configured to sense parameters of the input voltage. For example, the controller112may include or be coupled to one or more sensors configured to sense a voltage level of the AC power received at the input102, one or more current sensors in each of the components104,108, and114, and so forth. When AC power provided to the input102is acceptable (for example, by having parameters, such as an input voltage value, that meet specified values, such as by falling within a range of acceptable input voltage values), the controller112controls components of the UPS100to operate in a normal mode of operation. In the normal mode of operation, AC power received at the input102is provided to the AC/DC converter104. The AC/DC converter104converts the AC power into DC power and provides the DC power to the one or more DC busses106. The one or more DC busses106distribute the DC power to the DC/DC converter108and to the DC/AC inverter114. The DC/DC converter108converts the received DC power and provides the converted DC power to the energy-storage-device interface110. The energy-storage-device interface110receives the converted DC power, and provides the converted DC power to the battery124to charge the battery124. The DC/AC inverter114receives DC power from the one or more DC busses106, converts the DC power into regulated AC power, and provides the regulated AC power to the output116to be delivered to a load. When AC power provided to the input102from the AC mains power source is not acceptable (for example, by having parameters, such as an input voltage value, that do not meet specified values, such as by falling outside of a range of acceptable input voltage values), the controller112controls components of the UPS100to operate in a backup mode of operation. In the backup mode of operation, DC power is discharged from the battery124to the energy-storage-device interface110, and the energy-storage-device interface110provides the discharged DC power to the DC/DC converter108. The DC/DC converter108converts the received DC power and distributes the DC power amongst the one or more DC busses106. For example, the DC/DC converter108may evenly distribute the power amongst the one or more DC busses106. The one or more DC busses106provide the received power to the DC/AC inverter114. The DC/AC inverter114receives the DC power from the one or more DC busses106, converts the DC power into regulated AC power, and provides the regulated AC power to the output116. The controller112may control aspects of the UPS100in addition to, or in lieu of, one or more of the acts discussed above. For example, the controller112may control and/or communicate one or more components or devices of the AC/DC converter104, the DC/DC converter108, and/or the DC/AC inverter114, such as switches, fuses, relays, current sensors, and so forth. The controller112may store information in, and/or retrieve information from, the memory and/or storage118. For example, the controller112may store information indicative of sensed parameters (for example, input-voltage values of the AC power received at the input102, converter-current values of current in the components104,108, and/or114, and so forth) in the memory and/or storage118. The controller112may further receive information from, or provide information to, the communication interfaces120. The communication interfaces120may include one or more communication interfaces including, for example, user interfaces (such as display screens, touch-sensitive screens, keyboards, mice, track pads, dials, buttons, switches, sliders, light-emitting components such as light-emitting diodes, sound-emitting components such as speakers, buzzers, and so forth configured to output sound inside and/or outside of a frequency range audible to humans, and so forth), wired communication interfaces (such as wired ports), wireless communication interfaces (such as antennas), and so forth, configured to exchange information with one or more systems, such as the external systems122, or other entities, such as human beings. The external systems122may include any device, component, module, and so forth, that is external to the UPS100, such as a server, database, laptop computer, desktop computer, tablet computer, smartphone, central controller or data-aggregation system, other UPSs, and so forth. As discussed above, power converters, such as the AC/DC converter104, DC/DC converter108, and/or DC/AC inverter114, may include multiple converter legs. The multiple converter legs may be interleaved together. Interleaving may advantageously reduce a ripple current in the respective power converter. An example power converter, which may be an example of any of the converters104,108, and/or114, is provided with respect toFIG.2. Specific examples may be provided with respect to the DC/AC inverter114for purposes of explanation. FIG.2illustrates a block diagram of a power converter200according to an example. The power converter200may be an example of aspects of the DC/AC inverter114described above with reference toFIG.1. It is to be appreciated that one or more components of the DC/AC inverter114may be omitted for purposes of clarity. The power converter200is an example of an interleaved power converter having two interleaved converter legs. It is to be appreciated that, in some examples, a power converter (such as the DC/AC inverter114) may include more than two interleaved converter legs. Accordingly, no limitation is implied by the power converter200having two converter legs, which is provided for purposes of explanation only. The power converter200includes an input202, an output204, a first converter leg206, a second converter leg208, and common output components210. The first converter leg206includes at least one converter switch212(“first converter switches212”) and at least one first-leg output component214(“first output components214”). The second converter leg208includes at least one converter switch216(“second converter switches216”) and at least one second-leg output component218(“second output components218”). The input202is coupled to the first converter switches212and the second converter switches216. In some examples, the input202may also be coupled to a power source (not illustrated). For example, where the power converter200is an example of the DC/AC inverter114, the input202may be coupled to the DC busses106. The first converter switches212are coupled to the input202at a switch input, are coupled to the first output components214at a switch output, and are communicatively coupled to the controller112. The first output components214are coupled to the first converter switches212are a first connection and are coupled to the common output components210at a second connection. In some examples, one or more of the first output components214are communicatively coupled to the controller112. The first converter leg206is coupled in parallel with the second converter leg208. The second converter switches216are coupled to the input202at a switch input, are coupled to the second output components218at a switch output, and are communicatively coupled to the controller112. The second output components218are coupled to the second converter switches216at a first connection, and are coupled to the common output components210at a second connection. In some examples, one or more of the second output components218are communicatively coupled to the controller112. The common output components210are coupled to the first output components214and the second output components218at a first connection, and are coupled to the output204at a second connection. The output204is coupled to the common output components210and is configured to be coupled to a load. For example, where the power converter200is an example of the DC/AC converter114, the output204may be coupled to the output116. Accordingly, the power converter200includes output components in each respective leg (for example, the first output components214and second output components218) and includes common output components common to both of the converter legs206,208(for example, the common output components210). In one example, the output components214,218may each include a filter choke, and the common output components210may include one or more filtering components (for example, one or more capacitors and/or inductors), relays, fuses, and so forth. Because power passing through the common output components210is a combination of the power from the converter legs206,208, the current through the common output components210may be greater than a current through either of the converter legs206,208. The common output components210may therefore require a higher current rating than the output components214,218require at least because the common output components210receive a higher current than each of the output components214,218when the output components214,218simultaneously provide a current to the common output components210. As discussed above, subjecting certain components to high currents may disadvantageously require large components that are difficult or impossible to mount on a PCB, amongst other disadvantages discussed above. The common output components210, for example, may disadvantageously require large components such as relays, fuses, filtering components, and so forth. Conversely, the output components214,218may not be as large at least because the output components214,218conduct about half of the current that the common output components210conduct. Accordingly, in some examples, one or more of the common output components210may be removed and implemented instead in the output components214,218, enabling a reduction in size of the components. An example is provided with respect toFIG.3. FIG.3illustrates a block diagram of a power converter300according to another example. The power converter300may be an example of aspects of the DC/AC inverter114described above with reference toFIG.1. It is to be appreciated that one or more components of the DC/AC inverter114may be omitted for purposes of clarity. The power converter300is another example of an interleaved power converter having two interleaved converter legs. It is to be appreciated that, in some examples, a power converter (such as the DC/AC inverter114) may include more than two interleaved converter legs. Accordingly, no limitation is implied by the power converter300having two converter legs, which is provided for purposes of explanation only. The power converter300includes an input302, an output304, a first converter leg306, and a second converter leg308. The first converter leg306includes at least one first converter switch310(“first converter switches310”) and at least one first-leg output component312(“first output components312”). The second converter leg308includes at least one second converter switch314(“second converter switches314”) and at least one second-leg output component316(“second output components316”). The input302is coupled to the first converter switches310and the second converter switches314. In some examples, the input302may also be coupled to a power source (not illustrated). For example, where the power converter300is an example of the DC/AC inverter114, the input302may be coupled to the DC busses106discussed above with respect toFIG.1. As discussed in greater detail below, the input302may include several input connections, each configured to be coupled to a respective DC bus of the DC busses106. The first converter switches310are coupled to the input302at a switch input, are coupled to the first output components312at a switch output, and are communicatively coupled to the controller112. The first output components312are coupled to the first converter switches310are a first connection and are coupled to the output304at a second connection. In some examples, one or more of the first output components312are communicatively coupled to the controller112. The first converter leg306is coupled in parallel with the second converter leg308. The second converter switches314are coupled to the input302at a switch input, are coupled to the second output components316at a switch output, and are communicatively coupled to the controller112. The second output components316are coupled to the second converter switches314at a first connection, and are coupled to the output304at a second connection. In some examples, one or more of the second output components316are communicatively coupled to the controller112. The output304is coupled to the first output components312and the second output components316at a first connection, and is configured to be coupled to a load. For example, where the power converter300is an example of the DC/AC inverter114, the output304may be coupled to the output116. FIG.4illustrates a process400of operating a power converter, such as the power converter300, according to an example. In various examples, at least part of the process400may be executed by the controller112. At act402, the process400begins. At act404, input power is received at the input302. For example, the input power may be received by the DC/AC inverter114via the DC busses106from a power source coupled to the input102and/or from the battery124. At act406, the controller112controls the first converter leg306and/or the second converter leg308to draw power from the input302. For example, the controller112may control the first converter switches310and/or the second converter switches314to draw power from the input302. In some examples, the controller112may control the first converter switches310to draw power from the input302180° out of phase with the second converter switches314. That is, control signals provided to the second converter switches314may be substantially similar or identical to control signals provided to the first converter switches310, but phase-shifted by 180°. In some examples, the controller112may phase-shift control signals provided to a multi-leg converter based on a number of legs. For example, the phase shift may be equal to ϕ=360°/n, where ϕ is the phase shift and n is the number of legs in the nuti-leg converter. In other examples, the controller112may implement other methods of determining a phase shift. For example, the controller112may control the first converter switches310to draw power from the input302out of phase with the second converter switches314by a number of degrees other than 180°. The converter switches310,314may be implemented according to any of various known converter topologies, and the controller112may control the converter switches310,314in accordance a known control scheme corresponding to the known converter topology. As discussed below, the controller112may control the converter legs306,308to draw a balanced amount of power from the input302such that a current difference between the converter legs306,308is minimized. At act408, the controller112controls the first converter switches310and/or the second converter switches314to provide output power to the output304via the first output components312and/or the second output components316, respectively. For example, the controller112may control the converter switches310,314to provide, via respective switch outputs, converted output power to the output components312,316. Converting the output power may include inverting DC power received at the input302to AC power for output at the output304. As discussed in greater detail below, the output components312,316may include one or more filtering components configured to filter the output power. The output components312,316may further include one or more relays and/or fuses. In various examples, the output components312,316provide power to the output304from the converter switches310,314where, for example, the relay(s) are closed and conducting and the fuse(s) are conductive (that is, not “blown”). As discussed in greater detail below, the output components312,316may include inductively coupled chokes configured to share current between the legs306,308, such as by sharing a ripple current between the legs306,308. As discussed above with respect to act406, the controller112may control the first converter switches310to draw power from the input302180° out of phase with the second converter switches314. Accordingly, the controller112may control the first converter switches310such that a voltage signal provided by the first converter switches310(and, therefore, the first converter leg306) to the output304is out of phase with a voltage signal provided by the second converter switches314(and, therefore, the second converter leg308) to the output304. At act410, the controller112receives a first current-sense signal from the first converter leg306and a second current-sense signal from the second converter leg308. In various examples, the first output components312include at least one first current sensor (not illustrated), such as a current transformer (CT), and the second output components316include at least one second current sensor, such as a CT (not illustrated). The at least one first current sensor may measure a current through the first converter leg306and provide a first current-sense signal to the controller112, and the at least one second current sensor may measure a current through the second converter leg308and provide a second current-sense signal to the controller112. In some examples, a sampling frequency and/or period of the at least one first current sensor and of the at least one second sensor are aligned or standardized such that the first current-sense signal and the second current-sense signal are aligned. At act412, the controller112determines a current difference between the converter legs306,308. In some examples, the current difference is an instantaneous current difference. In other examples, the current difference is determined over a period of time, such as a complete switching cycle of signals provided to the converter switches310,314. The controller112may be configured to minimize the current difference such that a current through the converter legs306,308is balanced (for example, equal to one another instantaneously or over a period of time). At act414, the controller112decreases the current difference determined at act410. Decreasing the current difference may include increasing a power draw of one of the converter legs306,308and/or decreasing a power draw of the other of the converter legs306,308. For example, if a current in the first converter leg306exceeds a current in the second converter leg308, the controller112may decrease the current difference by controlling the first converter switches310to draw less power from the input302, and/or by controlling the second converter switches314to draw more power from the input302. The process400then returns to act404. Accordingly, the power converter300is configured to receive input power at the input302, convert the input power at least with the converter switches310,314, and provide the converted output power to the output304. A schematic diagram of one example of the power converter300is illustrated with respect toFIG.5. FIG.5illustrates a schematic diagram of the power converter300according to an example. The power converter300ofFIG.5includes a first input302aand a second input302b(collectively, the input302), the output304, the first converter leg306having the first converter switches310and the first output components312, and the second converter leg308having the second converter switches314and the second output components316. The first converter switches310include a first converter switch500and a second converter switch502. The second converter switches314include a third converter switch504and a fourth converter switch506. The first output components312include a first inductor508, a first capacitor510, a second inductor512(the inductors508,512and the first capacitor510collectively being a “first filter”), a first relay514, and a first fuse516. The second output components316include a third inductor518, a second capacitor520, a fourth inductor522(the inductors518,522and the second capacitor520collectively being a “second filter”), a second relay524, and a second fuse526. In some examples, the first output components312and the second output components316each include one or more current sensors (for example, CTs) (not illustrated) configured to measure a respective current through the converter legs306,308and send a respective current-sense signal to the controller112. In various examples, each of the second inductor512and the fourth inductor522may be referred to as a “choke.” The first input302ais coupled to the first converter switch500and the third converter switch504, and is configured to be coupled to a power source or medium therefor, such as by being coupled to the DC busses106. For example, the first input302amay be coupled to a positive-voltage DC bus, or another type of bus, of the DC busses106. The second input302bis coupled to the second converter switch502and the fourth converter switch506, and is configured to be coupled to a power source or medium therefor, such as by being coupled to the DC busses106. For example, the second input302bmay be coupled to a negative-voltage, neutral, common, or other type of DC bus of the DC busses106. The first converter switch500is coupled to the first input302aat a first connection, the second converter switch502and the first inductor508at a second connection, and is communicatively coupled to the controller112at a control connection. The second converter switch502is coupled to the first converter switch500and the first inductor508at a first connection, is coupled to the second input302bat a second connection, and is communicatively coupled to the controller112at a control connection. The third converter switch504is coupled to the first input302aat a first connection, is coupled to the fourth converter switch506and the third inductor518at a second connection, and is communicatively coupled to the controller112at a control connection. The fourth converter switch506is coupled to the third converter switch504and the third inductor518at a first connection, is coupled to the second input302bat a second connection, and is communicatively coupled to the controller112at a control connection. The first inductor508is coupled to the first converter switch500and the second converter switch502at a first connection, and is coupled to the first capacitor510and the second inductor512at a second connection. The first capacitor510is coupled to the first inductor508and the second inductor512at a first connection, and is coupled to a second reference node530at a second connection. The second inductor512is coupled to the first inductor508and the first capacitor510at a first connection, and is coupled to the first relay514at a second connection. In some examples, the second inductor512is inductively coupled to the fourth inductor522. For example, the second inductor512may be inductively coupled to the fourth inductor522to allow ripple current to flow between the converter legs306,308, to reduce the ripple current in capacitors510,520of power converter300. In some examples, the first inductor508and the third inductor518may also be at least partially inductively coupled to each other to reduce ripple currents in each converter leg. For example, a ripple current in the first inductor508may induce an induced current in the third inductor518, and a ripple current in the third inductor518may induce an induced current in the first inductor508. The first relay514is coupled to the second inductor512at a first connection, is coupled to the first fuse516at a second connection, and is communicatively coupled to the controller112. The first fuse516is coupled to the first relay514at a first connection, and is coupled to the output304at a second connection. The third inductor518is coupled to the third converter switch504and the fourth converter switch506at a first connection, and is coupled to the second capacitor520and the fourth inductor522at a second connection. The second capacitor520is coupled to the third inductor518and the fourth inductor522at a first connection, and is coupled to the second reference node530at a second connection. The fourth inductor522is coupled to the third inductor518and the second capacitor520at a first connection, and is coupled to the second relay524at a second connection. The second relay524is coupled to the fourth inductor522at a first connection, is coupled to the second fuse526at a second connection, and is communicatively coupled to the controller112. The second fuse526is coupled to the second relay524at a first connection, and is coupled to the output304at a second connection. The output304is coupled to the first fuse516and the second fuse526, and is configured to be coupled to a load or a medium thereto, such as the output304, which may be coupled to a load. It is to be appreciated that the foregoing connections are examples, and that other configurations are within the scope of the disclosure. In some examples, a position of the first relay514may be swapped with a position of the first fuse516in the first converter leg306. Similarly, a position of the first inductor508, the first capacitor510, and the second inductor512may be swapped with the first relay514and/or the first fuse516in the first converter leg306. Similar principles apply to the second converter leg308. Furthermore, although in some examples the output components312,316may be coupled between the converter switches310,314and the output304(for example, between the switch output of the converter switches310,314and the output304), in other examples, the output components312,316may be coupled between the input302and the output components312,316. For example,FIG.6illustrates a block diagram of a power converter600according to another example. The power converter600includes substantially similar components as the power converter300described above with reference toFIG.3, and like components are labeled accordingly. However, the power converter600may be an example of the AC/DC converter104and/or the DC/DC converter108. Components of the power converter600may therefore be arranged differently than those of the power converter300, notwithstanding the fact that, in some examples, the components of the power converter600are substantially the same as those of the power converter300. In some examples, components of the power converter600, such as one or more components of the output components312,316, may differ from those of the power converter300as discussed below. For example, the power converter600includes the input302, the output304, the first converter leg306, and the second converter leg308. The first converter leg306includes the first converter switches310and the first output components312. The second converter leg308includes the second converter switches314and the second output components316. However, components of the power converter600are arranged differently than those of the power converter300. For example, the input302is coupled to the first output components312and the second output components316, the first converter switches310are coupled between the first output components312and the output304, and the second converter switches314are coupled between the second output components316and the output304. In an example in which the power converter600is implemented as the AC/DC converter104, the input302may be coupled to the input102, and the output304may be coupled to the DC busses106. In an example in which the power converter600is implemented as the DC/DC converter108, the input302may be coupled to the energy-storage-device interface110, and the output304may be coupled to the DC busses106. The process400may be executed in connection with the power converter600in a substantially similar manner as with the power converter300, although certain acts of the process400may differ. For example, at act408, the controller112may control the converter switches310,314to convert power received at the input302to converted power to provide at the output304, but the conversion may include converting AC power to DC power, such as where the power converter600is an example of the AC/DC converter104, and/or may include converting DC power to converted DC power, such as where the power converter600is an example of the DC/DC converter108, rather than inverting DC power to AC power. However, other acts of the process400, such as decreasing the current difference between the legs306,308at act414, may be substantially similar or identical. FIG.7illustrates a schematic diagram of the power converter600according to an example. The power converter600ofFIG.7may be an example implementation of the AC/DC converter104, which may operate as a PFC. The power converter600ofFIG.7includes the input302, a first output304aand a second output304b(collectively, the output304), the first converter leg306, the second converter leg308, the first converter switches310, the first output components312, the second converter switches314, the second output components316, the converter switches500-506, the first inductor508, the first capacitor510, the second inductor512, the first relay514, the first fuse516, the third inductor518, the second capacitor520, the fourth inductor522, the second relay524, and the second fuse526. However, the components of the power converter600ofFIG.7may be interconnected differently than the components of the power converter300ofFIG.5. The input302is coupled to the first fuse516and the second fuse526, and is configured to be coupled to a power source. For example, where the power converter600ofFIG.7is an example of the AC/DC converter104, the input302may be coupled to the input102, which may be coupled to a power source. The first fuse516is coupled to the input302at a first connection, and is coupled to the first relay514at a second connection. The first relay514is coupled to the first fuse516at a first connection, and is coupled to the second inductor512at a second connection. In some examples, the first relay514is coupled to the controller112at a control connection. The second inductor512is coupled to the first relay514at a first connection, and is coupled to the first inductor508and the first capacitor510at a second connection. The first inductor508is coupled to the first capacitor510at a first connection, and is coupled to the first converter switch500and the second converter switch502at a second connection. The first capacitor510is coupled to the first inductor508and the second inductor512at a first connection, and is coupled to the second reference node530at a second connection. The second fuse526is coupled to the input302at a first connection, and is coupled to the second relay524at a second connection. The second relay524is coupled to the second fuse526at a first connection, and is coupled to the fourth inductor522at a second connection. In some examples, the second relay524is coupled to the controller112at a control connection. The fourth inductor522is coupled to the second relay524at a first connection, and is coupled to the third inductor518and the second capacitor520at a second connection. The third inductor518is coupled to the fourth inductor522and the second capacitor520at a first connection, and is coupled to the third converter switch504and the fourth converter switch506at a second connection. The first output304ais coupled to the first converter switch500and the third converter switch504, and is configured to be coupled to a load or a medium thereto. For example, the first output304amay be coupled to a positive-voltage DC bus, or another type of bus, of the DC busses106. The second output304bis coupled to the second converter switch502and the fourth converter switch506, and is configured to be coupled to a load or a medium thereto. For example, the second output304bmay be coupled to a negative-voltage, neutral, common, or other type of DC bus of the DC busses106. The first converter switch500is coupled to the first output304aat a first connection, is coupled to the first inductor508and the second converter switch502at a second connection, and is communicatively coupled to the controller112. The second converter switch502is coupled to the first inductor508and the first converter switch500at a first connection, is coupled to the second output304bat a second connection, and is communicatively coupled to the controller112at a control connection. The third converter switch504is coupled to the first output304aat a first connection, is coupled to the third inductor518and the fourth converter switch506at a second connection, and is communicatively coupled to the controller112at a control connection. The fourth converter switch506is coupled to the third converter switch504and the third inductor518at a first connection, is coupled to the second output304bat a second connection, and is communicatively coupled to the controller112at a control connection. As discussed above, the components of the power converter600ofFIG.7may be substantially similar to the components of the power converter300ofFIG.5. However, the components of the power converter600ofFIG.7may be interconnected differently inasmuch as a position of the converter switches310,314may be switched with a position of the output components312,316, respectively. As discussed above, the controller112may control the converter switches500-506differently, such as by controlling the converter switches500-506to convert AC power received at the input302to converted DC power to provide to the output304. Furthermore, whereas the power converter300ofFIG.5may include the first input302a, the second input302b, and the output304, the power converter600may include the input302, the first output304a, and the second output304b. In other examples, the power converter600may be implemented as another converter of the UPS100, such as the DC/DC converter108.FIG.8illustrates a schematic diagram of the power converter600according to another example. The power converter600ofFIG.8may be an example implementation of the DC/DC converter108, which may operate as a voltage-level converter. The power converter600ofFIG.8includes the input302, the first output304a, the second output304b, the first converter leg306, the second converter leg308, the first converter switches310, the first output components312, the second converter switches314, the second output components316, the converter switches500-506, the first inductor508, the first capacitor510, the first fuse516, the third inductor518, the second capacitor520, and the second fuse526. The components of the power converter600ofFIG.8may be interconnected differently than the components of the power converter300ofFIG.5and the power converter600ofFIG.7. The input302is coupled to the first fuse516and the second fuse526, and is configured to be coupled to a power source. For example, where the power converter600ofFIG.8is an example of the DC/DC converter108, the input302may be coupled to the energy-storage-device interface110, which may be coupled to a power source. The first fuse516is coupled to the input302at a first connection, and is coupled to the first inductor508and the first capacitor510at a second connection. The first inductor508is coupled to the first capacitor510at a first connection, and is coupled to the first converter switch500and the second converter switch502at a second connection. The first capacitor510is coupled to the first inductor508and the first fuse516at a first connection, and is coupled to the second reference node530at a second connection. The second fuse526is coupled to the input302at a first connection, and is coupled to the third inductor518and the second capacitor520at a second connection. The third inductor518is coupled to the second fuse526and the second capacitor520at a first connection, and is coupled to the third converter switch504and the fourth converter switch506at a second connection. The first output304ais coupled to the first converter switch500and the third converter switch504, and is configured to be coupled to a load or a medium thereto. For example, the first output304amay be coupled to a positive-voltage DC bus, or another type of bus, of the DC busses106. The second output304bis coupled to the second converter switch502and the fourth converter switch506, and is configured to be coupled to a load or a medium thereto. For example, the second output304bmay be coupled to a negative-voltage, neutral, common, or other type of DC bus of the DC busses106. The first converter switch500is coupled to the first output304aat a first connection, is coupled to the first inductor508and the second converter switch502at a second connection, and is communicatively coupled to the controller112. The second converter switch502is coupled to the first inductor508and the first converter switch500at a first connection, is coupled to the second output304bat a second connection, and is communicatively coupled to the controller112at a control connection. The third converter switch504is coupled to the first output304aat a first connection, is coupled to the third inductor518and the fourth converter switch506at a second connection, and is communicatively coupled to the controller112at a control connection. The fourth converter switch506is coupled to the third converter switch504and the third inductor518at a first connection, is coupled to the second output304bat a second connection, and is communicatively coupled to the controller112at a control connection. As discussed above, the components of the power converter600ofFIG.8may be similar to the components of the power converter600ofFIG.7. However, in the power converter600ofFIG.8, the first output components312do not include the first relay514and the second inductor512, and the second output components316do not include the second relay524and the fourth inductor522. Furthermore, the controller112may control the converter switches500-506differently, such as by controlling the converter switches500-506to convert DC power received at the input302(for example, from the battery124via the energy-storage-device interface110) to converted DC power to provide to the output304. Accordingly, it is to be appreciated that various interleaved power converters are provided having multiple converter legs. Each converter leg may include one or more components, such as one or more filtering components, relays, fuses, and so forth. Because the components in each leg conduct current from only one converter leg, the components receive less current than, for example, had the components been implemented in a common configuration in which the components are coupled to all of the converter legs. Accordingly, a current rating of each of the components may be reduced. Using the power converter300ofFIG.5as an example, each of the converter legs306,308may conduct an RMS current of approximately116A. As used herein, “approximately116A” may include between115-117A in one example; between113-119A in another example; between110-120A in another example; or other ranges in other examples. Rather than implementing a single set of output components coupled to both converter legs306,308rated to receive approximately232A of current, the output components312,316may only need to be rated to receive approximately116A of current each. For example, the fuses516,526and relays514,524may be implemented with components having a current rating of160A, which enables the fuses516,526and relays514,524to be implemented as PCB-mounted components, whereas it may be difficult or impossible to acquire a PCB-mounted fuse or relay rated to receive232A of current. Additionally, in the case of the fuses516,526, a clearing energy of the fuses516,526may be reduced as compared to, for example, a common fuse. Continuing with the foregoing example of implementing the fuses516,526as fuses rated at160A, an I2t clearing energy of each fuse is 16 kA2/s. Conversely, a common fuse rated at315A exhibits an I2t clearing energy of 82 kA2/s. By implementing the fuses516,526with lower current ratings than a common fuse, the clearing energy is reduced significantly and thus poses less danger to damaging adjacent components in the event of fuse being blown. As discussed above, the inductors512,522may be inductively coupled together to share a ripple current between the converter legs306,308. Such current sharing may reduce a ripple current in the capacitors510,520. In other examples, it may be advantageous to omit the inductive coupling of inductors512,522to improve internal-resonance performance, which may vary based on design requirements. In other examples, it may be advantageous to omit the inductors510,520entirely. As discussed above, the power converters300,600may be implemented in connection with any of the converters104,108, and/or114. In some examples, the power converters300,600may be implemented in all of the converters104,108, and114. For example, the power converter300may be implemented in the DC/AC inverter114, and the power converter600may be implemented in the AC/DC converter104as a PFC, and in the DC/DC converter108. In other examples, fewer than all of the converters104,108, and114may be implemented in connection with the power converters300,600. In some examples, a single common component may be replaced with one or more parallel-connected components. For example, a single common fuse may be replaced with two or more hard-parallel-connected fuses, and/or a single common relay may be replaced with two or more hard-parallel-connected relays. However, it may be difficult or impossible to obtain UL certification with hard-parallel-connected components. For example, if one parallel-connected fuse is blown, the remaining parallel-connected fuse(s) may be subjected to an excessively high voltage spike that prevents UL certification. Examples discussed above may therefore be advantageous inasmuch as no hard paralleling is implemented. Moreover, current sharing is provided in some examples, and balancing between the converter legs306,308is achieved by the controller112detecting a current difference between the converter legs306,308and decreasing the current difference. Various controllers, such as the controller112, may execute various operations discussed above. Using data stored in associated memory and/or storage, the controller112also executes one or more instructions stored on one or more non-transitory computer-readable media that may result in manipulated data. In some examples, the controller112may include one or more processors or other types of controllers. In one example, the controller112is or includes at least one processor. In another example, the controller112performs at least a portion of the operations discussed above using an application-specific integrated circuit tailored to perform particular operations in addition to, or in lieu of, a general-purpose processor. As illustrated by these examples, examples in accordance with the present disclosure may perform the operations described herein using many specific combinations of hardware and software and the disclosure is not limited to any particular combination of hardware and software components. Examples of the disclosure may include a computer-program product configured to execute methods, processes, and/or operations discussed above. The computer-program product may be, or include, one or more controllers and/or processors configured to execute instructions to perform methods, processes, and/or operations discussed above. Having thus described several aspects of at least one embodiment, it is to be appreciated various alterations, modifications, and improvements will readily occur to those skilled in the art. Such alterations, modifications, and improvements are intended to be part of, and within the spirit and scope of, this disclosure. Accordingly, the foregoing description and drawings are by way of example only. | 55,354 |
11863075 | DETAILED DESCRIPTION OF THE EMBODIMENTS FIG.3shows a block diagram of the multi-phase boost converting apparatus of the present disclosure. A multi-phase boost converting apparatus10of the present disclosure includes a multi-phase boost converter102and a passive lossless snubber104, wherein the passive lossless snubber104includes a first resonant capacitor C1, a second resonant capacitor C2, an input-end first unidirectional conduction component D3, an input-end second unidirectional conduction component D4, an output-end first unidirectional conduction component D5, an output-end second unidirectional conduction component D6and a resonant inductor Ls. The multi-phase boost converter102includes a first inductor L1, a second inductor L2, a first transistor switch Q1, a second transistor switch Q2, a first diode D1, a second diode D2, a switch controller106, an output end108, an input end110, an input-end capacitor Cinand an output-end capacitor Co, while the first transistor switch Q1has a first parasitic capacitor Coss1and the second transistor switch Q2has a second parasitic capacitor Coss2, wherein the components mentioned above are electrically connected to each other. For the ease of the explanations, the present disclosure assumes that the above-mentioned components are ideal, and the forward bias voltage of the diodes is all zero volts. The input-end first unidirectional conduction component D3, the input-end second unidirectional conduction component D4, the output-end first unidirectional conduction component D5and the output-end second unidirectional conduction component D6are, for example but not limited to, diodes; the first transistor switch Q1and the second transistor switch Q2are, for example but not limited to, metal oxide semiconductor field effect transistors; the switch controller106is, for example but not limited to, a pulse width modulation signal controller. According to different load power requirements, the operation types of the multi-phase boost converting apparatus10of the present disclosure can be divided into a half-type operation and a full-type operation, wherein the half-type operation includes eight working states (namely, the half-type first working state to the half-type eighth working state) while the full-type operation also includes eight working states (namely, the full-type first working state to the full-type eighth working state). First, the half-type operation of the present disclosure is described in detail as follows: Please refer toFIG.3again; when the switch controller106is configured to transmit a pulse width modulation signal112to the first transistor switch Q1to drive the first transistor switch Q1and a duty cycle of the pulse width modulation signal112is less than 50%, or when the switch controller106is configured to transmit the pulse width modulation signal112to the second transistor switch Q2to drive the second transistor switch Q2and the duty cycle of the pulse width modulation signal112is less than 50%, the multi-phase boost converting apparatus10is configured to sequentially operate in a half-type first working state, a half-type second working state, a half-type third working state, a half-type fourth working state, a half-type fifth working state, a half-type sixth working state, a half-type seventh working state and a half-type eighth working state. FIG.1-1toFIG.1-8respectively show the multi-phase boost converting apparatus in the half-type first working state to the half-type eighth working state of the present disclosure, wherein the dashed arrows are the directions of the currents, and for the sake of brevity, some components and symbols which are shown inFIG.3are omitted inFIG.1-1toFIG.1-8, while the symbol ON next to the transistor switches means that the transistor switch is turned on, and the symbol OFF means that the transistor switch is turned off. Please refer toFIG.1-1andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type first working state, the switch controller106is configured to turn on the first transistor switch Q1and keep turning off the second transistor switch Q2, and the first inductor L1is configured to be excited by an input-end voltage Vinof the input end110to store a first electric energy in a first magnetic field form, and a first inductor current iL1flowing through the first inductor L1increases gradually, and the resonant inductor Lsand the first resonant capacitor C1are configured to be charged by the input-end voltage Vinand to resonate, and the input-end first unidirectional conduction component D3is configured to enable the resonant inductor Lsand the first resonant capacitor C1to be configured to resonate for a half cycle and then stop resonating, so that a first resonant capacitor voltage vC1of the first resonant capacitor C1is twice the input-end voltage Vin, and so that a resonant inductor current iLsflowing through the resonant inductor Lsis zero, and then the multi-phase boost converting apparatus10is configured to operate in the half-type second working state. Please refer toFIG.1-2andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type second working state, the switch controller106is configured to keep turning on the first transistor switch Q1and keep turning off the second transistor switch Q2, and the first inductor L1is configured to continue to be excited by the input-end voltage Vin, and the first inductor current iL1continues to increase, and then the multi-phase boost converting apparatus10is configured to operate in the half-type third working state. Please refer toFIG.1-3andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type third working state, the switch controller106is configured to turn off the first transistor switch Q1and keep turning off the second transistor switch Q2, and the first parasitic capacitor Coss1is configured to be charged by the first inductor current iL1from zero volts, so that a first drain-source voltage vds1of the first transistor switch Q1increases gradually, and the first resonant capacitor C1is configured to discharge, so that the output-end first unidirectional conduction component D5is configured to be forward-biased conducted, and the first resonant capacitor voltage vC1is discharged from twice the input-end voltage Vinto zero volts, and the first drain-source voltage Vds1plus the first resonant capacitor voltage vC1is equal to an output-end voltage Voof the output end108, and when the first resonant capacitor voltage vC1is discharged to zero volts, the first diode D1is configured to be forward-biased conducted, and then the multi-phase boost converting apparatus10is configured to operate in the half-type fourth working state. Please refer toFIG.1-4andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type fourth working state, the switch controller106is configured to keep turning off the first transistor switch Q1and keep turning off the second transistor switch Q2, and the first diode D1is configured to continue to be forward-biased conducted by the first inductor current iL1, and the resonant inductor current iLsis zero, and an input-end first unidirectional conduction current iD3flowing through the input-end first unidirectional conduction component D3is zero, and an input-end second unidirectional conduction current iD4flowing through the input-end second unidirectional conduction component D4is zero, and a first resonant capacitor current iC1flowing through the first resonant capacitor C1is zero, and a second resonant capacitor current iC2flowing through the second resonant capacitor C2is zero, and an output-end first unidirectional conduction current iD5flowing through the output-end first unidirectional conduction component D5is zero, and an output-end second unidirectional conduction current iD6flowing through the output-end second unidirectional conduction component D6is zero (namely, no current flows through the components of the passive lossless snubber104), and the first electric energy stored in the first magnetic field form by the first inductor L1is transmitted to the output end108in a current form, and the first inductor current iL1decreases gradually, and then the multi-phase boost converting apparatus10is configured to operate in the half-type fifth working state. Please refer toFIG.1-5andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type fifth working state, the switch controller106is configured to turn on the second transistor switch Q2and keep turning off the first transistor switch Q1, and the second inductor L2is configured to be excited by the input-end voltage Vinto store a second electric energy in a second magnetic field form, and a second inductor current iL2flowing through the second inductor L2increases gradually, and the resonant inductor Lsand the second resonant capacitor C2are configured to be charged by the input-end voltage Vinand to resonate, and the input-end second unidirectional conduction component D4is configured to enable the resonant inductor Lsand the second resonant capacitor C2to be configured to resonate for the half cycle and then stop resonating, so that a second resonant capacitor voltage vC2of the second resonant capacitor C2is twice the input-end voltage Vin, and so that the resonant inductor current iLsis zero, and then the multi-phase boost converting apparatus10is configured to operate in the half-type sixth working state. Please refer toFIG.1-6andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type sixth working state, the switch controller106is configured to keep turning on the second transistor switch Q2and keep turning off the first transistor switch Q1, and the second inductor L2is configured to continue to be excited by the input-end voltage Vin, and the second inductor current iL2continues to increase, and then the multi-phase boost converting apparatus10is configured to operate in the half-type seventh working state. Please refer toFIG.1-7andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type seventh working state, the switch controller106is configured to turn off the second transistor switch Q2and keep turning off the first transistor switch Q1, and the second parasitic capacitor Coss2is configured to be charged by the second inductor current iL2from zero volts, so that a second drain-source voltage vds2of the second transistor switch Q2increases gradually, and the second resonant capacitor C2is configured to discharge, so that the output-end second unidirectional conduction component D6is configured to be forward-biased conducted, and the second resonant capacitor voltage vC2is discharged from twice the input-end voltage Vinto zero volts, and the second drain-source voltage vds2plus the second resonant capacitor voltage vC2is equal to the output-end voltage Vo, and when the second resonant capacitor voltage vC2is discharged to zero volts, the second diode D2is configured to be forward-biased conducted, and then the multi-phase boost converting apparatus10is configured to operate in the half-type eighth working state. Please refer toFIG.1-8andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the half-type eighth working state, the switch controller106is configured to keep turning off the second transistor switch Q2and keep turning off the first transistor switch Q1, and the second diode D2is configured to continue to be forward-biased conducted by the second inductor current iL2, and the resonant inductor current iLsis zero, and the input-end first unidirectional conduction current iD3is zero, and the input-end second unidirectional conduction current iD4is zero, and the first resonant capacitor current iC1is zero, and the second resonant capacitor current iC2is zero, and the output-end first unidirectional conduction current ins is zero, and the output-end second unidirectional conduction current iD6is zero (namely, no current flows through the components of the passive lossless snubber104), and the second electric energy stored in the second magnetic field form by the second inductor L2is transmitted to the output end108in the current form, and the second inductor current iL2decreases gradually. FIG.4ashows a part of waveform diagrams of the multi-phase boost converting apparatus in the half-type first working state to the half-type eighth working state of the present disclosure.FIG.4bshows the other part of waveform diagrams of the multi-phase boost converting apparatus in the half-type first working state to the half-type eighth working state of the present disclosure. Please refer toFIG.1-1toFIG.1-8andFIG.3at the same time; besides the above-mentioned component symbols, the first transistor switch Q1has a first gate-source voltage vgs1, a current flowing through the first transistor switch Q1is called a first drain-source current ids1, the second transistor switch Q2has a second gate-source voltage vgs2, a current flowing through the second transistor switch Q2is called a second drain-source current ids2, a current flowing through the first diode D1is called a first diode current iD1, a current flowing through the second diode D2called a second diode current iD2, the first inductor L1has a first inductor voltage vL1, the second inductor L2has a second inductor voltage vL2, the resonant inductor Lshas a resonant inductor current vLs, a peak current of the first inductor current iL1is a first inductor peak current iL1_pk, a peak current of the second inductor current iL2is a second inductor peak current iL2_pk, a valley current of the first inductor current iL1is a first inductor valley current iL1_vly, a valley current of the second inductor current iL2is a second inductor valley current iL2_vly, the resonant inductor Lshas a resonant inductor inductive reactance XLs, the first resonant capacitor C1has a first resonant capacitor capacitive reactance XC1, the second resonant capacitor C2has a second resonant capacitor capacitive reactance XC2, the half-type first working state is between a zero timing point t0and a first timing point t1, the half-type second working state is between the first timing point t1and a second timing point t2, the half-type third working state is between the second timing point t2and a third timing point t3, the half-type fourth working state is between the third timing point t3and a fourth timing point t4, the half-type fifth working state is between the fourth timing point t4and a fifth timing point t5, the half-type sixth working state is between the fifth timing point t5and a sixth timing point t6, the half-type seventh working state is between the sixth timing point t6and a seventh timing point t7, and the half-type eighth working state is between the seventh timing point t7and the zero timing point t0. Then, the full-type operation of the present disclosure is described in detail as follows: Please refer toFIG.3again; when the switch controller106is configured to transmit the pulse width modulation signal112to the first transistor switch Q1to drive the first transistor switch Q1and the duty cycle of the pulse width modulation signal112is greater than or equal to 50%, or when the switch controller106is configured to transmit the pulse width modulation signal112to the second transistor switch Q2to drive the second transistor switch Q2and the duty cycle of the pulse width modulation signal112is greater than or equal to 50%, the multi-phase boost converting apparatus10is configured to sequentially operate in a full-type first working state, a full-type second working state, a full-type third working state, a full-type fourth working state, a full-type fifth working state, a full-type sixth working state, a full-type seventh working state and a full-type eighth working state. FIG.2-1toFIG.2-8respectively show the multi-phase boost converting apparatus in the full-type first working state to the full-type eighth working state of the present disclosure, wherein the dashed arrows are the directions of the currents, and for the sake of brevity, some components and symbols which are shown inFIG.3are omitted inFIG.2-1toFIG.2-8, while the symbol ON next to the transistor switches means that the transistor switch is turned on, and the symbol OFF means that the transistor switch is turned off. Please refer toFIG.2-1andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type first working state, the switch controller106is configured to turn on the first transistor switch Q1and keep turning on the second transistor switch Q2, and the first inductor L1is configured to be excited by an input-end voltage Vinof the input end110to store a first electric energy in a first magnetic field form, and the second inductor L2is configured to be excited by the input-end voltage Vinto store a second electric energy in a second magnetic field form, and a first inductor current iL1flowing through the first inductor L1increases gradually, and a second inductor current iL2flowing through the second inductor L2increases gradually, and the resonant inductor Lsand the first resonant capacitor C1are configured to be charged by the input-end voltage Vinand to resonate, and the first inductor L1is configured to be continuously excited by the input-end voltage Vinto cause the first inductor current iL1to continuously increase, and the second inductor L2is configured to be continuously excited by the input-end voltage Vinto cause the second inductor current iL2to continuously increase, and then the multi-phase boost converting apparatus10is configured to operate in the full-type second working state. Please refer toFIG.2-2andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type second working state, the switch controller106is configured to keep turning on the first transistor switch Q1and turn off the second transistor switch Q2, and the first inductor L1is configured to continue to be excited by the input-end voltage Vin, and the first inductor current iL1continues to increase, and the resonant inductor Lsand the first resonant capacitor C1are configured to be charged by the input-end voltage Vinand to resonate, and the second parasitic capacitor Coss2is configured to be charged by the second inductor current iL2from zero volts, so that a second drain-source voltage vds2of the second transistor switch Q2increases gradually, and the second resonant capacitor C2is configured to discharge, and the second drain-source voltage vds2plus a second resonant capacitor voltage vC2of the second resonant capacitor C2is equal to an output-end voltage Voof the output end108, and then the multi-phase boost converting apparatus10is configured to operate in the full-type third working state. Please refer toFIG.2-3andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type third working state, the switch controller106is configured to keep turning off the second transistor switch Q2and keep turning on the first transistor switch Q1, and the first inductor L1is configured to continue to be excited by the input-end voltage Vin, and the first inductor current iL1continues to increase, and the resonant inductor Lsand the first resonant capacitor C1are configured to be charged by the input-end voltage Vinand to resonate, and the input-end first unidirectional conduction component D3is configured to enable the resonant inductor Lsand the first resonant capacitor C1to be configured to resonate for a half cycle and then stop resonating, so that a first resonant capacitor voltage vC1of the first resonant capacitor C1is twice the input-end voltage Vin, and so that a resonant inductor current iLsflowing through the resonant inductor Lsis zero, and the second inductor current iL2discharges the second resonance capacitor C2, and when the second resonant capacitor voltage vC2is discharged from twice the input-end voltage Vinto zero volts, the second diode D2is configured to be forward-biased conducted, and the second drain-source voltage vds2plus the second resonant capacitor voltage vC2is equal to the output-end voltage Vo, and then the multi-phase boost converting apparatus10is configured to operate in the full-type fourth working state. Please refer toFIG.2-4andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type fourth working state, the switch controller106is configured to keep turning off the second transistor switch Q2and keep turning on the first transistor switch Q1, and the second diode D2is configured to continue to be forward-biased conducted by the second inductor current iL2, and the resonant inductor current iLsis zero, and an input-end first unidirectional conduction current iD3flowing through the input-end first unidirectional conduction component D3is zero, and an input-end second unidirectional conduction current iD4flowing through the input-end second unidirectional conduction component D4is zero, and a first resonant capacitor current iC1flowing through the first resonant capacitor C1is zero, and a second resonant capacitor current iC2flowing through the second resonant capacitor C2is zero, and an output-end first unidirectional conduction current iD5flowing through the output-end first unidirectional conduction component D5is zero, and an output-end second unidirectional conduction current iD6flowing through the output-end second unidirectional conduction component D6is zero (namely, no current flows through the components of the passive lossless snubber104), and the second electric energy stored in the second magnetic field form by the second inductor L2is transmitted to the output end108in a current form, and the second inductor current iL2decreases gradually, and the first inductor L1is configured to be excited by the input-end voltage Vin, and the first inductor current iL1increases, and then the multi-phase boost converting apparatus10is configured to operate in the full-type fifth working state. Please refer toFIG.2-5andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type fifth working state, the switch controller106is configured to turn on the second transistor switch Q2and keep turning on the first transistor switch Q1, and the first inductor L1is configured to be excited by the input-end voltage Vinto store the first electric energy in the first magnetic field form, and the second inductor L2is configured to be excited by the input-end voltage Vinto store the second electric energy in the second magnetic field form, and the first inductor current iL1increases gradually, and the second inductor current iL2increases gradually, and the resonant inductor Lsand the second resonant capacitor C2are configured to be charged by the input-end voltage Vinand to resonate, and then the multi-phase boost converting apparatus10is configured to operate in the full-type sixth working state. Please refer toFIG.2-6andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type sixth working state, the switch controller106is configured to keep turning on the second transistor switch Q2and turn off the first transistor switch Q1, and the second inductor L2is configured to continue to be excited by the input-end voltage Vin, and the second inductor current iL2continues to increase, and the first parasitic capacitor Coss1is configured to be charged by the first inductor current iL1from zero volts, so that a first drain-source voltage vds1of the first transistor switch Q1increases gradually, and the first resonant capacitor C1is configured to discharge, and the first drain-source voltage vds1plus the first resonant capacitor voltage vC1is equal to the output-end voltage Vo, and then the multi-phase boost converting apparatus10is configured to operate in the full-type seventh working state. Please refer toFIG.2-7andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type seventh working state, the switch controller106is configured to keep turning on the second transistor switch Q2and keep turning off the first transistor switch Q1, and the second inductor L2is configured to continue to be excited by the input-end voltage Vin, and the second inductor current iL2continues to increase, and the resonant inductor Lsand the second resonant capacitor C2are configured to be charged by the input-end voltage Vinand to resonate, and the input-end second unidirectional conduction component D4is configured to enable the resonant inductor Lsand the second resonant capacitor C2to be configured to resonate for the half cycle and then stop resonating, so that the second resonant capacitor voltage vC2is twice the input-end voltage Vin, and so that the resonant inductor current iLsis zero, and the first inductor current iL1discharges the first resonance capacitor C1, and when the first resonant capacitor voltage vC1is discharged from twice the input-end voltage Vinto zero volts, the first diode D1is configured to be forward-biased conducted, and then the multi-phase boost converting apparatus10is configured to operate in the full-type eighth working state. Please refer toFIG.2-8andFIG.3at the same time; when the multi-phase boost converting apparatus10is configured to operate in the full-type eighth working state, the switch controller106is configured to keep turning on the second transistor switch Q2and keep turning off the first transistor switch Q1, and the first diode D1is configured to continue to be forward-biased conducted by the first inductor current iL1, and the first inductor current iL1is transmitted to the output end108to be demagnetized, and the first inductor current iL1decreases gradually, and the second inductor L2is configured to be continuously excited by the input-end voltage Vinto cause the second inductor current iL2to continuously increase, and the resonant inductor current iLsis zero, and the input-end first unidirectional conduction current iD3is zero, and the input-end second unidirectional conduction current iD4is zero, and the first resonant capacitor current iC1is zero, and the second resonant capacitor current iC2is zero, and the output-end first unidirectional conduction current iD5is zero, and the output-end second unidirectional conduction current iD6is zero (namely, no current flows through the components of the passive lossless snubber104). FIG.5ashows a part of waveform diagrams of the multi-phase boost converting apparatus in the full-type first working state to the full-type eighth working state of the present disclosure.FIG.5bshows the other part of waveform diagrams of the multi-phase boost converting apparatus in the full-type first working state to the full-type eighth working state of the present disclosure. The full-type first working state is between a zero timing point t0and a first timing point t1, the full-type second working state is between the first timing point t1and a second timing point t2, the full-type third working state is between the second timing point t2and a third timing point t3, the full-type fourth working state is between the third timing point t3and a fourth timing point t4, the full-type fifth working state is between the fourth timing point t4and a fifth timing point t5, the full-type sixth working state is between the fifth timing point t5and a sixth timing point t6, the full-type seventh working state is between the sixth timing point t6and a seventh timing point t7, and the full-type eighth working state is between the seventh timing point t7and the zero timing point t0. The advantage of the present disclosure is to use a snubber with a simple structure to reduce the switching loss of the multi-phase boost converter and to reduce the electromagnetic interference. The present disclosure can absorb spikes and slow down the rising slope of the switch cross voltage after the switch of the multi-phase boost converter is turned off, so as to reduce the electromagnetic interference emission intensity caused by the high voltage slope, so as to reduce the switching loss when the switch is turned off (namely, the overlap area of the switch voltage and the switch current on the voltage and current waveforms). The passive lossless snubber104of the present disclosure only includes four diodes, one inductor and two capacitors to achieve the advantage of the present disclosure mentioned above, and the first resonant capacitor C1, the second resonant capacitor C2, the input-end first unidirectional conduction component D3, the input-end second unidirectional conduction component D4, the output-end first unidirectional conduction component D5, the output-end second unidirectional conduction component D6and the resonant inductor Lsincluded in the passive lossless snubber104do not participate in the processing of the main power, nor are they in the power transmission path, so that the passive lossless snubber104only needs a very low component power rating; therefore, the present disclosure can reduce the component volume and the additional cost. According to the experimental data, under the same peripheral component parameters and full load efficiency, compared with the traditional RCD snubber, the present disclosure can reduce the switching loss and the electromagnetic interference greatly. | 29,972 |
11863076 | DETAILED DESCRIPTION OF THE INVENTION To make the purpose, technical solutions and advantages of the present invention more clear and understandable, the present invention is described in detail below through embodiments in combination with accompanying drawings. It shall be understood that the specific embodiments described herein are merely used to interpret the present invention rather than limiting the present invention. First Embodiment An embodiment of the present invention provides a switching frequency adjustment method for an isolated converter, which adjusts a switching frequency according to a load so as to reduce the switching loss. A specific adjustment method is shown in a flow chart ofFIG.5, which includes the following steps: Step1: the isolated converter is started, and an initial switching frequency is set to a first frequency value (such as 55 kHz); Step2: a load percentage is detected, the load percentage is compared with a preset threshold (such as 50% of rated power), and in the present invention, the load percentage is preferably load power; and Step3: the switching frequency of the isolated converter is adjusted based on a comparison result between the load percentage and the preset threshold, wherein when the load percentage is not greater than the preset threshold, the switching frequency increases to a first frequency value, and when the load percentage is greater than the preset threshold and in a stable state, the switching frequency decreases to a second frequency value. In the present invention, the stable state refers to that the load percentage is kept at a constant value or basically kept at the constant value (for example, floating between 90% and 110% of the constant value) within a specific period of time (such as 2 s). However, the limitation to the stable state is only exemplary and not restrictive. Specifically, whether the load percentage is greater than the preset threshold or not is determined. If no, whether the current switching frequency is less than the first frequency value or not is further determined; if the switching frequency is less than the first frequency value, the switching frequency increases to the first frequency value; if the switching frequency is not less than the first frequency value (generally equal to the first frequency value), the switching frequency is not adjusted, and the load percentage is detected continuously; and that is, if the load percentage is not greater than the preset threshold, a higher switching frequency (the first frequency value) is kept so as to prevent the magnetic saturation of the transformer of the isolated converter, that is, kept within a maximum allowable magnetic induction intensity of the transformer. If yes, whether the load percentage is in the stable state is further determined; if the load percentage is unstable, the load percentage is detected continuously; if the load percentage is in the stable state, whether the current switching frequency is greater than the second frequency value (such as 40 kHz) is further determined; if the switching frequency is greater than the second frequency value, the switching frequency decreases to the second frequency value; if the switching frequency is not greater than the second frequency value (generally equal to the second frequency value), the switching frequency is not adjusted, and the load percentage is detected continuously; that is, if the load percentage is greater than the preset threshold and in the stable state, a lower switching frequency (the second frequency value) is kept, so that the temperature of the isolated converter is not greater than the maximum allowable temperature; and the maximum temperature is the maximum temperature that elements of the isolated converter can bear. In the present invention, the higher first frequency value and the lower second frequency value are selected according to specific product models and application scenarios, preferably in a range of 1-500 kHz. In the present embodiment, in order to avoid the problem that an output load just falls on a load switching point (for example, 50% of the rated load), which causes the system to repeatedly adjust the switching frequency, a hysteresis loop of load switching can be added to solve the problem. For example, a hysteresis loop of 10% of the rated load is set: When the load percentage is less than 45%, a higher switching frequency is adopted. When the load percentage is 45%-55%, the current switching frequency is kept unchanged. When the load percentage is greater than 55%, the switching frequency decreases gradually to the lower switching frequency. That is, the aforementioned preset threshold is set to a preset threshold range; when the load percentage is less than a minimum value of the preset threshold range, the higher switching frequency is adopted; when the load percentage is greater than a maximum value of the preset threshold range, the lower switching frequency is adopted; and when the load percentage is just within the preset threshold range, the original switching frequency is kept unchanged, and the switching frequency is not adjusted. Furthermore, see the change of a transformer excitation curve when the switching frequency increases as shown inFIG.6and the change of a transformer excitation curve when the switching frequency decreases as shown inFIG.7. When the switching frequency increases, the transformer excitation curve is changed from a curve1to a curve2inFIG.6; the excitation curve shrinks, which may not cause the transformer saturation, so that the switching frequency may increase rapidly, for example, the current frequency hops to the first frequency value (that is, the switching frequency changes directly from the current frequency to the first frequency value). When the switching frequency decreases, the transformer excitation curve is changed from the curve1to a curve3inFIG.7; due to the expansion of the excitation curve, if the sudden change of the frequency may cause the saturation of the transformer, preferably, the switching frequency decreases gradually; and for example, the switching frequency decreases gradually at a specific interval, so that the excitation curve changes smoothly, and the saturation of the transformer is avoided. The specific interval may be a fixed value, for example, the switching frequency may decrease gradually to the second frequency value at an interval of 1 kHz; and the specific interval may also be a variable value, for example, the switching frequency may decrease gradually to the second frequency value randomly at 1 kHz, 2 kHz and 3 kHz. Second Embodiment The second embodiment provides a switching frequency adjustment device for an isolated converter. Referring toFIG.8, the switching frequency adjustment device includes a detection module1, a comparison module2and an adjustment module3; the detection module1is used to detect a load percentage and transmit a detection result to the comparison module2; the detection module1may be a detection module arranged in the isolated converter, and may also be a detection module arranged outside the isolated converter; a preset threshold is pre-stored in the comparison module2, and the comparison module compares the received load percentage with the preset threshold and transmits a comparison result to the adjustment module3; the adjustment module3adjusts a switching frequency of a switching transistor S1or S2based on the received comparison result, and the switching frequency of the switching transistor S1or S2may be fed back to the adjustment module3; when the load percentage is not greater than the preset threshold, the switching frequency increases to a first frequency value; and when the load percentage is greater than the preset threshold and in the stable state, the switching frequency decreases to a second frequency value. Specifically, the adjustment module determines whether the load percentage is greater than the preset threshold or not. If no, whether the current switching frequency is less than the first frequency value or not is further determined; if the switching frequency is less than the first frequency value, the switching frequency increases to the first frequency value; if the switching frequency is not less than the first frequency value (generally equal to the first frequency value), the switching frequency is not adjusted, and the load percentage is detected continuously; and that is, if the load percentage is not greater than the preset threshold, a higher switching frequency (the first frequency value) is kept. If yes, whether the load percentage is in the stable state is further determined; if the load percentage is unstable, the load percentage is detected continuously; if the load percentage is in the stable state, whether the current switching frequency is greater than the second frequency value (such as 40 kHz) is further determined; if yes, the switching frequency decreases to the second frequency value; if the switching frequency is not greater than the second frequency value (generally equal to the second frequency value), the switching frequency is not adjusted, and the load percentage is detected continuously; and that is, if the load percentage is greater than the preset threshold and in the stable state, a lower switching frequency (the second frequency value) is kept. Preferably, a preset threshold range is pre-stored in the comparison module2; when the load percentage is less than a minimum value of the preset threshold range, the higher switching frequency is adopted; when the load percentage is greater than a maximum value of the preset threshold range, the lower switching frequency is adopted; and when the load percentage is just within the preset threshold range, the original switching frequency is kept unchanged, and the switching frequency is not adjusted. Preferably, the adjustment module3reduces the switching frequency of the switching transistor S1or S2gradually at a specific interval. A specific example of the switching frequency adjustment device for the isolated converter is given below. First Example See a structural schematic diagram of a circuit of the first example of the switching frequency adjustment device of the present invention shown inFIG.9. A micro-controller MCU receives a comparison result between a load percentage and a preset threshold and outputs a digital voltage signal through a digital interface IO based on the comparison result; the digital voltage signal controls an on-off state (including an on state, an amplification state and an off state) of a triode Q1through a time delay circuit20and then adjusts an external resistor R4and an external resistor R5arranged outside an oscillator of a pulse width modulation chip (PWM IC) and further adjusts an oscillation frequency of the PWM IC; the oscillation frequency corresponds to an output frequency of the PWM IC; the PWM IC is preferably connected to the switching transistors S1and S2through a driving circuit so as to adjust the switching frequency thereof; and the switching frequency is consistent with the output frequency of the PWM IC. Specifically, the time delay circuit20includes resistors R1and R2connected in series, a diode D1and a resistor R3connected in series, and a capacitor C1; a branch formed by the resistors R1and R2is connected in parallel with a branch formed by the diode D1and the resistor R3; a node between the resistor R1and the diode D1is connected to an IO interface of the MCU; a node between the resistors R1and R2is grounded through the capacitor C1; a node between the resistors R2and R3is connected to a base electrode of the triode Q1; an emitting electrode of the triode Q1is grounded, and a collecting electrode is connected to an RT pin of the PWM IC through a resistor R4; and a CT pin of the PWM IC is grounded through a capacitor C6; and a node between the resistor R4and the RT pin is grounded through a resistor R5. A specific control process of the switching frequency adjustment device of the example is as follows: (1) When an JO output signal of the MCU changes from low level to high level, the D1is turned on, the high-level voltage is provided to the base electrode of the Q1through the D1, the Q1is saturated and turned on, and the R4and the R5are equivalent to parallel connection. At the time, because the RT pin of the PWM IC has a small resistance value, the output frequency of the PWM IC is relatively high. At the same time, the high level outputted by the MCU charges the C1through the R1, and the voltage of the C1increases gradually from low level to high level so as to prepare for the output of the MCT changing from high level to low level. (2) When the IO output signal of the MCU changes from high level to low level, the high level of the C1is discharged through the R1, the C1changes gradually from high level to low level, and the Q1passes through an amplification region from saturation turn-on and changes gradually to an off region. At the time, because the RT pin of the PWM IC has a large resistance value that is R5, the output frequency of the PWM IC is relatively low. In this process, the time when the output frequency of the PWM IC changes gradually from high level to low level is decided by the discharging time of the C1through the R1, that is, the switching frequency decreases gradually by a charging/discharging curve of an RC circuit, such as the charging/discharging curve of the RC circuit shown inFIG.10. It may be seen from the example that: (1) When the level of the digital voltage signal of the MCU changes, the output frequency of the PWM IC changes synchronously. (2) By adjusting parameters of the time delay circuit, especially the R1and the C1, the changing speed of the voltage of the C1may be changed, thereby changing the working state of the triode Q1. If the collecting electrode (C electrode) and emitting electrode (E electrode) of the triode are equivalent to a resistor, the resistance value of the equivalent resistor changes with the voltage of the C1. From the perspective of the PWM IC, the resistance value of a parallel resistor of an oscillation resistor connected with the PWM IC changes with the voltage of the C1, and finally, the output frequency of the PWM IC changes with the voltage of the C1. By adjusting the parameters of the R1and C1, the changing speed of the voltage of the C1may be changed, so that the changing speed of the output frequency of the PWM IC may be changed, thereby realizing the rapid increase and gradual decrease of the output frequency of the PWM IC. 3) By adjusting the resistance value of the R4, the resistance value of the RT pin connected with the PWM IC may be changed, so that the changing range of the output frequency of the PWM IC may be adjusted. Those skilled in the art can understand that in the example, the comparison module may be contained in the micro-controller MCU, which receives the load percentage information and compares the load percentage with the preset threshold. Second Example See a structural schematic diagram of a circuit of the second example of the switching frequency adjustment device of the present invention shown inFIG.11. Based on a comparison result between a load percentage and a preset threshold, an output of the MCU outputs a variable analog voltage signal (a DA output) through an analog signal module (a DA module); preferably, the driving capacity is enhanced by the buffering of a follower Op amp, and then an on-off state (including an on state, an amplification state and an off state) of a triode Q1is controlled by a resistor R1, then external resistors R4and R5outside an oscillator of a pulse width modulation chip PWM IC are adjusted, and then an oscillation frequency of the PWM IC is adjusted; the oscillation frequency corresponds to an output frequency of the PWM IC; the PWM IC is preferably connected to switching transistors S1and S2through a driving circuit so as to adjust the switching frequency thereof; and the switching frequency is consistent with the output frequency of the PWM IC. The analog signal module may be contained in the MCU, and may also be arranged outside the MCU. Specifically, in the example, the analog signal module (the DA module), a software control unit for controlling an analog signal, a preferred follower and the resistor R1form a time delay circuit; and the changing speed of an output signal of the DA module is controlled by the software control unit, thereby realizing the rapid increase and gradual decrease of the output frequency of the PWM IC. Furthermore, in the example, the following circuit of the resistor R1is consistent with that of the first example, that is, an emitting electrode of the triode Q1is grounded, a collecting electrode is connected to an RT pin of the PWM IC through the resistor R4, a CT pin of the PWM IC is grounded through a capacitor C6, and a node between the resistor R4and the RT pin is grounded through the resistor R5. A specific control process of the switching frequency adjustment device of the example is as follows: (1) When a DA output signal of the MCU changes from low level to high level rapidly, the Q1is saturated and turned on, and the R4and the R5are equivalent to parallel connection. At the time, because the RT pin of the PWM IC has a small resistance value, the output frequency of the PWM IC is relatively high. (2) When the DA output signal of the MCU changes gradually from high level to low level, the Q1passes through an amplification region from saturation turn-on and gradually changes to an off region. At the time, because the RT pin of the PWM IC has a large resistance value that is R5, the output frequency of the PWM IC is relatively low. In this process, the changing speed of the output frequency of the PWM IC gradually from high level to low level depends on the changing speed of the DA signal controlled by the software unit. It may be seen from the example that: (1) When the level of the DA output of the MCU changes, the output frequency of the PWM IC changes synchronously. (2) By adjusting the time delay circuit, the rapid increase and gradual decrease of the output frequency of the PWM IC can be realized. 3) By adjusting the resistance value of the R4, the resistance value of the RT pin connected with the PWM IC may be changed, so that the changing range of the output frequency of the PWM IC may be adjusted. The time delay circuit of the present invention is not limited to the circuit structure in the example and may have other variations, as long as the rapid increase and gradual decrease of the oscillation frequency of the PWM IC can be realized. Besides the above two examples in which the switching frequency is adjusted by hardware circuits, a digital control chip may also be adopted to realize the adjustment of the switching frequency of the S1and S2, for example, the switching frequency may be adjusted by changing a periodic register value of a PWM generation module in the MCU. 1) When the switching frequency needs to increase rapidly, the periodic register value of the PWM generation module is directly changed to a target value. 2) When the switching frequency needs to decrease gradually, the periodic register value of the PWM generation module can be changed gradually to the target value by a loop instruction. The changing range of the switching frequency may be changed by changing the target value of the periodic register. The changing speed of the switching frequency may be changed by changing the circulation times and the changing percentage of the periodic register value. To reflect the effect of the present invention, a product Callisto HV RT 3 kVA of EATON Company was tested by the inventor. By adopting the method of the prior art, in order to prevent the saturation of the transformer, the switching frequency was set to 55 kHz, a full-load battery was discharged to cause temperature rise, and the temperature of MOSFET rose to be greater than 140° C. and was still unstable, so the test can only be stopped by shutting down, and the predicted maximum temperature may reach 155° C. By adopting the solution of the present invention for test again, the maximum temperature of the MOSFET was less than 110° C., so that the temperature rise problem of the transformer can be solved effectively. Although the present invention is already described by preferred embodiments, the present invention is not limited to the embodiments described herein, but also includes various changes and variations made without departing from the scope of the present invention. | 20,736 |
11863077 | DETAILED DESCRIPTION This description relates generally to electronic circuits, and more particularly to a power supply system with active clamping. The power supply system includes an input stage, an output stage, a transformer interconnecting the input and output stages, and a switching controller. The input stage includes a set of input switches that are alternately activated to provide a primary current through a primary winding of the transformer. As described herein, the term “activate”, as describing a transistor, refers to providing sufficient bias (e.g., gate-source voltage for a field-effect transistor (FET)) to operate the transistor device in saturation mode. Similarly, the term “deactivate”, as describing a transistor, refers to removing bias to operate the transistor device in cutoff mode. As an example, the input stage can be arranged as a full-bridge input stage that includes a set of input four switches, with alternate pairs of the input switches being activated to provide the primary current through the primary winding of the transformer. The output stage includes a set of output switches that are alternately activated to rectify a secondary current that is induced in the secondary winding of the transformer and is provided to an output to provide an output voltage to a load. As an example, the output stage can be arranged as a full-bridge output stage that includes a set of four output switches, with alternate pairs of the output switches being activated to rectify the secondary current provided from the secondary winding of the transformer. The input and output switches are activated by input switching signals and output switching signals, respectively, that are provided from the switching controller. The output stage also includes an active clamping circuit. As an example, the active clamping circuit includes a clamping switch and a capacitor arranged in series across the output. The clamping switch can be activated responsive to a clamp switching signal to generate a clamping current that can mitigate ringing in the output stage, thereby mitigating energy inefficiencies caused by voltages being exhibited across the output switches of the output stage. Also, the clamping switch can be activated during a dead-time between activation of a first input switch (or first pair of input switches) and a second input switch (or second pair of input switches) of the input stage to enable zero-volt switching (ZVS) on the input switches. As described herein, ZVS is defined as activating a respective one of the switches (e.g., input or output switches) with a drain-source voltage VDSof approximately zero volts (e.g., +/− approximately 5%). As a first example, the power supply system can operate in continuous conduction mode (CCM), so the clamping switch can be activated at a predetermined time that is approximately half a ringing period of the primary current after deactivation of each of the input switches to facilitate ZVS. As a second example, the power supply system can operate in discontinuous conduction mode (DCM), so the clamping switch can be activated at a predetermined time that is prior to the activation of each of the input switches, along with concurrent activation of one of the output switches (or one of the pairs of output switches) to facilitate ZVS. As a result, the power supply system can operate more efficiently by mitigating ringing in the output stage and by implementing ZVS in the input stage. FIG.1is a block diagram of a power supply system100. The power supply system100can be configured to generate an output voltage VOUTbased on an input voltage VIN. The power supply system100can be implemented in any of a variety of direct current (DC) power-providing applications. The power supply system100includes an input stage102, an output stage104, a transformer106, and a switching controller108. The input stage102includes a set of input switches that are alternately activated responsive to a respective set of input switching signals Sir to provide a primary current IPRIthrough a primary winding LPRIof the transformer106. As an example, the input stage102can be arranged as a full-bridge input stage that includes a set of input four switches, with alternate pairs of the input switches being activated to provide the primary current IPRIthrough the primary winding LPRIof the transformer106. The output stage104includes a set of output switches that are alternately activated by a respective set of output switching signals SOUTto rectify a secondary current ISECthat is induced in the secondary winding LSECof the transformer106and is provided to an output to provide the output voltage VOUTto a load. As an example, the output stage104can be arranged as a full-bridge output stage that includes a set of four output switches, with alternate pairs of the output switches being activated to rectify the secondary current ISECprovided from the secondary winding LSECof the transformer106. In the example ofFIG.1, the output stage104further comprises an active clamping circuit110(“ACTIVE CLAMP”). The active clamping circuit110can include a clamping switch (e.g., transistor) and a capacitor in series across the output stage104. The active clamping circuit110, responsive to activation of the clamping switch by a clamp switching signal SAC, can conduct a clamping current to the output of the output stage104to mitigate ringing of the secondary current ISECin the output stage104. As an example, the ringing can occur based on resonant oscillation between the parasitic capacitance of the output switches in the output stage104and an output inductor (e.g., and a resonant inductor of the transformer106, as well). The ringing can thus be exhibited during a dead-time between activation of the input switches in the input stage102, thereby providing power inefficiencies based on a drain-source voltage VDSbeing exhibited on the output switches in the output stage104. Thus, the clamping current can include a portion of the ringing secondary current ISECto clamp the drain-source voltage VDS, thereby providing for a more energy efficient operation of the power supply system100. As an example, the switching controller108can be arranged in or as part of an integrated circuit (IC). The switching controller108is configured to generate the input switching signals Sir and the output switching signals SOUT. In the example ofFIG.1, the switching controller108receives the output voltage VOUTas an input, so the switching controller108can control the activation of the input switches in the input stage102based on the input switching signals Si and the activation of the output switches in the output stage104based on the output switching signals SOUTin response to the output voltage VOUT(e.g., in a pulse-width modulation (PWM) scheme). Also, the switching controller108is configured to provide the clamp switching signal SACto the clamping switch of the active clamping circuit110. The switching controller108is configured to activate the clamping switch of the active clamping circuit110during a dead-time between activation of a first input switch (or first pair of input switches) and a second input switch (or second pair of input switches) of the input stage102to facilitate zero-volt switching (ZVS). As a first example, the power supply system100can operate in continuous conduction mode (CCM). Thus, to facilitate ZVS, the clamping switch of the active clamping circuit110can be activated at a predetermined time that is approximately half a ringing period of the primary current IPRIafter deactivation of each of the input switches of the input stage102. In the example ofFIG.1, the primary current IPRIis provided to the switching controller108, such as to achieve peak current mode control or to use as an indication of an operating mode of the power supply system100(e.g., in CCM or in discontinuous conduction mode (DCM)). As a second example, the power supply system can operate in DCM, so the clamping switch can be activated at a predetermined time that is prior to the activation of each of the input switches, along with concurrent activation of one of the output switches (or one of the pairs of output switches) in the output stage104to facilitate ZVS. As a result, the power supply system can operate more efficiently by mitigating ringing in the output stage104and, based on activating the clamping switch during the activation dead-time of the input switches of the input stage102, by providing ZVS of the input switches in the input stage102. FIG.2is a block diagram of a power supply system200. The power supply system200can be configured to generate an output voltage VOUTbased on an input voltage VIN. The power supply system200can be used to implement the power supply system100in the example ofFIG.1. Therefore, reference is to be made to the example ofFIG.1in the following description of the example ofFIG.2. The power supply system200includes an input stage202and a transformer204. In the example ofFIG.2, the transformer204includes a primary winding LPRIand a secondary winding LSECthat are inductively coupled, and includes a resonant inductor LRESthat can represent a resonant inductance of the primary winding LPRI. The input stage202includes a first input switch N1, a second input switch N2, a third input switch N3, and a fourth input switch N4that are formed as a full-bridge. In the example ofFIG.2, the input switches N1, N2, N3, and N4are demonstrated as N-channel metal-oxide semiconductor field-effect transistors (MOSFETS). The first input switch N1is arranged as a first high-side switch that interconnects the input voltage VINand a first terminal206that is coupled to the resonant inductor LRESof the transformer204. The second input switch N2is arranged as a first low-side switch that interconnects a low voltage rail (e.g., a ground terminal) and the first terminal206. The third input switch N3is arranged as a second high-side switch that interconnects the input voltage VINand a second terminal208that is coupled to the primary winding LPRIof the transformer204. The fourth input switch N4is arranged as a second low-side switch that interconnects the low voltage rail and the second terminal208. In the example ofFIG.2, the terminal206has a voltage VAand the second terminal208has a voltage VB, so the primary winding LPRIand the resonant inductor LREShave a voltage VAB. The first and fourth input switches N1and N4and the second and third input switches N2and N3are alternately activated to provide the primary current IPRIin opposing polarities through the primary winding LPRIof the transformer204. During a first time duration, the first input switch N1is activated responsive to a first input switching signal SIN1and the fourth input switch N4is activated responsive to a fourth input switching signal SIN4. Therefore, during the first time duration, the primary current IPRIis provided from the input voltage VIN, through the first input switch NI, through the resonant inductor LRES, through the primary winding LPRIin a first polarity, through the fourth input switch N4, to the low-voltage rail. As an example, the input switching signals SIN1and SIN4can be staggered to provide for a staggered activation of the respective input switches N1and N4to control the primary current IPRIthrough the primary winding LPRI. During a second time duration, the second input switch N2is activated responsive to second input switching signal SIN2and the third input switch N3is activated responsive to third input switching signal SIN3. Therefore, during the second time duration, the primary current IPRIis provided from the input voltage VIN, through the third input switch N3, through the primary winding LPRIin a second polarity opposite the first polarity, through the resonant inductor LRES, through the second input switch N2, to the low-voltage rail. As an example, the input switching signals SIN1and SIN4can be staggered to provide for a staggered activation of the respective input switches N1and N4to control the primary current IPRIthrough the primary winding LPRI. As an example, the input switching signals SIN1, SIN2, SIN3, and SIN4can be provided from a switching controller (e.g., the switching controller108). The first input switching signal SIN1and the second input switching signal SIN2can be separated by a switching dead-time during which neither of the respective input switches N1and N2are activated. Similarly, the third input switching signal SIN3and the fourth input switching signal SIN4can be separated by a switching dead-time during which neither of the respective input switches N3and N4are activated The power supply system200also includes an output stage210. The output stage210includes a first output switch N5, a second output switch N6, a third output switch N7, and a fourth output switch N8that are formed as a full-bridge rectifier. In the example ofFIG.2, the output switches N5, N6, N7, and N8are demonstrated as N-FETs. The first output switch N5interconnects a terminal212and a terminal214that is coupled to the secondary winding LSECof the transformer204. The terminal212is also coupled to an output inductor LOUTthat is configured to conduct an output current IOUT. The second output switch N6interconnects the terminal214and a low-voltage rail (e.g., a ground terminal). The third output switch N7interconnects the terminal212and a terminal216that is coupled to the secondary winding LSECof the transformer204. The fourth output switch N8interconnects the terminal216and the low voltage rail. The first and fourth output switches N5and N8and the second and third output switches N6and N7are alternately activated to conduct the secondary current ISECfrom the secondary winding LSECto the output inductor LOUT. During a first time duration, the first output switch N5and the fourth output switch N8are activated concurrently responsive to first output switching signal SOUT1. Therefore, during the first time duration, the secondary current ISECis provided as a first rectifier current ISR1from the low voltage rail, through the fourth output switch N8, through the secondary winding LSECin a first polarity, and through the first output switch N5to the terminal212. During a second time duration, the third output switch N7and the second output switch N6are activated concurrently responsive to second output switching signal SOUT2. As an example, the output switching signals SOUT1and SOUT2can be provided from a switching controller (e.g., the switching controller108). Therefore, during the second time duration, the secondary current ISECis provided as a second rectifier current ISR2from the low voltage rail, through the second output switch N6, through the secondary winding LSECin a second polarity opposite the first polarity, through the third output switch N7, to the terminal212. As an example, the first and second time durations of the output stage210can approximately coincide with first and second time durations of the input stage202, respectively. The secondary current ISECcan thus be provided through the output inductor LOUTto provide an output voltage VOUTacross an output capacitor COUT. The output voltage VOUTcan thus power a load (not shown). In the example ofFIG.2, the output stage210further comprises an active clamping circuit218that includes a clamping switch N9and a capacitor CACin series between the terminal212and the low-voltage rail. The active clamping circuit218, in response to activation of the clamping switch N9by a clamp switching signal SAC, can conduct a clamping current ICLto the terminal212to mitigate ringing of the secondary current ISECin the output stage210. As an example, the ringing can occur based on resonant oscillation between the parasitic capacitance of the output switches N5, N6, N7, and N8and the output inductor LOUTin combination with the resonant inductor LRESin the transformer204. The ringing can thus be exhibited during a dead-time between activation of the input switches N1and N2, thereby providing power inefficiencies based on a drain-source voltage VDSbeing exhibited on the output switches N5, N6, N7, and N8in the output stage210. Thus, the clamping current ICLcan include a portion of the ringing secondary current ISEC(e.g., the active clamping circuit218can provide a current path for a portion of the ringing secondary current ISEC) to clamp the drain-source voltage VDS, thereby providing for a more energy efficient operation of the power supply system200. Also, as described in greater detail herein, the clamping switch N9can be activated during the dead-time between activation of the pair of the input switches N1and N2to facilitate ZVS of the input switches N1and N2. The operation of the power supply system200can be based on load conditions. As an example, the power supply system200can operate in CCM based on providing the output voltage VOUTto a heavier load. Thus, to facilitate ZVS, the clamping switch of the active clamping circuit218can be activated at a predetermined time that is approximately half a ringing period of the primary current IPRIafter deactivation of the input switches (e.g., the input switches N1and N2) of the input stage202. FIG.3is a timing diagram300. The timing diagram300demonstrates a number of the signals, currents, and voltages associated with the power supply system200in the example ofFIG.2operating in CCM. Therefore, reference is to be made to the example ofFIG.2in the following description of the example ofFIG.3. The timing diagram300includes the clamping current ICL, the output current IOUT, the primary current IPRI, the first rectifier current ISR1through the output switches N5and N8, a second rectifier current ISR2through the output switches N6and N7, the input switching signals SIN1and SIN2, and the voltage VAS across the primary winding LPRIof the transformer204. Prior to a time to, the input switching signal SIN1was asserted (e.g., logic high) to activate the input switch N1, while the input switching signal SIN2is de-asserted (e.g., logic low) to deactivate the input switch N2. Therefore, the primary current IPRIflows through the input switch N1, through the primary winding LPRI, and through the input switch N4(e.g., shortly after activation of the input switch N1) Thus, the primary current IPRIis demonstrated as being provided in the first polarity (e.g., less than −4 amps in the example ofFIG.3). The voltage VABis thus demonstrated as having an amplitude of approximately zero, as resulting from an approximate equal amplitude of both the voltages VAand VB. Concurrently, the secondary current ISECis induced in the secondary winding LSECfrom the primary winding LPRI, and is therefore provided as the second rectifier current ISR2, and thus the output current IOUT(decreasing in amplitude as it is provided to the load). At a time t1, the input switching signal SIN1is de-asserted (e.g., logic low) to deactivate the input switch N1. Also at the time t1, the input switching signal SIN2remains de-asserted (e.g., logic low) to deactivate the input switch N2. Therefore, the time t1is the beginning of a switching dead-time of the input switches N1and N2. At the time t1, the primary current IPRIbegins ringing (e.g., resonant oscillation among the parasitic capacitance of the input switches N1and N2, the resonant inductor LRES, the output inductor LOUT, and the parasitic capacitance of the output switches N5, N6, N7, and N8), and thus beings to increase. At the time t1, the secondary current ISECbegins ringing, and thus the rectifier currents ISR1and ISR2begin to oscillate based on the parasitic capacitance of the output switches N5, N6, N7, and N8and the output inductor LOUT. Also at the time t1, the voltage VABdecreases sharply as the voltage VBincreases relative to the voltage VA. At a time t2, the clamp switching signal SACis asserted to activate the clamping switch N9, and the clamping current ICLbegins to flow. As an example, the time t2can be associated with an approximate peak of the ringing of the primary current IPRI, and can thus be one-half of the ringing period of the primary current IPRI. As an example, the time t2can be set based on the switching controller108monitoring the amplitude of the primary current IPRI, and thus associating the time t2, and therefore the activation of the clamping switch N9by the clamp switching signal SAC, to the approximate ringing peak of the primary current IPRI. For example, the time t2can be statically (e.g., open-loop) set during fabrication and testing of the switching controller108with respect to a given power supply system200. As another example, the switching controller108can set the time t2at each switching cycle in a closed-loop feedback manner. By activating the clamping switch N9at the time t2, the primary winding LPRIdoes not dissipate as much magnetic energy, and thus maintains additional magnetic energy. As a result, the ringing of the primary current IPRIis mitigated after the time t2, resulting in a more rapid shift of the relative amplitudes of the voltages VAand VB. Accordingly, the voltage VBdecreases more rapidly to facilitate ZVS. In other words, because the ringing of the primary current IPRIis mitigated (and thus settles) after the time t2, the oscillation of the voltage VABis likewise mitigated after the time t2. At a time t3, the voltage VABdecreases to a minimum amplitude as the voltage VBincreases to an approximate maximum amplitude and the voltage VAdecreases to approximate zero. At a time t4, the input switching signal SIN2is asserted (e.g., logic high) to activate the input switch N2, while the input switching signal SIN1remains de-asserted (e.g., logic low) to deactivate the input switch N1. Therefore, the primary current IPRIflows through the input switch N3, through the primary winding LPRI, and through the input switch N2(e.g., shortly after activation of the input switch N3). Because the voltage VAS is at a minimum amplitude resulting from an approximate zero amplitude of the voltage VA, the activation of the input switch N2can be provided at approximate zero volts of the voltage VA. Accordingly, the input switch N2can be activated in a ZVS manner. The ZVS process can be repeated in approximately the same manner after de-assertion of the input switching signal SIN2, and thus deactivation of the input switch N2. As described above, the example ofFIG.3is described with respect to CCM operation of the power supply system200. As another example, the power supply system200can operate in DCM based on providing the output voltage VOUTto a lighter load. Thus, to facilitate ZVS, the clamping switch of the active clamping circuit218can be activated at a predetermined time that is just prior to activation of the input switches (e.g., the input switches N1and N2) of the input stage202. FIG.4is a timing diagram400. The timing diagram400demonstrates a number of the signals, currents, and voltages associated with the power supply system200in the example ofFIG.2operating in DCM. Therefore, reference is to be made to the example ofFIG.2in the following description of the example ofFIG.4. The timing diagram400includes the clamping current ICL, the output current IOUT, the primary current IPRI, the first rectifier current ISR1through the output switches N5and N8, a second rectifier current ISR2through the output switches N6and N7, the input switching signals SIN1and SIN2, and the voltage VAS across the primary winding LPRIof the transformer204. Prior to a time to, the input switching signal SIN2was de-asserted (e.g., logic low) to deactivate the input switch N2, while the input switching signal SIN1was likewise de-asserted to deactivate the input switch N1. Therefore, at the time to, the power supply system200is in a switching dead-time with respect to the input switches N1and N2. Because the power supply system200is operating in DCM, the primary current IPRIhas an amplitude of approximately zero, as well as the rectifier currents ISR1and ISR2and the output current IOUT. At a time t1, the clamp switching signal SACis asserted to activate the clamping switch N9, and the clamping current ICLbegins to flow. A short time later, at a time t2, the input switching signal Sim is asserted (e.g., logic high) to activate the input switch N1. Therefore, the activation of the clamp switching signal SACis provided at a time just prior to activation of one of the input switch (e.g., the input switch N1with respect to the times t1and t2). Also, concurrently with the activation of the clamping switch N9at the time t1, the output switches N6and N7can be activated by the output switching signals SOUT1. Therefore, the output switches N6and N7can be activated slightly earlier than the input switch N1. The activation of the clamping switch N9and the output switches N6and N7can facilitate injection of the clamping current ICLfrom the secondary winding LSECto the primary winding LPRI. In other words, prior to activation of the input switch N1at the time t2, the secondary current ISEC(comprised primarily of the clamping current ICL) is injected from the secondary winding LSECto the primary winding IPRIto increase the primary current IPRI, as demonstrated generally at402by the non-zero amplitude spike in the primary current IPRI. Therefore, additional magnetic energy is provided in the primary winding LPRIby the current induced from the secondary winding LSEC. As a result, the relative amplitudes of the voltages VAand VBshift more rapidly. Accordingly, the voltage VBincreases from the time t1more rapidly to facilitate ZVS. The time t2can therefore represent a time when the voltage VABis at an approximate minimum or maximum, and therefore when one of the voltages VAand VBhas an amplitude of approximately zero volts. As an example, the time between the times t1and t2can be optimized based on the amplitude of the primary current IPRIand the amplitude of the voltage VAB. For example, the time t1can be set prior to the time t2based on the switching controller108monitoring the amplitude of the primary current IPRI, and thus associating the time t2, and therefore the activation of the clamping switch N9by the clamp switching signal SAC, at an approximate time that the voltage VABachieves a maximum or minimum amplitude. Thus, the primary current IPRIdoes not increase or decrease for too long to result in inefficient operation (e.g., with respect to generating the output current IOUT). For example, the time t1can be statically (e.g., open-loop) set during fabrication and testing of the switching controller108with respect to a given power supply system200. As another example, the switching controller108can set the time t1at each switching cycle in a closed-loop feedback manner. At a time t3, which is representative of the end of the output inductor LOUTcharging period, the clamp switching signal SACis de-asserted to deactivate the clamping switch N9. For example, the clamp switching signal SACcan be de-asserted from approximately 100 nS up to the end of output inductor LOUTcharging period at t3. The rectifier current ISR2begins to decrease, as well as the output current IOUT. Also, the voltage VABbegins to decrease. At a time t4, the input switching signal SIN1is de-asserted (e.g., logic low) to deactivate the input switch N1. Therefore, the time t4is the beginning of a switching dead-time of the input switches N1and N2. The timing diagram400thus repeats in the opposite phase, similar to as described above, in which the clamping switch N9is activated just prior to the input switch N2. Accordingly, as described herein, the active clamping circuit218can be implemented not only to mitigate ringing in the output stage210, but by activating the clamping switch N9in the active clamping circuit218during the dead-time of the input switches N1and N2, the power supply system200can implement ZVS for more energy efficient operation relative to typical power supply systems. The power supply system200can be configured to implement ZVS in either CCM operation or DCM operation. As a result, the power supply system200can provide for more efficient operation using ZVS of the input switches N1and N2of the input stage202regardless of load conditions. In this description, the term “couple” may cover connections, communications, or signal paths that enable a functional relationship consistent with this description. For example, if device A generates a signal to control device B to perform an action, then: (a) in a first example, device A is directly coupled to device B; or (b) in a second example, device A is indirectly coupled to device B through intervening component C if intervening component C does not substantially alter the functional relationship between device A and device B, so device B is controlled by device A via the control signal generated by device A. In this description, a device that is “configured to” perform a task or function may be configured (e.g., programmed and/or hardwired) at a time of manufacturing by a manufacturer to perform the function and/or may be configurable (or reconfigurable) by a user after manufacturing to perform the function and/or other additional or alternative functions. The configuring may be through firmware and/or software programming of the device, through a construction and/or layout of hardware components and interconnections of the device, or a combination thereof. Furthermore, a circuit or device that is described herein as including certain components may instead be configured to couple to those components to form the described circuitry or device. For example, a structure described herein as including one or more semiconductor elements (such as transistors), one or more passive elements (such as resistors, capacitors, and/or inductors), and/or one or more sources (such as voltage and/or current sources) may instead include only the semiconductor elements within a single physical device (e.g., a semiconductor die and/or integrated circuit (IC) package) and may be configured to couple to at least some of the passive elements and/or the sources to form the described structure, either at a time of manufacture or after a time of manufacture, such as by an end-user and/or a third-party. Modifications are possible in the described embodiments, and other embodiments are possible, within the scope of the claims. | 30,478 |
11863078 | DETAILED DESCRIPTION The subject matter described herein will now be discussed with reference to several example embodiments. These embodiments are discussed only for the purpose of enabling those skilled persons in the art to better understand and thus implement the subject matter described herein, rather than suggesting any limitations on the scope of the subject matter. The term “comprises” or “includes” and its variants are to be read as open terms that mean “includes, but is not limited to.” The term “or” is to be read as “and/or” unless the context clearly indicates otherwise. The term “based on” is to be read as “based at least in part on.” The term “one embodiment” and “an embodiment” are to be read as “at least one embodiment.” The term “another embodiment” is to be read as “at least one other embodiment.” Unless specified or limited otherwise, the terms “mounted,” “connected,” “supported,” and “coupled” and variations thereof are used broadly and encompass direct and indirect mountings, connections, supports, and couplings. Furthermore, “connected” and “coupled” are not restricted to physical or mechanical connections or couplings. In the description below, like reference numerals and labels are used to describe the same, similar or corresponding parts in the Figures. Other explicit and implicit definitions may be included below. In a conventional solution, a DC-to-DC converter is configured to comprise an additional power stage electrically coupled to a secondary side, in order to implement a wide range of an output voltage and a constant output power. However, in the solution, since the additional power stage is provided, efficiency for the DC-to-DC converter is degraded and thus a power density is reduced. In view of the above, embodiments of the present disclosure provide an improved solution for DC-to-DC conversion, which provides a novel circuit topology and a corresponding control scheme to achieve the wide range of the output voltage and the constant output power without providing an additional power stage. According to the embodiments, at least two transformers are provided, and a controller is configured to controls currents flowing in primary windings of the transformers. When the currents in the primary windings of the transformers are controlled to flow in a first current-flow pattern, secondary windings of the transformers are electrically coupled in serial with each other to output a high voltage. When the currents in the primary windings of the transformers are controlled to flow in a second current-flow pattern, the secondary windings of the transformers are electrically coupled in parallel with each other to output a low voltage. In this way, the wide range of the output voltage and the constant output power are achieved while securing the high power density with fewer components. Hereinafter, the embodiments of the present disclosure will be described in detail with reference to the accompanying drawings. FIG.1illustrates a block diagram of an apparatus100for DC-to-DC conversion according to an embodiment of the present disclosure. Referring toFIG.1, the apparatus100comprises a first transformer102, a second transformer104, and a controller106. The controller106is configured to control a first current flowing in a primary winding of the first transformer102and a second current flowing in a primary winding of the second transformer104. The controller106controls flow directions of the first and second currents. The controller106controls the first current and the second current to flow in a same direction or in different directions. Herein, the same direction means that both the first current and the second current flow from or into dotted terminals. In addition, the different directions means that the first current flows from the dotted terminal while the second current flows into the dotted terminal, or means that the first current flows into the dotted terminal while the second current flows from the dotted terminal. Moreover, the controller106is configured to control the first current and the second current to flow in a first pattern, such that a secondary winding of the first transformer102and a secondary winding of the second transformer104are electrically coupled in serial with each other to output a first voltage. The controller106is further configured to control the first current and the second current to flow in a second pattern different from the first pattern, such that the secondary winding of the first transformer102and the secondary winding of the second transformer104are electrically coupled in parallel with each other to output a second voltage lower than the first voltage. In some embodiments, the first current and the second current flow in the same direction in the first pattern, and the first current and the second current flow in the different directions in the second pattern. Alternatively, the first current and the second current flow in the different directions in the first pattern, and the first current and the second current flow in the same direction in the second pattern. In some embodiments, the apparatus100outputs the high voltage in the first pattern for the current flow directions and outputs the low voltage in the second pattern for the current flow directions. In this way, the apparatus100may achieve the wide range of the output voltage without the additional power stage. Moreover, the secondary winding of the first transformer102and the secondary winding of the second transformer104are electrically coupled in serial to output a relatively low current and a relatively high voltage, and in parallel to output a relatively high current and a relatively low voltage. In this way, the apparatus100may achieve the constant output power with fewer components. Referring toFIG.1, in some embodiments, the apparatus100may further comprise a first switching circuit108and a second switching circuit110. The first switching circuit108may be electrically coupled to a first supply IN1, and the second switching circuit110may be electrically coupled to a second supply IN2. In some embodiments, each of the first supply IN1and the second supply IN2may be a DC supply. In some embodiments, the first switching circuit108may be electrically coupled between the first transformer102and the controller106, and the second switching circuit110may be electrically coupled between the second transformer104and the controller106. In some embodiments, the controller106may control the first current flowing in the primary winding of the first transformer102by supplying a first control signal to the first switching circuit108, and control the second current flowing in the primary winding of the second transformer104by supplying a second control signal to the second switching circuit110. In some embodiments, the first control signal comprises a first set of drive signals for corresponding switching elements of the first switching circuit108, and the second control signal comprises a second set of drive signals for corresponding switching elements of the second switching circuit110. Referring toFIG.1, in some embodiments, the apparatus100may further comprise a rectifier circuit112and a filter114. The rectifier circuit112may be electrically coupled to the secondary windings of the first transformer102and the second transformer104, and the filter114is electrically coupled between the rectifier circuit112and an output terminal OUT. The apparatus110may output a desired DC supply at the output terminal OUT through the rectifier circuit112and the filter114. In some embodiments, when the first and second currents of the primary windings flow in the first pattern, the rectifier circuit112allows output currents of the secondary windings to flow in a manner such that the secondary windings of the first transformer102and the second transformer104are electrically coupled in serial to output the high voltage. In some embodiments, when the first and second currents of the primary windings flow in the second pattern, the rectifier circuit112allows the output currents of the secondary windings to flow in a manner such that the secondary windings of the first transformer102and the second transformer104are electrically coupled in parallel to output the low voltage. Hereinafter, several circuit implementations of the apparatus100according to the embodiments of the present disclosure will be described in detail. FIG.2illustrates a diagram of circuit implementation of the apparatus100according to embodiments of the present disclosure. Referring toFIG.2, the first transformer102comprises a first winding T1A at the primary side and a second winding T1B at a secondary side, and the second transformer104comprises a third winding T2A at the primary side and a fourth winding T2B at the secondary side. Moreover, the apparatus100comprises an input terminal IN, a reference potential terminal VSS, and an output terminal OUT. It should be noted that, the controller is not illustrated inFIG.2, but the controller supplies drive signals to control terminals of transistors Q1to Q4. In some embodiments, the apparatus100may comprise a first LLC topology circuit and a second LLC topology circuit. The first LLC topology circuit comprises the first winding T1A of the first transformer102, a first inductor Lr1, and first and second capacitors Cr1and Cr2. In addition, the second LLC topology circuit comprises the third winding T2A of the second transformer104, a second inductor Lr2, and third and fourth capacitors Cr3and Cr4. In some embodiments, the first inductor Lr1is electrically coupled in serial with the first winding T1A, and the second inductor Lr2is electrically coupled in serial with the third winding T2A. In some embodiments, the apparatus100may further comprise a first half-bridge circuit and a second half-bridge circuit as the switching circuits. The first half-bridge circuit comprises a first transistor Q1and a second transistor Q2electrically coupled in series with each other between the input terminal IN and the reference potential terminal VSS. The first and second transistors Q1and Q2are electrically coupled with each other at a first intermediate node M1. In addition, the second half-bridge circuit comprises a third transistor Q3and a fourth transistor Q4electrically coupled in series with each other between the input terminal IN and the reference potential terminal VSS. The third and fourth transistors Q3and Q4are electrically coupled with each other at a second intermediate node M2. In some embodiments, the first winding T1A of the first transformer102is electrically coupled to the first intermediate node M1via the first inductor Lr1, and the third winding T2A of the second transformer104is electrically coupled to the second intermediate node M2via the second inductor Lr2. In some embodiments, the first winding T1A is electrically coupled to the input terminal IN via the first transistor Q1and to the reference potential terminal VSS via the second transistor Q2. In some embodiments, the third winding T2A is electrically coupled to the input terminal IN via the third transistor Q3and to the reference potential terminal VSS via the fourth transistor Q4. In some embodiments, the first and second capacitors Cr1and Cr2are electrically coupled in series with each other between the input terminal IN and the reference potential terminal VSS. The first and second capacitors Cr1and Cr2are electrically coupled with each other at a third intermediate node M3. In addition, the third and fourth capacitors Cr3and Cr4are electrically coupled in series with each other between the input terminal IN and the reference potential terminal VSS. The third and fourth capacitors Cr3and Cr4are electrically coupled with each other at a fourth intermediate node M4. In some embodiments, the first winding T1A of the first transformer102is further electrically coupled to the third intermediate node M3, and the third winding T2A of the second transformer104is further electrically coupled to the fourth intermediate node M4. In some embodiments, the first inductor Lr1is electrically coupled between the first intermediate node M1and the first winding T1A, and the second inductor Lr2is electrically coupled between the second intermediate node M2and the third winding T2A. It should be noted thatFIG.2is merely an example implementation without suggesting any limitations as to the scope of the present disclosure. For example, in alternative embodiments, the first inductor Lr1is electrically coupled between the third intermediate node M3and the first winding T1A, and the second inductor Lr2is electrically coupled between the fourth intermediate node M4and the third winding T2A. Moreover, a first current transformer may be electrically coupled to the first winding T1A, and a second current transformer may be electrically coupled to the third winding T2A. Referring toFIG.2, in some embodiments, the apparatus100may comprise first to sixth rectifying elements D1to D6that constitute the rectifier circuit. The first and second rectifying elements D1and D2are electrically coupled in series with each other between the output terminal OUT and the reference potential terminal VSS, and electrically coupled with each other at a first node N1. The third and fourth rectifying elements D3and D4are electrically coupled in series with each other between the output terminal OUT and the reference potential terminal VSS, and electrically coupled with each other at a second node N2. The fifth and sixth rectifying elements D5and D6are electrically coupled in series with each other between the output terminal OUT and the reference potential terminal VSS, and electrically coupled with each other at a third node N3. In some embodiments, each of the rectifying elements D1to D6is a diode. Cathodes of the first, third and fifth rectifying elements D1, D3and D5are electrically coupled to the output terminal OUT, and anodes of the first, third and fifth rectifying elements D1, D3and D5are electrically coupled to the first node N1, the second node N2and the third node N3, respectively. In addition, anodes of the second, fourth and sixth rectifying elements D2, D4and D6are electrically coupled to the reference potential terminal VSS, and cathodes of the second, fourth and sixth rectifying elements D2, D4and D6are electrically coupled to the first node N1, the second node N2and the third node N3, respectively. It is to be noted that the rectifying elements D1to D6can be implemented by other suitable devices than diodes. For example, in alternative embodiments, one or more of the rectifying elements D1to D6can be implemented by MOS transistors, silicon carbide transistors, IGBTs, and etc. In some embodiments, the second winding T1B of the first transformer102comprises a first terminal coupled to the first node N1and a second terminal coupled to the third node N3. In addition, the fourth winding T2B of the second transformer104comprises a third terminal coupled to the third node N3and a fourth terminal coupled to the second node N2. In some embodiments, the first terminal is a dotted terminal of the second winding T1B and the third terminal is a dotted terminal of the fourth winding T2B, or the second terminal is a dotted terminal of the second winding T1B and the fourth terminal is a dotted terminal of the fourth winding T2B. In other embodiments, the first terminal is a dotted terminal of the second winding T1B and the fourth terminal is a dotted terminal of the fourth winding T2B, or the second terminal is a dotted terminal of the second winding T1B and the third terminal is a dotted terminal of the fourth winding T2B. In the embodiment as shown inFIG.2, the dotted terminal of the second winding T1B and the dotted terminal of the fourth winding T2B are electrically coupled to the first node N1and the third node N3, respectively. In other embodiments, the dotted terminal of the second winding T1B and the dotted terminal of the fourth winding T2B may be electrically coupled to the third node N3and the second node N2, respectively. In both cases, when the output currents in the second and fourth windings T1B and T2B flow in the same direction, the second and fourth windings T1B and T2B are electrically coupled in series with each other. When the output currents in the second and fourth windings T1B and T2B flow in the different directions, the second and fourth windings T1B and T2B are electrically coupled in parallel with each other. It is to be noted that althoughFIG.2illustrates that the dotted terminal of the second winding T1B is electrically coupled to the first node N1and the dotted terminal of the fourth winding T2B is electrically coupled to the third node N3, the scope of the present disclosure are not limited thereto. In other embodiments, any of the terminals of the second winding T1B may be electrically coupled to the first node N1, and any of the terminals of the fourth winding T2B may be electrically coupled to the third node N3, for example. In alternative embodiments, the doted terminal of the second winding T1B and the dotted terminal of the fourth winding T2B may be both electrically coupled to the third node N3, or may be electrically coupled to the first node N1and the second node N2, respectively. In such cases, the second and fourth windings T1B and T2B are electrically coupled in series with each other when the output currents in the second and fourth windings T1B and T2B flow in the different directions, and in parallel with each other when the output currents in the second and fourth windings T1B and T2B flow in the same direction. Moreover, althoughFIG.2illustrates that the dotted terminal of the first winding T1A is electrically coupled to the first inductor Lr1and the dotted terminal of the third winding T2A is electrically coupled to the second inductor Lr2, it is merely an example implementation without suggesting any limitations as to the scope of the present disclosure. For example, in other embodiments, any of the terminals of the first winding T1A may be electrically coupled to the first inductor Lr1, and any of the terminals of the third winding T2A may be electrically coupled to the second inductor Lr2. Still in reference toFIG.2, in some embodiments, the apparatus100may further comprise an output capacitor Co as the filter. The output capacitor Co may be electrically coupled between the output terminal OUT and the reference potential terminal VSS. Moreover, in some embodiments, the apparatus100may further comprise an input capacitor Ci electrically coupled between the input terminal IN and the reference potential terminal. FIG.3illustrates drive signals for the transistors Q1to Q4of the apparatus100as shown inFIG.2. As shown inFIG.3, the first and third transistors Q1and Q3are driven by the controller in a same phase, and the second and fourth transistors Q2and Q4are driven by the controller in a same phase. Moreover,FIGS.4A and4Billustrate current flow patterns for the apparatus100as shown inFIG.2. Referring toFIGS.3and4A, if the first and third transistors Q1and Q3are turned on and the second and fourth transistors Q2and Q4are turned off by the controller with the respective drive signals, the first current in the first winding T1A and the second current in the third winding T2A are controlled to flow into the dotted terminals in a first direction at the primary sides. Accordingly, the output currents in the second and fourth windings T1B and T2B flow from the dotted terminals in the same direction at the secondary sides, and the second and fourth windings T1B and T2B are electrically connected in serial with each other to output the high voltage. In the current flow pattern illustrated inFIG.4A, the output currents flow through the first and fourth rectifying elements D1and D4. Moreover, as shown inFIGS.3and4B, if the first and third transistors Q1and Q3are turned off and the second and fourth transistors Q2and Q4are turned on by the controller with the respective drive signals, the first current in the first winding T1A and the second current in the third winding T2A are controlled to flow from the dotted terminals in a second direction at the primary sides. Accordingly, the output currents in the second and fourth windings T1B and T2B flow into the dotted terminals in the same direction at the secondary sides, and the second and fourth windings T1B and T2B are electrically connected in serial with each other to output the high voltage. In the current flow pattern illustrated inFIG.4B, the output currents flow through the second and third rectifying elements D2and D3. It is to be noted that althoughFIG.3shows that the first and third transistors Q1and Q3are turned on substantially at the same time by the drive signals, and the second and fourth transistors Q2and Q4are turned on substantially at the same time by the drive signals, it is merely an example implementation without suggesting any limitations as to the scope of the present disclosure. For example, in other embodiments, there may be a phase shift between the drive signal for the first transistor Q1and the drive signal for the third transistor Q3, or between the drive signal for the second transistor Q2and the drive signal for the fourth transistor Q4. FIG.5illustrates drive signals for the transistors Q1to Q4of the apparatus100as shown inFIG.2. As shown inFIG.5, the first and fourth transistors Q1and Q4are driven by the controller in a same phase, and the second and third transistors Q2and Q3are driven by the controller in a same phase.FIGS.6A and6Billustrate current flow patterns for the apparatus100as shown inFIG.2. As shown inFIGS.5and6A, if the first and fourth transistors Q1and Q4are turned on and the second and third transistors Q2and Q3are turned off by the controller with the respective drive signals, the first current in the first winding T1A is controlled to flow into the dotted terminal in the first direction and the second current in the third winding T2A is controlled to flow from the dotted terminal in the second direction at the primary sides. Accordingly, the output current in the second winding T1B flows from the dotted terminal and the output current in the fourth winding T2B flows into the dotted terminals at the secondary sides, and the second and fourth windings T1B and T2B are equivalent to be electrically connected in parallel with each other to output the low voltage. In the current flow pattern illustrated inFIG.6A, the output current in the second winding T1B flows through the first and sixth rectifying elements D1and D6, and the output current in the fourth winding T2B flows through the third and sixth rectifying elements D3and D6. Moreover, as shown inFIGS.5and6B, if the first and fourth transistors Q1and Q4are turned off and the second and third transistors Q2and Q3are turned on by the controller with the respective drive signals, the first current in the first winding T1A is controlled to flow from the dotted terminal in the second direction and the second current in the third winding T2A is controlled to flow into the dotted terminal in the first direction at the primary sides. Accordingly, the output current in the second winding T1B flows into the dotted terminal and the output current in the fourth winding T2B flows from the dotted terminals at the secondary sides, and the second and fourth windings T1B and T2B are equivalent to be electrically connected in parallel with each other to output the low voltage. In the current pattern illustrated inFIG.6B, the output current in the second winding T1B flows through the second and fifth rectifying elements D2and D5, and the output current in the fourth winding T2B flows through the fourth and fifth rectifying elements D4and D5. It is to be understood that the control schemes as shown inFIGS.4A,4B,6A and6Bare merely examples without suggesting any limitations as to the scope of the present disclosure. As described above, in some alternative embodiments, the doted terminal of the second winding T1B and the dotted terminal of the fourth winding T2B may be both electrically coupled to the third node N3, or may be electrically coupled to the first node N1and the second node N2, respectively. In such embodiments, the current flow patterns are opposite to those as illustrated inFIGS.4A,4B,6A and6Bunder the control of the drive signals ofFIGS.3and5. More specifically, when the output currents in the second and fourth windings T1B and T2B flow in the different directions, the second and fourth windings T1B and T2B are electrically coupled in series with each other; and when the output currents in the second and fourth windings T1B and T2B flow in the same direction, the second and fourth windings T1B and T2B are electrically coupled in parallel with each other. According to the embodiments of the present disclosure, the high voltage is output when the secondary windings of the transformers are electrically connected in series, and the low voltage is output when the secondary windings of the transformers are electrically connected in parallel. In this way, the wide range of the output voltage with the constant output power is achieved without additional power stage, and the designs for the transformers can be optimized to be best. Thus, the apparatus100has improved efficiency and is able to provide high power density with fewer components. Hereinafter, several alternative circuit implementations for the apparatus100will be described. Generally speaking, in those alternative embodiments, circuit topologies at the primary sides of the transformers can be varied. FIGS.7A,7B and7Cillustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. The implementations as shown inFIGS.7A,7B and7Care similar to that as discussed above with reference toFIG.2except for the primary sides of the transformers. In the following, descriptions on the similar circuit topology, control scheme and current flow pattern will be omitted. In the embodiment shown inFIG.7A, the apparatus100further comprises an intermediate reference terminal INT, a first input capacitor Ci1electrically coupled between the input terminal IN and the intermediate reference terminal INT, and a second input capacitor Ci2electrically coupled between the intermediate reference terminal INT and the reference potential terminal VSS. In some embodiments, the first and second transistors Q1and Q2are electrically coupled at the first intermediate node M1in series with each other between the input terminal IN and the intermediate reference terminal INT. In addition, the third and fourth transistors Q3and Q4are electrically coupled at the second intermediate node M2in series with each other between the intermediate reference terminal INT and the reference potential terminal VSS. Moreover, the first and second capacitors Cr1and Cr2are electrically coupled at the third intermediate node M3in series with each other between the input terminal IN and the intermediate reference terminal INT. In addition, the third and fourth capacitors Cr3and Cr4are electrically coupled at the fourth intermediate node M4in series with each other between the intermediate reference terminal INT and the reference potential terminal VSS. Referring toFIGS.7B and7C, the apparatus100further comprises a first diode Dp1electrically coupled in parallel with the first capacitor Cr1, a second diode Dp2electrically coupled in parallel with the second capacitor Cr2, a third diode Dp3electrically coupled in parallel with the third capacitor Cr3, and a fourth diode Dp4electrically coupled in parallel with the fourth capacitor Cr4. In the embodiment shown inFIG.7B, an anode of the first diode Dp1is coupled to the third intermediate node M3, and a cathode of the first diode Dp1is coupled to the input terminal IN. The anode of the second diode Dp2is coupled to the reference potential terminal VSS, and a cathode of the second diode Dp2is coupled to the third intermediate node M3. The anode of the third diode Dp3is coupled to the fourth intermediate node M4, and a cathode of the third diode Dp3is coupled to the input terminal IN. The anode of the fourth diode Dp4is coupled to the reference potential terminal VSS, and a cathode of the fourth diode Dp4is coupled to the fourth intermediate node M4. In the embodiment shown inFIG.7C, an anode of the first diode Dp1is coupled to the third intermediate node M3, and a cathode of the first diode Dp1is coupled to the input terminal IN. The anode of the second diode Dp2is coupled to the intermediate reference terminal INT, and a cathode of the second diode Dp2is coupled to the third intermediate node M3. The anode of the third diode Dp3is coupled to the fourth intermediate node M4, and a cathode of the third diode Dp3is coupled to the intermediate reference terminal INT. The anode of the fourth diode Dp4is coupled to the reference potential terminal VSS, and a cathode of the fourth diode Dp4is coupled to the fourth intermediate node M4. The drive signals for the transistors Q1to Q4as shown inFIGS.3and5as well as the current flow patterns as illustrated inFIGS.4A,4B,6A and6Bmay be applied to the circuit implementations ofFIGS.7A,7B and7C. FIGS.8A and8Billustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. The implementations as shown inFIGS.8A and8Bare similar to those as discussed above with reference toFIGS.2and7except for the primary sides of the transformers. In the following, descriptions on the similar circuit topology and current flow pattern will be omitted. In the embodiment shown inFIG.8A, the apparatus100may comprise a first LLC topology circuit and a second LLC topology circuit. The first LLC topology circuit comprises the first winding T1A of the first transformer102, the first inductor Lr1, and a first capacitor Cr1. In addition, the second LLC topology circuit comprises the third winding T2A of the second transformer104, the second inductor Lr2, and a second capacitor Cr2. In some embodiments, the first inductor Lr1is electrically coupled in serial with the first winding T1A, and the second inductor Lr2is electrically coupled in serial with the third winding T2A. In some embodiments, the first inductor Lr1may be included in the first transformer102. In some embodiments, the second inductor Lr2may be included in the second transformer104. In some embodiments, the apparatus100may comprise a first full-bridge circuit and a second full-bridge circuit as the switching circuits. The first full-bridge circuit comprises the first transistor Q1and the second transistor Q2electrically coupled at the first intermediate node M1in series with each other between the input terminal IN and the reference potential terminal VSS. In addition, the second half-bridge circuit comprises the third transistor Q3and the fourth transistor Q4electrically coupled at the second intermediate node M2in series with each other between the input terminal IN and the reference potential terminal VSS. Moreover, the first full-bridge circuit further comprises a fifth transistor Q5and a sixth transistor Q6electrically coupled at the third intermediate node M3in series with each other between the input terminal IN and the reference potential terminal VSS. In addition, the second full-bridge circuit further comprises a seventh transistor Q7and an eighth transistor Q8electrically coupled at the fourth intermediate node M4in series with each other between the input terminal IN and the reference potential terminal VSS. In some embodiments, the first winding T1A of the first transformer102is electrically coupled to the first intermediate node M1via the first capacitor Cr1and the first inductor Lr1, and the third winding T2A of the second transformer104is electrically coupled to the second intermediate node M2via the second capacitor Cr2and the second inductor Lr2. In some embodiments, the first winding T1A of the first transformer102is further electrically coupled to the third intermediate node M3, and the third winding T2A of the second transformer104is further electrically coupled to the fourth intermediate node M4. In some embodiments, the first winding T1A is electrically coupled to the input terminal IN via the first transistor Q1or the fifth transistor Q5and to the reference potential terminal VSS via the second transistor Q2or the sixth transistor Q6. In some embodiments, the second winding T2A is electrically coupled to the input terminal IN via the third transistor Q3or the seventh transistor Q7and to the reference potential terminal VSS via the fourth transistor Q4or the eighth transistor Q8. In the embodiment shown inFIG.8B, the apparatus100further comprises the intermediate reference terminal INT, the first input capacitor Ci1electrically coupled between the input terminal IN and the intermediate reference terminal INT, and the second input capacitor Ci2electrically coupled between the intermediate reference terminal INT and the reference potential terminal VSS. In some embodiments, the first and second transistors Q1and Q2are electrically coupled at the first intermediate node M1in series with each other between the input terminal IN and the intermediate reference terminal INT. The third and fourth transistors Q3and Q4are electrically coupled at the second intermediate node M2in series with each other between the intermediate reference terminal INT and the reference potential terminal VSS. The fifth and sixth transistor Q5and Q6are electrically coupled at the third intermediate node M3in series with each other between the input terminal IN and the intermediate reference terminal INT. In addition, the seventh and eighth transistors Q7and Q8are electrically coupled at the fourth intermediate node M4in series with each other between the intermediate reference terminal INT and the reference potential terminal VSS. FIGS.9A and9Billustrate drive signals for the transistors Q1to Q8of the apparatus100as shown inFIGS.8A and8B. As shown inFIG.9A, the first, third, sixth and eighth transistors Q1, Q3, Q6and Q8are driven by the controller in a same phase, and the second, fourth, fifth and seventh transistors Q2, Q4, Q5and Q7are driven by the controller in a same phase. Referring toFIGS.8A,8B and9A, if the first, third, sixth and eighth transistors Q1, Q3, Q6and Q8are turned on and the second, fourth, fifth and seventh transistors Q2, Q4, Q5and Q7are turned off by the controller with the respective drive signals, the first current in the first winding T1A and the second current in the third winding T2A are controlled to flow into the dotted terminals in the first direction at the primary sides. Accordingly, the output currents in the second and fourth windings T1B and T2B flow from the dotted terminals in the same direction at the secondary sides, and the second and fourth windings T1B and T2B are electrically connected in serial with each other to output the high voltage. Moreover, if the first, third, sixth and eighth transistors Q1, Q3, Q6and Q8are turned off and the second, fourth, fifth and seventh transistors Q2, Q4, Q5and Q7are turned on by the controller with the respective drive signals, the first current in the first winding T1A and the second current in the third winding T2A are controlled to flow from the dotted terminals in the second direction at the primary sides. Accordingly, the output currents in the second and fourth windings T1B and T2B flow into the dotted terminals in the same direction at the secondary sides, and the second and fourth windings T1B and T2B are electrically connected in serial with each other to output the high voltage. Further, as shown inFIG.9B, the first, fourth, sixth and seventh transistors Q1, Q4, Q6and Q7are driven by the controller in a same phase, and the second, third, fifth and eighth transistors Q2, Q3, Q5and Q8are driven by the controller in a same phase. Referring toFIGS.8A,8B and9B, if the first, fourth, sixth and seventh transistors Q1, Q4, Q6and Q7are turned on and the second, third, fifth and eighth transistors Q2, Q3, Q5and Q8are turned off by the controller with the respective drive signals, the first current in the first winding T1A is controlled to flow into the dotted terminal in the first direction and the second current in the third winding T2A is controlled to flow from the dotted terminal in the second direction at the primary sides. Accordingly, the output current in the second winding T1B flows from the dotted terminal and the output current in the fourth winding T2B flows into the dotted terminals at the secondary sides, and the second and fourth windings T1B and T2B are equivalent to be electrically connected in parallel with each other to output the low voltage. Moreover, if the first, fourth, sixth and seventh transistors Q1, Q4, Q6and Q7are turned off and the second, third, fifth and eighth transistors Q2, Q3, Q5and Q8are turned on by the controller with the respective drive signals, the first current in the first winding T1A is controlled to flow from the dotted terminal in the second direction and the second current in the third winding T2A is controlled to flow into the dotted terminal in the first direction at the primary sides. Accordingly, the output current in the second winding T1B flows into the dotted terminal and the output current in the fourth winding T2B flows from the dotted terminals at the secondary sides, and the second and fourth windings T1B and T2B are equivalent to be electrically connected in parallel with each other to output the low voltage. It should be noted thatFIGS.8A and8Bare merely example implementations without suggesting any limitations as to the scope of the present disclosure. For example, in alternative embodiments, the doted terminal of the second winding T1B and the dotted terminal of the fourth winding T2B may be both electrically coupled to the third node N3, or may be electrically coupled to the first node N1and the second node N2, respectively. In such embodiments, the second and fourth windings T1B and T2B are electrically coupled in series with each other when the output currents in the second and fourth windings T1B and T2B flow in the different directions, and in parallel with each other when the output currents in the second and fourth windings T1B and T2B flow in the same direction. FIGS.10A and10Billustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. The implementations as shown inFIGS.10A and10Bare similar to those as discussed above with reference toFIGS.2and7except that the first and second inductors Lr1and Lr2are omitted at the primary sides of the transformers and an output inductor Lf is added at the secondary sides of the transformers. In the following, descriptions on the similar circuit topology, control scheme and current flow pattern will be omitted. Referring toFIGS.10A and10B, in some embodiments, the first winding T1A is electrically directly coupled to the first intermediate node M2, and the third winding T2A is electrically directly coupled to the second intermediate node M2. Moreover, the output inductor Lf is electrically coupled between the output terminal OUT and a common node of cathodes of the first, third and fifth rectifying elements D1, D3and D5. In some embodiments, the output capacitor Co together with the output inductor Lf serve as the filter, and the output voltage is provided form the output terminal OUT between the output inductor Lf and the output capacitor Co. FIGS.11A and11Billustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. The implementations as shown inFIGS.11A and11Bare similar to those as discussed above with reference toFIGS.10A and10Bexcept for the primary sides of the transformers. In the following, descriptions on the similar circuit topology, control scheme and current flow pattern will be omitted. In the embodiment shown inFIG.11A, the fifth and sixth transistors Q5and Q6are electrically coupled at the third intermediate node M3in series with each other between the input terminal IN and the reference potential terminal VSS. In addition, the seventh and eighth transistors Q7and Q8are electrically coupled at the fourth intermediate node M4in series with each other between the input terminal IN and the reference potential terminal VSS. Moreover, in the embodiment shown inFIG.11B, the fifth and sixth transistors Q5and Q6are electrically coupled at the third intermediate node M3in series with each other between the input terminal IN and the intermediate reference terminal INT. In addition, the seventh and eighth transistors Q7and Q8are electrically coupled at the fourth intermediate node M4in series with each other between the intermediate reference terminal INT and the reference potential terminal VSS. FIGS.12A and12Billustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. The implementations as shown inFIGS.12A and12Bare similar to those as discussed above with reference toFIGS.11A and11Bexcept that the first and second inductors Lr1and Lr2are added at the primary sides of the transformers and there is a phase shift between the drive signals. In the following, descriptions on the similar circuit topology and current flow pattern will be omitted. Referring toFIGS.12A and12B, the first inductor Lr1is coupled between the first intermediate node M1and the first winding T1A, and the second inductor Lr2is coupled between the second intermediate node M2and the third winding T2A. FIGS.13A and13Billustrate drive signals for the transistors Q1to Q8of the apparatus100as shown inFIGS.12A and12B. Referring toFIG.13A, there is a phase shift between the drive signals for the first and third transistors Q1and Q3and the drive signals for the sixth and eighth transistors Q6and Q8, and there is a phase shift between the drive signals for the second and fourth transistors Q2and Q4and the drive signals for the fifth and seventh transistors Q5and Q7. That is, a turn-on timing of the sixth and eighth transistors Q6and Q8is delayed with respect to a turn-on timing of the first and third transistors Q1and Q3, and a turn-on timing of the fifth and seventh transistors Q5and Q7is delayed with respect to a turn-on timing of the second and fourth transistors Q2and Q4. Moreover, referring toFIG.13B, there is a phase shift between the drive signals for the first and fourth transistors Q1and Q4and the drive signals for the sixth and seventh transistors Q6and Q7, and there is a phase shift between the drive signals for the second and third transistors Q2and Q3and the drive signals for the fifth and eighth transistors Q5and Q8. That is, a turn-on timing of the sixth and seventh transistors Q6and Q7is delayed with respect to a turn-on timing of the first and fourth transistors Q1and Q4, and a turn-on timing of the fifth and eighth transistors Q5and Q8is delayed with respect to a turn-on timing of the second and third transistors Q2and Q3. FIGS.14A and14Billustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. In the embodiment shown inFIG.14A, the first transformer102comprises the first winding T1A and the second winding T1B, and the first winding T1A may comprise a first sub winding T1A1and a second sub winding T1A2. In addition, the second transformer104comprises the third winding T2A and the fourth winding T2B, and the third winding T2A may comprise a third sub winding T2A1and a fourth sub winding T2A2. In some embodiments, the apparatus100may comprise first to fourth transistors Q1to Q4which constitute the switching circuit. The first transistor Q1is electrically coupled between the reference potential terminal VSS and the first sub winding T1A1. The second transistor Q2is electrically coupled between the reference potential terminal VSS and the second sub winding T1A2. The third transistor Q3is electrically coupled between the reference potential terminal VSS and the third sub winding T2A1. In addition, the fourth transistor Q4is electrically coupled between the reference potential terminal VSS and the fourth sub winding T2A2. In some embodiments, the first sub winding T1A1comprises a dotted terminal electrically coupled to the input terminal IN and an opposite terminal electrically coupled to the first transistor Q1. The second sub winding T1A2comprises a dotted terminal electrically coupled to the second transistor Q2and an opposite terminal electrically coupled to the input terminal IN. The third sub winding T2A1comprises a dotted terminal electrically coupled to the input terminal IN and an opposite terminal electrically coupled to the third transistor Q3. In addition, the fourth sub winding T2A2comprises a dotted terminal electrically coupled to the fourth transistor Q4and an opposite terminal electrically coupled to the input terminal IN. Moreover, in the embodiment shown inFIG.14B, the apparatus100may further comprise the intermediate reference terminal INT. The first transistor Q1is electrically coupled between the intermediate reference terminal INT and the first sub winding T1A1. The second transistor Q2is electrically coupled between the intermediate reference terminal INT and the second sub winding T1A2. The third transistor Q3is electrically coupled between the reference potential terminal VSS and the third sub winding T2A1. In addition, the fourth transistor Q4is electrically coupled between the reference potential terminal VSS and the fourth sub winding T2A2. In some embodiments, the first sub winding T1A1comprises a dotted terminal electrically coupled to the input terminal IN and an opposite terminal electrically coupled to the first transistor Q1. The second sub winding T1A2comprises a dotted terminal electrically coupled to the second transistor Q2and an opposite terminal electrically coupled to the input terminal IN. The third sub winding T2A1comprises a dotted terminal electrically coupled to the intermediate reference terminal INT and an opposite terminal electrically coupled to the third transistor Q3. In addition, the fourth sub winding T2A2comprises a dotted terminal electrically coupled to the fourth transistor Q4and an opposite terminal electrically coupled to the intermediate reference terminal INT. FIGS.15A and15Billustrate drive signals for the transistors Q1to Q4of the apparatus100as shown inFIGS.14A and14B. Referring toFIG.15A, the first and third transistors Q1and Q3are driven by the controller in a same phase with the corresponding drive signals, and the second and fourth transistors Q2and Q4are driven by the controller in a same phase with the corresponding drive signals. Referring toFIGS.14A,14B and15A, if the first and third transistors Q1and Q3are turned on and the second and fourth transistors Q2and Q4are turned off by the controller with the respective drive signals, the first current flows in the first sub winding T1A1of the first winding T1A and flows into the dotted terminal, and the second current flows in the third sub winding T2A1of the third winding T2A and flows into the dotted terminals. Accordingly, the output currents in the second and fourth windings T1B and T2B flow from the dotted terminals in the same direction at the secondary sides, and the second and fourth windings T1B and T2B are electrically connected in serial with each other to output the high voltage. Moreover, if the first and third transistors Q1and Q3are turned off and the second and fourth transistors Q2and Q4are turned on by the controller with the respective drive signals, the first current flows in the second sub winding T1A2of the first winding T1A and flows from the dotted terminal, and the second current flows in the fourth sub winding T2A2of the third winding T2A and flows from the dotted terminals. Accordingly, the output currents in the second and fourth windings T1B and T2B flow into the dotted terminals in the same direction at the secondary sides, and the second and fourth windings T1B and T2B are electrically connected in serial with each other to output the high voltage. Further, referring toFIG.15B, the first and fourth transistors Q1and Q4are driven by the controller in a same phase with the corresponding drive signals, and the second and third transistors Q2and Q3are driven by the controller in a same phase with the corresponding drive signals. Referring toFIGS.14A,14B and15B, if the first and fourth transistors Q1and Q4are turned on and the second and third transistors Q2and Q3are turned off by the controller with the respective drive signals, the first current flows in the first sub winding T1A1of the first winding T1A and flows into the dotted terminal, and the second current flows in the fourth sub winding T2A2of the third winding T2A and flows from the dotted terminals. Accordingly, the output current in the second winding T1B flows from the dotted terminal and the output current in the fourth winding T2B flows into the dotted terminals at the secondary sides, and thus the second and fourth windings T1B and T2B are equivalent to be electrically connected in parallel with each other to output the low voltage. Moreover, if the first and fourth transistors Q1and Q4are turned off and the second and third transistors Q2and Q3are turned on by the controller with the respective drive signals, the first current flows in the second sub winding T1A2of the first winding T1A and flows from the dotted terminal, and the second current flows in the third sub winding T2A1of the third winding T2A and flows into the dotted terminals. Accordingly, the output current in the second winding T1B flows into the dotted terminal and the output current in the fourth winding T2B flows from the dotted terminals at the secondary sides, and thus the second and fourth windings T1B and T2B are equivalent to be electrically connected in parallel with each other to output the low voltage. FIGS.16A and16Billustrate diagrams of other circuit implementations of the apparatus100according to embodiments of the present disclosure. In some embodiments, the apparatus100may comprise a first DC-to-DC converter circuit1602electrically coupled to the first winding T1A of the first transformer102and a second DC-to-DC converter circuit1604electrically coupled to the third winding T2A of the second transformer104. Referring toFIG.16A, the first DC-to-DC converter circuit1602is electrically coupled in series between the input terminal IN and the reference potential terminal VSS. In addition, the second DC-to-DC converter circuit1604is electrically coupled in series with each other between the input terminal IN and the reference potential terminal VSS. Moreover, referring toFIG.16B, the first DC-to-DC converter circuit1602is electrically coupled in series between the input terminal IN and the intermediate reference terminal INT. In addition, the second DC-to-DC converter circuit1604is electrically coupled in series with each other between the intermediate reference terminal INT and the reference potential terminal VSS. At the secondary sides of the circuit implementations ofFIGS.16A and16B, when the output currents in the second and fourth windings T1B and T2B are controlled to flow in the same direction, the second and fourth windings T1B and T2B are electrically coupled in series with each other to output the high voltage. Moreover, when the output currents in the second and fourth windings T1B and T2B are controlled to flow in the different directions, the second and fourth windings T1B and T2B are electrically coupled in parallel with each other to output the low voltage. It should be noted thatFIGS.14A,14B,16A and16Bare merely example implementations without suggesting any limitations as to the scope of the present disclosure. For example, in alternative embodiments, the doted terminal of the second winding T1B and the dotted terminal of the fourth winding T2B may be both electrically coupled to the third node N3, or may be electrically coupled to the first node N1and the second node N2, respectively. In such embodiments, the second and fourth windings T1B and T2B are electrically coupled in series with each other to output the high voltage when the output currents in the second and fourth windings T1B and T2B are controlled to flow in the different directions, and in parallel with each other to output the low voltage when the output currents in the second and fourth windings T1B and T2B are controlled to flow in the same direction. It should be noted that the circuit implementation of the filter114is not limited to those as shown inFIGS.2and10A.FIGS.17A,17B and17Cillustrate diagrams of other circuit implementations of the filter114of the apparatus100according to the embodiments of the present disclosure. In the alternative embodiment shown inFIG.17A, the output inductor Lf is electrically coupled between the output capacitor Co and a common node of the anodes of the second, fourth and sixth rectifying elements D2, D4and D6. In the alternative embodiment shown inFIG.17B, a first output capacitor Co1is electrically coupled between a common node of the cathodes of the first, third and fifth rectifying elements D1, D3and D5and a common node of the anodes of the second, fourth and sixth rectifying elements D2, D4and D6. The output inductor Lf is electrically coupled between the common node of the cathodes of the first, third and fifth rectifying elements D1, D3and D5and a second output capacitor Co2. In addition, the second output capacitor Co2is electrically coupled between the output inductor Lf and the common node of the anodes of the second, fourth and sixth rectifying elements D2, D4and D6. In the alternative embodiment shown inFIG.17C, a first output capacitor Co1is electrically coupled between a common node of the cathodes of the first, third and fifth rectifying elements D1, D3and D5and a common node of the anodes of the second, fourth and sixth rectifying elements D2, D4and D6. The output inductor Lf is electrically coupled between the common node of the anodes of the second, fourth and sixth rectifying elements D2, D4and D6and a second output capacitor Co2. In addition, the second output capacitor Co2is electrically coupled between the output inductor Lf and the common node of the cathodes of the first, third and fifth rectifying elements D1, D3and D5. FIG.18illustrates a flowchart of a method1800for DC-to-DC conversion according to an embodiment of the present disclosure. Referring toFIG.18, the method1800comprises blocks1802and1804. At block1802, the method1800comprises controlling the first current in the first winding T1A of the first transformer102and the second current in the third winding T2A of the second transformer104to flow in the first pattern, such that the second winding T1B of the first transformer102and the fourth winding T2B of the second transformer104are electrically coupled in serial to output the first voltage. In some embodiments, controlling the first current and the second current to flow in the first pattern comprises controlling the first current and the second current to flow in the same direction. At block1804, the method1800comprises controlling the first current and the second current to flow in the second pattern different from the first pattern, such that the second winding T1B of the first transformer102and the fourth winding T2B of the second transformer104are electrically coupled in parallel to output the second voltage lower than the first voltage. In some embodiment, controlling the first current and the second current to flow in the second pattern comprises controlling the first current and the second current to flow in the different directions. Alternatively, in other embodiments, controlling the first current and the second current to flow in the first pattern comprises controlling the first current and the second current to flow in the different directions, and controlling the first current and the second current to flow in the second pattern comprises controlling the first current and the second current to flow in the same direction. In some embodiments, controlling the first current comprises supplying the first control signal to the first switching circuit108electrically coupled to the first winding T1A of the first transformer102. Moreover, controlling the second current comprises supplying the second control signal to the second switching circuit110electrically coupled to the third winding T2A of the second transformer104. In some embodiments, the first and second currents are controlled to flow in the first direction by turning on a first high-side transistor Q1electrically coupled to the first winding T1A of the first transformer102and a second high-side transistor Q3electrically coupled to the third winding T2A of the second transformer104, and by turning off a first low-side transistor Q2electrically coupled to the first winding T1A of the first transformer102and a second low-side transistor Q4electrically coupled to the third winding T2A of the second transformer104. Moreover, the first current and the second current are controlled to flow in the second direction opposite to the first direction by turning off the first high-side transistor Q1and the second high-side transistor Q3and turning on the first low-side transistor Q2and the second low-side transistor Q4. In some embodiments, the first current is controlled to flow in the first direction and the second current is controller to flow in the second direction by turning on the first high-side transistor Q1and the second low-side transistor Q4and turning off the first low-side transistor Q2and the second high-side transistor Q3. Moreover, the first current is controlled to flow in the second direction and the second current is controller to flow in the first direction by turning off the first high-side transistor Q1and the second low-side transistor Q4and turning on the first low-side transistor Q2and the second high-side transistor Q3. In embodiments of the present disclosure, a computer-readable medium may be provided. The computer-readable medium comprises computer-readable instructions stored thereon. The computer-readable instructions cause a device to perform the method as described above when executed by the device. According to the embodiments of the present disclosure, the flow directions of the currents in the primary windings of the transformers are controlled, and thus the flow directions of the output currents in the secondary windings of the transformers are controlled. When the output currents in the secondary windings of the transformers are controlled to flow in a pattern such that the secondary windings of the transformers are electrically coupled in series with each other, the high voltage is output by the DC-to-DC conversion. Moreover, when the output currents in the secondary windings of the transformers are controlled to flow in another pattern such that the secondary windings of the transformers are electrically coupled in parallel with each other, the low voltage is output by the DC-to-DC conversion. In this way, the wide range of the output voltage and the constant output power may be achieved without an additional power stage, and the design of the individual transformer may be optimized. Therefore, the efficiency and the power density are improved with fewer components. While several details are contained in the above discussions, these should not be construed as limitations on the scope of the subject matter described herein, but rather as descriptions of features that may be specific to particular embodiments. The previous description of the disclosure is provided to enable any person skilled in the art to make or use the disclosure. Various modifications to the disclosure will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other variations without departing from the spirit or scope of the disclosure. Thus, the disclosure is not intended to be limited to the examples and designs described herein, but to be accorded the widest scope consistent with the principles and novel features disclosed herein. | 60,697 |
11863079 | Embodiments of the present disclosure and their advantages are best understood by referring to the detailed description that follows. It should be appreciated that like reference numerals are used to identify like elements illustrated in one or more of the figures. DETAILED DESCRIPTION An improved isolated switching power converter with secondary-side control is provided. The advantages include improved transient load response as well as improved integration with secondary functions such as synchronous rectification and zero-voltage switching. In addition, an improved secondary-to-primary communication is provided. The secondary-side controller regulates an output voltage by controlling the switching of a primary-side power switch responsive to both an error signal such as between the output voltage and a desired output voltage and an emulation of the primary-side current conducted through the primary winding of the transformer. The secondary-side controller controls both the power switch on time and the power switch off time through a control signaling through at least one isolation capacitor. An example flyback converter100is shown inFIG.1. A secondary-side controller U2includes a terminal coupled to an output voltage rail to sense an output voltage Vout. Secondary-side controller U2includes an emulation circuit105that emulates a primary-winding current conducted in a primary winding W1of a transformer T. The primary-winding current conducts to ground through a power switch transistor M1. A drain (D) of power switch transistor M1connects to a bottom terminal of primary winding W1. An upper terminal of primary winding W1connects to an input voltage rail that carriers a rectified input voltage. Secondary-side controller U2controls the cycling of power switch transistor M1through on and off control signaling conducted through at least one isolation capacitor CFB. A primary-side controller U1responds to a cycle-on control signal received over the isolation capacitor by switching on the power switch transistor M1. While the primary winding current conducts through the power switch transistor M1, secondary-side controller U2keeps a synchronous rectifier (SR) switch transistor off to prevent a secondary winding current from flowing through a secondary winding W2of the transformer T. The primary-side controller U1also responds to a cycle-off control signal received over the isolation capacitor CFB by switching off the power switch transistor M1. Secondary-side controller U2may then switch on the SR switch transistor so that the secondary-winding current may flow into the output voltage rail and charge an output capacitor Cout with the output voltage Vout. Secondary-side controller U2includes a source terminal (SR_S) for sensing a source voltage Vsr_s of the SR switch transistor. This source voltage ideally is at the same potential as ground for secondary-side controller U2. In addition, secondary-side controller U2includes a gate (G) terminal for driving the gate of the SR switch transistor to control whether the SR switch transistor is switched on or off. Secondary-side controller U2also includes a SR D terminal to sense the drain voltage (Vsr_d) of the SR switch transistor. The emulation circuit105uses the output voltage Vout and the drain voltage Vds to emulate the primary winding current. Based on the error between the output voltage Vout and a desired Vout and also the emulated primary winding current, secondary-side controller U2determines the timing of the cycle-on and cycle-off control signals. In addition, secondary-side controller U2may sense the output current such as through terminals to a sense resistor Rsense to assist in the timing of the cycle-on- and cycle-off control signals. Secondary-side controller U2may control the cycling of the power switch transistor M1using any suitable modulation technique such a pulse-width modulation or pulse frequency modulation. The cycle-on and cycle-off control signaling from the secondary-side controller U2to the primary-side controller U1is further illustrated inFIG.2Afor an implementation using a single isolation capacitor CFB. On the secondary side, the secondary-side control and feedback functionality briefly pulses a secondary-side set signal (Sec_set) when it is desired to switch on the power switch transistor M1. This pulsing is shown occurring at a time t0in a signal plot shown inFIG.3. In response to the secondary-side set signal, an isolation capacitor driver in the secondary-side controller U2pulses a voltage of a first plate (node A) of the isolation capacitor CFB. This pulsing of the node A voltage causes a voltage of a second plate (node B) of the isolation capacitor to briefly spike high at time t0. A CFB receiver in the primary-side controller U1detects this spiking of the node B voltage as the cycle-on control signal. In response to the spiking of the node B voltage, the CFB receiver asserts a primary-side set signal (Pri_set) that causes a gate driver in the primary-side controller U1to switch on the power switch transistor M1. The control and feedback functionality in the secondary-side controller U2then briefly pulses a secondary-side reset signal (Sec_reset) at a time t1in response to a determination that the power switch transistor M1should be cycled off. The isolation capacitor driver then discharges the node A voltage at time t1, which causes the node B voltage to briefly pulse negatively. The CFB receiver responds to this negative pulsing of the node B voltage by briefly pulsing a primary-side reset signal to cause the gate driver to discharge the gate voltage of the power switch transistor M1. The gate voltage (gate signal) of the power switch transistor M1thus stays charged from time t0to a time t1. An analogous signaling through the isolation capacitor CFB is then repeated so that the power switch transistor M1may be cycled on at a time t2to a time t3. To improve the noise immunity, a differential signaling may be used as shown inFIG.2B. The isolation capacitor CFB discussed with regard toFIG.2Ais replaced by a positive isolation capacitor CFB_P and a negative isolation capacitor CFB_N. These isolation capacitors are then driven differentially in an analogous fashion as discussed with regard to the single-ended driving of isolation capacitor CFB. An embodiment of the primary winding current emulation that uses a secondary-side sense resistor R_sec is shown inFIG.4. Referring again to flyback converter100ofFIG.1, note that the primary-side controller U1does not need an auxiliary winding to control the cycling of the power switch transistor due to the secondary-side control. Similarly, the primary-side controller does not need a sense resistor voltage from a primary-side sense resistor coupled between the source of the power switch transistor M1and ground to control the cycling of the power switch transistor due to the secondary-side control But these components are traditional in primary-side control of the power switch transistor M1and are thus shown for conceptual purposes inFIG.4. With regard to this traditional primary-side control, it would be conventional for a primary-side controller to sense the primary-side sense resistor voltage to determine when to switch off the power switch transistor M1. The behavior of the primary-side sense resistor voltage depends upon whether discontinuous conduction mode (DCM) or continuous conduction mode (CCM) operation is active. In DCM, the primary-side sense resistor voltage rises from zero when the power switch transistor M1is switched on because there is no primary winding current flowing at that time. But in CCM, the primary winding current has a non-zero initial value (Iinitial) when the power switch transistor M1is switched on. The peak primary-side sense resistor voltage occurs when the power switch transistor M1is about to be turned off. The on-time period for the power switch transistor may be represented by a variable T. The peak primary-side sense resistor voltage for CCM operation may be shown to equal [(Vin/L)*T+Iinitial)]*Rsense, where Vin is the rectified input voltage, L is the magnetizing inductance of the primary winding, and Rsense is the resistance of the primary-side sense resistor. Note that this equation is also applicable to DCM since Iinitial is zero in that case. The question thus becomes one of how to emulate this peak primary-side sense resistor voltage in the secondary-side controller U2. At the secondary-side before the power switch transistor M1is cycled on, the current through the secondary-side sense resistor R_sec and Iinitial are related through the turn ratio of the transformer, which in this embodiment is assumed to be N. To emulate this initial current Initial, the secondary-side controller U2may thus sample a voltage Vrsec across the secondary-side resistor R_sec just prior to switching on the power switch transistor M1. After scaling Vrsec by a factor −m to account for the turn ratio, the secondary-side controller U2may pre-charge a capacitor C to a voltage of Vrsec*(−m) to model the effects of the initial current Iinitial. For example, the secondary-side controller U2may include a switch S1that closes to couple the voltage Vrsec*(−m) to the capacitor C and then opens prior to the power switch transistor M1on time. When the power switch transistor M1cycles on, a difference between the SR switch transistor drain voltage (Vsr_d) and the output voltage Vout is proportional to the input voltage Vin. This difference voltage may be impressed across a resistor R (not illustrated) to form a current K*(Vsr_d−Vout)/R, where K is a proportionality factor. This current may then charge the capacitor C so that the current is integrated over time to emulate the peak primary winding current (which is proportional to the peak primary-side sense resistor voltage). A voltage Cramp across the capacitor C will thus ramp up according to the equation of Cramp=(K/RC)*Vin/T+(−mVrsec). This Cramp voltage emulates or provides an equivalent of the primary-side sense resistor voltage. By monitoring the Cramp voltage, the secondary-side controller U2may thus determine when the desired peak primary winding current has been reached and command for the power switch transistor M1to be cycled off accordingly. The secondary-side sense resistor R_sec introduces loss. To improve efficiency, it may be eliminated as shown inFIG.5. In such an implementation, the voltage Vsr_ds across the SR switch transistor may be sampled during CCM operation just before the power switch transistor M1is switched on. In CCM operation, the primary winding initial current Iinitial is proportional to the voltage Vsr_ds at this sampling time. The voltage Vsr_ds may then be scaled by the factor −m to pre-charge the capacitor C through switch S1as discussed with regard toFIG.4. During DCM operation, the SR switch transistor is off prior to when the power switch transistor M1is cycled on. In response to the SR switch transistor being already switched off prior to the power switch transistor M1on time, the secondary-side controller U2may provide a zero initial voltage to the capacitor C. With Cramp set to the initial voltage, the capacitor C may integrate the current (Vsr_d−Vout)/R as discussed with regard toFIG.4. The Cramp voltage to emulate the peak sense resistor voltage may thus be defined using the equation of Cramp=(K/RC)*Vin/T+(−m)Vsr_ds for embodiments in which the current Initial is estimated without a secondary-side sense resistor. Note that the sampling of the Vsr_ds voltage may be implemented in a variety of fashions. For example, the Vsr_ds voltage may be sampled while the SR switch transistor is on, scaled with the factor (−m) and held so as to be applied to the capacitor C after the SR switch transistor switches off. Note thatFIGS.4and5are merely illustrative and that other implementations may be used to emulate the primary winding current. For example, the two components (K/RC)*Vin/T and (−m)Vsr_ds need not be combined but may be applied separately to different nodes of the control loop to emulate the primary winding current. A circuit implementation600for the emulation of the primary winding current without the use of a secondary-side sense resistor is shown inFIG.6. The output voltage Vout is converted to a current by a resistor R to drive the drain of a diode-connected NMOS transistor M4that is in a current-mirror configuration with an NMOS transistor M3. The gate of transistor M3is thus connected to the gate of the diode-connected transistor M4. Both the sources of transistors M3and M4are connected to ground. The transistor M3will thus conduct a mirrored version of the current (Vout)/R. A drain of transistor M3connects to a drain of a diode-connected transistor M2having a source connected to ground. The drain voltage Vsr_d of the SR switch transistor is converted into a current (Vsr_d)/R by another resistor R to drive the drain of diode-connected transistor M2. The transistor M2will thus conduct a difference current of (Vsr_d−Vout)/R. Diode-connected transistor M2forms a current mirror with another NMOS transistor M1that has its source connected to ground and a gate connected to a drain/gate of diode-connected transistor M2. Transistor M1will thus conduct a mirrored version of the current (Vsr_d−Vout)/R. The drain of transistor M1connects to a drain of a diode-connected PMOS transistor P1having a source connected to a power supply voltage node. Transistor P1forms a current mirror with another PMOS transistor P3having a drain coupled to ground through a first plate of capacitor C. Transistor P3will thus conduct a mirrored version of the current (Vsr_d−Vout)/R to charge capacitor C with the Cramp voltage as discussed earlier. To provide the initial charge on the capacitor C, a differential amplifier605amplifies the Vsr_ds voltage across the SR switch transistor to charge the first plate of capacitor C1by a coupling through an NMOS transistor M6. Transistor M6functions as discussed with regard to switch S1. To switch off transistor M6, diode-connected transistor P1also forms a current mirror with a PMOS transistor P2. When the power switch transistor M1is cycled on, transistor P2will thus conduct a mirrored version of the current (Vsr_d−Vout)/R into a drain of an NMOS transistor M5that forms a current mirror with diode-connected transistor M4. Depending upon the relative sizes of transistors M5and P2, the drain voltage of transistor M5will thus begin to rise when the power switch transistor M1is cycled on. An inverter601couples between the drain of transistor M5and the gate of transistor M6so that transistor M6is cycled off when the power switch transistor M1is cycled on. Prior to being switched off, transistor M6will conduct responsive to the output of differential amplifier605such that the capacitor C is pre-charged with the initial (−m)Vsr_ds voltage. A flyback converter700with the secondary-side control disclosed herein is shown in more detail inFIG.7. Although not needed for the secondary-side regulation, a primary-side sense resistor Rpri is included so that primary-side controller U1may monitor for an overcurrent condition using an overcurrent protection circuit750. Depending upon the regulation scheme, secondary-side controller U2may include an oscillator (e.g., included within a control logic circuit725) that determines the on-time of the power switch transistor M1. A CFB driver730in secondary-side controller U2and a CFB receiver735in primary-side controller U1function as discussed regarding the differential signaling embodiment ofFIG.2B. Secondary-side controller U2includes a primary-winding current emulator705that emulates the primary winding current as discussed herein. In addition, secondary-side controller U2includes an error amplifier715that generates an error signal depending upon the whether a constant voltage (CV) or a constant current (CC) mode of operation is active. In CV, the error amplifier715compares the output voltage Vout to a reference or desired output voltage to generate the error signal. In CC, the error amplifier715compare the output current as sensed over a secondary-side sense resistor Rsns. Error amplifier715may be configured to integrate and/or filter the error signal so that it may be mapped into a threshold signal representing a threshold current. A comparator720compares an emulated signal representing the emulated primary winding current to the threshold signal representing the threshold current. When this comparison indicates that the emulated primary winding current has risen in amplitude to equal the threshold current, the control logic circuit725may generate the secondary-side reset signal so that CFB driver730may signal the primary-side controller U1accordingly to switch off the power switch transistor M1. The comparison in comparator720may also be adjusted according to a slope compensation710to suppress sub-harmonic oscillation and improve control loop stability. In primary-side controller U1, gate control logic740responds to the CFB receiver735to drive gate driver711. Should an over-current condition exist, gate control logic740may switch off the power switch transistor M1accordingly. Those of some skill in this art will by now appreciate that many modifications, substitutions and variations can be made in and to the materials, apparatus, configurations and methods of use of the devices of the present disclosure without departing from the scope thereof. In light of this, the scope of the present disclosure should not be limited to that of the particular embodiments illustrated and described herein, as they are merely by way of some examples thereof, but rather, should be fully commensurate with that of the claims appended hereafter and their functional equivalents. | 17,872 |
11863080 | DETAILED DESCRIPTION The circuitry of this disclosure comprises a single-ended primary-inductor converter (SEPIC) configured to perform both buck and boost converter functions to convert a DC supply voltage to supply a variety of DC loads. The SEPIC converter may be arranged as an active-clamped isolated SEPIC DC-DC power converter (ACISC) to configured to operate efficiently over a wide input voltage range. The selected output voltage for the DC electronic loads may be within the input voltage range. The transformer in the ACISC converter may also provide galvanic isolation to isolate an output terminal of the circuit from an input terminal of the power converter circuit. The circuitry arrangement and component selection of the ACISC of this disclosure cause the power converter to operate in resonant DCM mode in the megahertz (MHz) frequency range, which may provide improved power density over other circuit arrangements. Specifically, the arrangement of the ACISC of this disclosure may include a capacitor located near the transformer, with the values for the transformer and capacitor selected to ensure the circuit operates in resonant discontinuous-conduction-mode (DCM) at a selected frequency. In discontinuous-conduction-mode (DCM), the current may fall to zero during part of the switching cycle. Continuous-conduction-mode (CCM) may refer to a mode where the current in the output rectifier, e.g., an output diode, never goes to zero between switching cycles. In some examples, DCM may offer higher efficiency than CCM, because of a reduced reverse recovery loss on the output diode and a softer turn on of the power switch for the power converter. DCM may also provide advantages including use of a smaller transformer for the same output, when compared to CCM operation, as well as a smaller input inductor, and therefore a reduced footprint. In some examples, the transformer in the circuit of this disclosure may be a planar transformer, to further reduce the size of the circuitry. FIG.1Ais a block diagram illustrating an overview of an example power converter according to one or more techniques of this disclosure. System10includes ACISC power converter30supplied by a direct current (DC) source VDC-IN12and configured to supply load18. System10also includes loop controller20which may monitor the output power, e.g., voltage and/or current to load18and output control signals to ACISC power converter30to maintain the output power within desired limits. In some examples, loop controller20may also monitor the input voltage VDC-IN12. Loop controller20may be implemented as any type of power converter loop controller, e.g., voltage mode controller, use an additional primary winding, opto-isolators and so on and may in some examples include processing circuitry, such as a microprocessor, logic circuitry, or some combination. In some examples loop controller20may be a separate integrated circuit connected to ACISC power converter30. In the example ofFIG.1A, ACISC power converter30includes boost stage14and asymmetrical half-bridge stage16. For the power converter circuit of this disclosure, boost stage14may share some components with asymmetrical half-bridge stage16and are not simply two cascaded stages. Loop controller20may be configured to adjust the duty cycle of switches in half-bridge stage16to compensate for changes in the voltage provided by VDC-IN12to maintain the desired output voltage to load18, which may be one or more DC sub-nets. In the example in which VDC-IN12is a battery, the voltage provided to ACISC power converter30may increase as VDC-IN12is charged and decrease over time as VDC-IN12discharges to provide power. The duty cycle output by loop controller20may increase as VDC-IN12voltage decreases, and vice-versa. As described above, ACISC power converter30may be configured to operate in resonant DCM and may include circuitry, not shown inFIG.1A, to provide galvanic isolation between input terminal22and output terminal24, such as a coupled inductor, e.g., a transformer. The circuitry of ACISC power converter30may include a topology and component values for capacitors, inductors and so on to ensure zero voltage switching for switches in system10. In this disclosure, zero voltage switching (ZVS) means that the voltage across the switch is zero volts during a switch transition from OFF to ON. As with any circuitry, “zero” may mean exactly zero volts or some tolerance about zero volts that is approximately equal to zero volts. The amount of the tolerance may vary based on a specific application. In the example of field effect transistor (FET) switches, the tolerance about zero volts may be based on a forward diode voltage of the body diode for the FET. When the voltage across the switch exceeds the body diode voltage drop, the body diode may conduct, but with a higher RDSthan the RDS-ONof the switch. In this manner, switching the FET when the voltage across the switch is less than the body diode voltage drop may result in improved switching efficiency. FIG.1Bis a block diagram illustrating an example application of the power converter of this disclosure in a system with at least two power supplies for low power subnets. In the example of system100, VDC-LOWbus110may supply one or more DC subnets and receive power from VDC-HIGHsupply104as well as from VDC-LOWbattery102. An arrangement similar to system100may be useful in a variety of applications including weather stations, vehicles such as unmanned aerial vehicles, urban air mobility vehicles, automobiles, and so on, as well as other types of applications. VDC-HIGHsupply104may provide power to one or more high voltage loads105, such as an electric motor, and to VDC-LOWbus110via VDC-HIGHto VDC-LOWconverter108. VDC-HIGHto VDC-LOWconverter108may be any type of DC-DC converter configured to reduce voltage to supply the loads of VDC-LOWbus110. In the example of an electric or hybrid automobile, VDC-HIGHsupply104may be a 48 V battery and VDC-LOWbattery102may be a twelve volt back-up battery configured to provide power to VDC-LOWbus110via DC-DC ASISC converter106, in the event of a problem with VDC-HIGHto VDC-LOWconverter108or VDC-HIGHsupply104. DC-DC ASISC converter106is an example of ACISC power converter30described above in relation toFIG.1Aand may have the same functions and characteristics. In the example ofFIG.1B, VDC-LOWbus110supplies several different loads through a variety of power supplies, which may also be called power converters. In other examples, ACISC converter106may supply power for more or fewer loads than shown in the example ofFIG.1B. For system100, load114is a microcontroller, which runs on a voltage of one volt. Converter112may be any type of DC-DC converter configured to reduce the power supplied from VDC-LOWbus110to the one volt needed for the microcontroller. Similarly sensors118and126may receive power from VDC-LOWbus110through 3.3V converter116and 7.5V converter124respectively. A communication load122, which may power a communication bus, like a CAN bus, or a wireless communication system such as Wi-Fi, BLUETOOTH or another type of communication system, may receive power via five volt converter120. FIG.2is a schematic diagram illustrating an example circuit arrangement to implement the ACISC according to one or more techniques of this disclosure. System200is an example of system10and ACISC converter106described above in relation toFIGS.1A and1B, respectively, and may have the same or similar functions and characteristics. For example, similar to system10ofFIG.1, system200is an ACISC power converter that integrates boost stage214and asymmetrical half-bridge stage216and includes loop controller220. The physical structure of the example system200may include transformer210, which may also be described as coupled inductors, or a primary coil232and secondary coil234with a specified turns ratio, e.g., the ratio of the number of turns in the primary compared to the secondary turns. Transformer210may be configured to provide galvanic isolation between input terminal236and output terminal238. In the example ofFIG.2, transformer210is modeled as leakage inductance Lpe240and magnetizing inductance Lme242. Physically, primary coil232may connect to resonance capacitor Cc202and to reference node246. As modeled, a first terminal of leakage inductance Lpe240connects resonance capacitor Cc202to magnetizing inductance Lme242. Lme242connects a second terminal of Lpe240to reference node246. Similarly the primary side of windings249, which in the example of system200, connects across Lme242between reference node246and the second terminal of Lpe240. The secondary side of windings249connects between secondary reference node250and the anode of output diode D3252. In some examples, transformer210may comprise single primary and single secondary winding. Such an arrangement may provide advantages, such as reduced footprint, cost and complexity over other types of power converter circuits with multiple windings. In other examples, transformer210may include a second primary winding, for example, as part of the loop control circuitry. Transformer210, in the example ofFIG.2, is modeled by an equivalent primary-side leakage inductance (Lpe240), an equivalent primary-side magnetizing inductance (Lme242), and an equivalent turns ratio (N). The equivalent turns ratio for windings249, labeled “N” in this disclosure, may be different than the physical turns ratio of the actual transformer and is based on characterizing the physical transformer. Modeling the transformer may include taking measurements of the physical transformer to determine an equivalent turns ratio, N, as well as the equivalent magnetizing inductance, Lme242, and other parameters to be used in modeling the circuitry of system200. In the example ofFIG.2, output diode D3252may rectify output voltage vo(t)244. In other examples, synchronous rectification circuitry, e.g., a switch and control circuitry, not shown inFIG.2, may replace output diode D3252and perform the rectification of output voltage vo(t)244. The voltage across output diode D3252is v3(t)258and the current through the secondary windings, and through D3252is i3(t)256. Output capacitor Cout254connects between the cathode of D3252and reference node250. The cathode of D3252connects to output terminal238, which outputs vo(t)244, and load239connects between output terminal238and secondary reference node250. The ACISC converter of system200includes main switch Q1256and clamping switch Q2258. Main switch Q1256, which may also be referred to as primary switch Q1256, connects between reference node246and switch node260. The voltage across Q1256is switch node voltage vsw(t)262. Switch node260may also be referred to as switching node260in this disclosure. Clamping switch Q2258connects switch node260to reference node246through capacitor Cs266. In some examples capacitor Cs266may be referred to as a snubber capacitor. Snubber circuitry may help protect the switch from the energy stored in the leakage inductance of the transformer, and may reduce the EMI caused by resonance between the leakage inductance of the primary inductor and the parasitic capacitance of the switch. System200may be considered a SEPIC converter with an added active clamping circuit provided by Q2258and Cs266. The active clamping function may help control high voltage spikes in the components to avoid overstress of the components. The voltage across capacitor Cs266is vs(t)264. In the example ofFIG.2, boost stage214includes input inductor Lg272, main switch Q1256, clamping switch Q2258and capacitor Cs266. In other words, main switch Q1256, clamping switch Q2258, and capacitor Cs266are shared between boost stage214and asymmetrical half-bridge stage216. In some examples, because the value of Cc202may be selected with a small value of capacitance, to allow for the resonant operation, the voltage ripple across Cc202may be large enough to have a measurable impact on the gain. Simply assuming the voltage across Cc202is the average voltage, e.g., Vc=Vg, may result in an error between calculated/simulation performance and actual performance. In some examples, a calculated value for the voltage gain, M, of the ACISC converter of system200may be shown by the following equation: M=VoVg=k1/[N(1-D)](1-D)(1+λ)+k1[k2+k3NVo] where: k1=fsωrosinαsinβ(1-cosαcosβ)tanα2 k2=1+Zr1-D2fsLme(1-γ)cotα2 k3=ZrIoN(1+γ)cotα2 γ=1+λλcotα2tanβ2 and where λ as the ratio between the equivalent leakage inductance Lpe and the equivalent magnetizing inductance Lme, e.g., λ=Lpe/Lme, β=ωroDTs, α=ωr(1−D)Ts, fsis the switching frequency, D is the duty cycle of the main switch, Ts is the switching period, Zr is the characteristic impedance, ωris the natural resonant frequency when D3252is ON, ωrois the natural resonant frequency when D3252is OFF, and Vo is the desired output voltage. ωr=1LpeCc Zr=LpeCc Zro=Lpe+LmeCc ωro=1(Lpe+Lme)Cc The input voltage supply is Vg268, which is an example of source VDC-IN12and VDC-LOWbattery102described above in relation toFIGS.1A and1B. Input supply Vg268provides power to input terminal236. Input current ig(t)270also enters the power converter circuit of system200at input terminal236and flows through input inductor Lg272to switch node260. Resonance current ig(t)274flows to switch node260through resonance capacitor Cc202and leakage inductance Lpe240. Magnetizing current im(t)278flows through magnetizing inductance Lme242in the model of system200. In the example ofFIG.2, loop controller220may monitor the output power, e.g., output voltage vo(t)244and/or current to load239. In the example ofFIG.2, load239is depicted as an active load. In other examples. load239may be replace by a resistive load Based on the output voltage, loop controller220may output control signals c1204and c2206to the control terminals of the switches of ACISC power converter200to maintain the output power within desired limits. In the example ofFIG.2, loop controller220includes processing circuitry276, analog to digital converter278, discrete time compensator282, and digital pulse width modulation driver284. In some examples, loop controller220may also include logic circuitry. In other examples, loop controller220may comprise other components and arrangements, including opto-couplers, and similar components. Loop controller220may adjust dead time for switches Q1256and Q2258to ensure zero voltage switching for switches Q1256and Q2258, where ZVS means that the voltage across the switch is zero volts during a switch transition from OFF to ON, as described above in relation toFIG.1A. In other words, loop controller220may drive the control terminals of main switch Q1256(c1(t)204) and clamping switch Q2258(c2(t)206) to operate the power converter circuit with dead times DT2218and DT1217. During the dead time interval, e.g., dead time duration, both main switch Q1256and clamping switch Q2258are OFF (not conducting through the main D-S channel). The dead time interval is such that a voltage at switch node260transitions between a first voltage magnitude (e.g., Vs) and a second voltage magnitude (approximately zero volts), and vice versa, within the dead time. ZVS may be desirable, e.g., to turn on the switch before voltage at the switching node increases enough to cause the body diode to conduct, because in some examples, such as examples with GaN switches, body diode conduction may result in a decrease in converter efficiency, when compared to turning on the switch, as described above. In other words, main switch Q1256connects switch node260reference node246and loop controller220may switch ON Q1256when the voltage at switch node260is the same as the reference node246, e.g., approximately zero volts. As described above the voltage at switch node260is approximately zero volts when the voltage is within a specified tolerance of zero volts, e.g., some specific value based on the application for system200, a body diode forward voltage or some other tolerance value. Similarly, loop controller220may switch Q2258when the voltage across switch Q2258is approximately zero volts, or within a specified tolerance of zero volts, as described above. For example, loop controller220may switch ON clamping switch Q2258when the voltage at switch node260equals a voltage across snubber capacitor Cs266. In the example ofFIG.2, switch node voltage vsw(t)262transitions between 0V and Vs=Vg/(1−D) and vice-versa, as noted above. Vs is the average voltage across the snubber capacitor Cs266. Because the voltage ripple across Cs266is negligible, in the example of system200, vs(t)264can be assumed to be equal to the average voltage Vs, during the switching cycle. As with the description of “equals zero” described herein, “equals a voltage” may also include being within a specified tolerance of the voltage, e.g., a percentage, a range of values, and so on. The values for the components, along with the circuit topology, and dead time regulation may ensure ZVS for resonant DCM operation. Input inductance Lg272charges and stores energy when main switch Q1256is ON. When main switch Q1256is OFF, input inductance Lg272provides the stored energy to capacitor Cs266. The value of Lg272may be selected to work with the selected values for Cc202, Lpe240, magnetizing inductance Lme242as well as parasitic inductance, capacitance and resistance of circuitry of system200to achieve ZVS during resonant DCM operation. The input current, ig(t)270flows through the input inductor Lg272and through either Q1256or Q2258during the switching cycle. To achieve zero voltage switching for Q1256, the net current entering switch node260, e.g., iL(t)≙ig(t)+ir(t) should be in the range as specified by the following equation: iL(Ts)<-Ceq,QtDTVs where iL(Ts) is the value of the net current into switch node260at the end of the period Ts286, Ceq.Qis the charge equivalent capacitance at switch node260, and Vs is the starting voltage at switching node260at the beginning of the period. Switch node260should transition from the starting voltage Vs to the reference node voltage249within the period t DT to achieve ZVS across a switch. In the example ofFIG.2, loop controller220may provide two dead-times. For switch Q1256, the first voltage level may be Vs and switching node260should transition to a second voltage level of approximately zero volts for Q1256to switch with zero volts across Q1256, and the desired dead time for this transition may be found using the equation above for iL(Ts). Therefore, loop controller220may be configured to set the dead time for the transition for Q1256in the switching cycle to be at least tDT. Similarly, for clamping switch Q2258, the first voltage level may be zero volts and the second voltage level may be Vs, and the dead time may be determined by rearranging the equation above before switching Q2258. In the example of system200, Vs=Vg/(1−D), where D is the duty cycle of main switch Q1256. Loop controller220may be configured to adjust the duration of the dead time depending on the selected operating region and the selected values for components in the circuitry of system200, as well as the charge equivalent capacitance for switch node260. The charge equivalent capacitance Ceq.Qmay include any parasitic capacitance associated with switches Q1256, Q2258and other parasitic capacitance connected to switch node260. In some examples, reducing the magnitude of the total rms current entering the switching node may improve efficiency of circuit200by reducing the conduction losses of switches Q1256and Q2258. Converter circuit200operates in resonance DCM and stays in resonance during the switching period. The voltage of switch node260varies based on Vg268, D and Cs266, which varies from 0V (at249) to the voltage across the snubber capacitor Cs266, e.g., vs(t)264. The shape and timing of the voltage curve at switch node260may depend on the selected values for inductance connected to the switching node, as well as the charge equivalent capacitance Ceq,Qof switch node260. For the resonant DCM, the dead time duration is controlled by loop controller220to allow the power converter portions of circuit200to autonomously transition the voltage of switching node260to achieve ZVS for both main switch Q1256and clamping switch Q2258. In some examples, loop controller220may be configured to regulate a dead time that is as short as feasible, but set long enough for the circuit to autonomously transition the voltage of the switching node from zero to Vs and vice versa to ensure ZVS. Some examples of values may be found in table 1 below. In other examples, an ACISC according to this disclosure may have different values depending on the application, load requirements, input voltage, desired gain and other factors: TABLE 1Example Converter ParametersParameterSymbolExample ValuesInput InductanceLg2.4μHEquivalent primary leakage inductanceLpe75.2nHEquivalent magnetizing inductanceLme1.2μHEquivalent transformer turns ratioN0.97Resonant capacitorCc47.8nFClamping/Snubber CapacitorCs1μFOutput capacitorCo10μFOperating frequencyfs2MHZ In the example ofFIG.2, the ACISC power converter may operate at frequencies greater than one megahertz (MHz). In some examples, the voltage transition time may be in a few nano seconds. The amount of current available at switch node260at the switching instant for a switch may also affect the voltage transition time and the amount of needed dead time to charge and discharge Ceq,Q. In some examples the ACISC power converter may operate at frequencies greater than 1.6 MHz, e.g., two MHz, three MHz and so on. For automobile applications, operating at frequencies greater than 1.6 MHz may avoid electromagnetic interference (EMI) with AM and FM band, which may be desirable for electric or hybrid automobiles, e.g., to reduce the cost and complexity of shielding, damping and so on for frequencies in the AM and FM band. As described above, the selection of the components, along with the topology of the circuit may impact parasitic factors for switch node260and affect the resonance behavior of system200. For portions of the switching cycle magnetizing inductance Lme242may affect the resonance, and therefore the voltage behavior of nodes in system200. When output diode D3252is OFF, magnetizing inductance Lme242participates in the resonance value because the magnetizing inductance Lme242is connected in series with the leakage inductance Lpe240, when output diode D3252is OFF. When D3252is ON, constant voltage across magnetizing inductance Lme242is −N*Vout, where N is the modeled equivalent transformer turns ratio, described above in relation toFIG.1. Therefore magnetizing inductance Lme242does not participate in resonance and only Cc202and leakage inductance Lpe240contribute to the resonance. When Q2258is ON, assuming a negligible ripple across Cs266then Cs266does not participate in the resonance because Cs266has a “constant” voltage when Q2258is ON. In other words, the resonance, which affects the timing and shape of the voltage waveform at switch node260, may change through the switching cycle. The switching speed of a MOSFET may be affected by the geometry of the transistor, e.g., the size. The smaller a transistor, the less capacitance the transistor may have between the gate (control terminal) and the source, which may result in faster switching speed compared to larger transistors. In this disclosure, switching describes changing the state of the transistor from closed, or an ON state to an open switch or OFF state, and vice versa. Also the output impedance of the source driving the base or the gate of the transistors may impact switching speed. Because the control terminal of a switch may include some amount capacitance to be charged or discharged, a large output impedance of the source may slow down the charging process and the switching speed. Charge is capacitance multiplied by voltage, (Q=CV), so switching speed may also depend on the gate capacitance and the voltage. In some examples, the semiconductor components of the ACISC of this disclosure may include Gallium Nitride (GaN) components. GaN is a high electron mobility semiconductor (HEMT) and GaN transistors may provide faster switching and smaller size than silicon MOSFETs. In some examples, the ACISC of this disclosure may allow faster switching than other circuit arrangements, and therefore higher efficiency, and smaller circuit board space needed to implement the circuit. GaN transistors may provide a low inductance, low resistance, smaller and low-cost packages, when compared to other types of semiconductor components. In addition, GaN transistors may offer higher performance in both hard switching and soft-switching applications, when compared to other types of components. In the example of an electric vehicle application, any improvement in efficiency may increase the time required between charging sessions, which may be beneficial to a user of an electric vehicle. The ACISC topology and operation of this disclosure may provide advantages compared to other types of SEPIC circuits. Some examples of isolated SEPIC rectifier may suffer from the disadvantage of low efficiency caused by hard switching (this is why other types of SEPIC circuits are not able to operate at such high switching frequencies), diode reverse recovery, transformer proximity losses, and losses due to energy stored in the transformer's leakage inductance. In contrast, the active clamp topology, and operation details, e.g., switching times, may provide advantages over other SEPIC circuits, including improved the efficiency of SEPIC and clamp the turn-off voltage spike on transformer210. FIG.3includes time graphs illustrating example circuit performance for the ACISC of this disclosure. The time graphs ofFIG.3show examples of some signals of the circuits described above in relation toFIGS.1A,1B and2. The voltages, currents and components listed inFIG.3correspond similar voltages, currents and components described above in relation toFIG.2. To show the trend of the circuit waveforms, the example ofFIG.3assumes that the loop controller drives the main switch and clamping switch in a complementary pattern, without a dead time, e.g., when Q1is ON, then Q2is OFF, as shown by Q1gate312and Q2gate314. In other examples, as described elsewhere in this disclosure, the loop controller may output the gate control signals Q1gate312and Q2gate314, which may include a dead time to allow the voltage to transition so that the switches may operate with ZVS. Input current ig(t)302is linear increasing from ig1304to ig2306and linear decreasing through ig3308back to ig1310. In the example ofFIG.3, D is the duty cycle of main switch Q1256and Ts is the switching period. The voltage of the switching node may vary based on the selected values for inductance connected to the switching node, as well as the charge equivalent capacitance Ceq,Q, of the switching node, as described above in relation toFIG.2. In some examples, the input current ig(t) may cross the average magnitude of input current described by the following equation: Ig=1η×VoIoVg where η is the efficiency of the ACISC converter, Vo is the desired output voltage, Io is the output current and Vg is the input voltage. The voltage v c (t)316across the resonance capacitor, e.g., Cc202ofFIG.2varies over the switching cycle and crosses the magnitude of the input voltage Vg. The resonance current ir(t)320passes through the resonance capacitance and the leakage inductance, e.g., Lpe240and Cc202ofFIG.2. Resonance current ir(t)320and magnetizing current im(t)322increase in a resonant manner. When the natural resonant frequency is much less than 2πfs(ωro<<2πfs), then the currents may increase almost linearly. Magnetizing current im(t)322crosses the average magnitude of magnetizing current described by the following equation: Im=1N×Io In the example ofFIG.3, resonance may always be present in each subinterval. In the subinterval from 0 to DTs, both Lpe and Lme (depicted inFIG.2) participate in the resonance because D3is OFF and currents ir(t)320and im(t)322increase in a resonant manner. The currents may increase almost linearly if the following condition is verified: ωro<<2*pi*fs, where ωro=1/sqrt{(Lpe+Lme)Cc}, also described above in relation toFIG.2. The output diode conducts (D3ON305) only during D2Ts324and not during the rest of the switching interval (D3OFF303and D3OFF307). Waveforms ir(t)320and im(t)322are equal during the first interval when the diode D3is OFF, because Lme is connected in series with Lpe. From DTs388to (D+D2)Ts only Lpe participates to the resonance because D3is ON so there is a constant voltage across Lme. The current ir(t)320, which is different from im(t)322shows the resonance behavior during D2Ts324shown inFIG.3. From (D+D2)Ts to Ts, both Lpe and Lme participate to the resonance because D3is OFF, as described above as well as in relation toFIG.2. If the resonant current ir(t)320, during its oscillation, becomes equal to the magnetizing current im(t)322before the end of the switching period, as shown inFIG.3, the rectifier diode D3turns OFF, thus making subinterval D3Ts326to appear, as inFIG.3. The shape of ir(t)320shown inFIG.3represents one example of resonant DCM mode. The shape of ir(t)320may have a different (but similar) trend if we choose different values for the components of the ACISC, e.g., as depicted inFIG.2. Note that because output diode D3only conducts for a portion of the (1−D)Ts interval then the converter operates in DCM. In other examples, if the output diode D3conducts for the entire (1−D)Ts interval (D2Ts=(1−D)Ts), the converter operates in CCM. FIG.4is a time chart illustrating example resonance and ZVS operation for an example power converter circuit according to one or more techniques of this disclosure. Similar toFIG.3, the time graphs ofFIG.4show examples of some signals of the circuits described above in relation toFIGS.1A,1B and2. The voltages, currents and components listed inFIG.4correspond similar voltages, currents and components described above in relation toFIGS.2and3. As described above in relation toFIG.3, input current ig(t)470is linear increasing and linear decreasing. The loop controller may control the switches to include dead time416and dead time418, as well as dead time415, in which both the main switch Q1and clamping switch Q2are switched OFF, as indicated by gate control signals Q1gate412and Q2gate414. Switch node voltage vsw(t)462fully transitions from a first voltage magnitude (e.g., Vs464) to a second voltage magnitude (e.g., zero volts) within the dead time periods416and418. This allows the main switch Q1to be switched ON with ZVS (402and406), because the voltage at the switch node approximately equals the voltage at the reference node, as described above in relation toFIG.2. As described in detail herein, the term “equals” may refer to equality to within a tolerance of value, or a situation where values are “approximately equal.” In other words, the switch voltage may equal zero within manufacturing, measurement and circuit performance tolerances. As shown byFIG.4, vsw(t)462varies about zero volts, but the variance is small compared to the other operating voltage magnitudes for vsw(t)462and may thus be within circuit performance tolerances of zero for this application. Similarly, dead time415(DT1) allows time for vsw(t)462to transition from a first voltage magnitude, zero volts in the example ofFIG.4, to a second voltage magnitude of approximately Vs464. As with the description of main switch Q1above,FIG.4shows vsw(t)462varying about Vs464during ZVS transition404, but may be within circuit performance tolerances of Vs464for this application. Therefore, as described above in relation toFIG.2, the voltage across clamping switch Q2may be considered zero volts during switching, and therefore provide efficiency and other performance advantages as described above. The current at the switch node includes a sum of currents injected into SW node from, for example, ig(t)470and ir(t)474. At the ZVS transitions400and404, both ig(t)470and ir(t)474are at high electrical current magnitude, when compared to ZVS transitions402and406. With more available current at the switch node, the circuit may charge (or discharge) the charge equivalent capacitance Ceq,Qof the switch node, as described above in relation toFIG.2. Therefore the voltage transitions from zero volts to, for example Vs464in the example ofFIG.4, may have a faster slew rate, e.g., a faster voltage transition time when compared to the voltage transitions from Vs464to zero volts. In other words, the switch node may have lower current, so more time may be needed from for the transition from high to low to charge and discharge the equivalent capacitance, Ceq,Q. As described above in relation toFIGS.1A-3, the ACISC of this disclosure may include components so that the switching node has time to autonomously complete the transition from zero volts to Vs and vice versa. The topology, component selection and loop controller settings create the condition that the switching node is within a tolerance of zero volts across the main switch, and the clamping switch before performing switching operation. In other words, the controller may be configured with the duration of dead times (DT2) as depicted by416, and418for main switch Q1and dead time (DT1) for clamping switch Q2as depicted by415, while the component selection and topology ensure the switching node voltage completes the transition within the dead time. FIGS.5A and5Bare graphs illustrating a comparison of efficiency for ACISC to a flyback power converter configured for similar low-power applications. The graphs ofFIGS.5A and5Bcompare the efficiency of an example GaN-based active-clamped isolated SEPIC converter operating at two MHz, with that of a 400 kHz silicon-based comparable flyback converter intended for the same application and having the same input/output voltage and rated current specifications. The comparison, carried out as a function of both input voltage Vg and output current Io, reveals a peak efficiency profile of the designed ACISC converter of ˜87.15% at Vo=12V and Io=600 mA, which is approximately 7.2% better than the comparable flyback converter, and the ACISC operates with a fivefold increase in switching frequency. Improved efficiency may be desirable for many applications, including system powered by battery power. Improved efficiency may result in longer operating life on a single charge and fewer recharge cycles when compared to systems with less efficient power converters. FIG.6is a flowchart illustrating an example operation of the ACISC of this disclosure. Unless otherwise noted, the blocks ofFIG.6will be described in terms ofFIG.2. The ACISC of this disclosure may receive, at input terminal236, power from Vg268within a predetermined voltage range (90). The voltage range may depend on the type of power source. A DC-DC converter may output power within a narrow voltage range when compared to a battery, which may output power across a wider range, depending on the depth of discharge (DoD) of the battery. For example, a nominal 12V lead-acid battery may output power between 14 V−10 V, and in some examples between 18 V−6 V. The topology of system200that may isolate output terminal238from input terminal236with galvanic isolation circuitry (92). The galvanic isolation circuitry, e.g., transformer210, may be modeled as including leakage inductance Lpe240and magnetizing inductance Lme242. The ACISC may operate in a resonant discontinuous conduction mode (DCM) based on inductance values of equivalent leakage inductance Lpe240, and equivalent magnetizing inductance Lme242as well as a capacitance of resonant capacitor Cc202. Capacitor Cc202is in series with the leakage inductance and the magnetizing inductance (94). Selecting main switch Q1256, clamping switch Q2258and the circuit topology depicted in system200may determine the value of leakage inductance Lpe240as well as the contributions to Ceq, Q at switch node260from the switches. Further selection of the value of resonant capacitor Cc202, as well as other components and circuit arrangement, may determine the resonance behavior of system200. For example, in system200, resonant capacitor Cc202connects switch node260to the galvanic isolation circuitry, transformer210. Also, input inductor Lg272connects input terminal236to switch node260and main switch Q1256connects to clamping switch Q2258at switch node260, all of which may contribute to the resonance behavior at different times in the switching cycle. As described above in relation toFIGS.2,3and4, when output diode D3is ON, then Lme242does not participate in the resonance because of the a “constant” voltage (=−N*Vo) across Lme242. When D3is OFF, the circuit does not transfer power to the output terminal238. Lme242participates in the resonance because it is connected in series with Cc202and Lpe240. Loop controller220may be configured to switch both main switch Q1256and clamping switch Q2258OFF during a dead time such that a voltage at the switch node transitions between a first voltage magnitude and a second voltage magnitude within the dead time, as described above in relation toFIG.4(96). In one or more examples, the functions described above may be implemented in hardware, software, firmware, or any combination thereof. For example, processing circuitry276ofFIG.2and loop controller20ofFIG.1may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on a tangible computer-readable storage medium and executed by a processor or hardware-based processing unit. Instructions may be executed by one or more processors, such as one or more digital signal processors (DSPs), general purpose microprocessors, application specific integrated circuit (ASIC), Field programmable gate array (FPGAs), or other equivalent integrated or discrete logic circuitry. Accordingly, the term “processor,” and “processing circuitry” as used herein, such as may refer to any of the foregoing structure or any other structure suitable for implementation of the techniques described herein. Also, the techniques could be fully implemented in one or more circuits or logic elements. The techniques of this disclosure may be implemented in a wide variety of devices or apparatuses, including a wireless handset, an integrated circuit (IC) or a set of ICs (e.g., a chip set). Various components, modules, or units are described in this disclosure to emphasize functional aspects of devices configured to perform the disclosed techniques, but do not necessarily require realization by different hardware units. Rather, as described above, various units may be combined in a hardware unit or provided by a collection of interoperative hardware units, including one or more processors as described above. One or more aspects of this disclosure may also be described in the following numbered clauses. Clause 1: A power converter circuit comprising a transformer configured to isolate an output terminal of the circuit from an input terminal of the circuit, wherein the transformer includes a magnetizing inductance and a leakage inductance; a resonance capacitor in series with the leakage inductance, wherein a value of capacitance for the resonance capacitor and a value for the leakage inductance configure the circuit to operate in resonant discontinuous conduction mode (DCM); a clamping switch; a main switch and a loop controller configured to: drive control terminals of the main switch and the clamping switch, operate the power converter circuit with a dead time: wherein both the main switch and the clamping switch are off, and such that a voltage at the switch node transitions between a first voltage magnitude and a second voltage magnitude within the dead time. Clause 2: The circuit of clause 1, wherein the main switch connects the switch node to a reference node, wherein the loop controller is configured to switch ON the main switch when the voltage at the switch node is approximately zero volts, wherein the voltage at the switch node is approximately zero volts when the voltage at the switch node is within a specified tolerance of zero volts. Clause 3: The circuit of any of clauses 1 and 2, wherein a resonant frequency of the resonant DCM is at least one megahertz (MHz). Clause 4: The circuit of any of clauses 1 through 3, wherein the clamping switch connects the switch node of the circuit to a reference node through a snubber capacitor. Clause 5: The circuit of clause 4, wherein the loop controller is configured to switch ON the clamping switch when the voltage at the switch node equals a voltage across the snubber capacitor. Clause 6: The circuit of any of clauses 1 through 5, wherein an output voltage at the output terminal is rectified by an output diode. Clause 7: The circuit of any of clauses 1 through 6, further comprising an input inductor, wherein a value of the input inductor is configured to achieve zero voltage switching (ZVS) for the main switch and the clamping switch. Clause 8: The circuit of any of clauses 1 through 7, wherein the main switch and the clamping switch are each implemented with a Gallium Nitride (GaN) metal-oxide-semiconductor field-effect transistor (MOSFET). Clause 9: A system comprising a power supply configured to produce a voltage within a predetermined voltage range; a power converter circuit includes a transformer configured to isolate an output terminal of the circuit from an input terminal of the circuit, wherein the transformer includes a leakage inductance and a magnetizing inductance, and wherein the input terminal is configured to receive from the power supply, the voltage within the predetermined voltage range; a resonance capacitor in series with the leakage inductance, wherein a value of capacitance for the resonance capacitor, a value for the magnetizing inductance and a value for the leakage inductance configure the circuit to operate in resonant discontinuous conduction mode (DCM); a clamping switch; a main switch; and a loop controller configured to: drive control terminals of the main switch and the clamping switch, operate the power converter circuit with a dead time: wherein both the main switch and the clamping switch are off, and such that a voltage at the switch node transitions between a first voltage magnitude and a second voltage magnitude within the dead time. Clause 10: The system of clause 9, wherein the clamping switch connects a switching node of the power converter circuit to a reference node through a snubber capacitor. Clause 11: The system of any of clauses 9 and 10, further comprising an input inductor, wherein a value of the input inductor is configured to achieve zero voltage switching (ZVS) for the main switch and the clamping switch. Clause 12: The system of any of clauses 9 through 11, wherein an output voltage at the output terminal is rectified by an output diode. Clause 13: The system of any of clauses 9 through 12, wherein the main switch and the clamping switch are each implemented with a Gallium Nitride (GaN) metal-oxide-semiconductor field-effect transistor (MOSFET). Clause 14: The system of any of clauses 9 through 13, wherein the power supply provides power to one or more electrical loads of a vehicle; and wherein the power converter circuit is configured to supply an output voltage at a specified magnitude without regard to a magnitude of the received voltage within the predetermined voltage range. Clause 15: The system of clause 14, wherein the vehicle is an automobile. Clause 16: A method comprising receiving, at an input terminal, power within a predetermined voltage range; isolating an output terminal from the input terminal with galvanic isolation circuitry, wherein the galvanic isolation circuitry comprises a leakage inductance; operating in a resonant discontinuous conduction mode (DCM) based on an inductance value of the leakage inductance, an inductance value for the magnetizing inductance and a capacitance of a resonant capacitor in series with the leakage inductance; wherein the resonant capacitor connects a switch node to the galvanic isolation circuitry, wherein an input inductor connects the input terminal to the switch node, wherein a main switch connects to a clamping switch at the switching node, and switching both the main switch and the clamping switch OFF during a dead time such that a voltage at the switch node transitions between a first voltage magnitude and a second voltage magnitude within the dead time. Clause 17: The method of clause 16, wherein the galvanic isolation circuitry comprises a transformer. Clause 18: The method of any of clauses 16 and 17, wherein the clamping switch connects the switch node of the power converter circuit to a reference node through a snubber capacitor. Clause 19: The method of any of clauses 16 through 18, wherein the main switch and the clamping switch are each implemented with a Gallium Nitride (GaN) metal-oxide-semiconductor field-effect transistor (MOSFET). Clause 20: The method of any of clauses 16 through 19, wherein an output voltage at the output terminal is rectified by a synchronous rectification circuit. Various examples of the disclosure have been described. These and other examples are within the scope of the following claims. | 45,647 |
11863081 | DETAILED DESCRIPTION The present disclosure proposes a Single/Multi-phase Power Management Integrated Circuit (PMIC) built on a silicon substrate, with coil layers processed on top of the Integrated Circuit (IC) metal layers. Each phase of the Multi-phase PMIC comprises a single output inductor. At least one of the power supply connection pads (Vss or Vdd) is placed in the center gap of the coil layers, to minimize parasitics, where the coil layers are created by electroplating thick copper/silver in a spiral resist template, formed by lithography. The key elements of the disclosure include an even number of spiral inductors (air and magnetic core based) each having a gap at its center. Using an even number of coils enables far field cancellation, resulting in lower electromagnetic interference (EMI). Further, layout of the inductors is such that they are all placed in a row in sets of two. The inductors in each set must have negative coupling, where two adjacent sets have positive coupling, to reduce EMI. The spiral inductors can be implemented as a racetrack (elongated spiral) design inFIG.1100, or a circular spiral design inFIG.2200.FIG.1element110illustrates the elongated spiral inductor, and120the gap at the center of the elongated spiral inductor.FIG.2element210illustrates the circular spiral inductor, and220the gap at the center of the circular spiral inductor. The coil layers of the disclosure are above the PMIC formed in and on the silicon substrate, and the supply interconnection pads (for Vss or Vdd) are placed in the center gap of the spiral inductor coils. With this construction, the parasitic interconnections between the spiral coil and supply pads for Vss or Vdd are minimized, resulting in lower losses and improved efficiency. Additionally, the gap at the center of the spiral coil reduces the path length for the flux to travel, hence increasing the inductance, and creating higher quality factor (Q-factor) for the device (in terms of inductance/DC resistance). FIG.3shows300, a schematic of a single or a multi-phase integrated voltage regulator (IVR) circuit, constructed with spiral inductor coils320, built on top of PMIC silicon330. The improved arrangement of the coils, with interconnect pads310at the center of the inductor coils, can reduce the parasitic resistance by up to 30-50% of the coil resistance, compared to other implementations of the inductor coil, built on top of PMIC silicon, without a gap at the center of the inductor coil. FIG.4shows400, the pad arrangement at the center of an spiral inductor coil420. The primary reason for the reduction in parasitic resistance in the disclosure is the possibility of connecting the supply rails430to the inside pad410of the inductor, with the shortest possible interconnect. In most inductor designs, the inductor pads are either on opposite sides (Solenoid or Stripline) or on the same side (Toroid or spiral), which means connections to supply pads need a longer interconnect path, thereby leading to a higher resistance. Additionally, for these designs, the presence of a magnetic core can potentially make it impossible to have a gap at the center of the device, which can minimize the interconnect path. In a comparison between a solenoid (magnetic core based) inductor and an air coil spiral inductor, both built on silicon, experimental results confirm a 5-10 mΩ additional interconnect resistance is needed for the case of the solenoid inductor connected to its supply pads. This is a significant issue, as the inductor resistance on it's own is 20 mΩ, and with the additional supply track, the total inductor resistance increases to 25-30 mΩ, or 30-50% of overall inductor resistance. The additional 5-10 mΩ resistance is not present in the air coil spiral inductor, as the pads can be placed in the center of the coil. Further, the additional supply track resistance will impact the overall Q-factor (in terms of AC loss performance merit) of the device at higher frequencies. The Q-factor of the inductor is given as2*π*Freq*Inductance/AC Resistance, where the additional interconnect path contributes to the overall AC resistance of the device, resulting in lower Q-factor of the device with reduced efficiency. FIG.5Ashows500, a cross-section of an integrated voltage regulator (IVR) circuit, with a Power Management Integrated Circuit (PMIC) and metal layers1-6(though other metal layer counts could also be used), with inductor metal layers (with air coils) built over the PMIC. It shows the complete cross-section of a typical PMIC in and on a silicon substrate, with an integrated coil processed on top of the IC metal layers. The530Metal layers are PMIC layers, and the520AP layer is the interconnection/rerouting layer to the inductor and capacitor from the PMIC. Finally, the510metal layers (PPI1, PPI2) on top represent the inductor layers, which in this case are a two-level spiral inductor, with a gap at the center for the supply pads. Note the inductor can be formed by a single metal layer as well. FIG.5Bshows550, how a magnetic core wrapping the windings is formed. The fabrication of the inductor takes place after the completion of all the metal layers required in the PMIC and is described below. FIGS.5C and5Dshow a circuit drawing showing a typical PMIC connected to an inductor, the inductor output, and VSS/VDD connections. The inductor and power switches of a buck configuration575inFIG.5C, and a boost configuration585inFIG.5D, are shown. For conventional switch-mode power supplies (SMPS), the power switches and control circuit are integrated within the PMIC. For the present invention, an air-core inductor L is integrated within the PMIC as well. FIG.6shows600, Step1after formation of the PMIC, and prior to the fabrication process of the inductor layers, where520is the AP layer for interconnecting/rerouting to the inductor and capacitor from the PMIC, and530is the PMIC metal layers. This step is for integrating the spiral inductor coil, by depositing a metal interconnection or rerouting AP layer, which connects the PMIC to passive elements, such as the inductors and capacitors. FIG.7shows700, Step2for depositing and patterning Copper (or other metals such as Silver PPI1) layers710, and the dielectric layer (PM1)715, in the fabrication process of the inductor layers. Following the deposition of the AP layer in Step1, Step2is used to deposit the first metal layer of the inductor (PPI1) and the supply pads. FIG.8shows800, Step3for depositing dielectric layers PM2-1, PM2-2816, in the fabrication process of the inductor layers. Step3is used to deposit the non-conducting dielectric layers PM2-1, PM2-2, and to isolate the first inductor metal layer PPI1, and then form openings for the inductor I/O pads and supply pads. FIG.9shows900, Step4for depositing Copper (or other metal PPI2) layers918and the dielectric layer (PM3)917, in the fabrication process of the inductor layers. Step4is used to deposit the second metal layer PPI2 of the inductor and supply pads, along with filling in the metal vias which connect the two metal layers PP1 and PP2. The top metal layer is isolated, using a second dielectric layer PM3. FIG.10shows1000, Step5for depositing the Under Bump Metallurgy (UBM)1021and solder ball1019, in the fabrication process of the inductor layers. Step5is the final step used to deposit the metallization on the I/O pads of the entire structure and bump the pads with solder balls. FIG.11is flow chart1100, of a method for constructing an Integrated Voltage Regulator, with an Integrated Air-Core Inductor. The steps include1110, forming an integrated voltage regulator, in and on a silicon substrate. The steps include1120, forming an integrated air-core inductor, over the integrated voltage regulator, where the air-core inductor has a spiral shape and gap at a center of the spiral shape. The steps also include1130, forming at least one supply connection pad of the integrated voltage regulator in the gap. The main advantage of one or more embodiments of the present disclosure include reduced interconnect parasitics and lower Electromagnetic Interference (EMI). The use of integrated air-core coils in the integrated voltage regulator allows the entire device to either be packaged or monolithically built with the application, that it powers, for example in a mobile processor. This enables significant reduction (if not elimination) of Power Distribution Network (PDN) circuitry, resulting in a smaller Bill of Materials (BOM) and overall improved efficiency. While particular embodiments of the present disclosure have been illustrated and described, it will be understood by those skilled in the art that various changes in form and details may be made without departing from the spirit and scope of the invention. | 8,827 |
11863082 | The reference symbols used in the drawings, and their meanings, are listed in summary form in the list of reference symbols. In principle, identical parts are provided with the same reference symbols in the figures. PREFERRED EMBODIMENTS OF THE INVENTION While the invention is susceptible to various modifications and alternative forms, specific embodiments thereof are shown by way of example in the drawings and will herein be described in detail. It should be understood, however, that the drawings and detailed description thereto are not intended to limit the invention to the particular form disclosed, but on the contrary, the intention is to cover all modifications, equivalents, and alternatives falling within the spirit and scope of the present invention as defined by the appended claims. Note, the headings are for organizational purposes only and are not meant to be used to limit or interpret the description or claims. Furthermore, note that the word “may” is used throughout this application in a permissive sense (i.e., having the potential to, being able to), not a mandatory sense (i.e., must).” The term “include”, and derivations thereof, mean “including, but not limited to”. The term “connected” means “directly or indirectly connected”, and the term “coupled” means “directly or indirectly connected”. As shown inFIG.1, a power distribution network1includes a first power source10, a first network11, a second power source12, a second network13, and a power converter14. For example, as shown inFIG.1, both of the first power source10and the second power source12are of AC power source, and the first network11and the second network13are of AC network distributing the AC power respectively to/from the first power source10and the second power source12. Telecommunication (e.g. GOOSE) between a controller144of the power converter14and each relay (IED) is arranged. The power converter14has a power conversion circuit140, and the power conversion140has a first terminal set141and a second terminal set142. The power conversion circuit140is configured to convert power input via one of the first terminal set141and the second terminal set142and output the converted power via the other of the first terminal set141and the second terminal set142. The power conversion circuit140of the power converter14uses an AC to AC power conversion circuit topology. The first terminal set141of the power conversion circuit140is electrically coupled to the first network11which in turn is electrically coupled to the first power source10, and the second terminal set142of the power conversion circuit140is electrically coupled to the second network13which in turn is electrically coupled to the second power source12. The power conversion circuit140can use a back-to-back converter system with two voltage source converters U1, U2electrically connected by a DC link C. and both of the two voltage source converters U1, U2can use power conversion circuits for converting between AC and DC power with their DC sides are electrically coupled by the DC link C. The skilled person should understand that as an alternative, a topology of direct AC to AC power converter, such as matrix converter, can be used for the power conversion circuit140. FIG.2illustrates the power converter according to an embodiment of present invention. As shown inFIG.2, the voltage source converter U1uses power semiconductor switches both turn-on and turn-off can be controlled, such as the insulated-gate bipolar transistor (IGBT). As a result, they can be used to make self-commutated converters. In such converters, the polarity of DC voltage is usually fixed and the DC voltage, being smoothed by a large capacitance of the DC link C, can be considered constant. The controllability gives many advantages, notably the ability to switch the IGBTs on and off many times per cycle in order to improve the harmonic performance. Being self-commutated, the converter no longer relies on synchronous machines in the AC system for its operation. The voltage source converter U1can use a six pulse bridge in which the power semiconductor switches use IGBTs with inverse-parallel diodes. With the help of the DC link C behaves like a DC smoothing capacitor, the voltage source converter can output two voltage levels at the AC output of each phase that correspond to the electrical potentials of the positive and negative DC terminals. Pulse-width modulation (PWM) can be used to improve the harmonic distortion of the converter. As an alternative, the voltage source converter U1can be built with some form of multilevel converter, most commonly the Modular Multilevel Converter (MMC), in which each valve consists of a number of independent converter submodules, each containing its own storage capacitor. The IGBTs in each submodule either bypass the capacitor or connect it into the circuit, allowing the valve to synthesize a stepped voltage with very low levels of harmonic distortion. Similarly, the other voltage source converter U2can use a six pulse bridge in which the power semiconductor switches use IGBTs with inverse-parallel diodes. With the help of the DC link C behaves like a DC smoothing capacitor, the voltage source converter can output two voltage levels at the AC output of each phase that correspond to the electrical potentials of the positive and negative DC terminals. Pulse-width modulation (PWM) can be used to improve the harmonic distortion of the converter. As an alternative, the voltage source converter U1can be built with some form of multilevel converter, most commonly the Modular Multilevel Converter (MMC), in which each valve consists of a number of independent converter submodules, each containing its own storage capacitor. The IGBTs in each submodule either bypass the capacitor or connect it into the circuit, allowing the valve to synthesize a stepped voltage with very low levels of harmonic distortion. Referring back toFIG.1, the power converter14further includes at least one measurement unit143. Measurement unit143in accordance with the invention is arranged at a measuring point of the first network11adjacent to one end thereof. Possible fault locations, within and outside of the protection zone L1, L2, L3, L4, are separated by circuit breakers S1, S2, S3, S4inserted in the first network11. As shown inFIG.1, the measuring unit143is arranged between one of the circuit breakers S1, S2, S3, S4and the first terminal set141of the power conversion circuit140. The measurement unit143can be a voltage sensor, a current sensor or a combination thereof. The measurement unit143can be used for measuring value of phase voltage at the measuring point of the first network11; besides, the measurement unit143can be used for measuring value of phase current at the measuring point of the first network11. The skilled person shall understand that as an alternative, the measurement unit143can be used for measuring line voltage and/or line current at the measuring point. With this configuration, a fault occurring on the first network11can be located. If it is desirable to locate a fault occurring on the second network13, the power converter14further includes another measurement unit145. Measurement unit145in accordance with the invention is arranged at a measuring point of the first network13adjacent to one end thereof. Possible fault locations, within and outside of the protection zone L5, L6, L7, L8, are separated by circuit breakers S5, S6, S7, S8inserted in the second network13. As shown inFIG.1, the measuring unit145is arranged between one of the circuit breakers S5, S6, S7, S8and the second terminal set142of the power conversion circuit140. The measurement unit145can be a voltage sensor, a current sensor or a combination thereof. The measurement unit145can be used for measuring value of phase voltage at the measuring point of the second network13; besides, the measurement unit145can be used for measuring value of phase current at the measuring point of the second network13. The skilled person shall understand that as an alternative, the measurement unit145can be used for measuring line voltage and/or line current at the measuring point. FIG.3illustrates fault location on the first network according to an embodiment of present invention. As shown inFIG.3, on occurrence of a fault at a point in the first network11, within one of the protection zones L1, L2, L3, L4, a disturbance in the form of travelling waves moves outwardly along the first network11from the fault-occurring point and is detected by the measurement143. For example, a phase to phase fault happens at point F within the protection zone L2, a voltage dip of corresponding phase lines will appear at the measurement unit143arranged at one end of the first network11to the first terminal set141of the power conversion circuit140. A controller144of the power converter14will control its output current, which can be regarded as the short circuit current contributed by the converter but controlled to a maximum limitation. The fault event is detected based the voltage dip amplitude and the short circuit current from the converter or other methods such as methods based on communication. In particular, the circuit breakers S1, S2will have fault current If1flows from the first power source10to fault location F at within the protection zone L2. The circuit breaker S2will trip immediately, while the circuit breaker S1is as backup protection, according to the TSOCP algorithm. The circuit breakers S3, S4will have fault current If2from the power converter14to the fault location F. The output voltage and current at the measuring point (i.e. UQ1and IQ1) at the side to the first terminal set141of the power converter14are measured by the measurement unit143and communicated to the controller144of the power converter14. The phase to phase fault F is identified by the voltage and current measurements, UQ1, IQ1since fault F will result in an obvious voltage dip and a short circuit current (limited to maximum current limitation of the voltage source converter U1). The fault event can be identified based on a condition if a change rate of the measurement of the voltage/current exceeds a threshold. If the fault component current can be detected, we will know that there is a short circuit fault occurring in the line. For example, define fault criteria as: {iA(k)-iA(k-N)>isetiB(k)-iB(k-N)>isetiC(k)-iC(k-N)>iset(1)Where, iA(k), iB(k)and iC(k)are the kth sampling point of Phase A, B and C respectively; iA(k-N), iB(k-N), and iC(k-N)are the (k-N)th sampling point of Phase A, B and C respectively; N is the number of sampled points per fundamental period, such as 20 ms for a 50 Hz grid system;isetis the predefined threshold current value for protection. If any of the three criteria is fulfilled, a short circuit fault can be identified. Similarly, the voltage changing rate can be also used for the phase to phase short circuit identification. FIG.5illustrates the operation of the power converter according to an embodiment of present invention. When short circuit between phase A and phase B is identified, an operation sequence will be ordered as shown in figure below, in which Q1g, Q2gand Q4gare the gate driver signals of Q1, Q2and Q4respectively. Here it is assumed that the converter U1is a three phase half bridge 2-level converter. Other converter type is also applicable by similar concept. At t0, the close signal of Saand Sbare ordered, phase reactor Laand Lbare by passed. At t2, Q4is switched on, Q1and Q2are switched on and off complementarily with high frequency. Consequently, a high frequency voltage pulse Vabis exerted between phase A and Phase B by U1. Once a fault event is identified by the controller144, the controller144controls the power conversion circuit140to generate a voltage/current waveform travelling along the first network11. Under such scenario, the power converter14performs as of an AC to AC power converter which converting power supplied from the second power source12to the voltage/current waveform travelling along the first network11. Once the fault event is identified, a detection waveform, for example which is a series of high frequency current pulses, will be generated by converter U1. In order to realize a steep current waveform edge, at least two of the phase reactors La, Lb or Lc is by passed by respective bypass switch Sa, Sb or Sc. The converter U1with bypass switch is shown inFIG.4. Because the solution is dependent on the travelling waves, which are transient, it is necessary to isolate these by removing the steady state current and voltage components from the signals provided by the transducers within the measurement unit143. This could be done in several ways, one satisfactory method, which has been used in the past, being continually to monitor the transducer outputs and to store the signals for a period corresponding to the duration of one whole cycle of the power-system (i.e. 0.02 or 0.01 s for a 50 Hz system). In the case, the stored quantities are subtracted from the signals during the next cycle to give zero output in each case, when the circuit conditions are unchanged. This process can be done by sampling the waves and storing them digitally. They can then be recovered in their original form for subtraction by using digital/analogue convertors. The output of non-zero quantities at the end of this process indicates the presence of a disturbance. The controller144is configured to identify the voltage/current waveform from the measurements of the measurement unit143and record a first timing when it identifies the waveform. As described above, under the control of the controller144, the power converter14generates the detection waveform and initiates travelling waves of voltage and current on the first network11, and they move outwardly along the first network11from the first terminal set141of the power converter14and are detected by the measurement143, and which in turn travel to the other end of the first network11. On arrival at the fault location F of the first network11the travelling waves are completely or partially reflected, depending on the circuits connected to the fault point F of the first network11. The echo voltage waveform reflected from fault location F will be measured by the measurement unit143and identified by the controller144from the instant measurement. The controller144records a second timing when it identifies the echo of the waveform. The time at which these voltages were present may be found from the variations which have been occurred in the line end voltages and currents and thus the time taken for the waves to travel from the fault to the line end may be determined. Because the velocity of the travelling waves is known the fault position can be found. A possible method of determining the position of a line fault is described below, a single phase circuit of length L being considered for simplicity. From the movement of the first voltage/current waveforms at the measuring measuring point, the voltage/current waveforms are injected into the first network11in response to a voltage dip detected by the measurement unit143. Based on the time difference Δt between the first timing and the second timing, the distance D between the power converter14and fault location F is calculated. Based on the value of D, the fault location is determined. In this example, the fault is located at line section L2, between breakers S2, S3. Finally, the breaker trip signal is sent to the circuit breaker S3via telecommunication in order to isolate the fault from the rest of lines at the power converter side. To this basic scheme several refinements are added to avoid possible malfunctions imposed by the following conditions. The power converter14further includes a by-passing device, being configured to short circuit a filter inserted between the first network and the first terminal set of the power conversion circuit; wherein: the controller is further configured to issue a command to close the by-passing device substantially in parallel with the generation of the voltage/current waveform. Though the present invention has been described on the basis of some preferred embodiments, those skilled in the art should appreciate that those embodiments should by no way limit the scope of the present invention. Without departing from the spirit and concept of the present invention, any variations and modifications to the embodiments should be within the apprehension of those with ordinary knowledge and skills in the art, and therefore fall in the scope of the present invention which is defined by the accompanied claims. | 16,728 |
11863083 | DETAILED DESCRIPTION Referring toFIG.1, a block diagram of an example of a modular rectifier assembly120is shown. The modular rectifier assembly120may be used with a driving apparatus110or a different driving apparatus110′. Each of the driving apparatuses110and110′ is, for example, variable speed drive (VSD), an adjustable speed drive (ASD), or a variable frequency drive (VFD). The modular rectifier assembly120is separate from and independent of the driving apparatuses110and110′ and may be used with driving apparatuses other than the apparatus110and110′. The modular rectifier assembly120may be used to upgrade, retrofit, and/or repair an existing driving apparatus or may be packaged and sold with a driving apparatus. The modular rectifier assembly120increases efficiency and performance while reducing costs by encouraging, enabling, and allowing reuse of existing equipment and by providing the operator or end user with the ability to select the installation location of the rectifier assembly120. FIG.2is a block diagram of a system200that includes a driving apparatus210. The driving apparatus210is another example of a driving apparatus that may use the modular rectifier assembly120. The driving apparatus210is electrically connected to an alternating current (AC) electrical power distribution network201. The driving apparatus210generates an AC driving signal204based on AC electrical power205from the network201, and the driving apparatus210provides the AC driving signal204to a load202. The electrical power distribution network201may be, for example, a multi-phase electrical power grid that provides electricity to industrial, commercial and/or residential customers. The AC electrical power distribution network201distributes AC electrical power that has a fundamental frequency of, for example, 50 or 60 Hertz (Hz). The distribution network201may have an operating three-phase line-to-line voltage of, for example, up to 690 volt (V) root mean square (RMS) for low voltage, and above 690V such as 10 kV for medium or high voltage. The network201may include, for example, one or more transmission lines, distribution lines, power distribution or substation transformers, electrical cables, and/or any other mechanism for transmitting electricity. The load202may be, for example, an induction machine, an induction motor, or a synchronous permanent magnet machine that operates at a speed and torque that is determined by the AC driving signal204. The driving apparatus210is any type of apparatus that is capable of producing the AC driving signal204. The driving apparatus210may be, for example, a variable speed drive (VSD), an adjustable speed drive (ASD), or a variable frequency drive (VFD). The driving apparatus210and the load202are used in an industrial process. The industrial process may be, for example, a conveying process; a heating, ventilation, and air conditioning (HVAC) process; a natural gas or oil refining process; a mining process; a lighting process; or a pumping process. The driving apparatus210includes an electrical system212. The electrical system212receives AC electrical power205from the distribution network201at an input node214and provides an the AC driving signal204via an output node209. When the rectifier assembly120is included in the driving apparatus210, the electrical system212includes the rectifier assembly120and the inverter240. The rectifier assembly120includes an electrical network222that converts the AC electrical power205into direct current (DC) electrical power213. The electrical network222may be, for example, a plurality of diodes that are arranged to form a rectifier, for example, a 6-pulse, 12-pulse, or 18-pulse rectifier. The inverter240converts the DC electrical power213into the AC driving signal204. As discussed in greater detail below, the rectifier assembly120is a modular assembly that is separate from and independent of the driving apparatus210and the inverter240. The rectifier assembly120is configured for use with the driving apparatus210and/or any other driving apparatus that uses a converter or rectifier to convert the AC electrical power205into the DC power213. The other driving apparatuses may have the same configuration as the driving apparatus210, or the other driving apparatuses may be different types or different models of driving apparatus. The rectifier assembly120may be used to, for example, retrofit, upgrade, and/or repair an existing driving apparatus. For example, the rectifier assembly120may be used to upgrade the original rectifier in an existing driving apparatus by simply replacing the original rectifier with the rectifier assembly120and without having to replace the entire driver apparatus210. Moreover, the modular nature of the rectifier assembly120allows an existing driving apparatus to be modified to suit the requirements of a particular application. Thus, the rectifier assembly120allows efficient reuse of existing equipment, reduces costs, and improves performance. The rectifier assembly120also allows the driving apparatus210to be used in a more flexible manner. For example, although the rectifier assembly120is electrically connected to the inverter240, the end user or other operator of the driving apparatus210may place the rectifier assembly120any place that is convenient for the application. For example, an end user may choose to mount the rectifier assembly120in a cabinet with the inverter240and associated equipment. Additionally, the end user may choose where in the cabinet to mount the rectifier assembly120. On the other hand, the modular nature of the rectifier assembly120allows the end user or other operator to place the rectifier assembly120away from the inverter240and other components of the driving apparatus210, for example, in an area that is separate from the inverter140and other components of the driving apparatus210. For example, the end user may choose to place the rectifier assembly120in a separate room as compared to the other components of the driving apparatus210. FIG.3Ais a schematic of a system300. The system300includes a modular rectifier assembly320. The modulator rectifier assembly320is an implementation of the modular rectifier assembly120ofFIGS.1and2. The system300includes an 18-pulse phase-shifting autotransformer380that is electrically connected to the modular rectifier assembly320. The distribution network201includes three electrical phases A, B, C. The phases A, B, C are time-varying or AC electrical signals (for example, voltage signals) that have substantially the same magnitude but are phase shifted 120° relative to each other. The phase-shifting autotransformer380receives the three-phase electrical power from the distribution network201and produces a nine-phase AC output at nodes1,2,3,4,5,6,7,8,9. The voltage at each of the nodes1through9is substantially the same magnitude but has a phase that is shifted by 40° relative to each adjacent node. Adjacent in this context refers to the numbering scheme for the nodes1through9. In the example ofFIG.3A, nodes1and2are adjacent nodes, for example. The phase-shifting auto transformer380includes transformer coils381(only one of which is labeled for simplicity) and a neutral point N. The nodes1through9of the 18-pulse phase-shifting autotransformer380are electrically connected to an electronic network322of the modular rectifier assembly320. In the example ofFIG.3A, the electronic network322of the modular rectifier assembly320is implemented as an 18-pulse rectifier. The modular rectifier assembly320includes diode modules323_1,323_2,323_3, each of which includes a respective electronic network322_1,322_2,322_3that includes six diodes arranged to form a 6-pulse rectifier. In other words, the electronic network322is a 18-pulse rectifier implemented with three six-pulse rectifiers. The diode modules323_1,323_2,323_3are electrically connected in parallel to each other across a DC bus324. FIG.3Bis a schematic that shows the electronic network322_1in greater detail. The electronic network322_1is a three-phase six-pulse bridge that includes six electronic switches. In the example ofFIGS.3A and3B, the six electronic switches are diodes D1through D6. Each diode D1through D6includes a cathode and an anode and is associated with a forward bias voltage. Each diode D1through D6allows current to flow in the forward direction (from the anode to the cathode) when voltage of the anode is greater than the voltage of the cathode by at least the bias voltage. When the voltage difference between the anode and the cathode is less than the forward bias voltage, the diode does not conduct current in the forward direction. The anode of the diode D1is connected to the cathode of the diode D4to form a first pair of diodes. Node1of the phase-shifting autotransformer380is connected to the first pair of diodes between the anode of the diode D1and the cathode of the diode D4. The anode of the diode D3is connected to the cathode of the diode D6to form a second pair of diodes. Node2of the phase shifting autotransformer380is connected to the second pair of diodes between the anode of the diode D3and the cathode of the diode D6. The anode of the diode D5is connected to the cathode of the diode D3to form a third pair of diodes. Node3of the phase-shifting autotransformer380is connected to the third pair of diodes between the anode of the diode D5and the cathode of the diode D2. The diodes D1through D6rectify the input currents from the nodes1,2,3, respectively, into a rectified current that is provided to the DC bus324. Each of the electronic networks322_2and322_3also includes six diodes D1through D6arranged in the same manner (three pairs of diodes that form a six-pulse rectifier). Each of the nodes4,5,6of the phase-shifting autotransformer380is electrically connected to one pair of diodes in the electronic network322_2. Each of the nodes7,8,9of the phase-shifting autotransformer380is electrically connected to one pair of diodes in the electronic network322_3. The electronic networks322_1,322_2,322_3provide a rectified current id to the DC bus324. Each electronic network322_1,322_2,322_3contributes equally to the DC current id. A capacitor network326is connected across the DC bus324. The rectified current id flows into the capacitor network326and is stored. The capacitor network326includes one or more capacitors that store and discharge electrical energy. The inverter340converts the DC power stored in the capacitor network326into the AC driver signal304that is provided to the load202. In this example, the AC driver signal304is a three-phase AC driver signal with phase components304u,304v,304w, each of which is provided to one of the three-phases of the load202. The inverter340includes a network of electronic switches SW1through SW6that are arranged to generate the driver signal304. Each of the switches SW1through SW6may be, for example, a power transistor. The inverter340may implement, for example, a pulse width modulation (PWM) technique to modulate the energy that is stored in the capacitor network326into the AC driver signal304. The PWM technique may be implemented based on any type of control algorithm, such as, for example, a 6-step electronic commutation, various field oriented controls, a space vector PWM, or a sinusoidal PWM. The switching of the electronic switches SW1through SW6is controlled such that the amplitude, frequency, and phase of the driver signal304is also controlled. The amplitude, frequency, and phase of the driver signal304determines the operating properties (for example the torque, speed, and/or direction) of the load302. The topology shown inFIGS.3A and3Bis provided as an example, and other topologies may be used. For example, the 18-pulse phase-shifting autotransformer380may be implemented in any manner known in the art, such as, for example, a symmetric 18-pulse differential delta configuration or an asymmetric 18-pulse differential delta configuration. Moreover, other types of phase shifting auto-transformers and rectification configurations may be used. For example, the phase-shifting autotransformer380may be implemented as a 12-pulse phase shifting auto-transformer that produces six phase outputs based on a three-phase input, and the rectifier assembly320may be implemented as a 12-pulse rectifier that includes 6 pairs of diodes. In yet another example, the system300may be implemented without the phase-shifting autotransformer. In these implementations, the rectifier assembly320is implemented as a three-pulse rectifier that includes three pairs of diodes arranged as shown inFIG.3B. In this implementation, each pair of diodes is electrically connected to one phase of the distribution network201. Referring toFIGS.4A-4C, various aspects of another example of a modular rectifier assembly420are shown. The modular rectifier assembly420is an implementation of the modular rectifier assembly120and320.FIG.4Ais an exterior side view of the rectifier assembly420in the X-Z plane, with hidden elements shown with dashed lines.FIG.4Bis a schematic of a diode module428a.FIG.4Cis a top view of the rectifier assembly420in the X-Y plane. The rectifier assembly420includes a mounting base430and a rectification module423that is mounted to the mounting base430. The rectification module423includes three diode modules428a,428b,428c, each of which includes a pair of diodes.FIG.4Bis a schematic that illustrates the electrical connections of the diode module428a. The diode module428aincludes diodes429_1and429_2. The diodes429_1,429_2are connected in series with the cathode of the diode429_2electrically connected to the anode of the diode429_1. The cathode of the diode429_1is connected to a conductor427a. The anode of the diode429_2is connected to a conductor427b. The conductor427ais electrically connected to a conductor425a, and the conductor427bis electrically connected to a conductor425b. The conductors425aand425bform a local DC bus425. The conductors427a,427b,425a, and425bare made of an electrically conducting material, such as, for example, a metal or a metal alloy. In the example ofFIGS.4A and4C, the conductors427aand427bare rods or bars that extend in the Z direction, and the conductors425aand425bare rods or bars that extend in the X direction. The conductors427a,427band the conductors425a,425bmay be made of a single piece of metal or may be multiple pieces or metal that are permanently joined by, for example, welding or soldering. Other configurations are possible. For example, the diode conductors427a,427bmay be wires rather than rigid bars or rods. Each of the diode modules428band428cinclude two diodes arranged and connected to the conductors425a,425bin the same manner as shown inFIG.4B. The diodes429_1and429_2are in the diode module428a. The diode module428ais made out of any durable material, such as, for example, a rugged polymer or a metal material. Referring toFIGS.4A and4C, the conductors427aand427bpass through the diode module428agenerally in the Z direction. The conductor427ais electrically connected to the conductor425a. The conductor427bis electrically connected to the conductor425b. In this way, the conductors427aand427belectrically connect the diode module428aacross the local DC bus425. The conductor427ais electrically connected to the conductor425a, for example, by soldering or welding, and the second diode conductor427bis electrically connected to the conductor425b, for example, by soldering or welding. The rectification module423also includes the diode modules428band428c, which are configured in the same manner as the diode module428a. The diode modules428band428care connected across the local DC bus425in parallel with each other and the diode module428a. The rectification module423includes a heat sink435on which the diode modules428a,428b,428care mounted. The diode modules428a,428b,428care thermally coupled to the heat sink435. For example, the diode modules428a,428b,428care thermally coupled to the h eat sink435by being in physical contact with at least part of the heat sink435. A heat sink is a passive heat exchanger that removes heat from a heat source that is thermally coupled to the heat sink. The heat sink435is used to remove heat from the diode modules428a,428b,428c. For example, the heat sink435may include a plate or block with openings or fasteners that hold the diode modules428a,428b,428c. The diode modules428a,428b,428cmay be, for example, attached to the heat sink435with an adhesive and/or with mechanical fasteners such as screws and/or other mechanical hardware devices. The heat sink435is made of a material with a relatively high thermal conductivity, allowing the heat sink435to conduct or draw heat from the modules428a,428b,428c. For example, the heat sink435may be made of a thermally conductive metal such as, for example, copper, aluminum, or an aluminum alloy. The mounting base430is made of sheet metal or another durable material. The mounting base430includes three openings432a,432b,432c. Each opening432a,432b,432cis sized to receive one rectification module such as the rectification module423. In the example ofFIGS.4A and4C, the heat sink435is received in the opening432aand is held to the mounting base430in the opening432ato attach the rectification module423to the mounting base430. For example, the heat sink435may be inserted into the opening432aand held by an interference fit between the mounting base430and the heat sink435. In another example, the heat sink435is attached to the mounting base430with an adhesive and/or with a mechanical fastener. Additional rectification modules may be installed in the rectifier assembly420. For example, one rectification module identical to the rectification module423may be installed into each of the openings432band432c. In such an implementation, the rectifier assembly420forms an 18-pulse rectification module. The rectification module423may include additional features. For example, in addition to the heat sink435, the rectification module423may include one or more additional heat exchange elements that remove heat from the diode modules428a,428b,428c. The additional heat exchange elements may be, for example, passive cooling elements such as additional heat sinks or openings that allow airflow, or active cooling elements such as fans, water-cooled heat exchangers, air-cooled heat exchangers, and/or thermoelectric cooling devices (such as peltier coolers). Referring also toFIG.4D, in operational use of the rectifier assembly420, the conductors425aand425bare electrically connected to a DC bus426of an electrical network411. The electrical network411may be, for example, the capacitor network326and the inverter340ofFIG.3A. For example, a first busbar421amay be electrically connected to the conductor425aof each rectification module423attached to the mounting base430, and a second busbar421bmay be electrically connected to the conductor425bof each rectification module423attached to the mounting base430. In this example, the first busbar421aand the second busbar421bare attached to the end of the respective conductors425aand425band extend along the X direction inFIGS.4Aand C. One phase of a AC power source (not shown inFIGS.4A-4C) is electrically connected to the diode module428abetween the first diode429_1and the second diode429_2at the node431(FIG.4B). The diode module428arectifies the AC current from the AC power source and provides a rectified current to the local DC bus425. Each diode module428b,428cis also electrically connected to one phase of a multi-phase power source. Additional instances of rectification modules may be added to the mounting base430, and those rectification modules are also electrically connected to the multi-phase power source and the DC bus426in the same manner. Referring toFIG.5, a perspective view of a rectification module523is shown. The rectification module523is configured for use with a modular rectifier assembly, such as the modular rectifier assembly120(FIG.1),420(FIGS.4A and4C),620(FIG.6A-6D), or920(FIGS.9A and9B). The rectification module523includes diode modules528a,528b,528c. Each of the diode modules528a,528b,528cincludes a pair of diodes that are connected in the configuration shown inFIG.4Band a housing that encloses the diodes. The diode modules528a,528b,528care mechanically mounted to a heat sink535and are thermally coupled to the heat sink535. For simplicity, only the housing of the diode module528ais labeled inFIG.5. The diode module528aincludes a housing533athat encloses a pair of diodes. The housing533ais made of a rugged material, such as, for example, sheet metal or steel. The housing533ais attached to a top portion535aof the heat sink535with screws534. For simplicity, only one of the screws534is labeled. Although the screws534are depicted inFIG.5, the housing533amay be attached to the top portion535ain another manner. For example, the housing533amay be glued to the top portion535awith a thermally conductive adhesive or a thermally conductive tape in addition to or instead of being screwed to the top portion535a. The heat sink535includes the top portion535a, which is generally planar. The heat sink535also includes a side portion535bthat is connected to the top portion535a. The side portion535bis generally planar and extends away from the top portion535aalong a direction that is perpendicular to a plane that includes the top portion535a. The heat sink535also includes another side portion (not shown) that is parallel to the side portion535b, is connected to the top portion535a, and extends away from the top portion535a. The heat sink535also includes a plurality of fins536that extend away from the underside (not shown) of the top portion535a. The underside of the top portion535ais opposite to the top portion535a. For simplicity, only one of the fins536is labeled inFIG.5. Each fin536is a plate-like structure that is separated from the other fins. The heat sink535is made of a material with a relatively high thermal conductivity, such as, for example, aluminum or copper. The housing533ais made of any durable material, such as, for example, sheet metal or steel. As noted above, the heat sink535is in thermal contact with the diode modules528a,528b,528c. The heat sink535transfers heat generated by the operation of the diodes in the diode modules528a,528b,528cto a fluid medium. In the example ofFIG.5, the fluid medium is air that surrounds the rectification module523. However, other implementations are possible. For example, the heat sink535may be used in combination with an active cooling element, such as, for example, a water channel that carries heat away from the heat skink535or a fan that encourages air flow through and/or in the vicinity of the heat sink535to thereby encourage heat removal. The diode pair in each module528a,528b,528cis electrically connected across a DC bus525formed by a first conductor525aand a second conductor525b. The first conductor525aand the second conductor525bare metal bars. The diode pair in the module528ais electrically connected to the DC bus525with diode conductors527aand527b(which are electrically connected to the diodes in the module528a). In the example shown, the diode conductors527aand527bare metallic screws that are secured to and electrically connected to the first and second conductors525aand525b, respectively, using nuts. The diode pair in the module528band the module528care also connected to the DC bus525in the same manner. In the example ofFIG.5, a U-shaped connection bracket538ais between the housing533aand the conductors525aand525b. The diode conductors527aand527bextend through the housing533aand the connection bracket538ato be secured to the conductors525aand525b, respectively. In some implementations, the connection bracket538ais not used. FIGS.6A-6Eshow various aspects of another example rectifier assembly620. The rectifier assembly620includes three instances of the rectification module523(FIG.5). The rectifier assembly620is an 18-pulse rectifier.FIG.6Ais a perspective view of the rectifier assembly620.FIG.6Bis a top view of the rectifier assembly620.FIG.6Cis a side view of the rectifier assembly620.FIG.6Dis a perspective view of a mounting base630.FIG.6Eis a perspective view of an air baffle670. Referring toFIGS.6A and6B, the rectifier assembly620includes the mounting base630, which holds three instances of the rectification module523. Referring also toFIG.6D, the mounting base630includes a plurality of sidewalls630athrough630e. In the example shown, each of the sidewalls630athrough630eis a substantially planar or plate-like structure. The sidewalls630athrough630eare made of any durable material that is suitable for the application. For example, the sidewalls630athrough630emay be made of sheet metal or steel. The first sidewall630ais a substantially planar structure that defines three openings632a,632b,632c. Each opening632a,632b,632cis the same size and shape. In the example shown, the openings632a,632b,632bare rectangular; however, the openings632a,632b,632bmay be other shapes. As shown inFIGS.6A and6B, one instance of the rectification module523is held in each opening632a,632b,632c. The mounting base630also includes the second sidewall630band the third sidewall630c. The second sidewall630band the third sidewall630care the same size and shape, extend in the Y-Z plane, and are parallel to each other. The second sidewall630bincludes an edge664and an edge666. The third sidewall630cincludes an edge665and an edge667. The edge666and the edge667extend along the Y direction. The edge664and the edge665extend along a direction that is at an angle α relative to the Y direction. The angle α may be, for example, less than 90 degrees (°), the angle α may be zero degrees, or the α may be between 10° and 70°. The mounting base630also includes a fourth sidewall630dthat extends in the X-Y plane between the second sidewall630band the third sidewall630c. The fourth sidewall630dis attached to the second sidewall630bat the edge666and to the third sidewall630cat the edge667. The fourth sidewall630ddefines three openings661athrough661c. Each opening661a,661b,661cis configured to hold a respective fan635a,635b,635c(FIGS.6A and6B). The openings661a,661b,661cpass through the fourth sidewall630dsuch that the fans635a,635b,635care able to draw air662(FIG.6C) from one side of the fourth sidewall630dto the other. In the example shown inFIGS.6A-6D, the openings661a,661b,661care circular. However, any shape that allows a fan to be received and held in the opening may be used. The first sidewall630aextends between the second sidewall630band the third sidewall630c. Specifically, the first sidewall630ais connected to the second sidewall630bat the edge664and to the third sidewall630cat the edge665. As discussed above, the edges664and665extend at the angle α relative to the Y direction, and the first sidewall630ais a planar structure. Thus, the first sidewall630ais angled at the angle α relative to the Y direction. Furthermore, and referring toFIG.6A, when the rectification modules523are mounted in the mounting base630, the modules523(and their respective components, such as the heat sink535) are also angled at the angle α relative to the Y direction. The mounting base630also includes a fifth sidewall630ethat extends in the X-Z plane and is connected to the first sidewall630a, the second sidewall630b, the third sidewall630c, and the fourth sidewall630d. The fifth sidewall630eseparates the first sidewall630aand the fourth sidewall630din the Z direction. This spacing provides space to accommodate the fans635a,635b,635c(FIGS.6A and6B). Furthermore, flanges663extend from the second sidewall630b, the third sidewall630c, and the fourth sidewall630d. The flanges663may be used to mount the mounting base630to a cabinet (such as shown inFIG.8) or to another structure Referring also toFIG.6C, the mounting base630also may include a sixth sidewall630f. The sixth sidewall630fextends in the X-Z plane and is connected to the first sidewall630a, the second sidewall630b, the third sidewall630c, and the fourth sidewall630d. In implementations that include the sixth sidewall630f, the mounting base630is similar to an enclosure and may be referred to as the mounting enclosure630. In implementations that include the sixth sidewall630f, a vent or opening668(FIG.6C) passes through the sixth sidewall630fto allow air662to exit the enclosure630. Some implementations do not include the sixth sidewall630f. In these implementations, air exits from the mounting base630through an open region668between the second and third side walls630band630c. FIGS.6A-6Cshow the rectifier assembly620in the assembled state. As shown inFIG.6C, when the fans635a,635b,635care operational, they draw air662from a region688through the mounting base630along a path669and out an open region or vent668between the second and third sidewalls630band630c. The heat sinks535(FIG.5) are not shown inFIG.6C, but, as discussed above, the heat sinks535are oriented at the angle α relative to the direction Y. The path669passes over, near, and/or through the spaces between the fins536(FIG.5). Thus, the air622flows near or through the heat sinks535and aids in the removal of heat from the heat sinks535. This allows for additional cooling and/or more effective cooling of the diode pairs in the diode modules528a,528b,528cand increases the amount of time during which the diodes operate within their rated thermal limit and/or prevents occurrences of thermal conditions under which the diodes are unable to operate optimally. Referring also toFIG.6E, in some implementations, the mounting base630includes the baffle670. Some implementations do not include the baffle670. The baffle670is used to help direct the moving air662drawn into the base630by the fans635a,635b,635ctoward the heat sinks535, thereby further enhancing the removal of heat from the heat sinks535. The baffle670includes a planar portion671and a flange672that extends from the planar portion671. The flange672allows the baffle670to be attached to the mounting base630. When attached to the mounting base630, the planar portion671is in the path669(FIG.6C). The flange672and the planar portion671are angled relative to each other such that the planar portion671directs the air662toward and through the heat sink535. In other words, the baffle670directs the air662in such a way that more heat is removed from the heat sink535. This allows the diodes in the diode modules528a,528b,528cto operate more efficiently and/or for a longer period of time. FIG.7Ashows simulated thermal data for the rectifier assembly620without the baffle670.FIG.7Bshows thermal data for the rectifier assembly620with the baffle670. The data shown inFIGS.7A and7Brepresents simulated temperature in two dimensions for a side cross section of the rectifier assembly620. In the examples ofFIGS.7A and7B, the solid lines with arrows represent the direction of air flow through the rectifier assembly620. The air is drawn into the rectifier assembly620by the fan635aat the bottom of the assembly620, and the air exits through the open region668. In the simulations, the inlet at635awas 60 square inches, and the cooling temperature at the inlet was less than 50° C. As shown by comparingFIGS.7A and7B, the air flow is more directed and concentrated in the simulation in which the baffle670is used (FIG.7B). Furthermore, the temperature is lower in the rectifier assembly620in the simulation in which the baffle670was used (FIG.7A). For example, the region labeled789had a temperature of about 115° C. in the simulation without the baffle760(FIG.7A) and a temperature of about 100° C. in the simulation with the baffle760(FIG.7B). Returning toFIGS.6A-6C, the rectifier assembly620includes an input interface639. The input interface639is on the sidewall630ein the example shown inFIGS.6A-6C. The input interface639includes a plurality of electrical connection points, each of which is electrically connected to one of the diode modules528a,528b,528cof each instance of the rectification module523. The electrical connection points also accept a connection to one phase of a multi-phase power source (such as the phase-shifting autotransformer380ofFIG.3A). In other words, the input interface639provides an interface for the source to electrically connect to the diode modules. To connect the rectifier assembly620to an inverter to form a driver apparatus, the conductors525aand525bon each instance of the rectification module523are electrically connected to a DC bus that is also connected to the inverter. For example, the conductors525aand525bof each rectification module523may be electrically connected to respective first and second bus bars (not shown) that are electrically connected to the DC bus of a driving apparatus. The first bus bar may be a metal bar that is in contact with the conductor525aof each rectification module523. The second bus bar may be a metal bar that extends in the X direction and is in contact with the conductor525bof each rectification module523. Referring toFIG.8, a system800is shown. The system800includes a cabinet885and the rectifier assembly620. A partial perspective view of the cabinet885is shown inFIG.8. The cabinet885is a hollow enclosed space that has a door or other access point. The cabinet885is used to house equipment, such as, for example, an inverter such as the inverter240, cables, monitoring systems, and computing equipment. In the example ofFIG.8, the cabinet885is used to enclose the rectifier assembly620. The cabinet885may or may not include additional equipment. The cabinet885includes an interior wall886. The interior wall886is substantially flat and defines a plurality of vents887. Each vent887includes a plurality of openings that pass through the interior wall886. In other words, the vents887allow air or other fluids to enter and leave the cabinet885. The flanges663of the rectifier assembly620are attached to the wall886to mount the rectifier assembly620to the wall886. The open region668(FIG.6C) of the rectifier assembly620faces the wall88and is aligned with one of the vents887such that air is able to flow through the assembly620. The fourth sidewall630dfaces away from the wall886. The fans635a,635b,635cface down toward a bottom888of the cabinet885. In operational use, the fans635a,635b,635cdraw air from the bottom888through the rectifier assembly620and exhaust the air through the vent887that aligns with the open region668of the assembly620. Orienting the rectifier assembly620with the fans635a,635b,635cpositioned to draw air from the region near the bottom888(as shown inFIG.8) may help to ensure that relatively cooler air is drawn into the rectifier assembly620. For example, relatively cool air generally sinks and is more likely to be near the bottom888. On the other hand, the orientation of the rectifier assembly620in the cabinet885depicted inFIG.8is an example, and other orientations are possible. For example, the rectifier assembly620may be rotated 90° or 180° relative to the configuration shown. Such other orientations may be desired if, for example, the bottom region888is unusually warm due to the presence of a particular piece of equipment or if the bottom region888is crowded with equipment such that little air is present in the bottom region888. Moreover, the rectifier assembly620may be mounted on structures other than the wall886. For example, the rectifier assembly620may be mounted on a cabinet door, or any substantially flat structure that is sufficiently strong to support the rectifier assembly620. Other implementations are within the scope of the claims. For example, the diode modules of the rectification modules used in a rectifier assembly may be arranged in any manner that is suitable for the application. Referring toFIG.9A, another implementation of a rectification module923that is used in a rectifier assembly920(FIG.9B) is shown. The rectification module923includes diode modules928a,928b,928c, each of which includes a pair of diodes electrically connected as shown inFIG.4B. The rectification module923also includes a local DC bus formed from conductors525aand525b. The diode modules928a,928b,928care electrically connected to the local DC bus525. Three instances of the rectification module523are mounted to a mounting base930to form the rectifier assembly930. Moreover, other implementations and uses are possible. For example, the any of the rectifier assemblies120,220,420,520,620, and920may be used to drive more than one electrical network such as the network411(FIG.4D). For example, the any of the rectifier assemblies120,220,420,520,620, and920may be used to drive a plurality of instances of networks (each including a capacitor network and an inverter) that receive DC electrical power from a bus electrically connected to the output of the rectifier assembly. In these implementations, the output of the rectifier assembly120,220,420,520,620, or920is electrically connected to a DC bus that powers more than one inverter such that the rectifier assembly120,220,420,520,620, or920becomes part of a driving apparatus that drives more than one AC motor or other AC load. | 37,285 |
11863084 | DETAILED DESCRIPTION One or more specific embodiments are described below. To provide a concise description of these embodiments, not all features of an actual implementation may be described in the specification. In the development of any such actual implementation, as in any engineering or design project, numerous implementation-specific decisions must be made to achieve the developers' specific goals, such as compliance with system-related and business-related constraints, which may vary from one implementation to another. Moreover, it should be appreciated that such a development effort might be complex and time consuming, but would nevertheless be a routine undertaking of design, fabrication, and manufacture for those of ordinary skill having the benefit of this disclosure. In the following description, for purposes of explanation, various details are set forth to provide a thorough understanding of the disclosed concepts. As part of this description, some of this disclosure's drawings represent structures and devices in block diagram form for sake of simplicity. In the interest of clarity, not all features of an actual implementation are described in this disclosure. Moreover, the language used in this disclosure has been selected for readability and instructional purposes, has not been selected to delineate or circumscribe the disclosed subject matter. Rather the appended claims are intended for such purpose. Various embodiments of the disclosed concepts are illustrated by way of example and not by way of limitation in the accompanying drawings in which like references indicate similar elements. For simplicity and clarity of illustration, where appropriate, reference numerals have been repeated among the different figures to indicate corresponding or analogous elements. In addition, numerous specific details are set forth in order to provide a thorough understanding of the implementations described herein. In other instances, methods, procedures and components have not been described in detail so as not to obscure the related relevant function being described. References to “an,” “one,” or “another” embodiment in this disclosure are not necessarily to the same or different embodiment, and they mean at least one. A given figure may be used to illustrate the features of more than one embodiment, or more than one species of the disclosure, and not all elements in the figure may be required for a given embodiment or species. A reference number, when provided in a given drawing, refers to the same element throughout the several drawings, though it may not be repeated in every drawing. The drawings are not to scale unless otherwise indicated, and the proportions of certain parts may be exaggerated to better illustrate details and features of the present disclosure. FIG.1Aillustrates an exemplary prior art multi-output AC/DC adapter100. Adapter100includes a main power stage102, which, in the illustrated example is a flyback converter but, in other embodiments or applications could be any suitable converter topology. Power stage102receives an input voltage Vin+, which may, for example, be received from an AC input751(FIG.7) connected via a rectifier752(FIG.7). An input capacitor CBk may serve smooth the rectified AC voltage. In the illustrated flyback converter configuration, a main switch S1may be switched by feedback loop104and controller106to alternately store energy in flyback transformer TX (when switch S1is closed) and discharge stored energy to the flyback stage output (voltage V0+) through the rectifier diode. Main switch S1may be a silicon, silicon carbide, or gallium nitride MOSFET, or any other suitable semiconductor switching device appropriate to the particular application. Output filter capacitor Co may serve to filter the output voltage, so as to reduce ripple seen by the loads on main power stage102. Feedback loop104compares the output voltage V0+ to a suitable reference and provides control signals to main switch S1via controller106to regulate the output voltage V0+ to a desired level. Operation of flyback converters (or other suitable topologies for main power stage102) is known to those skilled in the art, and, for sake of brevity will not be repeated here. However, any of a variety of flyback converter configurations, including primary resonant flyback converters, active clamp flyback converters, etc. could be used as appropriate for a given embodiment or application. Adapter100also includes a plurality of regulator stages112a-112d, one for each output. For conciseness only stages112aand112dare illustrated, but additional stages112band112care implied and may be substantially similar to the illustrated stages. Also, more or fewer regulator stages could be provided depending on the number of DC outputs desired. Each regulator stage112a-112dincludes a converter that regulates the output voltage V0+ from main power stage102to the level required for each output, i.e., Vo1-Vo4. In the illustrated example, each regulator stage112a-112dis a buck converter including a high side switch114h, a low side switch114l, an output filter capacitor Co1, and a power deliver switch118(discussed in greater detail below). Switches114hand114lmay be silicon, silicon carbide, or gallium nitride MOSFETs, or any other suitable semiconductor switching device appropriate to the particular application. Thus, main power stage102may be configured to produce a regulated output voltage V0+ that is greater than or equal to the largest output voltage Vo1-Vo4required by a respective device to be connected to such outputs. In other embodiments, one or more of regulator stages112a-112dcould be another converter topology, such as a boost converter or buck-boost converter, in which case the regulated output voltage of main power stage102could be less than a required output voltage. In any case, operation of such regulator stages is known to those skilled in the art and, for sake of brevity, will not be repeated here. In some embodiments, adapter100may implement the Universal Serial Bus Power Delivery (“USB-PD”) standard, such that a device connected to any one of outputs Vo1-Vo4may negotiate a suitable output voltage, e.g., 5V, 9V, 15V, 20V, etc. Additionally, adapter100may include, in the respective regulator stages112a-112d, power delivery switches118. Power delivery switches118may be silicon, silicon carbide, or gallium nitride MOSFETs, or any other suitable semiconductor switching devices appropriate to the particular application. These switches may be used to selectively disconnect/disable a respective output stage when its operation is not required or in the event of a fault (such as a short circuit failure of high side switch114hthat would otherwise permanently connect output Vo1to main power stage102's output voltage). However, these power delivery switches118may be omitted, as illustrated in converter101ofFIG.1B, which is substantially similar to converter100, except that regulator stages113a-113domit power delivery switches118. The two exemplary adapters100and101may suffer from various disadvantages depending on the power requirements of the respective loads connected to outputs Vo1-Vo4and/or the total power requirement. First, in adapter100, three additional switching devices114h,114l, and118are required per additional output, together with an additional magnetic element116. The same applies to adapter101, although only two additional switches (114h,114l) per output are required. If each output is intended to provide the full output power of the adapter, then each of these switches will be relatively large, and expensive. Otherwise, if only certain outputs are intended to carry the full rated power, then the user must know which output to use when full power is required and attach devices accordingly. Neither situation may be optimal. FIG.2illustrates an exemplary multi-output AC/DC adapter200that can address these issues. Adapter200includes a main power stage202, which, in the illustrated example is a flyback converter but, in other embodiments or applications could be any suitable converter topology. Main power stage202receives an input voltage Vin+, which may, for example, be received from an AC input701(FIG.7) connected via a rectifier702(FIG.7). An input capacitor CBk may serve smooth the rectified AC voltage. In the illustrated flyback converter configuration, a main switch S1may be switched by feedback loop204and controller206to alternately store energy in flyback transformer TX (when switch S1is closed) and discharge stored energy to the flyback stage output (voltage V0+) through the rectifier diode. Main switch S1may be a silicon, silicon carbide, or gallium nitride MOSFET, or any other suitable semiconductor switching device appropriate to the particular application. Output filter capacitor Co may serve to filter the output voltage, so as to reduce ripple seen by the loads on main power stage202. Feedback loop204compares the output voltage V0+ to a suitable reference and provides control signals to main switch S1via controller206to regulate the output voltage V0+ to a desired level. Operation of flyback converters (or other suitable topologies for main power stage202) is known to those skilled in the art, and, for sake of brevity will not be repeated here. However, any of a variety of flyback converter configurations, including primary resonant flyback converters, active clamp flyback converters, etc. could be used as appropriate for a given embodiment or application. Adapter200also includes a plurality of regulator stages212a-212d, one for each output. For conciseness only stages212aand212dare illustrated, but additional stages212band212care implied and may be substantially similar to the illustrated stages. Also, more or fewer regulator stages could be provided depending on the number of DC outputs desired. Each regulator stage212a-212dincludes a chopper circuit that regulates the intermediate output voltage derived from the secondary winding of flyback transformer TX to the level required for each output, i.e., Vo1-Vo4. In the illustrated example, each regulator stage212a-212dis a chopper circuit including a rectifier diode213, a chopper switch214, a chopper controller215, an output filter capacitor Co1, and a power delivery switch218. Chopper switch214and power delivery switch218may be silicon, silicon carbide, or gallium nitride MOSFETs, or any other suitable semiconductor switching device appropriate to the particular application. Thus, main power stage202may be configured to produce a regulated output voltage V0+ that is greater than or equal to the largest output voltage Vo1-Vo4required by a respective device to be connected to such outputs. Each regulator stage (e.g., chopper stage212a) includes a rectifier diode213that serves as “gatekeeper” to the stage. That is, the diode prevents back-feeding the main power stage202's output from the respective outputs of the adapter. Additionally, each chopper stage may include a corresponding chopper controller215. This controller may operate chopper switch214with a duty cycle selected to ensure that the corresponding output voltage Vo1+ is regulated to an appropriate value. For example, chopper controller215can compare the output voltage Vo1+ to a suitable reference voltage, with the difference between the two (the error signal) being compared to a ramp signal to generate a PWM switching signal applied to the gate of chopper switch214. For the example USB-PD applications, low voltage switching devices (e.g., 30V rated) may be used for chopper switches214. Additionally, adapter200may implement the USB-PD standard, such that a device connected to any one of outputs Vo1-Vo4may negotiate a suitable output voltage, e.g., 5V, 9V, 15V, 20V, etc. To that end, each chopper controller215may be connected to a controller220. Controller220may be implemented using any suitable combination of analog circuitry, digital circuitry, and/or programmable controllers or processors configured to operate as further described herein. Such circuitry may be implemented as any combination of discrete circuitry, integrated circuits, application specific integrated circuits (ASICs), field programmable gate arrays (FPGAs), and the like. Controller220may then serve to: (1) negotiate a USB-PD contract (including, e.g., output voltage, current, and power requirements) with the respective devices connected to DC outputs Vo1+-Vo4+; (2) configure feedback loop204to cause main power stage202to produce an output voltage V0+ that is greater than or equal to the largest required output voltage Vo1+-Vo4+; and (3) configure each chopper controller215to operate a corresponding chopper switch214with a duty cycle that reduces the main power stage output voltage/chopper stage input voltage V0+ to the appropriate output voltage level Vo1+-Vo4+. Controller220may configure main power stage202to generate the required output voltage V0+ by determining/selecting the reference signal provided to feedback loop204, as described in greater detail below with reference toFIG.4. Similarly, controller220may configure each chopper stage to generate the required output voltage Vo1+-Vo4+ by altering the reference signal provided to the chopper controller feedback loop. Additionally, adapter200may include, in the respective regulator stages212a-212d, power delivery switches218. These switches may be used to selectively disconnect/disable a respective output stage when its operation is not required or in the event of a fault. However, these power delivery switches218may be omitted, as illustrated in converter201ofFIG.3. Converter201is substantially similar to converter200, except that regulator stages213a-213domit power delivery switches218. In such configurations, chopper switches214may be used to disconnect/isolate an output as appropriate. Either of converters200or201can address the above-mentioned deficiencies of prior art multi-output adapters by providing for a reduced number of switches per output, i.e., as few as one switch per additional output in converter201. As a result, each switch may be sized to allow for the full output power to be delivered to each output. In such cases, controller220(e.g., via control logic221, discussed below) should be configured such that when power contracts are negotiated for the respective outputs, the total power capacity of the adapter is not exceeded. Suitable output current and/or power limiting may be provided to the respective regulator stages202a-212d, for example, by providing suitable signals to chopper controllers215. FIG.4further illustrates various aspects of adapter201, particularly with respect to controller220. Controller220may include internal control logic221, which may operate as described above (and further below with reference toFIG.6) to control main power stage202and chopper regulator stages213a-213dto generate the respective output voltages Vo1+-Vo4+. Thus, control logic221may negotiate power contracts with the respective output loads, set the reference voltage for main power stage202feedback loop204, and set the reference voltages for chopper controllers215(as well as selectively enabling/disabling chopper controllers215, as appropriate). Because each of regulator stages213a-213dis a chopper, it can only decrease the input voltage V0+ that it receives from main power stage202. Thus, control logic221must be configured to, after negotiating the respective output power contracts, provide a reference voltage to main power stage202's feedback loop204that corresponds to the highest negotiated voltage. Thus, controller220may include suitable circuitry for generating internal reference voltages corresponding to the available output levels (e.g., a 5V, 9V, 15V, and 20V) for USB-PD applications. Various reference voltage generation techniques are known, and thus are not repeated in detail here. Also illustrated inFIG.4are feedback loop switches SM1-SM4, which may be part of controller220. Feedback switches SM1-SM4may be a silicon, silicon carbide, or gallium nitride MOSFET, or any other suitable semiconductor switching device appropriate to the particular application. Feedback switches SM1-SM4may be operated by control logic221to provide a suitable voltage feedback signal to feedback loop204of main power stage202. More specifically, control logic221may be configured to selectively energize the one of switches SM1-SM4corresponding to the highest negotiated output voltage Vo1+-Vo4+. As a result, the highest output voltage will be provided as the feedback signal to main power stage202's feedback loop204, which will cause its output voltage V0+ to correspond to the highest output voltage required. Controller220can also provide the appropriate reference voltage to chopper controller stages215, allowing them to reduce V0+ to a level suitable for their respective outputs. Additionally, switching of feedback switches SM1-SM4could be omitted or delayed, as the intrinsic body diode of switches SM1-SM4would allow for the highest output voltage of Vo1+-Vo4+ to be coupled to feedback loop204. Similarly, feedback switches SM1-SM4could be replaced with diodes, which would also allow for the highest output to be passed to feedback loop204. Additionally, one or more components of regulator stages213a-213dcould be incorporated into controller220. For example, chopper control circuits215could be integrated with controller220. Similarly, main stage feedback loop204and switch controller206could be integrated with controller220. In some embodiments, the power switches themselves, including one or more of chopper switches214, power delivery switches218(fromFIG.2), and main switch S1could be integrated with controller220to form a single integrated circuit capable of implementing a multi-output power supply as described herein. Likewise, the various rectifier devices of main power stage202and regulator (chopper) stages213a-213dcould also be integrated into such a single integrated circuit. Although, as a practical matter, it may be desirable to integrate all of the various control circuitry into controller220, while leaving the power stage switches and diodes separate, allowing for one integrated controller to be easily used to provide different converter power levels by coupling to power devices having suitable ratings and capacities. FIG.5illustrates voltage plots530corresponding to an exemplary operating sequence of a multi-output AC-DC adapter as described above. Prior to time T0, the converter may be off. At time T0, the converter may be turned on, for example by plugging in the adapter to an AC power source. Thus at time T0controller220may begin operating main power stage202to generate an output voltage V0(voltage trace531) corresponding to a 5V level. As no loads are yet connected, this 5V level provides the bias level required for controller operation and basic switching functionality. At time T1, a first load may be connected to the first output. This first load may negotiate a 5V power delivery contract, meaning that voltage V1, trace532, increases to 5V. Then, at time T2, a second load may be connected to the second output. This second load may also negotiate a 5V power delivery contract, meaning that voltage V1, trace533, increases to 5V. Then at time T3, the load connected to the first output may renegotiate to a higher voltage contract, e.g., 20V. This may be because the initially connected load now has an increased power requirement, or because a new load has been connected. In either case, both the main power stage output voltage V0and first stage output V1can correspondingly increase to 20V. Subsequently, at time T4, the load connected to the second output may renegotiate to a higher voltage contract, e.g., 15V. This may also be because the initially connected load now has an increased power requirement, or because a new load has been connected. In either case, because the new voltage level is still below the main power stage level, no change to the main power stage output voltage is required. Similarly, at time T5, a load may be connected to third output, initially negotiating a 5V contract for output voltage V3, plotted with curve534. As above, because this negotiated level is below the current output voltage of main stage202, no change to those voltage are required. Subsequently, at time T6, the load connected to the first output may renegotiate its power contract to the 5V level. As a result, controller220can cause main power stage202to drop its output voltage level to the 15V level required by the load connected to the second output, which is now the highest output voltage. Then, at time T7, the third load may renegotiate to a higher voltage contract, e.g., 9V; however, because the main power stage is already providing 15V, no changes to its output are required. Similarly, at time T8, a fourth load may be connected to the fourth output, negotiating a 5V contract for V4, plotted by curve535. Because the main power stage is already generating a 15V output, no further change is required. The above-described sequence is merely one example of a possible operating sequence meant to provide a concrete illustration of the application of the control logic.FIG.6illustrates a flowchart640that more generally describes the operating sequence employed by controller220. Beginning at block641, controller220can determine the output voltage of each641. As one example, this may be determined by virtue of the USB-PD contract negotiation performed by controller220. Then, in block642, controller220can regulate the main power stage202to produce the highest output voltage of the respective stages. This can be performed as described above by providing suitable reference and feedback signals to the feedback loop204of main power stage202. Also, in block643, controller220can regulate the respective regulator (chopper) stages to corresponding output voltages. This can be performed as described above by providing suitable reference signals to the respective stages. If more than one output has the highest voltage, the controller may either parallel their outputs to the main power stage feedback loop204or can select either of them. FIG.7illustrates a rectifier circuit700including an AC input701coupled to a rectifier701. Rectifier701can produce an output voltage Vin+ that can be provided to the converter circuits described above with respect toFIGS.1A,1B,2,3, and4. The foregoing describes exemplary embodiments of multi-output AC/DC converters. Such systems may be used in a variety of applications but may be particularly advantageous when in conjunction with multiple personal electronic devices, such as notebook computers, tablet computers, smartphones, and various accessories, such as wireless earphones, styluses, and the like. Although numerous specific features and various embodiments have been described, it is to be understood that, unless otherwise noted as being mutually exclusive, the various features and embodiments may be combined in various permutations in a particular implementation. Thus, the various embodiments described above are provided by way of illustration only and should not be constructed to limit the scope of the disclosure. Various modifications and changes can be made to the principles and embodiments herein without departing from the scope of the disclosure and without departing from the scope of the claims. Additionally, it is well understood that the use of personally identifiable information should follow privacy policies and practices that are generally recognized as meeting or exceeding industry or governmental requirements for maintaining the privacy of users. In particular, personally identifiable information data should be managed and handled so as to minimize risks of unintentional or unauthorized access or use, and the nature of authorized use should be clearly indicated to users. The techniques presented and claimed herein are referenced and applied to material objects and concrete examples of a practical nature that demonstrably improve the present technical field and, as such, are not abstract, intangible or purely theoretical. Further, if any claims appended to the end of this specification contain one or more elements designated as “means for [perform]ing [a function] . . . ” or “step for [perform]ing [a function] . . . ”, it is intended that such elements are to be interpreted under 35 U.S.C. 112(f). However, for any claims containing elements designated in any other manner, it is intended that such elements are not to be interpreted under 35 U.S.C. 112(f). | 24,699 |
11863085 | DESCRIPTION OF SPECIFIC EMBODIMENT OF THE INVENTION FIG.1ashows an exemplary embodiment of a converter assembly according to the invention for converting a DC voltage Vdcfrom a DC voltage source, e.g. a battery, a fuel cell or a DC voltage intermediate circuit, into a 3-phase AC voltage (N=3) with the phases L1, L2, L3for connection to an AC voltage network. For this purpose, a connected inverter unit1is provided. This comprises an active bridge inverter with six half bridges2,2′,2a,2a′,2b,2b′, wherein in each case a phase L1, L2, L3is supplied via two half bridges (M=2). The half bridges in each case comprise two electronically switchable semiconductor switches, which are connected to an electronic control unit3. In this exemplary embodiment, the semiconductor switches are designed as SiC switches and have a high dielectric strength. The control unit3switches the semiconductor switches in a pulse width modulation process with a frequency of around 33 kHz in order as far as possible to be able to form the ideal sinusoidal shape for each of the phases. Furthermore, the control unit3is designed to activate those half bridges pairs which supply the same phase in a phase-offset manner, in such a way that the current of this phase is divided substantially equally between the two half bridges. For example, the control unit3first activates the first half bridge2for a certain period of time tonand then the half bridge2′ for an identical period of time ton. This halves the power transmitted per half bridge and doubles the frequency of the PWM process per phase. Consequently, the ripple in the output current decreases and interfering feedback into the DC voltage intermediate circuit is also reduced. In this exemplary embodiment, the outputs of two half bridges which supply the same phase are interconnected via interleaving chokes4,4′,4a,4a′,4b,4b′. The interleaving chokes are current-compensated and wound on a common iron core for each phase. This allows a particularly ripple-free operation of the converter assembly. In order to damp electrical common-mode interference, the phases L1, L2, L3are connected to a respective winding5,5′,5″ of a common-mode choke10with a common magnetic core. This compensates common-mode interference in the phases. A first resistance-damped capacitor circuit6is provided at the output of the interleaving chokes4,4′,4a,4a′,4b,4b′ which, in conjunction with the leakage reactance (transverse reactance) of the interleaving chokes, forms a first LC filter stage8. A second resistance-damped capacitor circuit7is provided at the output of the common-mode chokes5,5′,5″ which, in conjunction with the leakage reactance (transverse reactance) of the windings5,5′,5″ of the common-mode choke10, forms a second LC filter stage9. The first and second capacitor circuits comprise capacitors in each case arranged in a star connection and provided with parallel resistors; the neutral point of the second capacitor circuit7can be earthed via a PEN or PE connector. In an exemplary embodiment, not shown, a damping resistor is arranged between the centre of the DC voltage intermediate circuit and the neutral point of the second capacitor circuit7. The intermediate circuit is thus stabilised with regard to common-mode interference (capacitively coupled to PEN), and the common-mode interference then only occurs in the form of an alternating signal at the neutral point of the first capacitor circuit. In this exemplary embodiment, the converter assembly is designed for a DC voltage of around 850 V, and the inverter unit1is designed to generate a 3-phase line voltage with an amplitude of 400 V and a phase current of 630 A at a frequency of 50 Hz. The DC voltage Vdcin the DC voltage intermediate circuit is symmetrically stabilised (not shown) in relation to the ground potential, for example+420 V/−420 V. This reduces earth currents and insulation stresses in downstream units. In the present exemplary embodiment, the interleaving chokes4,4′,4a,4a′,4b,4b′ are designed with non-bifilar upright windings with around nine windings per leg on a nanocrystalline C-core with high relative magnetic permeability (μr of around 40,000), a core cross-section of around 17 cm2and a very narrow air gap of around 150 μm. The inductance of each individual winding is around 500 pH, the coupling factor is 0.97, the longitudinal inductance around 7.5 pH and the transverse inductance around 1.94 mH. The assigned first capacitor circuit6has a capacitance of around 30 μF per phase, so that the cut-off frequency of the low pass formed by the first filter arrangement8assumes a value of around 67 kHz: f=1LC=17.5µH·30µF=66.67kHz This corresponds to around 2 times the switching frequency of 33 kHz, so that the interference can be effectively filtered through the switching processes. In the present exemplary embodiment, the common-mode chokes5,5′,5″ in each case comprise around 4 windings on a nanocrystalline C-core with high relative magnetic permeability (μr of around 40,000) and a core cross-section of around 14 cm2. The inductance of each individual winding is around 1.8 mH, the leakage reactance around 3.5 pH at a frequency of around 48 kHz. The assigned second capacitor circuit7has a capacitance of around 11 μF per phase, so that the cut-off frequency of the low pass formed by the second filter arrangement9assumes a value of around 161 kHz: f=1LC=13.5µH·11µF=161.16kHz This staggered arrangement of two low-pass filters enables efficient filtering of high-frequency interference without the need for additional EMC filter components. FIG.1bshows another exemplary embodiment of a converter assembly according to the invention for converting a DC voltage Vdcfrom a DC voltage source into a 3-phase AC voltage (N=3) with the phases L1, L2, L3for connection to an AC voltage grid. In this exemplary embodiment, rather than two separate capacitor circuits, a combined capacitor circuit11is provided. This interacts with the interleaving chokes4,4′,4a,4a′,4b,4b′ as well as with the windings5,5′,5″ of the common-mode choke10to form two schematically indicated filter stages8,9. When dimensioning the two filter stages, it must be ensured that the capacitor circuit11is effective both for the cut-off frequency of the first LC filter stage8and also for the cut-off frequency of the second LC filter stage9; consequently, the elements cannot be dimensioned independently of each other as in the exemplary embodiment according toFIG.1a. Otherwise, this exemplary embodiment corresponds to the exemplary embodiment according toFIG.1a. FIG.1cshows a further exemplary embodiment of a converter assembly according to the invention for converting a DC voltage Vdcfrom a DC voltage source into a single-phase AC voltage (N=1) with a phase L and a connected neutral conductor N. Both the phase L and the neutral conductor N are provided via a connected inverter unit1with in each case two half bridges (M=2). Again, rather than two separate capacitor circuits, a combined capacitor circuit11is provided. Otherwise, this exemplary embodiment corresponds to the exemplary embodiment according toFIG.1b. The invention is not limited to the described exemplary embodiment, but includes all converter assemblies according to the following claims, and in particular their use in test stands for vehicles. LIST OF REFERENCE SYMBOLS 1inverter unit2,2′,2a,2a′,2b,2b′ half bridge3control unit4,4′,4a,4a′,4b,4b′ interleaving choke5′,5″ winding6first capacitor circuit7second capacitor circuit8first filter stage9second filter stage10common-mode choke11combined capacitor circuit | 7,646 |
11863086 | DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS Definitions As used herein, including in the claims, unless otherwise indicated in context, the following terms shall have the following meanings. A cylinder (or circular cylinder) is a curvilinear surface, not necessarily a solid. A cylinder is the locus of points traced by a finite-length line segment rotated about an axis, where the line segment is co-planar with the axis, but the line segment is not perpendicular to the axis. The line segment may be straight, curved or formed of a plurality of straight and/or curved sub-segments. If the line segment is parallel to the axis, the cylinder is a conventional right cylinder. If, however, the line segment is not parallel to the axis, the cylinder may be tapered, i.e., shaped like a cone or a portion of a cone. A circular hollow cylinder (or cylindrical shell) is a three-dimensional region bounded by two circular cylinders having the same axis, two parallel sides and two parallel (not necessarily equal diameter) annular bases perpendicular to the cylinders' common axis. Commutation is a process of switching electric current and/or voltage in motor phases to generate motion. Brushed motors have physical brushes to achieve this process twice or more per rotation, while brushless direct current (BLDC) electric motors do not. Due to the nature of their design, BLDC motors can have any number of pole pairs for commutation. Similarly, electrostatic motors can have any number of electrodes for commutation. An electrode is an electrical conductor through which electric current and/or charge enters or leaves an object, substance or region. An electret is a material that retains a permanent or semi-permanent electric charge after exposure to a strong electric field. A dielectric (or dielectric material) is an electrical insulator that can be polarized by an applied electric field. An electrical conductor is a material having an electrical resistivity less than about 10−6Ω-m. An electrical insulator is a material having an electrical resistivity greater than about 10−6Ω-m. A partial vacuum is a region with a gaseous pressure less than about 40 Pa. A partial vacuum is a dielectric. A fluid is any liquid, gas, supercritical fluid or multiphase mixture of liquid, gas and/or supercritical fluid that has a suitably high dielectric breakdown strength, permittivity and/or low viscosity. Cylindrical Electrostatic Motor Embodiments of the present invention provide electrostatic motors that improve upon the efficiency, weight and cost of conventional electromagnetic electric motors and electric motor-driven systems. These embodiments operate more efficiently over a wider range of speeds than conventional electric motors and known electrostatic motors, weigh less and cost less. By operating more efficiently than conventional motors, machines powered by these embodiments are less costly to operate, because they consume less power. The improved efficiency of these embodiments, in combination with their low weight, enable vehicles to travel further and/or bear a larger payload, since a lower volume and mass of batteries, or other energy storage devices, is required. The lower cost and weight of these embodiments is due to the replacement of expensive and heavy materials used in conventional electromagnetic motors, such as magnets, copper wire and electrical steels, with inexpensive, lightweight electret materials and/or thin, conductive electrodes. These motors can be used to power a wide range of machines that now rely on electromagnetic electric motors. The list of potential applications includes, but is not limited to: vehicles (including primary drivetrain and auxiliary motors for motor vehicles, drones, etc.), aircraft, watercraft (including boats and underwater vehicles), electric tools, robots, manufacturing/material handling equipment, construction equipment, HVAC (heating, ventilation, and air conditioning) equipment, toys and medical devices. The motors are also capable of operating as generators, so they have applications in electricity generation, energy scavenging and hybrid motor/generators. FIG.1is an isometric diagram, andFIG.2is a perspective cut-away diagram, of an assembled electrostatic motor100, according to an embodiment of the present invention.FIG.1shows a case102, front end cap104of the case102, shaft106and axis108.FIG.2shows a cylindrical rotor202with a plurality of alternatingly charged electrets, represented by electrets204and206, disposed on an outside surface of the rotor202. The rotor202rotates about the axis108. A plurality of electrodes, represented by electrodes208,210,212and214, is disposed on an inside cylindrical surface of a dielectric substrate215to form a stator216. Ones of the electrets204-206of the rotor202register with, counterface, are coaxial with, and spaced apart from, the electrodes208-214of the stator216to define a cylindrical shell between the electrets204-206and the electrodes208-214, although the number of electrets204-206need not equal the number of electrodes208-214. In some embodiments, the cylindrical shell may be on the order of 100s of μm thick. In other embodiments, the cylindrical shell may be thicker or thinner than 100s of μm thick. A dielectric fluid (not shown inFIG.2), or a partial vacuum, fills the cylindrical shell. Force generated by any electric motor is defined by the well-known Lorentz force equation (1): F=qv×B+qE(1) In magnetic motors, the value of the term qv depends on an electric current in a coil of wire, and the value of the term B depends on the strength of a magnetic field from a permanent magnet or an electromagnet. In an electrostatic motor, the value of the term qE depends on the strengths of a static charge (q) and an electric field (E). FIG.3is an exploded isometric view of the electrostatic motor100.FIG.3shows the case102, front end cap104and shaft106, front and rear bearings300and302, respectively, rear end cap304, the dielectric substrate215on which are formed or disposed drive electrodes306(corresponding to electrodes210-224inFIG.2), such as in slots308, and a rotor assembly310. Collectively, the dielectric substrate215and the electrodes306form the stator216. FIG.4is an exploded isometric view of the rotor assembly310, including the shaft106, an electret cylinder400, which includes alternatingly polarized electrets402(or alternatively electrically conductive bands, or alternatively both electrets and electrically conductive bands), and front and rear end caps404and406, respectively. FIG.5is an end view, andFIG.6is a cross-sectional view, of the electrostatic motor100.FIG.7is an enlarged view of a portion ofFIG.6, as indicated by an arrow600. Much of the motor100can be empty, as indicated at602, thereby reducing its weight, compared to conventional electromagnetic motors. A cylindrical shell700is defined between the electrets402and the drive electrodes306. FIG.8is a side view, andFIG.9is a cross-sectional view, of the electrostatic motor100. A portion900ofFIG.9is shown enlarged inFIGS.12and13.FIG.10is a cross-sectional view of the electrostatic motor100, similar toFIG.9, but with additional detail.FIG.10illustrates a rotor1000and a stator1002with a dielectric fluid1004therebetween.FIG.11is an enlarged view of a portion1006ofFIG.10, showing forces, exemplified by repulsion and attraction forces1100and1102, respectively, generated within the motor100. Dimensions of the electrets and electrodes are exaggerated for clarity. FIG.12is an enlarged view of a portion ofFIG.9showing motor components, andFIG.13is the enlarged view of the portion ofFIG.9showing forces and resulting rotation of the rotor assembly310. InFIG.13, attractive and repelling forces between drive electrodes and electrets (or rotor electrodes) are indicated at1300and a direction of net rotation of the motor100, caused by torque resulting from the forces1300, is indicated by an arrow1302. Rotor with Electrodes Embedded in Bulk Electret Material to Facilitate Charging In some embodiments, the rotor includes electrically conductive material embedded in the bulk electret material to form embedded charging electrodes. These electrodes facilitate contact charging, without need for external charging fixtures, thus simplifying the process for manufacturing an electret rotor. Furthermore, the rotor surface can be submerged in dielectric fluid during charging, which prevents air breakdown near the rotor surface or accumulation of charged particles on the rotor, providing further advantages.FIG.14is an enlarged view of a portion900ofFIG.9, which shows a cross-section through the motor, showing regions of conductive material within the rotor that form embedded charging electrodes1400. Optionally, additional regions of electrically conductive material (not shown) may be disposed on the outer surface of the rotor, radially registered with, but radially spaced apart from, the embedded charging electrodes1400. Embodiments with both outer and embedded charging electrodes provide choices for charging the electrets. Charging the electrets can be performed by applying a voltage across circumferentially adjacent charging electrodes on the outside of a rotor, across circumferentially adjacent charging electrodes on the inside of a rotor and across radially adjacent charging electrodes, i.e., one charging electrode on the outside of the rotor and one charging electrode on the inside of the rotor. Charging the electrets according to these choices yields different charge patterns and charge densities. To embed electrodes within a bulk electret material, a layer of conductive material may be deposited on the surface of a cylinder of electret material using known methods, such as sputtering or chemical vapor deposition (CVD). The conductive material is patterned, such as through shadow masking, to form discrete electrodes. Next, a layer of electret material is deposited on top of the electrodes deposited in the previous step, burying most of the conductive material. The electret material may be deposited by over-molding, chemical vapor deposition or any other suitable method. The final layer of electret material is prevented from covering a small region of each embedded electrode, such as by masking or other suitable method, such that portions of the embedded electrodes remain physically accessible for connection to a voltage source. The electret material is then polarized by connecting adjacent, embedded electrodes to positive and negative poles of a voltage source, respectively, which thereby creates an electric field within the electret material between the adjacent embedded electrodes, causing that portion of material to become polarized. In some embodiments, charging electrodes1400of every other electret are electrically connected together within the rotor, forming two circuits, one circuit for electrets that are to be positively charged, and the other circuit for electrets that are to be negatively charged. Nevertheless, each electret is considered to have a respective charging electrode1400. In the embodiment shown inFIGS.1-13, electrets are disposed on the surface of the rotor assembly310, and electrodes306(208-214) are disposed on the inside surface of the stator216. However, in some other embodiments, the electrets are disposed on the inside surface of the stator216, instead of or in addition to the electrodes306(208-214), and other electrodes are disposed on the surface of the rotor assembly310. In these embodiments, electrical signals are sent to the rotor assembly310, rather than the stator216, such as via brushes or slip rings, to commutate the motor100. In the embodiment shown inFIGS.1-13, the rotor assembly310is disposed within the stator216, and the rotor assembly310is a cylinder with an outside surface that counterfaces an inside cylindrical surface of the stator216. The rotor assembly310spins within the stator216. However, in some other embodiments, the rotor assembly310spins outside the stator216. In these other embodiments, the stator is configured as a cylinder with electrodes and/or electrets on its outside surface, and the rotor defines an inside cylindrical surface that counterfaces the stator. In general, electrostatic motors generate torque by electrostatic attraction or repulsion between charged surfaces. Motors, according to embodiments of the present invention, include a cylindrical rotor202(FIG.2) with an outer surface made of a plurality of polarized electret materials and/or conductive electrodes, represented by electrets204-206. The motors also include a case102with a cylindrical arrangement of conductive electrodes, represented by electrodes208-214that surround the rotor202. The electrodes208-214collectively form a stator216. A thin cylindrical shell of dielectric fluid (not shown inFIG.2), but shown at1004inFIGS.10and11, is situated between the outer surface of the rotor202and the stator216. In some embodiments, as little as a few milliliters of dielectric fluid fill the space between the outer surface of the rotor202and the stator216. Some embodiments use electret rotors. In these embodiments, electrets204-206(402) and drive electrodes208-214(306) are arranged such that by applying positive and negative voltages to the drive electrodes306, electrically charged regions402on the rotor102are electrostatically attracted to, or repelled from, nearby electrodes306, causing a net torque to be applied to the rotor, causing it to rotate, as schematically illustrated inFIG.11. The electrodes208-214(306) are manufactured onto a substrate material, such as alumina, with a high dielectric constant to prevent discharges (arcing) between the electrodes. By varying the polarities of voltages on the drive electrodes208-214(306) over time, a sustained clockwise or counterclockwise torque is applied to the rotor1002, causing continuous rotation of the shaft. The magnitude of the torque depends on the intensities of the electric fields that the electrodes208-214(306) apply to the rotor1002, charge on the electrets204-206(402), number of counterfacing electrode/electret pairs (although there need not necessarily be equal numbers of electrets on the rotor as electrodes on the stator) and radius at which the electrets are disposed from the center of the shaft. Optionally, each electrode208-214(306) is surrounded by an electrically, floating-potential conductive guard ring or field plate, exemplified by ring1106, to shape the electric field of the electrode306, as is known in the art. The angle subtended by each electrode (equivalently, the circumferential width of each electrode) can be made very small, on the order of 1 Thus, the stator may have a very large number of drive electrodes. The number of drive electrodes can thus greatly exceed the number of polarized regions of electret material, thus a very large number of phases may be independently activated to reduce torque ripple and improve motor efficiency. An external power supply (not shown) applies voltages to the drive electrodes, and a switching circuit (not shown) modulates the voltages, i.e., commutates the motor, depending on the real-time angular position of the rotor, such that a net torque is consistently applied in a desired orientation. The power supply and switching circuit energize the electrodes to commutate the motor in a manner analogous to a conventional brushless direct current motor. The angular position of the rotor may be sensed using an electrostatic method described herein, or by conventional angular position sensors, such as optical encoders, Hall effect sensors or the like. The switching circuit may include semiconductor devices, such as field-effect transistors, bipolar transistors, commutator brushes and other related components capable of switching high voltages, as is well known in the art. In an alternate embodiment, the electrets are either replaced or supplemented with conductive electrodes (“rotor electrodes”), and the rotor electrodes are charged by an external voltage source. Dielectric Fluid Between Rotor and Stator In general, large electric fields are not stable, because air becomes conductive, i.e., ionizes (breaks down), in an electric field exceeding approximately 3 MV/m. The dielectric fluid1004(FIG.10) in some motor embodiments enables very high electric fields to be applied to the rotor from the electrodes without breakdown, enabling much higher torque than would be possible with an air gap between the electrodes and the electrets/rotor electrodes. A low viscosity dielectric fluid is preferred, such as an alkane hydrocarbon fluid or fluorocarbon fluid, since low viscosity enables high shaft rotation speeds with low fluid frictional losses. However, a high viscosity dielectric fluid may be suitable where relatively low shaft speed is acceptable, such as in a robot arm. Suitable dielectric fluids may include alkanes, perfluorocarbons, purified water, silicone oil, mineral oil and other chemicals known in the art. Suitable nanoparticles may be added to the dielectric fluid to increase electrical breakdown strength. In an alternate embodiment, the dielectric fluid is a high partial vacuum established within the motor case, such as by a vacuum pump. A vacuum-compatible rotary seal, such as a ferrofluidic seal or end-face mechanical seal, as known in the art, enables the rotating motor shaft to penetrate the case without loss of vacuum. A high partial vacuum can replace the dielectric fluid because it has a very high electric field breakdown strength. In some embodiments, the partial vacuum has a pressure below about 1 mTorr (0.13 Pa). Specific vacuum requirements depend on electrode (or electret) spacing and drive voltage and may be calculated by reference to Paschen's law. Such a partial vacuum provides an efficiency advantage because fluid friction energy losses that would result from dielectric fluid are eliminated. In a related embodiment, a dielectric fluid is installed in the cylindrical shell between the rotor and case, but air in the remainder of the empty space within the case is replaced by a partial vacuum or an inert gas, such as nitrogen. Replacement of the air is advantageous for maintaining purity of the dielectric fluid, which may otherwise become contaminated by atmospheric gases within the air. Multiple Concentric Cylindrical Arrangement In another alternate embodiment, multiple concentric cylindrical arrangements, exemplified by cylinders2700and2702, of polarized electret material are attached to one or more discs, such as disc2704, that are attached to the shaft106, as shown inFIGS.26-28. One or more other discs, exemplified by disc2706, are attached to the case and support electrodes that are disposed in multiple concentric cylindrical arrangements, exemplified by cylinders2708,2710and2712. The electrodes extend from these discs2708-2712to apply electric fields to the multiple electret cylinders2700-2702. It should be noted that electrodes and electrets may be disposed on one or both surfaces of each cylinder2708-2712and2700-2702, for example as shown at2714and2716. This configuration provides more torque than with a single electret cylinder and is suited to applications where a large diameter is available for a motor, but length of the motor is limited. The disadvantage of this approach is that the weight of the motor is increased. FIG.15is an isometric view,FIG.16is a side view, andFIG.17is a cross-sectional view of the electret cylinder400of the rotor assembly310.FIG.18is an enlarged view of a portion1700of the electret cylinder400. As shown inFIG.18, in another embodiment, the electret rotor is coated in a thin layer of electrically conductive material1800, which concentrates electret charge near the surface of the rotor and also concentrates the electric field emanating from electrode pairs at the surface of the rotor. By concentrating both the electric field from the electrodes and the charge from the electrets close to the rotor surface, a higher torque can be applied to the rotor than if the conductive coating is absent. Furthermore, the electrically conductive material1800prevents direct physical contact between the dielectric fluid1004and the electrets402. Such contact can cause an accumulation of charge that damages the electrets402. In some embodiments, the layer of electrically conductive material1800is segmented by electrically insulative material, exemplified by electrical insulators1802. Optionally, the electrically conductive material1800, or ones of the segments of electrically conductive material, is connected to a voltage supply to recharge depleted electrets. Dielectric Fluid Filtration Performance of the dielectric fluid may degrade over time by infiltration of particles, bubbles and/or dissolved atmospheric gases. To exclude these contaminants, a rotary seal may be installed at the end(s) of the case, where the shaft protrudes. Suitable rotary seals include ferrofluidic seals and end-face mechanical seals. Another method for avoiding performance loss due to particle, bubbles and/or dissolved gases is to enable the dielectric fluid to be replenished from a reservoir of purified dielectric fluid. The dielectric fluid reservoir may be pressurized with an atmosphere of inert gas, such as nitrogen, and routed to the interior of the motor. When it becomes necessary to replace the fluid, a mechanical or electronically actuated valve may be opened to drain the contaminated dielectric fluid, and to allow the fluid stored in the reservoir to displace the contaminated fluid. The reservoir is preferably a detachable cartridge, which can be replaced as necessary to maintain optimal dielectric fluid quality. In another embodiment, a large pressure is externally applied to the dielectric fluid. When pressurized, the dielectric fluid can support a larger electric field without breakdown than when no pressure is applied. This beneficial effect increases approximately linearly with increased pressure above atmospheric pressure. The pressure may be generated mechanically, such as with a compressed spring that applies force to a piston situated in fluid communication with the dielectric fluid, or with a pressurized, inert gas, such as nitrogen, that applies pressure to a surface of the dielectric fluid. An additional benefit of this configuration is that the pressurized fluid acts as a hydrostatic bearing, which helps stabilize and lower friction on the rotor, and maintain a very small gap between the rotor and the stator. One advantage of the small gap between the rotor and the stator is that the motor can produce a large torque, relative to other configurations that would require a larger gap. Another advantage of the small gap is that the motor can be operated with lower voltages than would be required if the gap were larger. In a related embodiment, the surface of the rotor is patterned with three-dimensional features, such as chevron shapes exemplified by chevron1500inFIG.15, that locally increase the pressure of the dielectric fluid when the rotor is in motion, as indicated by an arrow1502, boosting the fluid's breakdown strength. Accumulation of charged particles on the surfaces of electrets may cancel the electret charges and degrade performance of the motor. Some embodiments include a charging and cleaning mode, in which a sufficiently large voltage is applied between each adjacent pair of electrodes. The large voltage creates a strong electric field gradient, which displaces entrapped particles from the surfaces of the electrets while also polarizing the electrets to a desired level of charge. Some embodiments include components to continuously, periodically or occasionally clean the contaminated dielectric fluid of particles, bubbles and/or dissolved gases. In some such embodiments, impellers extending from the rotor cause the dielectric fluid to be pumped through a filter, such as a porous membrane or an electrostatic or magnetic filter. An advantage of an electrostatic filter is that the motor's electric power supply may be sufficient to power the electrostatic filter. In both of the above embodiments, the impellers, which form an internal pump, may be replaced by a conventional, secondary pump, such as a peristaltic pump, scroll pump or other suitable fluid pump. Real-Time Angular Measurement of Rotor/Shaft Some embodiments detect real-time angular position of the rotor assembly310, without a need for additional sensors, such as optical encoders, magnetic sensors, resolvers or the like. It is useful to detect the rotor position, for accurate rotor positioning and speed control, and doing so without additional sensors simplifies the motor and reduces its cost, size and weights. In these embodiments, to measure the shaft106angle, the rotor is extended such that a small portion of each charged electret band, or rotor electrode, extends past the ends of the drive electrodes. A secondary arrangement of two or more sensing electrodes is attached to the interior of the case, near the extended portion of the rotor, and is electrically isolated from the electrodes that are used to apply torque to the rotor. As the rotor spins, the motion of the electrets causes electrical charges to be induced on the sensing electrodes. The physics of electrostatic induction causes the magnitudes of these charges to vary in relation to the angle of the shaft, and the charges can be measured using standard voltage measurement circuits, as are known in the art. The angular position of the shaft can be inferred by the magnitude of these measured voltages. Geometry of Cylindrical Electrostatic Motor Embodiments of the present invention include a particular geometric arrangement of electrets402, or rotor electrodes, and drive electrodes306that is distinct from previous motors. In particular, disposing the electrets402and/or electrodes on a cylindrical surface, rather than on a surface of a disk or on pegs, provides advantages. For example, all the electrets402and/or electrodes are disposed a radial distance (for example, the radius of the rotor's electret cylinder400) from the shaft106. Therefore, electrostatic forces1300acting on these electrets402and/or electrodes act through a moment arm equal to the radius to apply torque to the shaft106. In disc-based electrostatic motors, charges acting on portions of the disc close to the shaft apply far less torque than charges acting on portions of the disc close to the circumference of the disc. Furthermore, electret rotor embodiments combine two elements that have not previously been combined in a rotary motor: (1) electrets402and (2) dielectric fluid1004. Both surfaces in contact with the dielectric fluid1004, i.e., the surface of the rotor1000and the surface of the stator1002, are smooth, relative to conventional electric motors, thus these smooth surfaces create much less fluid drag than conventional motors, leading to improved efficiency of the electrostatic motor. The volume of dielectric fluid1004used to fill the gap700between the rotor and stator is very small, relative to other motors (such as the motor described in U.S. Pat. Publ. No. 2016/0344306) that use dielectric fluid, owing to the small distance between the rotor1000and the stator1002. Thus, the fluid weight is also small, the cost of the fluid is low compared to motors that require a larger volume of fluid, and the fluid is easier to dispose of. The embodiments described herein were arrived at with the aid of advanced computer modeling tools, such as finite element analysis, which were not available to earlier generations of electrostatic motor designers, and required multiple design iterations and experimentation with geometry and material properties. Furthermore, the electret rotor embodiments were inspired in part by our knowledge and experience with state-of-the-art electret materials and manufacturing methods, which were not known to earlier designers, such as Jefimenko and Walker, and also our analysis of state-of-the-art, high-breakdown dielectric fluids. Several textbooks have “taught away” from pursuing electrostatic motors at scales larger than the micro-scale, arguing that the breakdown strength of air presents a fundamental limitation, discouraging inquiry into the subject. Methods and Apparatus for Charging Defined Regions on a Cylindrical Electret Another aspect of the present invention involves apparatus and methods for contact charging defined regions on a cylindrical electret. Conventional contact charging is a well-known method for polarizing flat electrets. However, conventional contact charging methods are inadequate for charging defined regions on a cylindrical electret.FIG.19illustrates a contact charging fixture1900that may be used to polarize a cylinder of electret material400with multiple, positively charged regions and multiple negatively charged regions. The cylinder of electret material400is inserted into a bore1902of the contact charging fixture1900, by advancing the cylinder400in a direction shown by an arrow1904. The contact charging fixture1900includes an arrangement of electrically conductive electrodes, exemplified by electrodes1906,1908and1910, attached to a non-conducting cylinder1912. The non-conducting cylinder1912maintains physical separation between positively charged and negatively charged portions of the conductive electrodes1906-1910, thus preventing electrical shorting between them. FIG.20is a view of the contact charging fixture1900, with the non-conductive cylinder1912removed for clarity. The number of charging electrodes, here exemplified by charging electrode2000, is equal to the number of regions on the electret cylinder400that will become polarized during contact charging. A first electrically conductive ring2002at one end of the charging fixture is an electrically common point for one group of charging electrodes. A second electrically conductive ring2004at the opposite end of the charging fixture is an electrically common point for a second group of charging electrodes, to be charged with opposite polarity to those charged with electrodes common to the first conductive ring2002. To charge the electret cylinder, one pole from an electrical power supply (not shown) is connected to the first ring2002, and the opposite pole is connected to the second ring2004. Electric fields are thus created within the electret cylinder between positive and negative charging electrodes2000, causing electret material within the electric fields to be polarized. It is advantageous to apply pressure from electrodes to electrets during contact charging. The contact charging fixture1900may apply pressure to the electret cylinder, for example by selectively constricting the fixture around the electret cylinder or by expanding the fixture before insertion of the electret cylinder and then, after insertion of the electret cylinder, elastically contracting around the electret cylinder. The charging fixture's diameter may be made compliant by removing portions of material2006from one or both end rings2002and2004. InFIG.20, the end rings2002and2004are shown to be completely severed to define gaps2006and2008in the end rings. Compliance may be also be increased without severing the rings, such as by selectively removing material, as is commonly done to create flexural springs using well-known methods. To constrict the charging fixture about an electret cylinder, the end ring diameters may, for example, be reduced by wrapping a tensionable strap around either end ring or both end rings or around the structure shown inFIG.20or around the non-conductive cylinder1912, or by bridging a gaps2006and2008with respective screws (not shown), such that rotation of the screws draws opposing faces of the gaps together. FIG.21is a side view of the contact charging fixture1900with an electret cylinder400(not visible) inserted therein.FIG.22is a cross-sectional view through both the contact charging fixture1900and the electret cylinder400.FIG.23is an enlarged view of a portion2200ofFIG.22. A charging electrode2000is shown in a charging configuration, with an inner, curved surface interface2300of the charging electrode2000in intimate contact with a region that will become an electret402(once polarized) on the electret cylinder400. The curved surface2300may be coated with a conductive liquid, such as mercury, or another conductive material, such as gallium or a eutectic alloy, preferably with a lower melting point than the charging electrode to promote transfer of charges to the electret by liquid charging. Liquid-contact charging is a method known to those with ordinary skill in the art for charging of flat electrodes. Although the contact chargingFIG.1900has been described for use in creating electrets on the outside surface of a cylindrical surface, the contact chargingFIG.1900can, with suitable modifications, be used to create electrets on the inside surface of a cylindrical surface. In this case, the non-conductive surface1912may be placed on the inside of the electrodes1906-1910, rather than on the outside, and electrodes1906-1910may be placed inside the cylinder to be processed. After the fixture1900is inserted into the cylinder, the fixture1900may be expanded to apply pressure on the inside of the cylinder. Alternatively, the fixture1900may be constricted before being inserted into the cylinder, and then allowed to expand once inside the cylinder to apply the pressure to the inside of the cylinder. The contact charging fixture1900described with reference toFIGS.19-23is configured to form all the electrets402on the cylindrical surface at once. However, in an alternative embodiment, some of the electrodes1906-1910are omitted, so as to form a subset of the electrets402in a single step. The pressure from the contact charging fixture1900may be released, the fixture1900may be rotated, relative to the cylinder, and then another set of electrets may be formed. In some circumstances, such as when the electrets402are particularly small, it may be advantageous to form the electrets402in pairs or other small groups, relative to the total number of electrets402, rather than all at once. Another contact charging fixture, an embodiment of which is shown inFIGS.29and30, enables forming groups of electrets on a cylindrical surface, without forming all of the ultimate electrets at once. This embodiment facilitates repeating the process of forming groups of electrets on successive locations on the cylindrical surface, including rotating the cylinder between forming each group of the electrets. FIG.29is an isometric diagram illustrating a contact charging fixture2900disposed above a dielectric cylinder2902in preparation for forming a pair of electrets, andFIG.30is an isometric diagram illustrating the contact charging fixture2900in intimate physical contact with the dielectric cylinder2902workpiece. The contact charging fixture2900includes two electrodes2904and2906, respectively, and an electrically insulative member2908separating the two electrodes2904and2906. The two electrodes2904and2906and the electrically insulative member2908between the two electrodes2904and2906collectively form an electrode assembly2910. End surfaces2912and2914of the electrodes2904and2906that contact the dielectric cylinder2902should form cylindrical sectors. The dielectric cylindrical workpiece2902has a longitudinal axis2916. Each electrode2904and2906has a respective longitudinal axis2918and2920. The longitudinal axes2918,2920of the two electrodes2904and2906and the longitudinal axis2916of the workpiece2902are all parallel to each other. The electrode assembly2910is translatable between two positions, represented byFIGS.29and30, respectively. The electrode assembly2910is translatable along an axis2922that is perpendicular to the longitudinal axis2916of the workpiece2902. In use, the two electrodes2904and2906are electrically coupled to an electrical power supply3000(FIG.30), and the electrode assembly2910, specifically the electrodes2904and2906, is brought into intimate physical contact with the surface of the dielectric cylinder2902, as shown inFIG.30, creating an electric field3002within the dielectric cylinder2902and thereby forming two electrets on a surface of the workpiece2902. Pressure should be applied by the contact charging fixture2900, specifically the electrodes2904and2906, on the surface of the dielectric cylinder2902. Once the electrets are formed, the electrical power supply3000may be disconnected and the electrode assembly2910is withdrawn from the surface of the dielectric cylinder2902, as shown inFIG.29. Then, the dielectric cylinder2902, or the contact charging fixture2900, is rotated, for example as indicated by an arrow3004, a distance to index the contact charging fixture2900, relative to the dielectric cylinder2902, to a position of the next set of electrets. The forming process is repeated for each subsequent set of electrets. Although the electrode assembly2910is shown with two electrodes2904and2906, the electrode assembly2910may include any number of electrodes. Thus, although described as forming two electrets at a time, the contact charging fixture2900may form any number of electrets at a time. As with the contact charging fixture1900(FIG.19), with suitable modifications, the contact charging fixture2900may be configured to form electrets on an inside surface of a cylinder. Another embodiment of the present invention is a method for corona-charging defined regions in an electret cylinder. Corona charging of flat electrets is known. However, as noted, charging curved electrets poses problems for conventional charging methods.FIG.24is an isometric diagram of a mask2400with an aperture2402, through which ions may be guided to corona-charge a region defined by the aperture2402in an electret cylinder or other curved surface.FIG.25is an isometric/schematic diagram of the mask2400, without the electret cylinder, but with other components of a corona charging system, according to an embodiment of the present invention. These components include an electrode2500, a conductive mesh2502, and an array of one or more sharp, electrically conductive needles2504(a “needle array”). The wall of an electret cylinder (not shown for clarity) is positioned between the electrode2500and the aperture2402. A first voltage source2506is connected to the needle array2504, causing air to ionize near the needle points. The potential between the electrode2500, which has opposite polarity to the needles2504, accelerates ions through the aperture2402and onto the electret cylinder. The mask2400prevents ions from reaching areas of the electret cylinder, other than a region exposed through the aperture2402. The mesh grid2502is connected to a second voltage source2508with the same polarity as the first voltage source2506used to charge the needles2504, but of a lower amplitude. The mesh grid2502promotes uniform distribution of charge on the electret cylinder. The charged region on the electret is given a positive or negative charge, depending on the polarity of voltage sources2506and2508, respectively. Reversing the polarity of both voltage sources2506and2508, or exchanging the relative positions of (a) the needle array2504and mesh grid2502and (b) the electrode2500, causes an electret region to be charged to the opposite polarity. Alternating regions of positive and negative charge may be created on the electret cylinder by: (1) charging a region of the electret cylinder with a first charge, then (2) rotating the electret cylinder, relative to the mask, so the aperture2402reveals an uncharged region of the electret cylinder, then (3) corona charging the uncharged region with a charge opposite the first charge and repeating the process for the remaining desired regions. The needle array2504, the mesh grid2502, the mask2400and the electrode2500are maintained in a fixture orientation, relative to one another. In an alternative embodiment for corona charging, multiple regions on the electret cylinder may be simultaneously charged. In this embodiment, the mask2400contains multiple apertures2402, and the electrode2500forms a complete cylinder, or at least has portions disposed under each aperture2402. A separate needle array2504is positioned above each aperture2402. All the needle arrays2504are made electrically common to the needle array voltage source2506. The mesh grid2502forms a complete cylinder, enclosing the mask2400, or at least has portions disposed between each needle array2504and its corresponding aperture2402. Activating the voltage sources causes a pattern of polarized regions to be created on the electret cylinder. An equal number of polarized regions of the opposite polarity may be created by: (1) charging a plurality of regions of the electret cylinder, then (2) disconnecting the voltage sources, (3) rotating the electret cylinder, relative to the mask, such that the apertures2402align with un-polarized regions of electret material, and then (4) reconnecting the voltage sources2506and2508, but with reversed polarity. In another embodiment, two sets of voltage sources, needle arrays, mesh grids and electrodes are used, with each set configured to charge the electret cylinder with an opposite charge. In this embodiment, the mask defines an aperture for each region of the electret cylinder that is to be charged. This embodiment does not require rotating the electret cylinder. Both polarities of regions on the electret cylinder may be charged simultaneously, or each polarity may be charged in turn. Contact charging, liquid contact charging and corona charging are described in Kao, Kwan Chi, “Dielectric Phenomena in Solids,” ISBN 9780123965625, Academic press, 2504. Imbedded Electrets Rather than form the electrets in the cylinder of electret material400, the electrets may be formed as separate strips or other suitable shapes and attached to the cylinder. The electrets may be attached to the surface of the cylinder, such as by a suitable adhesive. Alternatively, the electrets may be press fit or interference fit into suitable grooves in the surface of the cylinder. Optionally, the electrets may have a cross-sectional shape, such as a trapezoid, that locks into a similar cross-sectional shape defined by the groove, when the electret is pressed into the groove. Hydrostatic or Aerostatic Bearing FIG.31is an isometric diagram of a hydrostatic or aerostatic (depending on whether the dielectric fluid1004is a liquid or a gas) bearing3100between a rotor1000and a stator1002of an electrostatic motor. For simplicity of explanation, the term hydrostatic, as used herein, including in the claims, means hydrostatic or aerostatic. The hydrostatic bearing3100defines a plurality of ports, represented by ports3102,3104and3106, extending through respective electrodes402.FIG.32is a cross-sectional view of a portion of the rotor1000and stator1002showing one fluid port3102of the hydrostatic bearing3100extending through one of the electrodes306. Dielectric fluid flows, as indicated by an arrow3200, as a result of rotation3202of the rotor1000into the cylindrical shell defined between the rotor1000and the stator1002. The dielectric fluid flow supports the rotor1000within the stator1002. While the invention is described through the above-described exemplary embodiments, modifications to, and variations of, the illustrated embodiments may be made without departing from the inventive concepts disclosed herein. For example, although specific parameter values, such as dimensions and materials, may be recited in relation to disclosed embodiments, within the scope of the invention, the values of all parameters may vary over wide ranges to suit different applications. Unless otherwise indicated in context, or would be understood by one of ordinary skill in the art, terms such as “about” mean within ±25%. As used herein, including in the claims, the term “and/or,” used in connection with a list of items, means one or more of the items in the list, i.e., at least one of the items in the list, but not necessarily all the items in the list. As used herein, including in the claims, the term “or,” used in connection with a list of items, means one or more of the items in the list, i.e., at least one of the items in the list, but not necessarily all the items in the list. “Or” does not mean “exclusive or.” Disclosed aspects, or portions thereof, may be combined in ways not listed above and/or not explicitly claimed. In addition, embodiments disclosed herein may be suitably practiced, absent any element that is not specifically disclosed herein. Accordingly, the invention should not be viewed as being limited to the disclosed embodiments. | 44,918 |
11863087 | DETAILED DESCRIPTION Example structures (e.g. devices, systems, or other apparatus) described herein facilitate electrically induced mechanical movement, which may accordingly provide one or more tactile effects (e.g. tactile feedback), and can in addition (or instead) be able to electrically detect external mechanical movement exerted thereon. Examples merely typify possible variations. Unless explicitly stated otherwise, structures (e.g. structural components, such as layers or nodules) are optional and may be combined or subdivided, and operations are optional and may vary in sequence or be combined or subdivided. In the following description, for purposes of explanation, numerous specific details are set forth to provide a thorough understanding of various example embodiments. It will be evident to one skilled in the art, however, that the present subject matter may be practiced without these specific details. Various example embodiments of the structures discussed herein may be or include a special electrostatic actuator (e.g. electrostatic actuator structure) or sensor that includes at least a first electrode and a second electrode. The electrostatic actuator may also include one or more electrostatic actuation layers, and at least one of said electrostatic actuation layers includes: a first substrate film, which is intrinsically conductive or semi conductive, or includes a first conductive electrode layer, the first conductive electrode layer being a part of the first electrode, a second substrate film, which is intrinsically conductive or semi conductive, or includes a second conductive electrode layer, the second conductive electrode layer being a part of the second electrode, at least one of the first and second conductive electrode layers being insulated (e.g. electrically) from the respective first and second substrate films, and a grid array that includes a plurality (e.g. multitude) of elastic support nodules, the plurality of elastic support nodules being configured (e.g. arranged) between the first substrate film and the second substrate film, such that there is a compression space between the first and second conductive electrode layers, the compression space being not entirely filled with solid material; and wherein the electrostatic actuator is configured to compress (e.g. by a certain percentage or by a certain distance) in response to a voltage difference between the first and second electrodes (e.g. in response to the voltage difference exceeding or otherwise transgressing a threshold voltage difference, such that the electrostatic actuator compresses by a certain percentage or by certain distance when a sufficient voltage difference is applied between at least the first electrode and the second electrode). Accordingly, the electrostatic actuator may be or include an actuator that comprises: a first substrate having a first conductive surface (e.g., functioning as a first electrode); a second substrate having a second conductive surface (e.g. functioning as a second electrode), the first and second conductive surfaces facing toward each other across a compression space between the first and second substrates; and a plurality of elastic nodules spanning the compression space and separating the first and second conductive surfaces, the compression space being less than fully filled with solid (e.g. elastic) material, the compression space being configured to compress (e.g. by certain percentage or by a certain distance) in response to a voltage difference between the first conductive surface and the second conductive surface (e.g. in response to the voltage difference exceeding or otherwise transgressing a threshold voltage difference). FIG.1is a cross-sectional diagram illustrating at least a portion of a single electrostatic actuation layer100of an electrostatic actuator (e.g., an electrostatic actuator structure), according to some example embodiments. The electrostatic actuation layer100illustrated inFIG.1includes a first substrate101(e.g. a first substrate film) and a second substrate102(e.g. a second substrate film). In the example embodiments shown inFIG.1, the first substrate101and the second substrate102each may have typically a thickness of 5 to 100 micrometers. The first substrate101may include electrically insulating material or be intrinsically conductive or semi conductive, according to various example embodiments. In the example embodiments shown inFIG.1, the first substrate101is a film that includes or otherwise provides a first conductive layer103(e.g. a first conductive electrode layer) applied on top of the first substrate101. The first conductive layer103may form all or part of a first electrode. Accordingly, the first substrate101can be described as having a first conductive surface, whether the first conductive surface is intrinsically conductive, intrinsically semi conductive, or intrinsically insulative but topped with the first conductive layer103. Similarly, as shown inFIG.1, the second substrate102is a film that includes or otherwise provides a second conductive layer104(e.g. a second conductive electrode layer) applied on top of the second substrate102. The second conductive layer104may form all or part of a second electrode. Accordingly, the second substrate102can be described as having a second conductive surface, whether the second conductive surface is intrinsically conductive, intrinsically semi conductive, or intrinsically insulative but topped with the second conductive layer104. According to various example embodiments, the first conductive layer103, the second conductive layer104, or both, are insulated. The first substrate101(e.g. with the insulated first conductive layer103) has a layer of electrically insulating elastomer material105(e.g. an electrically insulating elastomer coating) applied on top of the insulated first conductive layer103of the first substrate101. Furthermore, there is a group (e.g. plurality or multitude) of elastic support nodules106arranged (e.g. in a grid array) and adhered on top of, or forming part of, the intrinsic structure of the layer of electrically insulating elastomer material105. In the example embodiments shown inFIG.1, the layer of electrically insulating elastomer material105may typically have a thickness of 1 to 30 micrometers. Furthermore, according to certain example embodiments, the elastic support nodules106each have a height of 5 to 100 micrometers and a diameter of 10 to 400 micrometers. In addition, according to some example embodiments, the elastic support nodules106each have an aspect ratio of height to maximum width, and the aspect ratio may have a maximum value of two. The layer of electrically insulating elastomer material105, the group of elastic support nodules106, or both may be applied using a suitable microfabrication technique (e.g. a deposition technique, such as a thin film deposition technique). The group of elastic support nodules106may be fully or partially made of a silicon-based organic polymer (e.g. polydimethylsiloxane (PDMS)), rubber (e.g. natural or synthetic), or any suitable combination thereof. Furthermore, the group of elastic support nodules106may be arranged less than 5 millimeters apart from each other in distance, and in some example embodiments, the inter-nodule distance is less than 500 micrometers. In certain example embodiments, the group of elastic support nodules106is arranged in a grid array, such as a spaced two-dimensional row-column grid array, as illustrated inFIG.2. In some alternative example embodiments, the group of elastic support nodules106is arranged in a spaced triangular grid array, as illustrated inFIG.3. After the group of elastic support nodules106has been applied to the first substrate101, the second substrate102with the second conductive layer104may be turned upside down and placed on top of the group of elastic support nodules106to form the electrostatic actuation layer100(e.g. a single electrostatic actuation layer that may be combined with one or more additional electrostatic actuation layers into a multi-layer electrostatic actuator structure). Before the second substrate102is placed on top of the group of elastic support nodules106, there may be a layer of adhesive applied to the second substrate102, to the group of elastic support nodules106, or to both. As the first substrate101and the second substrate102have been stacked one above the other (e.g. to form the electrostatic actuation layer100), the distance between the first conductive layer103and the second conductive layer104, which may be the distance between the first and second electrodes, in the electrostatic actuation layer100may be less than 1000 micrometers and, in some example embodiments, less than 200 micrometers. In the electrostatic actuation layer100shown inFIG.1, the distance between the first and second conductive layers103and104may be 80 micrometers. In the electrostatic actuation layer100, the group of elastic support nodules106provides a compression space between the first and second conductive layers103and104. In many example embodiments, the compression space is not entirely filled with solid material (e.g. the compression space is less than fully filled with solid material, such as solid elastic material). In the example embodiments shown inFIG.1, the compression space may be 80 micrometers thick, and the layer of electrically insulating elastomer material105may be 20 micrometers thick, thus leaving a space gap of 60 micrometers. This space gap in the compression space may be filled with fluid (e.g. air, nitrogen, or a dielectric liquid, such as dielectric hydraulic fluid) in places where the elastic support nodules106are not present. In the example embodiments shown inFIG.1, the electrostatic actuation layer100(e.g. one of multiple electrostatic actuation layers within the electrostatic actuator) may also include a group of limiting nodules, which may be arranged in a grid array (e.g. similar to the row-column grid array illustrated inFIG.2or the triangular grid array illustrated inFIG.3). This group of limiting nodules may be arranged between or among the elastic support nodules106and between the first substrate101and the second substrate102, for limiting the compression of the compression space and thereby limiting the compression of the electrostatic actuation layer100overall. In the example embodiments shown inFIG.1, the conductive surfaces of the first and second conductive layers103and104on both sides of the space gap, the elastic support nodules106, or both may be inherently hydrophobic, hydrophobically or superhydrophobically coated, hydrophobically or superhydrophobically treated, or any suitable combination thereof. The electrostatic actuation layer100is configured (e.g. arranged) to compress when a sufficient voltage difference is applied between at least the first conductive layer103(e.g. functioning as a first electrode) and the second conductive layer104(e.g. functioning as a second electrode). Hence, an electrostatic actuator (e.g. an electrostatic actuator structure) that includes one or more electrostatic actuation layers (e.g. electrostatic actuation layer100) may be configured to compress in response to such a voltage difference between the first conductive layer103and the second conductive layer104(e.g. between respectively first and second conductive surfaces thereof) exceeding a threshold voltage difference (e.g. a predetermined threshold voltage difference). Accordingly, an electrostatic actuator that includes one or more of such electrostatic actuation layers may be configured to be compressed in response to application of such a voltage difference. According to various example embodiments, the electrostatic actuator may be included (e.g. embedded) as part of a flexible or elastic substrate. For example, the electrostatic actuator itself may be intrinsically flexible, elastic, or both, and may be included in such a flexible or elastic substrate. The electrostatic actuation layer100shown inFIG.1may have an overall thickness in the range of approximately 15 micrometers to approximately 500 micrometers, including the thickness of the first and second conductive layers103and104and, in some example embodiments, the thickness of an adhesive layer between the elastic support nodules106and the second conductive layer104. FIG.4is a cross-sectional diagram illustrating at least a portion of a single electrostatic actuation layer200of an electrostatic actuator (e.g., an electrostatic actuator structure), according to some example embodiments. The electrostatic actuation layer200illustrated inFIG.4includes a first substrate201(e.g. a first substrate film) and a second substrate202(e.g. a second substrate film). In the example embodiments shown inFIG.4, the first substrate201may have a thickness of 20 micrometers, and the second substrate202may have a thickness of 50 micrometers. The first substrate201may be a film that includes or otherwise provides a first conductive layer203(e.g. a first conductive electrode layer) applied on top of the first substrate201. The first conductive layer203may form all or part of a first electrode. The second substrate202may be a film that includes or otherwise provides a second conductive layer204(e.g. a second conductive electrode layer) applied on top of the second substrate202. The second conductive layer204may form all or part of a second electrode. According to various example embodiments, the first conductive layer203, the second conductive layer204, or both, are insulated. The first substrate201(e.g. with the insulated first conductive layer203) has a layer of electrically insulating elastomer material205(e.g. an electrically insulating elastomer coating) applied on top of the insulated first conductive layer203of the first substrate201. Furthermore, there is a group (e.g. plurality or multitude) of elastic support nodules206arranged (e.g. in a grid array) and adhered on top of, or forming part of, the intrinsic structure of the layer of electrically insulating elastomer material205. In the example embodiments shown inFIG.4, the layer of electrically insulating elastomer material205may have a thickness of 20 micrometers. Furthermore, according to certain example embodiments, the elastic support nodules206each have a height of 80 micrometers. In addition, according to some example embodiments, the elastic support nodules206each have an aspect ratio of height to maximum width, and the aspect ratio may have a maximum value of two. The layer of electrically insulating elastomer material205, the group of elastic support nodules206, or both may be applied using a suitable microfabrication technique (e.g. a thin film deposition technique). The group of elastic support nodules206may be fully or partially made of a silicon-based organic polymer (e.g. PDMS), rubber (e.g. natural or synthetic), or any suitable combination thereof. Furthermore, the group of elastic support nodules206may be arranged less than 10 millimeters apart from each other in distance, and in some example embodiments, the inter-nodule distance is less than 2 millimeters. In certain example embodiments, the group of elastic support nodules206is arranged in a grid array, such as a spaced two-dimensional row-column grid array, as illustrated inFIG.2. In some alternative example embodiments, the group of elastic support nodules206is arranged in a spaced triangular grid array, as illustrated inFIG.3. As shown inFIG.4, the second substrate202may be microfabricated (e.g. etched) to provide wells208in the second substrate202. The wells208may be arranged to match the group of elastic support nodules206applied to the first substrate201. The wells208may be microfabricated using a suitable microfabrication technique (e.g. an anisotropic wet etching technique). The microfabricated second substrate202with the second conductive layer204may be turned upside down and placed on top of the group of elastic support nodules206, such that the wells208coincide with the group of elastic support nodules206to form the electrostatic actuation layer200(e.g. a single electrostatic actuation layer that may be combined with one or more additional electrostatic actuation layers into a multi-layer electrostatic actuator structure). In the electrostatic actuation layer200, the elastic support nodules206may reside in the wells208. Before the second substrate202is placed on top of the group of elastic support nodules206, there may be adhesive207applied to the wells208, to the group of elastic support nodules206, or to both. In the example embodiments shown inFIG.4, the depth of the wells208may be 90 micrometers. Other suitable depths may be used, depending on the thickness of the second substrate202to be microfabricated. As the first substrate201and the second substrate202have been stacked one above the other (e.g., to form the electrostatic actuation layer200), the distance between the first conductive layer203and the second conductive layer204, which may be the distance between the first and second electrodes, in the electrostatic actuation layer200may be less than 1000 micrometers and, in some example embodiments, less than 20 micrometers. In the electrostatic actuation layer200, the distance between said first and second conductive layers203and204may be 80 micrometers. Due to the wells208of the second substrate202coinciding (e.g. matching) with the group of elastic support nodules206on the first substrate201, the distance between the first and second conductive layers203and204may be considerably less than the height of the group of elastic support nodules206. In the electrostatic actuation layer200, the group of elastic support nodules206provides a compression space between the first and second conductive layers203and204. In many example embodiments, the compression space is not entirely filled with solid material (e.g. the compression space is less than fully filled with solid material, such as solid elastic material). In the example embodiments shown inFIG.4, the compression space may be 80 micrometers thick, and the layer of electrically insulating elastomer material205may be 20 micrometers thick, thus leaving a space gap of 60 micrometers. This space gap in the compression space may be filled with fluid (e.g. air, nitrogen, or a dielectric liquid, such as dielectric hydraulic fluid) in places where the elastic support nodules206are not present. In the example embodiments shown inFIG.4, the electrostatic actuation layer200(e.g. one of multiple electrostatic actuation layers within an electrostatic actuator) may also include a group of limiting nodules, which may be arranged in a grid array (e.g. similar to the row-column grid array illustrated inFIG.2or the triangular grid array illustrated inFIG.3). This group of limiting nodules may be arranged between or among the elastic support nodules206and between the first substrate201and the second substrate202, for limiting the compression of the compression space and thereby limiting the compression of the electrostatic actuation layer200overall. In the example embodiments shown inFIG.4, any one or more of the conductive surfaces of the first and second conductive layers203and204on either or both sides of the space gap, the elastic support nodules206, or both may be inherently hydrophobic, hydrophobically or superhydrophobically coated, hydrophobically or superhydrophobically treated, or any suitable combination thereof. The electrostatic actuation layer200is configured (e.g. arranged) to compress when a sufficient voltage difference is applied between at least the first conductive layer203(e.g. functioning as a first electrode) and the second conductive layer204(e.g. functioning as a second electrode). Hence, an electrostatic actuator (e.g. electrostatic actuator structure) that includes one or more electrostatic actuation layers (e.g. electrostatic actuation layer200) may be configured to compress in response to such a voltage difference between the first conductive layer203and the second conductive layer204(e.g. between respectively first and second conductive surfaces thereof) exceeding a threshold voltage difference (e.g. a predetermined threshold voltage difference). Accordingly, an electrostatic actuator that includes one or more of such electrostatic actuation layers may be configured to be compressed in response to application of such a voltage difference. According to various example embodiments, the electrostatic actuator may be included (e.g. embedded) as part of a flexible or elastic substrate. For example, the electrostatic actuator itself may be intrinsically flexible, elastic, or both, and may be included in such a flexible or elastic substrate. The electrostatic actuation layer200shown inFIG.4may have an overall thickness in the range of approximately 15 micrometers to approximately 500 micrometers, including the thickness of the first and second conductive layers203and204and, in some example embodiments, the thickness of the adhesive207applied between the wells208and the elastic support nodules206. FIG.5is a cross-sectional diagram illustrating at least a portion of an electrostatic actuator (e.g. electrostatic actuator structure) with four electrostatic actuation layers, according to some example embodiments. The illustrated electrostatic actuator includes four electrostatic actuation layers211-214. In the example embodiments shown inFIG.5, each of the electrostatic actuation layers211-214includes a corresponding first substrate (e.g. first substrate201) with a corresponding first conductive layer (e.g. first conductive layer203), and the first conductive layer may be part of a first electrode230. Similarly, each of the electrostatic actuation layers211-214includes a corresponding second substrate (e.g. second substrate202) with a corresponding second conductive layer (e.g. second conductive layer204), and the second conductive layer may be part of a second electrode240. Moreover, the first conductive layer may be insulated (e.g. from its first substrate, from another conductive layer, or from both), the second conductive layer may be insulated (e.g. from its second substrate, from another conductive layer, or from both), or both. Each of the electrostatic actuation layers211-214may further include a grid array that includes a group of elastic support nodules (e.g. elastic support nodules206), and the group of elastic support nodules may be arranged between their corresponding first and second substrates, such that there is arranged a compression space between the first and second conductive layers (e.g. first and second conductive layers203and204). In many example embodiments, the compression space is not entirely filled with solid material (e.g. the compression space is less than fully filled with solid material, such as solid elastic material). According to various example embodiments, the electrostatic actuator that includes the illustrated electrostatic actuation layers211-214may further include a high voltage driver250(e.g. a high voltage driver with a flyback-mode boost converter). In the example embodiments shown inFIG.5, the electrostatic actuation layers211-214are stacked one above the other, such that similar structural elements of the electrostatic actuation layers211-214coincide at least partially (e.g. coincide fully). For example, the elastic support nodules (e.g. elastic support nodules206) of one electrostatic actuation layer211may be fully or partially aligned over the elastic support nodules of another electrostatic actuation layer212. Similarly, the wells (e.g. wells208) of one electrostatic actuation layer211may be fully or partially aligned over the wells of another electrostatic actuation layer212. In alternative example embodiments, the electrostatic actuation layers211-214may be imbricatedly stacked (e.g. like overlapping roof tiles). The electrostatic actuator shown inFIG.5is configured to compress when a sufficient voltage difference is applied between the first electrode230and the second electrode240(e.g. compress in response to such a voltage difference exceeding a threshold voltage difference, such as a predetermined threshold voltage difference). Due to the coinciding, adhered, and joint structure of the electrostatic actuation layers211-214, the compression effect of the electrostatic actuator structure is increased substantially as a function of the number of coinciding electrostatic actuation layers. With the help of the adhered and joint structure of the electrostatic actuation layers211-214, the appearance of holes or inter-layer gaps is avoided, and the potential reduction of the compression effect due to the potential inter-pillar swallowing for air compression is covered. Furthermore, potential inter-layer bouncing or non-uniform separation (e.g. due to inter-layer pulling forces resulting from the compression of each layer) is also avoided. In the example embodiments shown inFIG.5, the electrostatic actuation layers211-216are stacked one above another (e.g. as a stack of electrostatic actuation layers), such that a bifunctional substrate (e.g. a bifunctional substrate film) forms both the first substrate of one electrostatic actuation layer (e.g. electrostatic actuation layer212) and the first substrate of an adjacent electrostatic actuation layer (e.g. electrostatic actuation layer211) among the electrostatic actuation layers211-216, one extending below the bifunctional substrate and the other extending above the bifunctional substrate. Alternatively, the bifunctional substrate may form both the second substrate of one electrostatic actuation layer (e.g. electrostatic actuation layer212) and the second substrate of an adjacent electrostatic actuation layer (e.g. electrostatic actuation layer211). Accordingly, such a bifunctional substrate can be considered as being or including (e.g. containing) the boundary between two adjacent electrostatic actuation layers (e.g. between the electrostatic actuation layers211and212), as well as being or including the junction of the two adjacent electrostatic actuation layers. The bifunctional substrate may include a mesh of elastomer material. Each of the electrostatic actuation layers211-214may have an overall thickness of approximately 365 micrometers. Accordingly, the thickness of the electrostatic actuator structure with the four electrostatic actuation layers211-214may be approximately four times 365 micrometers, thus resulting in a total thickness of approximately 1.46 millimeters. The electrostatic actuator structure may be hermetically sealed. The described compressing nodule structure, together with hermetic sealing, allows the compression of the electrostatic actuation layers211-214as a pump (e.g. a pneumatic pump or a hydraulic pump). In addition to the increased actuation produced by the combined compression of the electrostatic actuation layers211-214, when hermetically sealed, the above-described elastic support nodules allow the compression of these layers as a pump (e.g. a pneumatic pump or a hydraulic pump). Fluid (e.g. gas or liquid) configured to flow into or out of the space gaps within the electrostatic actuation layers211-214can be used to actuate various elastic structures that are affected by the pressure of the fluid. This effect may be used to create a textured surface or to actuate some part of a system via pneumatic or hydraulic means. FIG.6is a cross-sectional diagram illustrating at least a portion of a single electrostatic actuation layer300that may form all or part of an electrostatic actuator (e.g. electrostatic actuator structure), according to some example embodiments. The illustrated single electrostatic actuation layer300includes a first substrate301(e.g. a first substrate film) and a second substrate302(e.g. a second substrate film). In the example embodiments shown inFIG.6, the first substrate301may have a thickness of 100 micrometers, and the second substrate302may have a thickness of 175 micrometers. The first substrate301may be a film that includes or otherwise provides a first conductive layer303(e.g. a first conductive electrode layer) applied on top of the first substrate301. The first conductive layer303may form all or part of a first electrode. The second substrate302may be a film that includes or otherwise provides a second conductive layer304(e.g. a second conductive electrode layer) applied on top of the second substrate302. The second conductive layer304may form all or part of a second electrode. According to various example embodiments, the first conductive layer303, the second conductive layer304, or both, are insulated. The first substrate301(e.g. with the insulated first conductive layer303) has a layer of electrically insulating elastomer material305(e.g. an electrically insulating elastomer coating) applied on top of the insulated first conductive layer303of the first substrate301. Furthermore, there is a group (e.g., plurality or multitude) of elastic support nodules306arranged (e.g. in a grid array) and adhered on top of, or forming part of, the intrinsic structure of said layer of electrically insulating elastomer material305. In the example embodiments shown inFIG.6, the layer of electrically insulating elastomer material305may have a thickness of 20 micrometers. Furthermore, according to certain example embodiments, the elastic support nodules306each have a height of 80 micrometers. In addition, according to some example embodiments, the elastic support nodules306each have an aspect ratio of height to maximum width, and the aspect ratio may have a maximum value of two (2). The layer of electrically insulating elastomer material305, the group of elastic support nodules306, or both may be applied using a suitable microfabrication technique (e.g. a thin film deposition technique). The group of elastic support nodules306may be fully or partially made of a silicon-based organic polymer (e.g. PDMS), rubber (e.g. natural or synthetic), or any suitable combination thereof. Furthermore, the group of elastic support nodules306may be arranged less than 10 millimeters apart from each other in distance, and in some example embodiments, the inter-nodule distance is less than 2 millimeters. In certain example embodiments, the group of elastic support nodules306is arranged in a grid array, such as a spaced two-dimensional row-column grid array, as illustrated inFIG.2. In some alternative example embodiments, the group of elastic support nodules306is arranged in a spaced triangular grid array, as illustrated inFIG.3. Furthermore, the electrostatic actuation layer300may include one or more fluid reservoirs309(e.g. gas reservoirs, such as air reservoirs) that reduce the force involved for overall compression of the electrostatic actuation layer300when fluid displacement is restricted by contour sealing, when fluid displacement is restricted by compression speed (e.g. depending on the hydrodynamic properties of the fluid), when the fluid compresses, when the fluid moves from a compression space (e.g. a space gap), or any suitable combination thereof. The second substrate302may be microfabricated (e.g. etched) to provide the fluid reservoirs309(e.g. gas reservoirs) in the second substrate302. The fluid reservoirs309may be arranged to reduce the compression ratio of fluid volume (e.g. gas volume, such as air volume). The fluid reservoirs309may be microfabricated using a suitable microfabrication technique (e.g. an anisotropic wet etching technique). The microfabricated second substrate302with the second conductive layer304may be turned upside down and placed on top of the group of elastic support nodules306to form the electrostatic actuation layer300. In the example embodiments shown inFIG.6, the depth of the fluid reservoirs309may be 90 micrometers. Other suitable depths may be used, depending on the thickness of the second substrate302to be microfabricated. The one or more fluid reservoirs309are in fluid communication with (e.g., connected to) one or more compression spaces in the electrostatic actuation layer300. In various example embodiments, the horizontal contour of these fluid reservoirs309(e.g. wells) can be ellipsoidal, and their centers can be located equidistant from each nodule in each group of four neighboring elastic support nodules (e.g. as shown in inFIG.2orFIG.3). Alternatively, the fluid reservoirs309can have an arbitrary shape and can be located outside of the pillar region (e.g. somewhere in the contour of the elastic support nodules layer306). FIG.7is a cross-sectional diagram illustrating at least a portion of an electrostatic actuator (e.g. electrostatic actuator structure) with six electrostatic actuation layers, according to some example embodiments. The illustrated electrostatic actuator includes six electrostatic actuation layers111-116. In the example embodiments shown inFIG.7, each of the electrostatic actuation layers111-116includes a corresponding first substrate (e.g. first substrate101) with a corresponding first conductive layer (e.g. first conductive layer103), and the first conductive layer may be part of a first electrode130. Similarly, each of the electrostatic actuation layers111-116includes a corresponding second substrate (e.g. second substrate102) with a corresponding second conductive layer (e.g. second conductive layer104), and the second conductive layer may be part of a second electrode140. Moreover, the first conductive layer may be insulated (e.g. from its first substrate, from another conductive layer, or from both), the second conductive layer may be insulated (e.g. from its second substrate, from another conductive layer, or from both), or both. In the example embodiments shown inFIG.7, the electrostatic actuation layers111-116are stacked one above another (e.g. as a stack of electrostatic actuation layers), such that a bifunctional substrate (e.g. a bifunctional substrate film) forms both the first substrate of one electrostatic actuation layer (e.g. electrostatic actuation layer112) and the first substrate of an adjacent electrostatic actuation layer (e.g. electrostatic actuation layer111) among the electrostatic actuation layers111-116, one extending below the bifunctional substrate and the other extending above the bifunctional substrate. Alternatively, the bifunctional substrate may form both the second substrate of one electrostatic actuation layer (e.g. electrostatic actuation layer112) and the second substrate of an adjacent electrostatic actuation layer (e.g. electrostatic actuation layer111). Accordingly, such a bifunctional substrate can be considered as being or including (e.g. containing) the boundary between two adjacent electrostatic actuation layers (e.g. between the electrostatic actuation layers111and112), as well as being or including the junction of the two adjacent electrostatic actuation layers. The bifunctional substrate may include a mesh of elastomer material. According to various example embodiments, the electrostatic actuator that includes the illustrated electrostatic actuation layers111-116may further include a high voltage driver150(e.g. a high voltage driver with a flyback-mode boost converter). Furthermore, any one or more of the substrates (e.g. a bifunctional substrate configured or functioning as both the first substrate of one electrostatic actuation layer and the second substrate of an adjacent electrostatic actuation layer) in the stack of electrostatic actuation layers111-116may include an embedded connection element. The electrostatic actuation layers111-116are shown inFIG.7as including embedded connection elements121-127, any one or more of which may take the example form of an embedded metal wire or other conductive filament. The embedded connection elements121-127each connect one or more of the conductive layers (e.g. first conductive layer103or second conductive layer104, one or more of which may be or include a conductive coating) to the high voltage driver150. Within the stack of electrostatic actuation layers111-116, the interior (e.g. non-exterior) embedded connection elements122-126connect the conductive electrode layers of the interior bifunctional substrates together and to the high voltage driver150, as shown inFIG.7. Each of the electrostatic actuation layers111-116may further include a grid array that includes a group of elastic support nodules (e.g. elastic support nodules106), and the group of elastic support nodules may be arranged between their corresponding first and second substrates, such that there is arranged a compression space between the first and second conductive layers (e.g. first and second conductive layers103and104). In many example embodiments, the compression space is not entirely filled with solid material (e.g. the compression space is less than fully filled with solid material, such as solid elastic material). FIG.8is a cross-sectional diagram illustrating at least a portion of an electrostatic actuator (e.g. electrostatic actuator structure) with six electrostatic actuation layers, according to some example embodiments. The illustrated electrostatic actuator includes six electrostatic actuation layers131-136. In the example embodiments shown inFIG.8, each of the electrostatic actuation layers131-136includes a corresponding first substrate (e.g. first substrate101) with a corresponding first conductive layer (e.g. first conductive layer103), and the first conductive electrode layer may be part of a first electrode (e.g. first electrode130). Similarly, each of the electrostatic actuation layers131-136includes a corresponding second substrate (e.g. second substrate102) with a corresponding second conductive layer (e.g. second conductive layer104), and the second conductive layer may be part of a second electrode (e.g. second electrode140). Moreover, the first conductive layer may be insulated (e.g. from its first substrate, from another conductive layer, or from both), the second conductive layer may be insulated (e.g. from its second substrate, from another conductive layer, or from both), or both. In the example embodiments shown inFIG.8, the electrostatic actuation layers131-136are stacked one above another (e.g. as a stack of electrostatic actuation layers), such that a bifunctional substrate (e.g. a bifunctional substrate film) forms both the first substrate of one electrostatic actuation layer (e.g. electrostatic actuation layer132) and the second substrate of an adjacent electrostatic actuation layer (e.g. electrostatic actuation layer131) among the electrostatic actuation layers131-136, one extending below the bifunctional substrate and the other extending above the bifunctional substrate. Accordingly, such a bifunctional substrate can be considered as being or including (e.g. containing) the boundary between two adjacent electrostatic actuation layers (e.g. between the electrostatic actuation layers131and132), as well as being or including the junction of the two adjacent electrostatic actuation layers. The bifunctional substrate may include a mesh of elastomer material. According to various example embodiments, the electrostatic actuator that includes the illustrated electrostatic actuation layers131-136may further include a high voltage driver (e.g. high voltage driver150, which may have a flyback-mode boost converter). Furthermore, any one or more of the substrates (e.g. a bifunctional substrate configured or functioning as both the first substrate of one electrostatic actuation layer and the second substrate of an adjacent electrostatic actuation layer) in the stack of electrostatic actuation layers131-136may include an embedded connection element. The electrostatic actuation layers131-136are shown inFIG.8as including embedded connection elements141-147, any one or more of which may take the example form of an embedded wire or other conductive filament. The embedded connection elements141-147each connect one or more of the conductive layers (e.g. first conductive layer103or second conductive layer104, one or more of which may be or include a conductive coating) to the high voltage driver (e.g. high voltage driver150), as shown inFIG.8. Each of the electrostatic actuation layers131-136may further include a grid array that includes a group of elastic support nodules (e.g. elastic support nodules106), and the group of elastic support nodules may be arranged between their corresponding first and second substrates, such that there is arranged a compression space between the first and second conductive layers (e.g. first and second conductive layers103and104). In many example embodiments, the compression space is not entirely filled with solid material (e.g. the compression space is less than fully filled with solid material, such as solid elastic material). FIG.9is a cross-sectional diagram illustrating at least a portion of an electrostatic actuator (e.g. electrostatic actuator structure) with six electrostatic actuation layers, according to some example embodiments. The illustrated electrostatic actuator includes six electrostatic actuation layers151-156. In the example embodiments shown inFIG.9, each of the electrostatic actuation layers151-156includes a corresponding first substrate (e.g. first substrate101) with a corresponding first conductive layer (e.g. first conductive layer103), and the first conductive layer may be part of a first electrode (e.g. first electrode130). Similarly, each of the electrostatic actuation layers151-156includes a corresponding second substrate (e.g. second substrate102) with a corresponding second conductive layer (e.g. second conductive layer104), and the second conductive layer may be part of a second electrode (e.g. second electrode140). Moreover, the first conductive layer may be insulated (e.g. from its first substrate, from another conductive layer, or from both), the second conductive layer may be insulated (e.g. from its second substrate, from another conductive layer, or from both), or both. In the example embodiments shown inFIG.9, the electrostatic actuation layers151-156are stacked one above another (e.g. as a stack of electrostatic actuation layers), such that a bifunctional layer of electrically insulating elastomer material forms both the first substrate of one electrostatic actuation layer (e.g. electrostatic actuation layer152) and the second substrate of an adjacent electrostatic actuation layer (e.g. electrostatic actuation layer151) among the electrostatic actuation layers151-156, one extending below the bifunctional layer and the other extending above the bifunctional layer. Accordingly, such a bifunctional layer of electrically insulating elastomer material can be considered as being or including (e.g. containing) the boundary between two adjacent electrostatic actuation layers (e.g. between the electrostatic actuation layers151and152), as well as being or including the junction of the two adjacent electrostatic actuation layers. The bifunctional substrate may include a mesh of elastomer material. Furthermore, in certain example embodiments, a bifunctional conductive layer (e.g. a bifunctional conductive electrode layer) is applied to the bifunctional layer of electrically insulating elastomer material and thus arranged to act as both a first conductive layer (e.g. first conductive layer103) and a second conductive layer (e.g. second conductive layer104) for a pair of adjacent electrostatic actuation layers (e.g. electrostatic actuation layers151and152), one extending below the bifunctional conductive layer and the other extending above the bifunctional conductive layer. According to various example embodiments, the electrostatic actuator that includes the illustrated electrostatic actuation layers151-156may further include a high voltage driver (e.g. high voltage driver150, which may have a flyback-mode boost converter). Furthermore, any one or more of the bifunctional layers (e.g. a bifunctional layer of electrically insulating elastomer material or a bifunctional conductive layer) in the stack of electrostatic actuation layers151-156may include an embedded connection element. The electrostatic actuation layers151-156are shown inFIG.9as including embedded connection elements161-167, any one or more of which may take the example form of an embedded wire or other conductive filament. The embedded connection elements161-167each connect one or more of the conductive layers (e.g. one or more bifunctional conductive layers, any one or more of which may be or include a conductive coating) to the high voltage driver (e.g. high voltage driver150), as shown inFIG.9. Each of the electrostatic actuation layers151-156may further include a grid array that includes a group of elastic support nodules (e.g. elastic support nodules106), and the group of elastic support nodules may be arranged between their corresponding first and second substrates, such that there is arranged a compression space between the first and second conductive layers (e.g. first and second conductive layers103and104). In many example embodiments, the compression space is not entirely filled with solid material (e.g. the compression space is less than fully filled with solid material, such as solid elastic material). Any combination of one or more of the above-described electrostatic actuation layers (e.g. electrostatic actuation layer100,200, or300) may be included in an electrostatic actuator (e.g. an electrostatic actuator structure), and such included electrostatic actuation layers may be stacked one above the other to form at least two stacks of electrostatic actuation layers, as illustrated inFIGS.10A,10B,11A, and11B. In the example embodiments shown inFIGS.10A,10B,11A, and11B, any one or more of the first and second conductive surfaces of the first and second conductive layers on either or both sides of the space gap, the elastic support nodules, or both, may be inherently hydrophobic, hydrophobically or superhydrophobically coated, hydrophobically or superhydrophobically treated, or any suitable combination thereof. FIGS.10A,10B,11A, and11Bare cross-sectional diagrams each illustrating an electrostatic actuator1000(e.g. electrostatic actuator structure), which may have multiple (e.g. at least two stacks) of electrostatic actuation layers (e.g. at least two separate stacks that each include multiple instances of the electrostatic actuation layer100,200, or300). In alternative example embodiments, the electrostatic actuator1000has a single stack, and the single stack has a cavity (e.g. a hole) that, when viewed in cross-section, has the appearance shown inFIG.10A,10B,11A, or11B. Each stack includes at least one fluid reservoir (e.g. fluid reservoir309, which may be a gas reservoir, such as an air reservoir) arranged between the at least two stacks of electrostatic actuation layers. Moreover, there may be an elastic surface layer arranged on top of the at least two stacks of electrostatic actuation layers and on top of said at least one fluid reservoir (e.g. covering both the at least two stacks and the at least one fluid reservoir). InFIGS.11A and11B, the electrostatic actuator1000also has an additional firm (e.g. rigid) structure on top of the elastic surface layer. The firm structure may facilitate a homogeneous (e.g. flat) area of actuation, collect actuation power from multiple bulges, protect flexible layers from the environment, or any suitable combination thereof.FIGS.10A and11Adepict the electrostatic actuator1000in a state of rest.FIGS.10B and11Bdepict the electrostatic actuator1000when the stacked electrostatic actuation layers are being compressed, such that the elastic surface layer on top of at least one fluid reservoir bulges accordingly as a result of the electrostatic actuation layers being compressed. InFIGS.11A and11B, the electrostatic actuator1000may include an additional grid array of limiting nodules, which may be arranged between the elastic surface layer and the firm structure, for facilitating the entrance of air between both layers in the expansion process from the configuration shown inFIG.11Ato the configuration shown inFIG.11B. FIGS.12A,12B, and12Care cross-sectional diagrams each illustrating at least a portion of a single layer (e.g. a single electrostatic actuation layer) within the electrostatic actuator1000, according to some example embodiments. In some example embodiments, such a single layer forms the entirety of an electrostatic actuation structure. In alternative example embodiments, such a single layer is one layer (e.g. a topmost or exterior layer) among multiple electrostatic actuation layers included in an electrostatic actuation structure. As shown inFIGS.12A,12B, and12C, the illustrated single layer has a flexible upper substrate. FIG.12Adepicts the single layer (e.g. the single electrostatic actuation layer) in a state of rest.FIG.12Bdepicts the single layer when being partially compressed (e.g. due to a voltage difference across its first and second electrodes transgressing (e.g. exceeding) a first threshold voltage difference).FIG.12Cdepicts the single layer when being fully compressed (e.g. due to the voltage difference across its first and second electrodes transgressing a second threshold voltage difference, which may be higher than the first threshold voltage difference). InFIGS.12B and12C, as the flexible upper substrate of the single layer is being compressed, fluid flows from the corresponding compression space (e.g. space) and creates one or more bulges in the flexible upper substrate. These bulges may occur at the positions of the elastic support nodules (e.g. elastic support nodules106). As noted above, in certain example embodiments, one or more electrostatic actuation layers (e.g. electrostatic actuation layer100,200, or300) within the electrostatic actuator1000(e.g. electrostatic actuator structure) may be stacked one above the other in an aligned manner. For example, the elastic support nodules (e.g. elastic support nodules106) of one electrostatic actuation layer may be fully or partially aligned over the elastic support nodules of another (e.g. adjacent) electrostatic actuation layer. Similarly, if present, the wells (e.g. wells208) of one electrostatic actuation layer may be fully or partially aligned over the wells of another (e.g. adjacent) electrostatic actuation layer. However, in certain alternative example embodiments, one or more electrostatic actuation layers (e.g. electrostatic actuation layer100,200, or300) within the electrostatic actuator1000may be imbricatedly stacked one above the other without such alignment (e.g. such that each node resides at the center of mass of a system formed by its closest four neighbor nodes, in the adjacent layer). For example,FIG.7illustrates a situation in which the elastic support nodules (e.g. elastic support nodules106) of one electrostatic actuation layer are not aligned (e.g. are completely unaligned) with the elastic support nodules of another (e.g. adjacent) electrostatic actuation layer. Likewise, if present, the wells (e.g. wells208) of one electrostatic actuation layer are not aligned with the wells of another (e.g. adjacent) electrostatic actuation layer. Generally speaking, electrostatic actuators with imbricatedly stacked electrostatic actuation layers may be less vulnerable to overall bending under the aggregated (e.g. compounded) forces, deformations, or both in the stacked electrostatic actuation layers. This may provide the benefit of maximizing the amplitude of compression in situations where there is some layer bending in the inter-node space. This may also provide the benefit of an overall structure in which the stack of electrostatic actuation layers compresses approximately uniformly, despite localized bending occurring in one or more individual layers within the overall structure. “Bending” in this context refers to a process by which one or more of the electrostatic actuation layers become curved, such that interstitial regions between the elastic support nodules (e.g. elastic support nodules106or206) experience more compression than regions near or at the elastic support nodules. In certain example embodiments, the electrostatic actuator1000(e.g. electrostatic actuator structure) includes a grid of relatively rigid tile structures and relatively malleable areas between the tile structures. This may provide the benefit of allowing the bending of the actuator surface (e.g. uppermost or exterior substrate) at the relatively malleable areas, while retaining local rigidity at or near the relatively rigid tile structures. The relatively rigid tile structures may facilitate compression while providing resistance to bending between the elastic support nodules when the voltage difference is applied to the first and second electrodes of the electrostatic actuation layers. In various example embodiments, instead of elastic support nodules (e.g. elastic support nodules106), the electrostatic actuator1000(e.g. electrostatic actuator structure) may include rigid supports (e.g. non-elastic support nodules), elastic layer materials, foam-filled structures, continuous supports, continuous limiting structures, web structures, bulging supports (e.g. attached to a top layer) or any suitable combination thereof. According to some example embodiments, one or more elastic support nodules may be replaced by sealed (e.g. hermetically sealed or non-hermetically sealed), gas-filled (e.g. air-filled) cells that function as springs when compressed. According to certain example embodiments, one or more elastic support nodules may be replaced by solid semi-foam, again to function as a spring between the first and second electrodes. As used herein, “solid semi-foam” refers to a solid foam in which air pockets (e.g. air bubbles) are not completely sealed, but rather are polymerized, resulting in holes in the solid walls between the air pockets. According to various example embodiments, the electrostatic actuator1000includes a three-dimensional (3D) printed or moulded grid made from one or more suitable polymers that function as springs when compressed. According to some example embodiments, one or more elastic support nodules may be replaced by constrained magnets arranged in repulsion (e.g. with similar poles facing each other) to function as springs (e.g. with higher spring constant values). InFIGS.12A,12B, and12C, the electrostatic actuator1000may include an additional grid array of limiting nodules, which may be arranged between the elastic surface layer and the firm structure, for facilitating the entrance of air between both layers in the expansion process from the configuration shown inFIG.12C, through the configuration shown inFIG.12B, to the configuration shown inFIG.12A. According to some example embodiments, the grid array of elastic support nodules is in direct contact with both the top and bottom insulated conductive electrode layers, without any additional layer of electrically insulating elastomer material. In certain example embodiments, the grid array of elastic support nodules forms part of the intrinsic structure of a bottom grid of elastomer material, located between the nodules and the bottom insulated conductive electrode layer. According to various example embodiments, the grid array of elastic support nodules is adhered to both top and bottom layers of electrically insulating elastomer material, or forms part of the intrinsic structure of both top and bottom layers of electrically insulating elastomer material. As shown inFIG.13andFIG.14, the electrostatic actuator1000may include non-uniform electrodes (e.g. a first or second electrode that does is not coextensive with the entire multitude of elastic support nodules). InFIG.13, the elastic support nodules in a first zone1310are arranged between a first electrode (e.g. first conductive layer103) and a second electrode (e.g. second conductive layer104), and the first and second electrodes are present within the first zone1310. However, inFIG.13, the elastic support nodules in a second zone1320are not arranged between any electrodes, and the first and second electrodes do not extend into the second zone1320. Accordingly, while the multitude of elastic support nodules may be uniform and homogeneous (e.g. arranged in a grid array), the electrostatic actuator1000can produce one or more non-uniform pressure patterns that result in three-dimensional formations (e.g., one or more ridges) on the surface of the electrostatic actuator1000. This is facilitated by the elastic surface material of the electrostatic actuator1000, fluid material flowing from the first zone1310to the second zone1320, or both, within the compression space. For example, such fluid flow may be resultant from the electrostatic pressure being lower in the second zone1320where electrodes are absent, compared to the first zone1310where electrodes are present. Similarly, inFIG.14, the elastic support nodules in a first zone1410are arranged between a first electrode (e.g. first conductive layer103) and a second electrode (e.g. second conductive layer104), and the first and second electrodes are present within the first zone1410. However, inFIG.14, the elastic support nodules in a second zone1420are not arranged between any electrodes, and the first and second electrodes do not extend into the second zone1420. As noted above, while the multitude of elastic support nodules may be uniform and homogeneous (e.g. arranged in a triangular array), the electrostatic actuator1000can produce one or more non-uniform pressure patterns that result in three-dimensional formations (e.g. one or more ridges) on the surface of the electrostatic actuator1000. This is facilitated by the elastic surface material of the electrostatic actuator1000, fluid material flowing from the first zone1410to the second zone1420, or both, within the compression space. For example, such fluid flow may be resultant from the electrostatic pressure being lower in the second zone1420where electrodes are absent, compared to the first zone1410where electrodes are present. As a result, the non-uniform electrodes may cause the surface of the electrostatic actuator1000to produce a three-dimensional mechanical oscillation pattern (e.g. a pattern of one or more three-dimensional ridges, bulges, depressions, or any suitable combination thereof). For example, fluid flow from a first zone (e.g. first zone1310or1410) to a second zone (e.g. second zone1320or1420) may cause the second zone to bulge as the first zone compresses, and such bulging may contribute to the production of the three-dimensional mechanical oscillation pattern. In some example embodiments, a bulging second zone pushes on a rigid surface layer that is mechanically coupled to the elastic surface layer and causes the rigid surface layer to move away from the remainder of the electrostatic actuator1000. For example, if the second substrate102is mechanically coupled to the rigid surface layer within the second zone1320, the bulging of the second zone1320pushes the rigid surface layer away from the first substrate101, at least within the second zone1320. Similarly, if the second substrate202is mechanically coupled to the rigid surface layer within the second zone1420, the bulging of the second zone1420pushes the rigid surface layer away from the first substrate201, at least within the second zone1420. As is well-known, an actuator operates at high voltages, creating potential differences between the layers causing them to be compressed. In a stacked structure, when the layers are very close and there is insufficient isolation, these different potentials can cause sparks between the electrodes. Prior arts have attempted to solve this problem by forming an electrode that only partly covers the top surface of the substrate upon which it is disposed, creating one gap between the electrode and the substrate edges. This increases the distance between the electrode edges from adjacent layers on the structure, thus minimizing the possibility of sparks occurring between the said layers. The present invention provides a structure which can also solve the problem of sparks occurring between successive layers of an actuator or sensor, while preserving the robustness of the stacked structure and using a simple manufacturing process. Refer toFIG.15A, which is a top view of an electrode1500according to an exemplary embodiment of the present invention. As shown inFIG.15A, the electrode1500contains a series of indented, i.e. scratched, grooved or etched lines1502, which follow the shape of the electrode and connect to the protruding end.FIG.15Bshows the underside of the electrode1500shown inFIG.15A, wherein the elastic pillars1504are disposed on the underside.FIG.15Cis a cross-sectional diagram of a flat active element (FAE)1500having the structure shown inFIG.15AandFIG.15B. As shown inFIG.15C, the etched lines1502fully cut through the conductive layer1501but do not penetrate the substrate1505. In some embodiments, the etched lines may penetrate the substrate but will not fully cut through it, as the aim is to provide isolation. This allows isolation between stacked layers. In the example shown inFIGS.15A,15B and15C, the electrode1500comprises two parallel etched lines. This is merely for illustrative purposes, however. In one embodiment, the electrode may have a single etched line, or may have multiple etched lines. Two etched lines is a preferred embodiment, as the number of etched lines may be limited by the size of the electrode. More than one etched line further ensures that isolation can be achieved even if one of the etched lines has a defect such that complete electrical isolation between both sides of the gap is not achieved. The distance between the etched lines will depend on the intended voltage used in the system. In some embodiments, these etched lines (gaps) may be filled by the elastic layer. This typically occurs by first etching the gaps and then creating the elastic layer on top of the electrode layer. This may also be achieved by a wet coating process. This provides additional insulation. In other embodiments, there may also be an additional insulator layer. By etching lines into the electrodes rather than forming electrodes with different dimensions than their respective substrates, all layers of the stacked actuator or sensor can have the same horizontal dimensions from the top view of the stacked actuator or sensor as shown inFIG.15A, making the entire structure more robust. Etching can be performed by any profiling tool, such as a CNC machine, a blade or a laser. No chemicals are required. In order to further increase stability between the stacked layers, the present invention provides various embodiments for joining the FAE layers together to form an actuator or sensor. Refer toFIG.16A, which is a diagram of a stacked structure1600according to a first exemplary embodiment. InFIG.16A, every alternate layer is coupled at the electrical contact end1607to a conductive paste or adhesive, wherein a flat conductive element such as a wire1606is coupled to the top layer and the top of the electrical contact end1607. Forming the stack using this method would require applying the conductive paste/adhesive to each layer before they are applied to the stack. As shown in the diagram, no deformation of any layer occurs as a result of this binding method. A large amount of conductive adhesive is required, however. A second embodiment of a stacked structure1630is therefore illustrated inFIG.16B. As shown in the diagram, each alternate layer has a connecting ‘arm’ of a different respective length. These ‘arms’ are bent towards each other and electrically coupled together with conductive paste or adhesive at an electrical contact end1617. The different lengths allow less conductive paste/adhesive to be used as compared to the embodiment shown inFIG.16A. As the conductive paste/adhesive can be applied to the underside of the connecting ‘arms’, as shown in the diagram, the conductive paste/adhesive can be applied after the stacked structure is formed. This method requires that each coupled FAE layer stem or arm be manufactured with a different length. A third embodiment of a stacked structure1650illustrated inFIG.16Cshows coupled FAE layers all of a same length, wherein the ‘arms’ of each coupled FAE layer are coupled together and each coupled layer also comprises an individual flat conductive element1670, all of which are coupled together to form the electrical connection between layers. Each individual conductive element is electrically coupled to a respective layer using a small amount of conductive paste or adhesive at an electrical contact end1627, as illustrated in the diagram. Compared to the embodiment shown inFIG.16A, only a small amount of conductive paste/adhesive is required, and this conductive paste/adhesive and the coupling of the flat conductive element1670to each individual layer can be applied before forming the stack. Complexity of production of the individual layers increases slightly, however, in comparison to the structure shown inFIG.16A. A further embodiment is proposed which simplifies the above stacking schemes while providing electrical connection between the layers. This embodiment uses a solid binding edge which has some similarity to how paperback books are bound and therefore can also be called a book binding edge, wherein an edge is formed on one or two sides of the electrode of an FAE layer and used for bonding with another FAE layer. Refer toFIG.17A, which is a diagram of a stacked structure1700with a book binding edge1720according to a first embodiment. Similar to the elastic nodule structure between the layers, the book binding edge1720also has a nodule structure. By applying adhesive to the nodules and pressing the layers together, the adhesive can penetrate and form connection between the layers. As shown in the diagram, the book binding edge1720can be formed at both ends for increased stability. In addition, one or both of the book binding edges may further comprise a flipped film1730which is formed on the outside of the book binding edge for added stability. FIG.17Bis a diagram of a stacked structure1750with a book binding edge1760/1761according to a second embodiment. In this embodiment, electrical connection between the layers is achieved via the book binding edge. In this structure, the electrodes are alternately divided, such that electrodes of one polarity are coupled together at one side of the stacked structure and electrodes of the other polarity are coupled together at the other side of the stacked structure. For example, the first, third and fifth layers are coupled together at the left edge1760and the second, fourth and sixth layers are coupled together at the right edge1761. Further, the electrode layers at each respective side have an exposed edge. The adhesive used to form the book binding edges at each side of the stacked structure is a conductive adhesive, such that it forms an electrical connection with the exposed electrode layers. This can further provide a single point of contact for an electrical contact such as a wire. As shown in the diagram, the electrodes at each side have a diagonal structure which exposes more of the edge, enabling a better electrical connection. In all stacked structures, there can be a problem with too much lateral displacement occurring between the layers. In practice, it is difficult to achieve a structure that is robust but does not prevent the intended vertical movement between layers, particularly when the number of layers becomes high. The invention therefore provides a number of embodiments which provide bonding methods for stacked structures that can prevent or limit lateral displacement between the FAE layers. Refer toFIG.18A, which is a diagram of a stacked structure1800comprising various FAE layers similar to that illustrated inFIG.15A,FIG.15BandFIG.15C. As shown inFIG.18A, alternate FAE layers correspond to different polarities, such that the stacked structure comprises two sets of ‘arms’ extending from the main stack. By fixing the arms using an adhesive method such as glue, adhesive tape or cement, or an external holding structure, the lateral displacement between the layers can be prevented. This is shown inFIG.18B, wherein a strip of adhesive tape1810covers the two sets of arms of the stacked structure. Rather than joining the stacked structure at the arms, the stacked structure can be joined at the main body using flexible strips which will deform (bend) when the actuator is in a compressed state. These flexible strips can be of various shapes as shown in the embodiments illustrated inFIG.18C,FIG.18DandFIG.18E, illustrating flexible strips1820,1830and1840, respectively. In a modification (not shown), both the arms and the main body of the stacked structure can use flexible strips to prevent lateral displacement. In the embodiments shown inFIG.18C,FIG.18DandFIG.18E, separation between layers in the vertical direction can also be prevented. In yet another modification of the above structure, the entire stacked structure can be enclosed in a solid casing1870comprising walls that may be fully rigid or somewhat flexible, as shown inFIG.18F. As illustrated in the diagram, the solid casing further (optionally) includes a lid1850which has a hollow centre that can allow access to the top of the stacked structure. The lid can optionally be attached to the solid casing to further prevent vertical displacement of the layers. In addition, a mounting structure1860can be applied to the stacked structure to prevent loss of functionality if the stacked structure at full compression cannot protrude through the hollow centre of the lid1850. Please note that this solid casing can be implemented separately or in combination with the embodiments shown inFIG.18B,FIG.18C,FIG.18DandFIG.18E. In a final embodiment, which can also be implemented in combination with the embodiments shown inFIG.18B,FIG.18C,FIG.18DandFIG.18E, the entire stacked structure can be disposed in a flexible hermetically sealed or semi-sealed pouch1900as illustrated inFIG.19AandFIG.19B. The flexibility of the pouch allows the compression of the actuator/sensor, while the hermetic sealing prevents accumulation of dust within the stacked structure, and further controls humidity. Semi-sealed refers to a structure that is essentially sealed but wherein the internal compartment may in some situations (such as notable pressure differences) slowly move towards ambient pressure. The pouch may protect the stack from external contamination or humidity accumulation. The shape of the pouch can be designed so that it is in close contact with the external surface of the stack when in a resting state, and that it deforms minimally to accommodate the air displaced when the stack is compressed or when variations in temperature or pressure induce expansion of the inner air. Importantly, the semi-sealed structure does not notably hinder the compression or expansion of the stack. As the driving signal is known, and the change in capacitance and the current drawn by the actuator can be monitored, it is possible to estimate the overall compression of the actuator. Furthermore, capacitance of the system can be monitored without a driving signal that would notably compress the system. In this case, changes in the system capacitance indicate compression of the system by external forces. Capacitance changes can also be monitored via various other signal proxies such as voltage changes, current draw changes or changes in oscillator frequencies in the system. These techniques are examples of methods enabling the sensor application of the stacked structures. The system may also be divided into subsystems, with each subsystem having its own monitoring circuitry. This allows for more localized monitoring of the physical changes in the system. The configuration of the electrode layers makes the capacitance of the stack very sensitive to small stack compressions. Therefore, functionality for compression sensing can be achieved by the constant monitoring of the capacitance of the stack. FIG.19Apresents a profile view of a stack contained in a flexible hermetically sealed pouch1900.FIG.19Billustrates the deformation of the pouch1900under compression whereby a same volume leads to a different pouch external shape to accommodate approximately the same fluid/air volume. The pouch1900can also accommodate changes in external air pressure due to its structure, providing minimal hindrance to the compression decompression cycles of the actuator or sensor. FIGS.19C and19Drepresent two examples for the use of flexible hermetically sealed pouches in arrays of stacked actuators.FIG.19Cillustrates an array1950, wherein parts of the pouch reach the base supporting surface between the different stacked actuators. In an embodiment, they may be adhered to the base supporting surface in order to provide increased robustness and to prevent possible local damage in some part of the pouch that could affect the whole array of actuators. In this form, the deformation of the pouch upon compression occurs laterally.FIG.19Dillustrates an example of a pouched array1970, which can be easier to manufacture and where the deformation of the pouch occurs vertically in the regions between the stacks. The invention provides a number of stacked structures which have good isolation between the layers. Holding structures such as binding edges or strip layers can increase the robustness by preventing lateral displacement of the layers in the stacked structure with respect to each other wherein the vertical movement of their active regions is not hindered. The pouch structure may also be implemented as a one-sided elastic film, or pre-shaped flexible film which facilitates the form of the stack, and which adheres to a substrate. For example, a pouch structure can be formed with an elastic substrate with a pressure sensitive adhesive that is larger in area than the sensor/actuator, and which is attached to a substrate or host system encircling the sensor/actuator. This method enables simultaneous placement of a sensor or actuator on a suitable medium such as to a printed circuit board, while also providing the sealing. Further, the electrical connection of the sensor/actuator may be accomplished using a conductive adhesive or anisotropic conductive adhesive or film at suitable contact points of the printed circuit board or substrate (1915inFIG.19E,FIG.19F,FIG.19GandFIG.19H), which simplifies the application or assembly process. For example, in this method the sensor is secured in its place and may also be electrically connected to the system at hand simultaneously. Examples for the implementations above are illustrated inFIG.19EandFIG.19F. InFIG.19E, the elastic pouch1902is attached to the surface of the hosting board or device (not shown) by placing its contour border on top of a closed band of a pressure sensitive adhesive1911that is adhered to the surface of the hosting board or device. In some cases, such as when using pouch materials with a low surface energy (for example, cured silicone), the adhesion between the pouch and the closed band of pressure sensitive adhesive may be insufficient for providing enough attachment force to sensors or actuators with the pouch structure1902as illustrated inFIG.19E. This risk may be minimized by attaching the sensor using an alternative structure1904, where the closed band of pressure sensitive adhesive1913is placed on top of the contour border of the pouch and extends slightly beyond this border, as illustrated inFIG.19F. A drawback of this alternative structure is that it occupies slightly more area at the surface of the hosting board or device. These figures show a simple adhesion method while the actual electrical connections may be accomplished via, for example, multiple contact points and pads from a sensor or actuator to the host system or printed circuit board. The pouch or sealing structure may also be formed in various other ways to achieve the objective of facilitating simple adherence to a desired surface. For example, the pouch may have edges tucked along sensor lines, and encircles the stacked structure from all sides except from the “bottom” adherence area1915, as illustrated inFIG.19GandFIG.19H. In this configuration, the area required by the sensor and the adherence area are essentially the same. InFIG.19G, the elastic pouch1906is attached to the surface of the hosting board or device (not shown) by placing the back (outer) side of its contour border on top of a closed band of pressure sensitive adhesive1911that is adhered to the surface of the hosting board or device. As stated above, in some cases, such as when using pouch materials with low surface energy (for example, cured silicone), the adhesion between the pouch and the closed band of pressure sensitive adhesive may be insufficient for providing enough attachment force to sensors or actuators with the pouch structure as illustrated inFIG.19G. This risk may be minimized by attaching the sensor using an alternative structure, where the closed band of pressure sensitive adhesive1913is placed on top of the inner side contour border of the pouch1908and makes physical contact at the same time with the surface of the hosting board or device (not shown), as illustrated inFIG.19H. One factor adding complexity and cost to manufacturing of the described sensor or actuators is that the sensor/actuator structure always comprises multiple stacked layers. All functional layers have some electrical connection and generally terminate in an electrical connector where all contact pads are in a single plane. An example of such a connector for sensors is a “zero insertion force” type connector. Normally, this requires electrically conducting through holes or other electrically connecting structures, which can be expensive and complicated to manufacture. An inventive method makes the sensor in such a manner that a single plain/sheet electrode pattern can be printed or etched. The single plain etching/coating can be cost effectively performed, for example by a roll-to-roll procedure. Further, elastic nodules can be imprinted on the same single sheet, for example from PDMS again using a roll-to-roll (web processing) manufacturing method. A film pattern as shown in the drawings is used, providing a multilayer structure made by appropriate cuts and folds that do not prevent the compressive expansion cycles. In some embodiments, at least a total of four electrode layers are needed due to at least two electrode layers and two shield layers being required, and they can be used for actuation or sensing or both. At least some part of the structure is folded multiple times to achieve a stacked structure, wherein all the layers are electrically connected via the folded part but wherein the layer structure can be dismantled to a single layer print pattern comprising the electrodes and elastic nodules. This structure may also contain internal air reservoirs. FIGS.20A and20Billustrate a flat multilayer structure formed using a roll-to-roll method. As is well-known in the art, roll-to-roll manufacturing takes a flexible rolled substrate and prints or etches various layers thereon by feeding the substrate from one roller to another along the web path. In this way, a number of initial flat multilayer structures can be formed on the substrate as shown inFIG.20A. Further layers, such as an insulation layer, can be formed over these structures as shown inFIG.20B. Then, these individual structures can be sequentially flipped (or folded or tucked), bent and compressed to form individual sensors or actuators. Holes or linear cuts or incisions, cut in the individual structures facilitate the bending of the sensor structure, and lines can be formed on the structure indicating areas which can be cut away to form the required shape. Further, adhesive layers are added to improve the robustness of the final sensor or actuator structure. The resultant structure is a capacitive pressure sensor, actuator or both, comprising a sensing or actuating area formed of two conductive layers on either side of an elastic layer, wherein the elastic layer is formed of elastic nodules as detailed previously. When pressure is mechanically applied to the sensing area when used as a sensor, or electrically induced into the actuating area when used as an actuator, air therein will be displaced. The present invention provides various embodiments allowing for air displacement in different ways, either by providing air compartments within the sensor or actuator, or by including open sides which displace the air to the outside. FIGS.21A-21Cillustrate the stages of a first embodiment of a capacitive pressure sensor formed using the above-described roll-to-roll method. The cuts and folds made in the multilayer structure are for enabling folding of the multilayer structure and do not prevent compressive expansion cycles.FIG.21Aillustrates a substrate2018having conductive areas2025formed thereon, wherein the patterns are formed corresponding to the desired sensor shape. The conductive area on the left-hand side of the substrate2018has elastic nodules2005formed thereon. FIG.21Billustrates an individual structure2100formed from the substrate shown inFIG.21Awith adhesive layers2028formed over various parts of the structure, and holes cut into the substrate2018to enable bending and to further distinguish a 1stbending from an initial layer, a 2ndbending and a 3rdbending of the structure2100. As illustrated in the diagram, the adhesive layer2028will cover some of the conductive lines and conductive areas. In addition, sections of the substrate2018have been cut away. As can be seen inFIG.21B, certain sections of the substrate are not covered by the adhesive layers. A section at the bottom of the structure2100is left exposed to form a connecting area. Further, on the initial layer and the first bending, a section below the conductive areas is left exposed. This area will form an air compartment once the entire structure has been flipped and bent. FIG.21Cillustrates the final flipped and bent sensor2100, wherein sections at either side of the conductive lines have been removed to form the desired sensor shape. As shown in the diagram, a sensing area is formed in the elastic layer2005, and an air reservoir compartment is formed below the sensing area. When the sensing area is compressed, displaced air can flow into the air reservoir compartment. A second embodiment of a sensor2200is illustrated inFIG.22AandFIG.22B, and includes air reservoirs in both the top and the bottom of the sensor. The sensor2200initially comprises the same conductive patterns2025as shown inFIG.21A. Adhesive layers2028are then formed over some of the conductive patterns2025as shown inFIG.22A, but leaving a connecting area at the bottom of the structure2200exposed. Additionally, an area of the substrate2018in the 2ndbending is exposed to form a top air reservoir, and an area of the substrate2018in the 3rdbending is exposed to form a bottom air reservoir. Finally, the initial layer includes two air holes2058which connect the top and bottom air reservoirs. FIG.22Billustrates the final flipped, bent and compressed sensor2200. As in the previous embodiment, excess sections at the side of the sensor2200are removed. When the sensing area is compressed, air will move from the sensing area into both the top and bottom air reservoir compartments via the holes2058. FIG.22Cillustrates a side on view of each bending stage of the sensor2200shown inFIG.22B. The initial layer comprises holes which are cut through the substrate2018and the conductive layer patterns2025. A top air reservoir is formed in the adhesive layer. The 1stbending folds the elastic layer onto the initial layer, such that the elastic nodules penetrate the gap in the adhesive layer. The 2ndbending then folds onto the 1stbending so that the adhesive layer on the 2ndbending contacts the adhesive layer on the initial layer, and the top air reservoir is formed over the conductive lines on the initial layer. Finally, the 3rdbending is folded onto the back of the initial layer, such that the holes allow air flow from the air reservoir in the initial layer, the air reservoir in the 2ndbending and the air reservoir in the 3rdbending. A stiffener can optionally be formed on the back of the 3rdbending, to facilitate electrical connection to a specific connection port. In a final embodiment, rather than including air reservoirs within the sensor, the sides of the sensor are left open so that displaced air can directly flow out. As shown inFIG.23A, the conducting area patterns2025formed on the substrate2018are slightly different from those in the previous two embodiments. As shown inFIG.23B, there are open sections rather than cut holes formed between the 1stbending, initial layer, 2ndbending and 3rdbending. Due to these open sides, the adhesive layer2028can cover more of the substrate2018, as the structure2300does not require air reservoirs. A stiffener2085is formed on the back of the 3rdbending, to facilitate electrical connection to specific connection ports. The final flipped, bent and compressed sensor2300, including the sensing area and connecting area, is shown inFIG.23C. Those skilled in the art will readily observe that numerous modifications and alterations of the device and method may be made while retaining the teachings of the invention. Accordingly, the above disclosure should be construed as limited only by the metes and bounds of the appended claims. | 86,134 |
11863088 | Reference signs in attached figures:100, vortex-induced vibration-based piezoelectricity and friction nanometer power generation combined energy collector;1, support frame;101, fixed plate;102, cantilever plate;103, connecting plate;2, piezoelectric plate;3, cylinder;301, semi-cylinder;302, first connecting bolt;4, solid-liquid type friction nanometer power generation assembly;401, induction electrode;402, insulating friction inner shell;403, sealing part;404, first through hole;405, second through hole;5, semi-cavity;6, right-angled groove;7, vertical groove;8, cross-shaped channel;9, first connecting through hole;10, first threaded hole;11, second connecting through hole;12, butt joint face;13, plane provided with the piezoelectric plate; and14, half of the induction electrode DETAILED DESCRIPTION OF THE EMBODIMENTS The following clearly and completely describes the technical scheme in the embodiments of the present disclosure with reference to the attached figures in the embodiments of the present disclosure. Apparently, the described embodiments are merely a part rather than all of the embodiments of the present disclosure. All other embodiments obtained by those skilled in the art based on the embodiments of the present disclosure without creative efforts shall fall within the protection scope of the present disclosure. The present disclosure aims to provide a vortex-induced vibration-based piezoelectricity and friction nanometer power generation combined energy collector. Energy collection of low-flow-speed water flow is achieved, the advantage of low-frequency energy collection of friction nanometer power generation is fully utilized, and the utilization efficiency of energy collection is improved. To make the foregoing objective, features and advantages of the present disclosure clearer and more comprehensible, the present disclosure is further described in detail below with reference to the attached figures and specific embodiments. As shown inFIG.1toFIG.4, the embodiment provides a vortex-induced vibration-based piezoelectricity and friction nanometer power generation combined energy collector100. The energy collector comprises a support frame1, a piezoelectric plate2, a cylinder3and a solid-liquid type friction nanometer power generation assembly4, wherein the support frame1comprises a fixed plate101, a cantilever plate102and a connecting plate103which are sequentially connected from top to bottom, the piezoelectric plate2is fixed on one side of the cantilever plate102, the cylinder3is mounted at the lower part of the connecting plate103, a mounting cavity is formed in the cylinder3, the solid-liquid type friction nanometer power generation assembly4comprises an outer shell, an insulating friction inner shell402and a sealing part403, the outer shell is fixed in the mounting cavity, the outer shell comprises two symmetrically arranged induction electrodes401, insulating layers are arranged between the butt joint faces12of the two induction electrodes401, the included angles between the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are not 90°. Specifically, the cantilever plate102extends along the vertical direction, and the two induction electrodes401are symmetrically arranged relative to the vertical plane. The wire of each induction electrode401can penetrate through the cylinder to stretch to the outside, the wire is used for connecting an externally connected load, and the piezoelectric plate2is also used for connecting the externally connected load. The insulating friction inner shell402is clamped between the two induction electrodes401. Clamping force is applied to the insulating friction inner shell402by the two induction electrodes401after mounting, namely, the insulating friction inner shell402and the two induction electrodes401cannot move relatively. A first through hole404is formed in the insulating friction inner shell402. The insulating friction inner shell402is filled with water. Water in the insulating friction inner shell402is not in a full state. The sealing part403is used for sealing the first through hole404, and then water in the insulating friction inner shell402is prevented from flowing out. Specifically, the sealing part403is in interference fit with the first through hole404. Specifically, the cylinder3comprises a first connecting assembly and two symmetrically arranged semi-cylinders301, the upper ends of the two semi-cylinders301are both mounted on the connecting plate103, the first connecting assembly is used for connecting the two semi-cylinders301, a semi-cavity5is formed in the inner side of each semi-cylinder301, and the two semi-cavities5are in butt joint to form the mounting cavity. Specifically, the first connecting assembly comprises a plurality of first connecting bolts302, a plurality of first connecting through holes9are formed in one semi-cylinder301, a plurality of first threaded holes10corresponding to the first connecting through holes9are formed in the inner side surface of the other semi-cylinder301, each first connecting bolt302penetrates through one first connecting through hole9to be mounted in one first threaded hole10, and then the connection of the two semi-cylinder301is achieved. In the embodiment, the energy collector further comprises a plurality of second connecting bolts, wherein a plurality of second connecting through holes11are formed in the connecting plate103, a plurality of second threaded holes are formed in the top surface of each semi-cylinder301, each second connecting bolt penetrates through one second connecting through hole11to be mounted in one second threaded hole, and then the connection of the connecting plate103and the cylinder3is achieved. In the specific embodiment, the semi-cavity5is a semi-spherical cavity, and the two semi-cavities5are in butt joint to form a spherical cavity. The induction electrode401is in the shape of a semi-spherical shell, and the insulating friction inner shell402is in the shape of a spherical shell. The inner diameter of the mounting cavity is the same as the outer diameter of the outer shell, and the inner diameter of the outer shell is the same as the outer diameter of the insulating friction inner shell402. Specifically, the lower surface of the connecting plate103is provided with a cross-shaped channel8, a vertical groove7is formed in the upper part of the semi-cavity5, the bottom of the vertical groove7communicates with the semi-cylinder5, two right-angled grooves6are symmetrically arranged on the two sides of the semi-cylinder5, the two ends of each right-angled groove6respectively communicate with the semi-cylinder5and the vertical groove7, the two vertical grooves7form a vertical channel and the four right-angled grooves6form two right-angled channels after the two semi-cylinders301are in butt joint, the top of the vertical channel communicates with the middle part of the cross-shaped channel8, and the two wires respectively pass through the two right-angled channels and stretch to the outside through the vertical channel and the cross-shaped channel8. The cross-shaped channel8comprises four horizontal channels, and the two wires stretch to the outside from the different horizontal channels. In the specific embodiment, the insulating friction inner shell402is made of an FEP material, the wall thickness of the insulating friction inner shell402is 50 microns, and the induction electrode401is made of copper. In the specific embodiment, the butt joint face12of each induction electrode401is provided with the insulating layer, namely, the annular butt joint face12of the induction electrode401in the shape of a semi-spherical shell is provided with the insulating layer. In the specific embodiment, the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are parallel to each other or coplanarly arranged. At the moment, the potential difference generated under the shaking of the cylinder3from left to right, and is in an optimal condition. As shown inFIG.3andFIG.4, the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are coplanarly arranged. The liquid surface is horizontal in a resting state. Due to the fact that the polarity of water is different from that of the FEP material, the surface of the insulating friction inner shell402is carried with negative charges and the water is carried with positive charges. The static electricity is balanced and no electric potential is generated at resting positions. When shaking is generated, the cylinder3and the cantilever plate102vibrate left and right, water deforms. When the liquid surface surges to the left side, the contact area between the water and the left side is increased, and the contact area between the water and the right side is decreased. Due to the fact that the water is carried with positive charges, when the contact area between the left side and the right side is increased, the positive charges on the left side are increased accordingly, and the positive charges on the right side are decreased accordingly, and potential difference is generated. When the liquid surface surges to the right side in the same way as above, current from right to left is generated, and the generated current flows into the externally connected load through the induction electrodes401and the wires. According to the embodiment, the condition that the included angles between the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are not 90° is met, so that when the cylinder3swings left and right, potential difference is generated between one half of the insulating friction inner shell402corresponding to the two induction electrodes401and the other half of the insulating friction inner shell402corresponding to the two induction electrodes401. Specifically, the induction electrodes401are bonded in the semi-cavities5. It needs to be noted that the one-to-one correspondence relation between the induction electrodes401and the semi-cavities5is not necessarily achieved. In order to meet the condition that the included angles between the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are not 90°, when the induction electrodes401are mounted, it is possible that the two sides of one induction electrode401are bonded in the two semi-cavities5. In the specific embodiment, the butt joint face12of the semi-cylinder301and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are vertical to each other. If the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are parallel to each other or coplanarly arranged, half of each of the induction electrodes401is bonded in each of the two semi-cavities5. Each of the two induction electrodes401includes two halves14bounded by a diameter D, which extends to the semi-circular hole, of the two halves14., one of the two halves14of one of the two induction electrodes401is arranged in the semi-cavity5of a corresponding one of the two semi-cylinders301, and one of the two halves14of another one of the two induction electrodes401is arranged in the semi-cavity of another one of the two semi-cylinders301, such that the butt joint faces12of the two induction electrodes401and the plane13, provided with the piezoelectric plate2, of the cantilever plate102are parallel to each other or coplanarly arranged. Specifically, the first through hole404is formed in the top of the insulating friction inner shell402. A semi-circular hole is formed in the upper part of the inner side of each induction electrode401. The two semi-circular holes are in butt joint to form a second through hole405. The sealing part403can penetrate through the second through hole405to stretch to the outside of the outer shell, and is arranged in the vertical channel. In the specific embodiment, the sealing part403is in the shape of a cylinder, and the first through hole404and the second through hole405are both circular holes. In the specific embodiment, the two ends of the cantilever plate102are respectively vertically mounted on the fixed plate101and the connecting plate103. In the embodiment, the fixed plate101is a rectangular plate, and the connecting plate103is a circular plate. When the energy collector is used, the energy connector is vertically fixed in water flow through the fixed plate101. When water flows through the cylinder3, the flow around a circular cylinder occurs, and a vortex street periodically sheds off. Periodical lifting force can be generated on the two sides of the cylinder3due to the influence of vortex shedding. The cylinder3vibrates through the lifting force. The vibration of the cylinder3can drive the liquid inside the solid-liquid type friction nanometer power generation assembly4to shake, so that friction is generated between the liquid and the insulating friction inner shell402, and then current is generated, and the purpose of energy collection is achieved. In the embodiment, the vibration generates based on the vortex-induced vibration caused by the flow around a circular cylinder, and the vortex-induced vibration can be generated at a relative low flow speed, so that the energy collection of the water flow at a low flow speed is achieved, power can be supplied for equipment with low energy consumption, such as sensors on wireless sensor networks, micro electro mechanical systems and unmanned aerial vehicles. Compared with electromagnetic power generator technology, the friction nanometer power generation method has the advantages in the aspect of flow energy collection, especially low-frequency flow energy collection. The friction nanometer power generation method has the characteristics of high energy conversion efficiency, ultrahigh output voltage, rich material sources, simple manufacturing technology, high cost performance, high biocompatibility and the like. Further, compared with other modes of friction nanometer power generation methods, the solid-liquid type friction nanometer power generation method has extremely low friction coefficients between solid and liquid, and has relatively large solid-liquid effective contact area, so that the method is high in efficiency, and each time of vibration can be effectively utilized. Meanwhile, the cantilever plate102deforms through the vibration of the cylinder3, and current is generated in the piezoelectric plate2through the deformation. Therefore, in the embodiment, piezoelectric energy collection is combined with friction nanometer energy collection, energy of low-flow-speed water flow is preferably utilized, the advantage of low-frequency energy collection of friction nanometer power generation is fully utilized, the regenerable low-speed flow energy stored in oceans, rivers and channels can be fully utilized, the utilization efficiency of water flow energy collection is improved, and the practicability is enhanced. Specific examples are used for illustration of the principles and implementation methods of the present disclosure. The description of the above-mentioned embodiments is used to help illustrate the method and the core principles of the present disclosure; and meanwhile, those skilled in the art can make various modifications in terms of specific embodiments and scope of application in accordance with the teachings of the present disclosure. In conclusion, the content of this specification shall not be construed as a limitation to the present disclosure. | 15,915 |
11863089 | DETAILED DESCRIPTION Before describing several exemplary embodiments of the disclosure, it is to be understood that the disclosure is not limited to the details of construction or process steps set forth in the following description. The disclosure is capable of other embodiments and of being practiced or being carried out in various ways. As used in this specification and the appended claims, the term “substrate” refers to a surface, or portion of a surface, upon which a process acts. It will also be understood by those skilled in the art that reference to a substrate can also refer to only a portion of the substrate unless the context clearly indicates otherwise. Additionally, reference to depositing on a substrate can mean both a bare substrate and a substrate with one or more films or features deposited or formed thereon One or more embodiments of the disclosure provide apparatus and/or methods for real-time monitoring of the high voltage output from a high voltage power supply. Some embodiments advantageously enable early detection of voltage drop. For example, in some embodiments, a voltage drop from the high voltage power supply indicates loss of chucking or chucking ability of an electrostatic chuck. Some embodiments advantageously allow for real-time immediate adjustment of the high voltage output. For example, if a voltage drop is in progress, the disclosure allows for adjustment of the output to prevent de-chucking of a substrate. Some embodiments of the disclosure provide high voltage measurement techniques using voltage transducers to measure high voltage output with high precision accuracy which can be incorporated to process chambers and Test Setups very easily. This enables live monitoring of high voltage output generated from Pedestal ESC inside the process chamber during chuck operation Some embodiments allow for the real-time measurement of pedestal ESC high voltage raw output to measure failures or drops in voltage. Early identification of ESC power supply output failures can be measured, allowing for early intervention, and reducing wafer scraps, improving overall yield, reducing chamber downtime and/or allowing for ESC power supply replacement before major failure. In some embodiments, a high voltage transducer with +15V & −15V DC Power supply is connected to the high voltage power supply. In an exemplary procedure, a pedestal ESC high voltage power supply output is connected to HV transducer input terminals (HV+, HV−). Apply +15V & −15V DC supply to power terminals of HV transducer. In the exemplary process, the ESC power supply produces high voltage up to +/−2 KV DC based on set voltage to chuck the wafers during the process inside the process chamber. The HV transducer of some embodiments continuously converts the output (2000V:10V ratio) voltage with respect to ESC power supply output from 0-2000V DC. The HV transducer output can be directly monitored online with multimeter or data logger continuously without any safety concerns since it provides maximum output of 10V for equivalent power supply maximum output of 2000V. Some embodiments advantageously provide a method in which any output deviation or output degradation from the pedestal ESC power supply output can be captured online and can be fixed immediately which will avoid the wafer damage as well as tool down time. FIG.1illustrates a system100comprising a high voltage measurement system110according to one or more embodiment of the disclosure. The skilled artisan will recognize that measurement system can be part of a process chamber hardware, part of the high voltage power supply, or a separate component, as illustrated. The system100illustrated comprises a high voltage power supply120, at least one load140(e.g., an electrostatic chuck, internal load), and the high voltage measurement system110. In some embodiments, the high voltage measurement system110is connected directly to the high voltage power supply120, in parallel electrical connection to the load. In some embodiments, the high voltage measurement system110is connected to the high voltage power supply through the load140. The load140can be one component or a combination of components that draw power from the high voltage power supply120. FIG.2illustrates a more detailed embodiment of a system100. The high voltage power supply120includes an input122and at least one output124. As used in this specification, multiple similar components may be numbered with a numeric prefix and a trailing letter. The use of the numeric prefix only, refers to all of the similar components beginning with the numeric prefix. For example, the output124of the high voltage power supply120refers to output124a, output124b, output124cand output124dunless otherwise indicated. The illustrated embodiment includes four outputs124labeled a-d. However, this is merely representative of one possible configuration and should not be taken as limiting the scope of the disclosure. In some embodiments, the high voltage power supply120has more or less than four outputs124. In some embodiments, there are in the range of two to eight outputs124. The input122of the high voltage power supply120of some embodiments is connected to a high voltage source123. The high voltage source123can be any suitable source known to the skilled artisan. In some embodiments, the high voltage source123provides voltage at the input122in the range of 50 V to 500 V. The high voltage power supply120converts the alternating current AC input voltage to a high voltage direct current DC voltage. In some embodiments, the high voltage power supply120provides an output power in the range of 0 V to ±2000 V. The skilled artisan will understand that the output from the high voltage power supply120includes two power lines in which the electrical potential difference is in the range of 0 V to ±2000 V. The skilled artisan will be familiar with high voltage power supplies and handling high voltage DC power. In some embodiments, the load140comprises at least one electrostatic chuck ESC140a. The electrostatic chuck140ais connected to the at least one output124aof the high voltage power supply120. In the embodiment illustrated inFIG.2, there are four electrostatic chucks140a,140b,140c,140d. However, the skilled artisan will recognize that there can be more or less than four loads140. In some embodiments, there are in the range of 1 to 12 electrostatic chucks. In some embodiments, there are the same number of electrostatic chucks as outputs on the high voltage power supply. A high voltage measurement system110is connected to the high voltage power supply120either directly, or through the electrostatic chuck140aor other loads140. The high voltage measurement system110includes at least one high voltage transducer112a. In some embodiments, there are the same number of high voltage transducers112as outputs124on the high voltage power supply120. In some embodiments, there are more high voltage transducers112than outputs124. In some embodiments, there are the same number of high voltage transducers112as loads140. In some embodiments, there are more high voltage transducers112than loads140. In some embodiments, the at least one transducer112of the high voltage measurement system110is connected in parallel to the at least one electrostatic chuck (load140). The high voltage measurement system110includes a controller150that is configured to measure an output from the high voltage transducer112. In some embodiments, the controller150is connected to the transducer112and is configured to measure one or more of voltage or current from the output114. In the illustrated embodiment, the controller150is outside of the high voltage measurement system110. However, this is a schematic arrangement that is merely representative of one possible configuration. In some embodiments, the controller150is internal to the high voltage measurement system110. The controller150can be a single control system or a combination of control systems. For example, the controller150of some embodiments comprises internal circuitry in the high voltage measurement system110that work with an external computer system. The controller150can be any suitable component that can control the high voltage power supply120and/or the high voltage measurement system110. For example, the controller150can be a computer including a central processing unit (CPU), memory, inputs/outputs (I/O), and support circuits. The controller150may control the system100directly, or via computers (or controllers) associated with particular processes and/or support system components. In one or more embodiments, the controller150may be one of any form of general-purpose computer processor that can be used in an industrial setting for controlling various chambers and sub-processors. The memory or computer readable medium of the controller may be one or more of readily available memory such as non-transitory memory (e.g. random access memory (RAM)), read only memory (ROM), floppy disk, hard disk, optical storage media (e.g., compact disc or digital video disc), flash drive, or any other form of digital storage, local or remote. The memory can retain an instruction set that is operable by the processor (CPU) to control parameters and components of the system. The support circuits are coupled to the CPU for supporting the processor in a conventional manner. These circuits include cache, power supplies, clock circuits, input/output circuitry and subsystems, and the like. One or more processes may be stored in the memory as software routine that, when executed or invoked by the processor, causes the processor to control the operation of the system100or individual components in the manner described herein. The software routine may also be stored and/or executed by a second CPU (not shown) that is remotely located from the hardware being controlled by the CPU. Some or all of the processes and methods of the present disclosure may also be performed in hardware. As such, the process may be implemented in software and executed using a computer system, in hardware as, e.g., an application specific integrated circuit or other type of hardware implementation, or as a combination of software and hardware. The software routine, when executed by the processor, transforms the general-purpose computer into a specific-purpose computer (controller) that controls the chamber operation such that the processes are performed. In some embodiments, the controller150has one or more configurations to execute individual processes or sub-processes to perform the method. The controller150can be connected to and configured to operate intermediate components to perform the functions of the methods. In some embodiments, the controller150is connected to and configured to control one or more of the high voltage power supply120or the high voltage measurement system110. The transducers112comprise an input113and an output114. The input113of the transducers112is configured to accept a high voltage from the high voltage power supply120and output114a low voltage signal correlated to the input voltage. In some embodiments, the high voltage power supply120has an output in the range of 0 V to ±2000 V connected to the input113of the transducer112, and the transducer112has an output114in the range of 0 V to ±50V, or in the range of 0 V to ±40 V, or in the range of 0 V to ±30 V, or in the range of 0 V to ±20 V or in the range of 0 V to ±10 V. In some embodiments, the output114of the transducers112provide a voltage that correlates with the output124of the high voltage power supply120with a 0.2% accuracy. Stated differently, the output of the transducer provides a correlated value to the output of the high voltage power supply that is within 0.2% relative to the actual high voltage signal. In some embodiments, the transducer112provides an output voltage in the range of 0% to 2.5%, or 0% to 2%, or 1.5%, or 0% to 1%, or 0% to 0.5% of the voltage at the input. In some embodiments, controller150is operatively connected to an interlock116with the high voltage power supply120. In some embodiments, the controller150of some embodiments is configured to turn off the high voltage power supply120through the interlock116when a decrease in voltage greater than a threshold value is measured from the transducer112. For example, a controller expecting to receive a reading of 10V from the transducer112a, receives instead a value of 5 V. If the 5V reading is different from the expected value (e.g., 10V) by a threshold amount (e.g., 1%), then controller150can disable high voltage power supply120through the interlock116. In some embodiments, the controller150is configured to adjust the high voltage power supply120to bring the measured value into the expected range and maintain the chucking of a wafer. Some embodiments of the disclosure are directed to methods for the live measurement of voltage from a high voltage power supply. A load is powered using the high voltage power supply. For example, an ESC is connected to the output of the high voltage power supply. The voltage from the output of the high voltage power supply is converted using a transducer in a high voltage measurement system. The measured converted voltage of some embodiments is logged in a system database. An interlock between the high voltage measurement system and the high voltage power supply is operated or suitably controlled if the measured converted voltage meets a threshold value to disable the high voltage power supply. Additional embodiments of the disclosure are directed to non-transitory computer readable medium including instructions, that, when executed by a controller of a processing tool, causes the processing tool to perform operations of: measuring an output of a transducer connected to a high voltage power supply; comparing the output to a predetermined threshold value; and operating an interlock between the high voltage power supply and the transducer to turn off the output from the high voltage power supply. In some embodiments, the non-transitory computer readable medium further includes instructions, that, when executed by the controller, causes the processing tool to perform operation of powering at least one electrostatic chuck using an output from the high voltage power supply. Reference throughout this specification to “one embodiment,” “certain embodiments,” “one or more embodiments” or “an embodiment” means that a particular feature, structure, material, or characteristic described in connection with the embodiment is included in at least one embodiment of the disclosure. Thus, the appearances of the phrases such as “in one or more embodiments,” “in certain embodiments,” “in one embodiment” or “in an embodiment” in various places throughout this specification are not necessarily referring to the same embodiment of the disclosure. Furthermore, the particular features, structures, materials, or characteristics may be combined in any suitable manner in one or more embodiments. Although the disclosure herein has been described with reference to particular embodiments, those skilled in the art will understand that the embodiments described are merely illustrative of the principles and applications of the present disclosure. It will be apparent to those skilled in the art that various modifications and variations can be made to the method and apparatus of the present disclosure without departing from the spirit and scope of the disclosure. Thus, the present disclosure can include modifications and variations that are within the scope of the appended claims and their equivalents. | 15,697 |
11863090 | DETAILED DESCRIPTION A method and system are disclosed for the generation of electrical energy for use in numerous applications. The method is general in its applications and can be applied to many electrically powered devices, such as portable tools, sensors, optical devices, lighting, heating, cooling, breathing apparatus, medical devices, timing devices, portable computers, cell phones, powered cooling or heating devices as well as other similar and larger stationary applications where a convenient and powerful supply of electrical energy is needed. The need for such a device and method is well documented. The Carver Voltaic Effect (CVE) is a kinetic physical effect that can be used to provide significant electrical power. The CVE can be described as the minute transient increase in the power of a single power transmission transient in electrical conductors or in energy transfers in materials through space. The term “kinetic” is used to describe the transitory nature of the effect. It can be detected during transitory events, such as fast voltage changes and some other phase and state changes in materials. Embodiments of the devices described herein are constructed to take advantage of this phenomena (i.e., the CVE) by the apparent conversion of thermal energy to electrical energy. The magnitude of the CVE is associated with large dV/dt values (changes in voltage with respect to time). Understanding the operation and manufacture of the device includes the recognition of the presence of an etalon in the output circuit and methods for the implementation and manufacture of the etalon are disclosed. InFIG.1, is a circuit100for converting thermal energy into electrical energy. A square wave generator105generates a square wave pulse train (continuous pulses) that enters a primary side of a coupled inductor110. The coupled inductor's secondary side is connected to a nonlinear resistive device, or as is sometimes called, a negative resistance device112, such as a thyristor. The negative resistance device112serves as a device to limit the current from the secondary to a certain value determined by its internal construction based upon the input voltage. It will not conduct meaningful current until the voltage exceeds a certain amount in the positive direction and will not conduct in the negative voltage until the voltage is more negative than a certain amount. For example, the two voltages may be +25V and −25V. Because of this voltage characteristic, the output of the secondary side of the coupled inductor is always certain to exceed +25V and −25 Volts provided sufficient power is available to overcome parasitic losses. The negative resistance device can be any device that can provide this type of action. Example devices include, but are not limited to, the following:1. Gas discharge lamps2. Spark gaps3. Zener diodes4. Thyristors5. Triacs6. Gunn diodes7. Diodes (all kinds)8. Silicon controlled rectifiers (SCR)9. Switching devices controlled by a logic circuit As the driving electronics for the transformer (or coupled inductor) cause the output of the secondary to swing from positive to negative, very fast transitions from the >25V to more negative than −25V will take place. These high dV/dt transients are then utilized to produce fast voltage swings desired for the CVE to be utilized. Thus, the larger the dV/dt (higher voltage, less time), the more pronounced the CVE. The square wave in combination with the negative resistance device112help to achieve this goal. In this example, the capacitor C1114and the inductor116form an oscillatory circuit that further amplifies the effects of the current with its voltage swings to produce useful output at C2118. The C2capacitor118is in turn connected to one or more rectification diodes, shown generally at120to produce both a positive and negative voltage output, V+ and V−, respectively. The oscillatory circuit formed by the capacitor114and inductor116can generate a signal oscillating at a frequency greater than the frequency of the square wave input signal. A thermal exchanger130provides a thermal conduction path for the materials to have a continual influx of thermal energy for conversion to electrical energy. The thermal exchanger can be any device used to inject heat into the circuit. In one example, a tube (e.g., a conductive tube or non-conductive tube) is used that is filled with material having a desired permittivity and permeability. Potential materials include air, water, methanol, ethanol, and acetamide (or a solution in liquids such as water or ethanol). Ferrite slurries can also be used. The material can be pumped or circulated through the tube using an external pump, not shown. Alternatively, the solid materials can be immobilized within the resonant cavity. Subsequently liquids can be pumped through the tube to provide heat exchange to the materials and the tube itself. The tube can be any desired length. For example, the tube can be 1 ft to 5 ft in length. The tube can be any desired shape in cross-section such as round, square, rectangular, elliptical, a flat-sided oval, or a custom shape. Any geometric shape can be used (e.g., an N-sided polygon or a folded shape). Whatever the cross-section, the tube can be elongated with a cavity therein through which fluid can pass. The tube can be an etalon as described herein. The etalon can also be a solid conductor. FIG.2shows a generic version of the circuit200. An optional driver210can be a continuous pulse generator that supplies a continuous stream of pulses with high dV/dt. This provides the starting impulse to the device. It can serve as the on/off switch to run the device and it can help control the frequency at which the device is operated. A dV/dt device220is shown.FIG.1showed the dV/dt device as a transformer or a coupled inductor110to indicate at least one way of generating a high dV/dt pulse or series of pulses. Alternatives to this could be a capacitor or capacitor array, a mechanical switch, or other spinning or rotation devices that bring an electrical (charge) or magnetic field (magnet) in proximity to another coil, capacitor, inductor, or another magnet or magnetic field. The CVE device may have one or more significant active devices incorporated within it. Examples are the negative resistance devices, such as a thyristor or Zener diode. The CVE emitter230is shown coupled to a thermal exchanger240. The thermal exchanger can, in turn, be coupled to a CVE receiver250. The rapid formation of a dV/dt charge on the emitter230leads to the production of a “wave” of energy from the emitter. In this antenna-like mode, the emitter may be in contact with a material other than a vacuum or air. The material may have the properties of having a different dielectric constant or magnetic permeability characterized by its relative permittivity or permeability. It may also be in contact with a conductive material. The emitter230and receiver250can be a wide variety of materials (e.g., copper, brass, bronze, stainless steel, graphene) that create impedance changes at the ends of the etalon chamber. Indeed, anything can be used, so long as it changes the permittivity, permeability, or both with respect to the material between the emitter and receiver. Thus, the emitter230couples the circuit to the thermal exchanger240(which can be an etalon) and transmits a signal to the thermal exchanger. The receiver250receives the signal once it passes through the thermal exchanger. The thermal exchanger240is shown as being between the CVE emitter and the CVE receiver. It may, in fact, be surrounding the emitter and the receiver. For example, where the thermal exchanger is a tube having a cavity therein, the emitter230and receiver250can be mounted in respective ends of the tube. The thermal exchanger provides the needed thermal conduction path for the materials to have a continual influx of thermal energy for conversion to electrical energy. The materials may also be electrically conductive. The thermal exchanger can be any device used to inject heat into the circuit. In one example, a tube (e.g., a conductive tube or non-conductive tube) is used that is filled with material having a desired permittivity and permeability. Potential materials include air, water, methanol, ethanol, and acetamide (or a solution in liquids such as water or ethanol). Ferrite slurries can also be used. The material can be pumped or circulated through the thermal exchanger using an external pump, not shown. Alternatively, the solid materials can be immobilized within the resonant cavity. Subsequently liquids can be pumped through the cavity to provide heat exchange to the materials and the cavity itself. Thus, the material can have a dual purpose of acting as a medium between the CVE emitter and CVE receiver and acting as a thermal exchanger having an external source that is circulated through the thermal exchanger. Electronic waves can be transmitted between the CVE emitter and CVE receiver and the permittivity and permeability of the materials contained therein can impact the resonant frequency. The CVE receiver250is shown coupled to the thermal exchanger. It may or may not be in contact (e.g., air gapped or spaced) with the thermal exchanger240. The receiver250, by electrical induction from the wave, electrical contact with the thermal exchanger, or by electrical contact with the emitter230has the increased energy provided by the CVE. The receiver harvests the converted heat into an electrical conduction path to either be utilized directly by a load260or to be conditioned by a conditioning circuit270. The load260can be any desired load and can have a resistive component (e.g., a light bulb). The conditioning circuit270are shown connected to the CVE receiver250. This circuit270is typically a circuit to convert the AC signal (or pulsed DC) into another frequency range or convert to a DC voltage or voltages. An example conditioning circuit can be a full bridge rectifier and capacitor. An electrical load280receives an output of the conditioning circuits270. The load may be anything that uses electrical energy. It is similar to the direct use of the electrical energy load260but it may require conditioning from module270. Module260is the direct use of the output of the CVE receiver250. This output has typical AC signal characteristics. Resistive loads would be acceptable for this type of electrical characteristic as either square or sinusoidal waves. FIG.3is a circuit300in which the negative resistance device is used in conjunction with the emission of the dV/dt wave as shown by connection to component320. A pulse generator310is coupled to an inductor or transformer312. The output of the secondary of the coupled inductor or transformer312is referenced to a voltage indicated by V340. A negative resistance device345is coupled to the inductor The emission of the wave from component320can be coupled to the receiving component350. The receiving component350can also be connected to a load360. The connection between the receiving component320and the receiving component350is shown by a dashed bidirectional arrow and can be a vacuum, air, or other dielectric materials either homogeneous or heterogenous. Conductive materials can also be used. FIG.4is a circuit400using an etalon for amplification. The dV/dT device410can be any pulse generator. Alternatively, as shown above, the dV/dT device can be a transformer coupled to a negative resistance device, as is shown inFIG.3. The combination of elements420,430comprise a resonance cavity similar to an etalon or Fabry-Perot interferometer. It can be similar to the description of the thermal exchanger130. It is shown without a load. It may be utilized without an attached load by either emission of electrically induced waves or by simply being a higher voltage source reference for reference applications. With a load (e.g. resistive) the etalon can produce amplified power from the dV/dt device by capturing the thermal energy between the emitter and the receiver and the coupling component itself, particularly but not exclusively, when resonance occurs. Activation frequencies can be used that are much lower than optical frequencies. In most cases, the lowest fundamental wavelength in the resonance cavity is very long compared to the relative sizes of the other components. In order to reduce the size of the resonance cavity, higher relative permittivity or permeability materials can be used to significantly reduce the length of the etalon involved. This area of the device is shown by the dotted double-headed arrow between components420and430. In the case of a high permittivity capacitors, relative permittivity in the ranges of 3 to ≥20,000 are not uncommon. Higher permittivity materials are known. These materials provide for a highly decreased etalon length by similar factors such as the square root of the inverse of the relative permittivity multiplied by the relative permeability. An etalon440is shown between the components420,430. The etalon (wave resonant cavity) chamber can be considered as one (or more) of the oscillator components. This particular etalon differs from a purely electrical conductivity element by involving emitted electrical waves rather than electrical current oscillation in a conductor. A hollow etalon also provides the ability to fill the resonance cavity with a material that has a permittivity (and/or a magnetic permeability) that is greater than vacuum or air. This increased permittivity/permeability decreases the fundamental oscillation length. Folding (or coiling) the length helps reduce the overall size. The etalon cavity may be where most of the heat conversion to electrical energy will take place. Fluid can be moved through the etalon's cavity. The fluid will be constantly cooled by the resonance of the dV/dt waves while the movement of the etalon fluid provides a way to effectively get heat into the resonance volume by carrying the heat from an external source. Or, simple heat conduction/convection into the resonance cavity volume can be used to provide the heat from an external heat source, possibly using a second fluid (e.g. water) or heat pipe. The etalon440is shown as a cylindrical tube, in this embodiment, with a cavity extending therethrough. A pump450is used to pump fluid through the etalon440. A heat sink460is used to extract heat from the environment (which can be an ambient environment or any temperature above or below ambient) and pass the heat to the fluid. The etalon can then convert the heat to electrical energy. The etalon can be filled with materials that have different permittivities and permeabilities, such as air, water, methanol, ethanol, and acetamide (e.g. in a solution of water or ethanol). Higher permittivity materials allow a lower drive frequency to be used and still be at resonance. The etalon can have a dual purpose of acting as an electrical coupling between the component420and the component430and also acting as a thermal exchanger. The emitter420and receiver430can be a wide variety of materials (e.g., copper, brass, bronze, stainless steel, graphene) that create impedance changes at the ends of the etalon chamber. Different electrical elements can also be used as the emitter420and receiver430, such as inductors and capacitors. Indeed, anything can be used, as long as it changes the permittivity, permeability, or both with respect to the material between the emitter and receiver. The load should be selected so as to have proper impedance matching with the source, as is well known in the laser, transmission, and antenna fields. FIG.5is a circuit500that is an additional schematic representation of the material510in between the etalon's reflective surfaces,520and530. The thermal energy material510is in the transmissive path and/or reflective path of the wave coming from the emitter or the reflected wave from the receiver. Due to the CVE, the power in the wave is augmented by each traverse of the wave between the surfaces. In this way the material510is cooled, since the energy required for the increase in energy in the wave is obtained from the thermal energy contained in the material itself due to the law of conservation of energy. To achieve resonance in a given cavity, the cavity's shape must be taken into account. Square or round shapes may be used as well as oval, elliptical, polygonal, and other geometrical shapes. Also, the material filling a resonance cavity plays a part in determining the frequency of resonance. It is known that increasing the permittivity or permeability of the material filling a given cavity changes its resonance to a lower frequency. In the case of the frequency of electrical waves, the resonant frequency of the cavity is related to the square root of the inverse of the relative permittivity multiplied by the relative permeability of the material vs a pure vacuum. Thus, higher permeability and higher permittivity materials can lead to reduced physical sizes of the etalon cavity. Higher permittivity materials (Thermal Energy Material) may be used to provide an etalon cavity that is substantially shorter (thereby smaller) than that with vacuum or air-filled cavity. Additionally, the material510may be thermally conductive to facilitate thermal transfer into the cavity from the environment or heat source. Liquid materials are attractive in that they can be circulated to facilitate heat transfer. Materials that can be used are those that are transmissive to the wave itself. Some materials (or mixtures, suspensions, or slurries thereof) that may be used but are not the limitation for use are as follows:1. Barium titanate2. Other Perovskite mixed metal titanates3. Ferrite4. Inorganic Oxides5. Air6. Organic alcohols7. Organic materials that may be transmissive to the wave8. Conductive metals9. Semiconductive materials10. Species of carbon (e.g. graphite, graphene, Fullerenes)11. Materials which themselves re-resonate at other frequencies (e.g. phosphors, rhodamine) via harmonic generation12. Water or water with dissolved salts, liquids, or other species suspended or homogeneous. Materials can be used to partially fill or fully fill the cavity to provide a pathway for thermal conduction to the etalon cavity. The load540can be any desired electrical load, such as a load having a resistive component. The dV/dt device550is similar to those described above. As an example of the device, the following set of components can be used.1. Transformer (coupled inductor), 10:1 ratio, 2 A current rating, 700 uH secondary inductance2. 0.01 uF, 1000 V ceramic capacitor3. 254 uH ferrite single inductor, 10 A inductor4. Copper tube (⅝″ OD×½″ ID×24 inches length)5. Powdered ferrite (125 mesh)6. Resistive load (110 Ohm, 100 W metal film resistor)7. 2 pc Copper wire (10 AWG×1″ long)8. Zener Diode (1N5388) Using the schematic shown inFIG.1, the copper tube is first packed with the ferrite powder. One piece each of the copper wire is inserted into each end of the tube and used to make connection to the remainder of the circuit. The transformer is driven by means of a pulsed current source at a frequency of 1 Hz to several GigaHertz. The exact frequency required can be tuned by maximizing the ratio of power produced to the power necessary to drive the transformer's primary. The secondary of the transformer is attached to one piece of the copper wire in the copper tube. The other end of the copper tube with the remaining wire is attached to a negative resistance device such as a Zener diode. The other end of the diode is attached to an inductor. The remaining connection is led back to the secondary of the transformer's output. Electrical energy can be obtained by attachment of a capacitor to almost any portion of the above secondary circuit as a tap to the voltage produced in the resonance circuit. The remaining lead on the capacitor can optionally connect to a rectifier circuit for further conversion to an AC, pulsed DC, or smoothed DC output by conventional means. FIG.6is a flowchart for generating power according to an embodiment. In process block610, a continuous stream of pulses is generated, such as by a pulse generator. The pulse generator can generate pulses having a dV/dt of 100V/μs or even 10,000 to 100,000 V/μs or higher. Specific use cases have used between 3 to 10V/μs. In some cases, 1V/μs can be used. In process block620, the continuous stream of pulses are applied to a tube having a cavity extending therethrough. The tube can be conductive and have fluid continuously pumping through the cavity (process block630). The fluid can be warmed by a heat sink or other heating element. The fluid can be cooled as it passes through the tube due to the CVE. At process block640, an electrical signal can be output from the tube having a greater power than was output by the pulse generator due to conversion of thermal energy of the fluid to electrical energy. In some embodiments, an oscillator can be used to generate pulses at a greater frequency than the pulse generator. The etalon described above is a “cavity” wherein an electrical wave can resonant to provide an increased voltage at its exit. To provide the required reflections of the electrical wave, an inductive element is provided on one end of the etalon and a capacitive element is provided on the other end. The inductive elements (L1and L2) alone or in combination provide a wave-reflective element on a first end of the etalon and the capacitor (C1) provides another reflective element on the opposing end. Additionally, the inductor(s) and capacitor(s) can provide for an oscillator circuit that is capable of being electrically driven to provide a current to the etalon resonant cavity that lies in series within the oscillatory circuit of inductors and capacitors. InFIG.7A, a circuit including a driving component710provides a signal to an etalon720that is amplified by reflections within the etalon to provide an increased amount of energy to a load730. In all figures the load is represented as a resistive element, but other components can be used. For example, a rectifier circuit can be used to convert the AC waveform into a DC voltage.FIG.7Bshows a variety of different driver circuits710can be used including a transformer tap, a capacitor tap and a center tap. InFIG.8, an etalon810has a resonant frequency that is in-phase with an external oscillator frequency. Although the fundamental frequency of the etalon does not have to exactly match an external oscillator frequency, it should provide at least an integer multiple of a drive frequency to optimally correlate with its internal fundamental frequency. However, in some cases, the driven frequency can be faster than the etalon's internal frequency. In the case of a single pulse from the “External Coupling Driver” (ECD)820, if the pulse has frequency content that is an integer multiple or other mode multiple of the etalon fundamental frequency (such as would be present in a square wave), then some amplification of the etalon's wave can take place. Different methods can be used to magnify and increase the Q (or finesse) of an electrical resonant cavity or circuit. InFIG.9, an etalon910can be driven with a single pulse from an External Coupling Driver920. The etalon910can fully resonant and amplify the pulse as much as possible before the transitory signal dies down due to power drawn from stray resistances, emissions, and the load. The induced voltage from the ECD at the point of connection to the oscillatory circuit can optionally be at a single point due to the phase shift of the signal upon its return, although ground or other reference voltages can be used as well. An inductor930and capacitor940are on opposite sides of the etalon910and can be swapped if desired. InFIG.10, a general scheme for driving an etalon1010is shown. The previous figures show an inductor and capacitor as the elements in an oscillator circuit. The L1and C1components can be generalized to be any partially reflective and partially transmissive component in the conductive circuit, as shown at1020. A variety of negative resistance devices fulfill this description as described above. The L1and C1components can be a variety of electrical components, such as zener diodes, gas plasm lamps, or similar devices described above. These components are show as simple open parentheses on both ends of the etalon. Other components can optionally be added that may provide the required combination of both transmission and reflection to make the etalon's partially reflective endpoints. In order to fulfill the demands of the etalon and the oscillatory circuit simultaneously, it is advantageous to have at least one variable passive component in the circuit. InFIG.11, a variable capacitor1120and a variable inductor1130are used in conjunction with an etalon1140. The variable capacitor1120can be controlled by an external voltage (e.g. varactor) or may be a mechanically actuated variable air capacitor. Other methods can be used. The variable inductor1130can have a mechanically movable magnetic core material. Either a fixed capacitor or fixed inductor can be swapped for one of the variable elements. Other inductors that provide for a variable inductance can be used as substitutes. The variable passive devices can be controlled using an analog comparator1150, digital microcontroller, or other electronic controls to control in a logical manner the output of the circuit at the load point. This is accomplished by controlling the drive frequency and pulse width of the “external coupling drive” (path A), inductance value of L2(path B), and controlling the value of C1(path C). FIG.12is one example method for starting and controlling the etalon resonance. In process block1210, a starting point is set for the inductance and capacitance (if variable passive components are used) by selection of their default values along with the default load (if adjustable). For example, inFIG.11, a starting point can be chosen for the inductor1130and capacitor1120, such as by using a default value determined based upon previous testing. In process block1220, a frequency and pulse width are initiated for the ECD1160. Generally, the voltage for the drive is fixed, although this voltage may be an additional parameter to adjust. In process block1230, the output power of the circuit is measured at the load input, such as at1170. Alternatively, the load can be coupled to the opposite side of the capacitor C1, as shown by optional line1172(shown in dashed). If the output power is adequate or within specification, then in decision block1240, process block1230is repeated. On the other hand, if decision block1240is answered in the negative, then the inductance (L2)1130or the capacitance (C1)1120are driven higher or lower to maximize the output to a desired value or to a maximum value. In process block1260, the output of the ECD is minimized by reducing the frequency of the pulses to a specified value (based upon the specifications of the particular unit defined by the factory). In process block1270, a pulse width of the ECD is minimized to provide the lowest wattage input while providing the desired output to the load. A loop is then performed where process blocks starting with1230are repeated. Variations of the circuits disclosed herein can be made without deviating from the underlying principles. For example, the inductors (L1and L2,FIG.8) are shown as electrical symbols with magnetic cores such as ferrite or iron. This symbol was chosen for its clarity, but it may also represent other inductors such as air-core inductors. The inductors L1and L2can be grouped into a single variable inductor. The “load” is shown as a resistor, but it may be any electrical energy consuming device that utilizes the output of this device. The load is connected to the opposite side of the circuit to the etalon from the drive point of the circuit. This load may be optionally connected at any point in the circuit. The position shown is adequate for most implementations of the circuit. The etalon may simply be a length of copper wire (or other conductor) wound in the shape of a torus, a tubing form made from a conductor, or a conductive packed tube with non-conductive materials. The shape may be a torus to provide compact density for long conductors of solid or tube conductors packed or unpacked. The winding may be elongated and/or wound to be a dense pack of loosely spaced elements to provide for better heat transfer. In each case, the heat transfer may be primarily through and into the etalon due to its size and shape. The etalon may be a conductive hollow tube that will serve as a wave-guide. It may be filled with nonconductive materials. The wave-guide is not part of the circuit, but serves to contain the wave. The ends of the filled tube are simply conductive materials that serve as antenna or “electrodes” for the circuit's etalon. Electrically, the conductive endpoints of the tube are connected to the oscillatory circuit, and the waveguide may be isolated from the circuit. In view of the many possible embodiments to which the principles of the disclosed invention may be applied, it should be recognized that the illustrated embodiments are only preferred examples of the invention and should not be taken as limiting the scope of the invention. Rather, the scope of the invention is defined by the following claims. We therefore claim as our invention all that comes within the scope of these claims. | 29,626 |
11863091 | DETAILED DESCRIPTION The inventive concept will be described more fully hereinafter with reference to the accompanying figures, in which embodiments of the inventive concept are shown. This inventive concept may, however, be embodied in many alternate forms and should not be construed as limited to the embodiments set forth herein. Accordingly, while the inventive concept is susceptible to various modifications and alternative forms, specific embodiments thereof are shown by way of example in the drawings and will herein be described in detail. It should be understood, however, that there is no intent to limit the inventive concept to the particular forms disclosed, but on the contrary, the inventive concept is to cover all modifications, equivalents, and alternatives falling within the spirit and scope of the inventive concept as defined by the claims. Like numbers refer to like elements throughout the description of the figures. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the inventive concept. As used herein, the singular forms “a”, “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprises”, “comprising,” “includes” and/or “including” when used in this specification, specify the presence of stated features, integers, steps, operations, elements, and/or components, but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or groups thereof. Moreover, when an element is referred to as being “responsive” or “connected” to another element, it can be directly responsive or connected to the other element, or intervening elements may be present. In contrast, when an element is referred to as being “directly responsive” or “directly connected” to another element, there are no intervening elements present. As used herein the term “and/or” includes any and all combinations of one or more of the associated listed items and may be abbreviated as “/”. Unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this inventive concept belongs. It will be further understood that terms used herein should be interpreted as having a meaning that is consistent with their meaning in the context of this specification and the relevant art and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein. It will be understood that, although the terms first, second, etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another. For example, a first element could be termed a second element, and, similarly, a second element could be termed a first element without departing from the teachings of the disclosure. Although some of the diagrams include arrows on communication paths to show a primary direction of communication, it is to be understood that communication may occur in the opposite direction to the depicted arrows. Some embodiments of the inventive concept arise from a realization that, in reduced voltage soft starters (RVSSs), metering and control applications (e.g, torque control) may be implemented using an estimate of motor voltage synthesized from line voltage and motor current, thus eliminating the need to directly sense the motor voltage. In some embodiments, the input line-to-line voltage is converted using an ABC-dq0 transformation. The output (e.g., a “d” output or equivalent thereof) of the transformation is processed to determine an RMS value that accounts for voltage drop in the line impedance upstream of the RVSS. The RMS value is scaled for conversion from line-to-line voltage to line-to-neutral voltage and provided to a dq0-ABC converter with a 30 degree phase shift with respect to the original ABC-dq0 conversion to produce sinusoidal line-to-neutral reference voltage signals. Motor phase current signals may then be used to “notch” (i.e., selectively correct for times when the RVSS switches are blocking or commutating) these reference voltage signals to generate line-to-neutral voltage signals representative of the actual line-to-neutral motor voltages. This processing may involve using window comparators to detect nulls in the phase currents and adjusting the line-to-neutral reference voltages accordingly so that they reflect the effects of switching by the SCRs of the RVSS. In some embodiments, the thresholds of the windows may be about 10% to about 15% of the rated full load current of the motor. The estimated motor line-to-neutral voltage signals may be used, for example, to develop electromagnetic torque estimates for motor torque control. They can also be used for metering and other purposes. FIG.1illustrates apparatus according to some embodiments. An RVSS includes a switching circuit110connected between an AC power source10and a motor20. The switching circuit may include pairs of antiparallel-connected silicon-controlled rectifiers (SCRs) for each motor phase. A bypass contactor130is configured to bypass the switching circuit110and directly couple the AC power source10to the motor20. A control circuit120controls the switching circuit110and the bypass contactor130. The control circuit120includes a motor voltage estimator circuit122which is configured to receive voltage signals Va_1, Vb_1, Vc_1representing the line voltage applied to the switching circuit110and current signals Ia, Ib, Icrepresenting motor currents, which are sensed by an input voltage sensor105and an output current sensor115, respectively. The motor voltage estimator circuit122responsively generates motor voltage signals Va_m, Vb_m, Vc_mrepresentative of line-to-neutral voltages at the motor20. Along with the motor current signals Ia, Ib, Ic, these may be applied to a torque control circuit124, which controls a driver circuit126that fires the SCRs of the switching circuit110. Referring toFIG.2, the torque control circuit124ofFIG.1may include a torque estimator circuit210that is configured to generate an instantaneous electromagnetic torque estimate signal τem. An RMS circuit220generates an RMS torque signal τem_rmsfrom the instantaneous electromagnetic torque estimate signal τem. The RMS torque signal τem_rmsmay be compared with a torque reference signal τemref by a summing circuit230to generate an error signal Δτ that is applied to torque compensator circuit240, which may responsively provide a command signal to the driver circuit126shown inFIG.1. FIG.3illustrates apparatus and operations of torque estimator circuit according to some embodiments. The line voltage signals Va_1, Vb_1, Vc_1are applied to an ABC/dq0 converter circuit310, which generates corresponding dq0 component signals d, q, 0 using a phase angle ω. The d component signal d is divided by the square root of 3 by a scaling circuit320and the result applied to a dq0/ABC converter circuit340, which generates pre-compensated line-to-neutral voltage signals V*a, V*b, V*cusing a phase angle ω* that is shifted by 30 degrees with respect to the phase angle ω by a phase shift circuit330. The line-to-neutral voltage signals line-to-neutral voltage signals V*a, V*b, V*care provided to a notch compensation circuit350, which selectively introduces notches into the line-to-neutral voltage signals V*a, V*b, V*cresponsive to the motor current signals Ia, Ib, Icto generate estimated motor voltage signals Va_m, Vb_m, Vc_mthat represent the line-to-neutral voltages of the motor20. FIG.4illustrates notch detection and compensation circuitry according to some embodiments. The motor current signals Ia, Ib, Icare provided to respective window comparator circuits410, which detect nulls in these currents based on whether these current signals are inside or outside of windows defined by upper and lower thresholds around zero. Comparison signals415generated by the window comparator circuits410are provided to compensator circuits420, which also receive the motor current signals Ia, Ib, Icand respective ones of the line-to-neutral voltage signals V*a, V*b, V*c. Responsive to the comparison signals415, receive the motor current signals Ia, Ib, Icand respective ones of the line-to-neutral voltage signals V*a, V*b, V*c, the compensator circuits420generate respective ones of the motor voltage signals Va_m, Vb_m, Vc_m. FIG.4is a flowchart illustrating representative operations for generating an estimated a-phase motor voltage signal Va_mvalue according to some embodiments. If the a-phase current signal Iais within the associated detection window, the estimated motor voltage signal Va_mis set (notched) to a value equal to 0.5 times the product of the b-phase current signal Iband a stator inductance Ls of the motor and the updated value for the a-phase motor voltage signal Va_mis provided to the torque estimator (blocks510,520,580). The stator inductance Ls may be based, for example, on a heuristic rule of thumb, a locked rotor code, or motor parameter identification. If the a-phase current signal Iais not within the window, the a-phase motor voltage signal Va_mmay be set to the current value of the pre-compensated line-to-neutral voltage signal V*a(blocks510,530). If either of the b-phase or c-phase current signals Ib, Icis within their corresponding null detection windows, the motor voltage signal Vam is set to a value equal to the pre-compensated line-to-neutral voltage signal V*acorrected by factors equal to the a-phase current signal Iamultiplied by the stator inductance Ls (blocks540,550,560,570). The estimated motor voltage signal Va_mvalue is provided to the torque estimator (block580). The b- and c-phase estimated motor voltage signals Vb_m, Vc_mmay be generated in a comparable manner. In particular, the b-phase estimated motor voltage signal Vb_mmay be set to a value equal to 0.5 times the product of the c-phase current signal Icfor nulls of the b-phase current Ib, and c-phase estimated motor voltage signal Vc_mmay be set to a value equal to 0.5 times the product of the a-phase current signal Iafor nulls of the c-phase current Ic. Corrections to the pre-compensated voltage signals V*a, V*b, V*care made for nulls in the other phase currents along the lines described with reference toFIG.5.FIG.6illustrates an estimated motor voltage600generated using such operations, in comparison to a measured motor voltage610and the line-to-line voltage620of the AC source. Apparatus described herein may be implemented using electrical circuitry that implements various control structures. It will be understood that such circuitry may include analog and digital circuitry. Components referred to herein as “circuits” may be implemented using analog circuitry, discrete logic, embedded data processing circuitry (e.g., microprocessors, microcontrollers and the like) and combinations thereof. For example, certain control circuitry may be implemented using a microprocessor or microcontroller that executes computer program instructions to provide circuitry that implements the control operations described. Such data processing circuitry may be used in conjunction with analog circuitry (e.g., peripheral devices such as analog/digital converters and analog filters) and/or discrete digital circuitry (e.g., logic gates, buffers and the like). Such control operations may also be implemented using similarly functioning analog control circuitry and/or discrete digital circuitry (e.g., logic gates, programmable logic devices, and the like) alone or in combination with data processing circuitry. Signals produced by such circuitry may include analog signals, logic signals and/or digital signals (values) stored in structures such as a register or memory associated with a data processing circuit, such as a microprocessor. Example embodiments herein with reference to block diagrams and/or flowchart illustrations. It is understood that a block of the block diagrams and/or flowchart illustrations, and combinations of blocks in the block diagrams and/or flowchart illustrations, can be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general purpose computer, special purpose computer, and/or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer and/or other programmable data processing apparatus, create means (functionality) and/or structure for implementing the functions/acts specified in the block diagrams and/or flowchart block or blocks. These computer program instructions may also be stored in a tangible or non-transitory computer-readable storage medium that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instructions which implement the functions/acts specified in the block diagrams and/or flowchart block or blocks. The computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer-implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions/acts specified in the block diagrams and/or flowchart block or blocks. Accordingly, example embodiments may be implemented in hardware and/or in software (including firmware, resident software, micro-code, etc.). Furthermore, example embodiments may take the form of a computer program product on a computer-usable or computer-readable storage medium having tangible, non-transitory computer-usable or computer-readable program code embodied in the medium for use by or in connection with an instruction execution system. In the context of this document, a computer-usable or computer-readable medium may be any medium that can contain, store, communicate, propagate, or transport the program for use by or in connection with the instruction execution system, apparatus, or device. The terms “tangible” and “non-transitory,” as used herein, are intended to describe a computer-readable storage medium (or “memory”) excluding propagating electromagnetic signals but are not intended to otherwise limit the type of physical computer-readable storage device that is encompassed by the phrase computer-readable medium or memory. For instance, the terms “non-transitory computer readable medium” or “tangible memory” are intended to encompass types of storage devices that do not necessarily store information permanently, including for example, random access memory (RAM). Program instructions and data stored on a tangible computer-accessible storage medium in non-transitory form may further be transmitted by transmission media or signals such as electrical, electromagnetic, or digital signals, which nay be conveyed via a communication medium such as a network and/or a wireless link. In the drawings and specification, there have been disclosed typical preferred embodiments of the invention and, although specific terms are employed, they are used in a generic and descriptive sense only and not for purposes of limitation, the scope of the invention being set forth in the following claims. | 15,702 |
11863092 | DETAILED DESCRIPTION Hereinafter, an embodiment for carrying out the present invention will be described in detail with reference to the accompanying drawings. FIG.1is a block diagram illustrating a configuration example of a motor control system using a control device according to an embodiment of the present invention. The motor control system includes: a motor2to be controlled that drives a load1; a host device3that generates a host torque command value for the motor2; a motor control unit4that controls driving of the motor according to a host torque command value Tmfrom the host device3and a state of the motor2; and a power conversion unit5that supplies a drive current according to control of the motor control unit4to the motor2. The motor2drives, for example, an electric vehicle, and is configured using, for example, a three-phase brushless motor. In addition, for example, in the case of the electric vehicle, the host device3is configured using a vehicle control unit (VCU) or the like that generates the host torque command value according to an accelerator opening degree, a current vehicle speed, or the like. Note that the motor2may be a motor or the like that drives an armor the like of a robot. In addition, the motor2may be configured using another motor such as a brushed DC motor. In addition, the power conversion unit5is configured using, for example, an inverter, and performs switching of a power supply voltage according to current control from the motor control unit4to supply the power supply voltage to the motor2. FIG.2is a block diagram illustrating a configuration example of the motor control unit4. The motor2includes a field coil that generates a magnetic field according to the drive current supplied from the power conversion unit5, a permanent magnet attached to a rotor, and the like, and generates a driving force according to the magnetic field generated by the field coil. In addition, the motor2includes a position detection unit2A that detects a position (angle) of the rotor. The motor control unit4includes: a feedforward unit10that obtains a feedforward torque command value TmFFaccording to the host torque command value Tmfrom the host device3; a speed calculation unit20that obtains an angular velocity ωmof the motor2from a detection output of the position detection unit2A; a feedback unit30that obtains a feedback torque command value TmFBaccording to the angular velocity ωmof the motor2; and a calculation unit40that obtains a torque command value Tmres. In addition, the motor control unit4includes: a vibration component removal unit (NF: Notch Filter)50that removes a vibration component in a predetermined band of the angular velocity ωmobtained by the speed calculation unit20; and a current control unit60that controls the drive current to be supplied to the motor2according to the torque command value Tmresand the like. In this motor control system, the current control unit60performs control based on the torque command value Tmres, obtained from the torque command value TmFFfrom the feedforward unit10and the torque command value TmFBfrom the feedback unit30, thereby performing non-interference control to suppress the vibration of the motor2. The feedforward unit10includes: a noise removal unit (NF: Notch Filter)11that removes a component in a predetermined band of the host torque command value Tmfrom the host device3; and an amplifier12. In addition, the feedback unit30further includes: a filter (HPF: High Pass Filter)31that passes components in the predetermined band or higher of the angular velocity ωmof the motor2obtained by the speed calculation unit20; and an amplifier32. For example, the NF50is configured using a notch filter having a characteristic of removing a signal in a specific band as illustrated inFIG.3. The NF50removes a predetermined vibration component from the angular velocity ωmobtained by the speed calculation unit20. Specifically, the NF50removes vibration components of bands fr−fwto fr+fwhaving a predetermined width (2fw) centered on a resonant frequency frof a mechanical section, which includes the motor2and the load1, from the angular velocity ωmobtained by the speed calculation unit20. The current control unit60includes: a current command calculation unit61that obtains a current command value igaccording to the torque command value Tmresand the angular velocity ωmof the motor2from which the vibration component in the predetermined band has been removed by the NF50; and a current control calculation unit62that controls the drive current for driving the motor2according to the current command value igand the angular velocity ωmof the motor2obtained by the speed calculation unit20. The current control calculation unit62performs pulse width modulation (PWM) control of the drive current by controlling the power conversion unit5according to the current command value ig, for example. In the motor control system configured as described above, when the host torque command value Tmis supplied from the host device3, the motor control unit4controls driving of the motor2according to the host torque command value Tmand the angular velocity ωmof the motor2. Specifically, the feedforward unit10obtains the feedforward torque command value TmFFaccording to the host torque command value Tmfrom the host device3, and supplies the feedforward torque command value TmFFto the calculation unit40. The speed calculation unit20obtains the angular velocity ωmof the motor2from a detection output of the position detection unit2A, and supplies the angular velocity ωmto the feedback unit30and the like. The feedback unit30obtains the feedback torque command value TmFBfrom the angular velocity ωmof the motor2and supplies the feedback torque command value TmFBto the calculation unit40. The calculation unit40supplies a difference between the feedforward torque command value TmFFand the feedback torque command value TmFBto the current command calculation unit61as the torque command value Tmres. The NF50removes a predetermined vibration component from the angular velocity ωmobtained by the speed calculation unit20and supplies the resultant to the current command calculation unit61. The current command calculation unit61obtains the current command value igfrom the torque command value Tmressupplied from the calculation unit40, the angular velocity ωmobtained by removing the predetermined vibration component and supplied via the NF50, and a power supply voltage value VDC, and supplies the current command value igto the current control calculation unit62. The current control calculation unit62controls the power conversion unit5according to the current command value igsupplied from the current command calculation unit61. Specifically, the current control calculation unit62controls a timing at which the power conversion unit5performs switching according to a current value irof the current supplied to the power conversion unit5, the current command value igfrom the current command calculation unit61, and the angular velocity ωmfrom the speed calculation unit20. As a result, the drive control according to the host torque command value Tmfrom the host device3and the angular velocity ωmof the motor2is executed. FIG.4is a view illustrating an example of the angular velocity ωmof the motor2obtained by the speed calculation unit20. As illustrated inFIG.4, the angular velocity ωmof the motor2includes a vibration component. If the current command value igis obtained by the current command calculation unit61directly using the angular velocity ωm, the current command value igalso includes a vibration component. In particular, if a component having a frequency close to the resonant frequency frof the mechanical section including the motor2, the load1, and the like is included, a strong vibration occurs due to resonance. For this reason, the vibration component is removed from the angular velocity ωmof the motor2by the NF50in this motor control system. As a result, the angular velocity ωmsupplied from the NF50to the current command calculation unit61is the angular velocity ωmfrom which the vibration component has been removed, for example, as illustrated inFIG.5. Since the current command value igis obtained by the current command calculation unit61using such an angular velocity ωm, it is possible to suppress the vibration in the drive control of the motor without impairing control responsiveness. In addition, in the motor control system, the current control calculation unit62performs current control for driving the motor2according to the current command value igobtained by the current command calculation unit61and the angular velocity ωmof the motor2obtained by the speed calculation unit20. In this manner, the control based on the actual angular velocity ωmof the motor2can be performed in the current control in the current control calculation unit62by using the angular velocity ωmof the motor2obtained by the speed calculation unit20, that is, the actual angular velocity ωmof the motor2from which the vibration component is not removed. As a result, it is possible to suppress generation of a vibration caused by inconsistency with the actual angular velocity ωmof the motor2, and to efficiently suppress the vibration. As described above, it is possible to remove the vibration component in the calculation of the current command value and to suppress the vibration in the drive control of the motor without impairing the control responsiveness by removing the vibration component from the angular velocity of the motor used for the non-interference control according to the present embodiment. In addition, the vibration component is removed from the angular velocity by the notch filter in the present embodiment, and thus, it is possible to reduce a delay in response caused by removal processing as compared with a case where a vibration component is removed using a moving average, a low-pass filter, or the like. In addition, the motor angular velocity used for the current control calculation directly uses a detection value of the motor angular velocity in the present embodiment. That is, the motor angular velocity used for the non-interference control can avoid inconsistency with the actual angular velocity of the motor, the accuracy of the non-interference control can be improved, and the vibration can be efficiently suppressed. The above-described embodiment is an example as a means for implementing the present invention, and should be appropriately modified or changed according to a configuration of a device or a system to which the present invention is applied or various conditions, and the present invention is not limited to the above-described embodiment. Features of the above-described preferred embodiments and the modifications thereof may be combined appropriately as long as no conflict arises. While preferred embodiments of the present disclosure have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the present disclosure. The scope of the present disclosure, therefore, is to be determined solely by the following claims. | 11,259 |
11863093 | DETAILED DESCRIPTION Example embodiments according to the present disclosure will be described in detail below with reference to the accompanying drawings.FIG.1is a block diagram illustrating an overall configuration of motor control devices according to the example embodiments of the present disclosure. A motor control device1illustrated inFIG.1includes a motor controller10that is configured or programmed to function as a drive controller of an electric motor15. The electric motor15is, for example, a three-phase brushless DC motor. The motor controller10is configured or programmed to include a torque estimation calculator30, a current command value calculator12, a PWM signal generator21, an external battery BT, an inverter23, and the like. The inverter23is a motor drive circuit that generates alternating current from power supplied from the battery BT through a power supply relay24. The alternating current is used for driving the electric motor15. The power supply relay24is configured to be able to cut off power supplied from the battery BT, and further can be configured as a semiconductor relay. The PWM signal generator21generates, based on voltage command values to be described later, ON/OFF control signals (PWM signals) for a plurality of semiconductor switching elements (FET1to FET6) constituting the inverter23. The semiconductor switching elements correspond to respective phases (phase a, phase b, phase c) of the electric motor15. The switching element (FET) is also called a power element. Examples of the switching element to be used include a metal-oxide semiconductor field-effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), and the like. The inverter23serving as the motor drive circuit supplies motor drive currents to the electric motor15. The motor drive currents are detected by a current detector25including current sensors (not illustrated) disposed for the respective phases. For example, the current detector25detects direct current flowing through a shunt resistor for motor drive current detection, using an amplifier circuit including an operational amplifier and the like. The current detector25provides output signals (current detection signals) to an analog-to-digital converter (ADC)27. The ADC27converts analog current values into digital values by an A/D conversion function to thereby obtain three-phase currents ia, ib, and ic. The three-phase currents ia, ib, and ic are input to a coordinate transformer28. The coordinate transformer28has a function of transformation from three phases to two phases, and calculates a current id on a d-axis and a current iq on a q-axis based on a rotation angle θ detected by a rotation angle sensor29and the three-phase currents ia, ib, and ic. That is, the coordinate transformer28calculates a d-axis current and a q-axis current based on an actual current. The torque estimation calculator30includes a first torque estimator33, a second torque estimator35, a torque weight adjuster (weighting adjuster)37, and the like. The torque weight adjuster37adjusts weightings respectively applied to a first torque estimation value output from the first torque estimator33and a second torque estimation value output from the second torque estimator35. The torque weight adjuster37adjusts a torque estimation value {circumflex over (T)} (T hat) of the electric motor15according to Expression (3) below. In Expression (3), T1represents the first torque estimation value, T2represents the second torque estimation value, and α (0≤α≤1) represents a weighting adjustment coefficient. [Mathematical Expression 3] {circumflex over (T)}=(1−α)T1+αT2(3) The first torque estimation value T1is expressed by Expression (4) below. Expression (4) is an arithmetic expression according to the cross product method. T1=Npp{Φiq+(Ld−Lq)idiq} (4) In Expression (4), Npprepresents the number of motor pole pairs, Ldand Lqrepresent motor inductances in a rotation vector coordinate system, Φ represents a coil interlinkage flux, and idand iqrepresent motor currents. The second torque estimation value T2is expressed by Expression (5) below. Expression (5) is an arithmetic expression according to an energy method, P represents motor input power, and ωmrepresents a motor rotational speed. T2=P/ωm(5) The motor input power P is estimated by an input power estimator31using Expression (6) below. P=vdid+vqiq−Pc−Pi(6) In Expression (6), vdand vqrepresent voltages applied to the motor in the rotation vector coordinate system, Pcrepresents a copper loss, and Pirepresents a loss (such as iron loss or shaft loss) other than the copper loss. The copper loss Pcis estimated by a copper loss estimator, not illustrated, using Expression (7) below. Pc=R(id2+iq2) (7) In Expression (7), R represents a coil resistance value of the electric motor15. Thus, accuracy of the motor torque estimation can be increased based on a result of estimating the motor input power and the copper loss. Note that, for the coil resistance value of the electric motor15, a means for detecting a coil temperature may be provided, and a coil resistance corrector41may correct a coil resistance using a coil temperature (Temp_coil) derived from the means for detecting a coil temperature. The coil resistance is thus corrected based on the coil temperature of the motor, and therefore, the accuracy of the motor torque estimation can be increased. Furthermore, a means for detecting a magnet temperature of a magnet may be provided in a rotor of the electric motor15, and a coil interlinkage flux corrector43may correct a coil interlinkage flux of the electric motor15using a magnet temperature (Temp_mag) derived from the means for detecting a magnet temperature. The coil interlinkage flux is thus corrected based on the magnet temperature of the motor, and therefore, the accuracy of the motor torque estimation can be increased. A description will now be given of a weighting adjustment coefficient in the torque estimation of the motor control device according to the present example embodiment. In Expression (3) (also referred to as a torque estimation equation) described above, the weighting adjustment coefficient α is a coefficient for adjusting a proportion between the first torque estimation value T1and the second torque estimation value T2, and is also called a torque weight. Based on a predetermined input, a coefficient calculator38calculates the weighting adjustment coefficient α according to some methods. For example, as a first method, as shown in Expression (8) below, a weighting coefficient α is calculated based on a ratio (proportion) of an output of the electric motor15. [MathematicalExpression4]α=T^ωmTmaxωm_max(8) In Expression (8), Tmaxrepresents a maximum value of a motor output, and Wm_maxrepresents a maximum value of the motor rotational speed. In Expression (3) for determining the torque estimation value {circumflex over (T)} (T hat), according to Expression (8), the torque weight adjuster37adjusts the weightings respectively applied to the first torque estimation value T1and the second torque estimation value T2, using the weighting coefficient α calculated based on output power of the electric motor15. Thus, a motor torque estimation can be performed in an entire region from a low-speed region to a high-speed region. As a second method, as shown in Expression (9) below, the coefficient calculator38calculates a weighting coefficient α based on a ratio of a rotational speed of the electric motor15. [MathematicalExpression5]α=ωmωm_max(9) In Expression (3) for determining the torque estimation value {circumflex over (T)} (T hat), the torque weight adjuster37adjusts the weightings respectively applied to the first torque estimation value T1and the second torque estimation value T2, using the weighting coefficient α calculated based on the rotational speed of the electric motor15. According to Expression (3), at a low motor rotational speed, the weighting coefficient α is adjusted to be small and a torque estimation utilizing the features of the cross product method can be performed, and at a high motor rotational speed, the weighting coefficient α is adjusted to be large and a torque estimation utilizing the features of the energy method can be performed. Thus, the weightings are adjusted using the weighting coefficient α calculated based on the rotational speed of the electric motor, and therefore, the motor torque estimation can be performed in the entire region from the low-speed region to the high-speed region. In a third method for calculating the weighting adjustment coefficient α, using Expression (10) below, a weighting coefficient α is calculated based on a ratio of a current flowing through a coil of the electric motor15. In Expression (10), imaxrepresents a maximum value of a motor current. [MathematicalExpression6]α=iimax(10) The torque weight adjuster37calculates a weighting coefficient α based on the current flowing through the coil of the electric motor15, and adjusts, using the resultant weighting coefficient α, weightings respectively applied to the first torque estimation value T1and the second torque estimation value T2. The weightings are adjusted using the weighting coefficient α based on the current flowing through the coil of the electric motor, and therefore, the motor torque estimation can be performed in the entire region from the low-speed region to the high-speed region. The current command value calculator12determines a difference (torque deviation) between instruction torque Tq which is externally input and the torque estimation value {circumflex over (T)} (T hat) adjusted by the torque weight adjuster37, and performs proportional-plus-integral control (PI control) on the resultant difference. The current command value calculator12performs predetermined current command calculation based on a torque value obtained by the PI control to determine a d-axis command current idrefwhich is a magnetic field component and a q-axis command current iqrefwhich is a torque component. A subtractor13acalculates a difference (denoted as Dq) between the q-axis command current iqrefand the q-axis current iq. A subtractor13bcalculates a difference (denoted as Dd) between the d-axis command current idrefand the d-axis current id. The difference Dq is input to a PI controller16a, and the difference Dd is input to a PI controller16b. The PI controller16aperforms PI (proportional-plus-integral) control for converging Dq to zero, and calculates a q-axis voltage command value vq that is a command value of a q-axis voltage. The PI controller16bperforms PI (proportional-plus-integral) control for converging Dd to zero, and thereby calculates a d-axis voltage command value vd that is a command value of a d-axis voltage. Thus, the PI controllers16aand16bserving as current controllers determine the d-axis and q-axis voltage command values so as to set a difference between the d-axis and q-axis current command values and a detected current value to zero. A coordinate transformer17calculates voltages applied to the motor, based on the d-axis and q-axis voltage command values vq and vd and the rotation angle of the electric motor15. That is, the q-axis voltage command value vq and the d-axis voltage command value vd are input to the coordinate transformer17having a function of transformation from two phases to three phases, and the coordinate transformer17converts, based on the rotation angle θ, the values vq and vd into voltage command values va*, vb*, and vc* which are respective voltage command values of the three phases. The voltage command values va*, vb*, and vc* thus converted are input to the PWM signal generator21. The PWM signal generator21generates drive signals (PWM signals) for the electric motor15based on the current command values. A description will now be given of a method for driving and controlling the electric motor by the motor control device according to the present example embodiment.FIG.2is a flowchart illustrating drive and control (operation examples) of the electric motor by the motor control device according to the present example embodiment. In step S11ofFIG.2, an angular velocity co of the electric motor15is calculated based on an electrical angle (rotation angle) θ detected by the rotation angle sensor29. In step S13, the motor current is detected. Here, as described above, the current detection signals from the current detector25are subjected to the A/D conversion in the ADC27to obtain the three-phase currents ia, ib, and icas digital values. In step15, three-phase to two-phase transformation and rotating coordinate transformation are performed in the coordinate transformer28to calculate a current idon a d-axis and a current iqon a q-axis (which are feedback currents) based on the rotation angle θ detected in step S11and the three-phase currents ia, ib, and icobtained in step S13. In step S17, the input power estimator31estimates the motor input power P using Expression (6) above. In subsequent step S19, the first torque estimator33calculates the first torque estimation value T1according to Expression (4) above. In step S21, the second torque estimator35calculates the second torque estimation value T2using Expression (5) above. That is, in step S19, the first torque estimation value is determined based on the coil interlinkage flux, the motor current, and the like, and in step S21, the second torque estimation value is determined based on the motor input power, the motor rotational speed, and the like. In step S23, the coefficient calculator38calculates the weighting adjustment coefficient α. The coefficient calculator38calculates the weighting adjustment coefficient α using any one of the above-described first to third methods. Here, for example, any one of the first to third methods may be fixed for use, or may be appropriately selected based on, for example, a driving state of the electric motor15. In step S25, according to Expression (3) above, using the weighting adjustment coefficient α calculated in step S23, the torque weight adjuster37adjusts the torque weights respectively applied to the first torque estimation value T1obtained in step S19and the second torque estimation value T2obtained in step S21, and the torque weight adjuster37calculates the torque estimation value {circumflex over (T)} (T hat). In the torque weight adjustment in step S25, for example, when the coefficient shown in Expression (9) above is used as the weighting adjustment coefficient α, the proportion of the torque estimation value T2according to the energy method is increased in the torque estimation value {circumflex over (T)} (T hat) obtained in Expression (3) as the motor output increases. As a result, a torque estimation error can be reduced, which is caused by magnetic saturation in a high-output region of the motor. When the coefficient shown in Expression (8) above or Expression (9) above is used as the weighting adjustment coefficient α, in the calculation of the torque estimation value according to Expression (3), the motor rotational speed corn can be eliminated by multiplication, division using the motor rotational speed can be avoided to simplify the calculation, and the torque estimation can be performed in the entire region. In step S27, q-axis and d-axis current command values and q-axis and d-axis voltage command values are calculated. Specifically, in the current command value calculator12, current command calculation is performed based on a difference between the instruction torque Tq and the torque estimation value {circumflex over (T)} (T hat) to calculate q-axis and d-axis current command values. After the d-axis and q-axis command currents are calculated, PI control is performed on a difference between the q-axis command current iqrefand the q-axis current iq to calculate a q-axis voltage command value vq which is a command value of the q-axis voltage. Furthermore, PI control is performed on a difference between the d-axis command current idrefand the d-axis current id to calculate a d-axis voltage command value vd which is a command value of the d-axis voltage. In step S29, the coordinate transformer17performs two-phase to three-phase transformation to determine voltage command values va*, vb*, and vc* which are respective voltage command values of the three phases, based on the q-axis voltage command value vq and the d-axis voltage command value vd which have been calculated in step S27, and the rotation angle θ. In step S31, the respective voltage command values va*, vb*, and vc* of the three phases determined in step S29are input to the PWM signal generator21. The PWM signal generator21generates drive signals (PWM signals) for the electric motor15based on the current command values. A description will now be given of a torque weight adjustment effect in the motor control device according to the present example embodiment.FIG.3illustrates an operating point based on the torque weight adjustment effect. Here, as illustrated inFIG.4, it is assumed that the electric motor accelerates from a stopped state and rotates at a constant rotational speed of 500 rpm. FIG.5illustrates an adjusted torque weight. The torque weight inFIG.5is an example of weighting using the weighting coefficient α calculated according to Expression (8) above. Then, the torque weight adjustment effect will be described by comparing the conventional torque weight adjustment with the torque weight adjustment in the motor control device according to the present example embodiment.FIG.6illustrates a result of a conventional example in which torque is estimated according to the above-described cross product method, andFIG.7illustrates a result of a conventional example in which torque is estimated according to the energy method. Meanwhile,FIG.8illustrates a result of a torque estimation in the motor control device according to the present example embodiment. As can be seen from comparison betweenFIG.6andFIG.8, a torque estimation error is greatly smaller in the torque estimation in the motor control device according to the present example embodiment than in the conventional torque estimation according to the cross product method. As can be seen from comparison betweenFIG.7andFIG.8, a torque estimation error at zero speed (0.5 sec) is smaller in the torque estimation in the motor control device according to the present example embodiment than in the conventional torque estimation according to the energy method. The reason therefor is as follows. In the conventional torque estimation according to the energy method illustrated inFIG.7, the estimation error rapidly increases when the motor starts to rotate (at substantially zero speed) as indicated by a sign A, and the torque estimation becomes unstable. In the case of the torque estimation in the motor control device according to the present example embodiment, the estimation error does not occur even when the motor starts to rotate as illustrated inFIG.8. The motor control device according to the present example embodiment can be mounted on, for example, an electric pump, home electrical appliances, various industrial devices, an electric power steering device, and the like. For example, when the motor control device according to the present example embodiment is mounted on an electric power steering device, a torque estimation error is reduced in motor drive and control by the motor control device, and a steering torque during steering assistance can be accurately estimated. At the same time, the torque generated from the electric motor assists rotation of a rotation shaft connected to a steering wheel, thereby assisting a driver in steering. Furthermore, the electric power steering device can be mounted on an electric power steering system. Even in this case, the torque estimation error is reduced in the motor drive and control by the motor control device, and the steering torque can be accurately estimated during steering assistance of the electric power steering system. In addition, the motor control device according to the present example embodiment can be mounted on a vehicle, such as an electric vehicle (EV) and a hybrid vehicle, using an electric motor as a drive source. In this case, it is possible to reduce the torque estimation error of the electric motor serving as a power source during both low-speed traveling and high-speed traveling of the vehicle. As described above, in the motor control device according to the present example embodiment, the weightings are adjusted using the weighting coefficient α calculated in accordance with the predetermined condition, the weightings being respectively applied to the two types of torque estimation values: the first torque estimation value estimated according to the cross product method in the first torque estimator; and the second torque estimation value estimated according to the energy method in the second torque estimator. With the combined use of such two different types of torque estimation methods, the weightings respectively applied to the calculated torque estimation values are adjusted, and therefore, a more accurate motor torque estimation value can be determined in both operation regions of a low-output region and a high-output region of the electric motor. As a result, in the motor control device, for example, a torque estimation utilizing the features of the cross product method is performed in driving the motor at low output (at a low speed), and a torque estimation utilizing the features of the energy method is performed in driving the motor at high output (at a high speed), and therefore, the torque estimation can be performed utilizing the features of each of the cross product method and the energy method. Thus, in the motor control device, the inverter circuit is controlled using the current command values calculated based on the torque estimation values and the pulse width modulation (PWM) signals generated based on the voltage command values, and therefore, inverter control of the electric motor can be performed using the torque estimation values with increased accuracy in estimation. Features of the above-described preferred example embodiments and the modifications thereof may be combined appropriately as long as no conflict arises. Additionally, a part or whole of the motor control device1and/or the functional units or blocks thereof as described above with respect to the various preferred embodiments of the present invention be implemented in one or more circuits or circuitry, such as an integrated circuit(s) or as an LSI (large scale integration). Each functional unit or block of the motor control device1may be individually made into an integrated circuit chip. Alternatively, part or whole of the functional units or blocks may be integrated and made into an integrated circuit chip. Additionally, the method of forming a circuit or circuitry defining the motor control device1is not limited to LSI, and an integrated circuit may be implemented by a dedicated circuit or a general-purpose processor or controller that is specifically programmed to define a special-purpose processor or controller. Further, if technology of forming an integrated circuit, which replaces LSI, arises as a result of advances in semiconductor technology, an integrated circuit formed by that technology may be used. Furthermore, a program which is operated in the motor control device1and/or other elements of various preferred embodiments of the present invention, is a program (program causing a computer to perform a function or functions) controlling a CPU, Control Unit, Controller, Control Circuit, Processor, Microprocessor, Processor Circuit, etc. in order to realize functions of the various preferred embodiments according to the present invention, including each of the various circuits or circuitry described herein and recited in the claims. Therefore, information which is handled by the motor control device1is temporarily accumulated in a RAM at the time of the processing. Thereafter, the information is stored in various types of circuitry in the form of ROMs and HDDs, and is read out by circuitry within, or included in combination with, the motor control device1as necessary, and modification or write-in is performed thereto. As a recording medium storing the program, any one of a semiconductor medium (for example, the ROM, a nonvolatile memory card or the like), an optical recording medium (for example, a DVD, an MO, an MD, a CD, a BD or the like), and a magnetic recording medium (for example, a magnetic tape, a flexible disc or the like) may be used. Moreover, by executing the loaded program, the functions of the various preferred embodiments of the present invention are not only realized, but the functions of preferred embodiments of the present invention may be realized by processing the loaded program in combination with an operating system or other application programs, based on an instruction of the program. While example embodiments of the present disclosure have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the present disclosure. The scope of the present disclosure, therefore, is to be determined solely by the following claims. | 25,493 |
11863094 | DETAILED DESCRIPTION Hereinafter, example embodiments of the present disclosure will be described in detail with reference to the attached drawings to allow those skilled in the art to easily execute the same. The present disclosure may, however, be implemented in many different forms and should not be construed as being limited to the example embodiments described herein. In addition, parts irrelevant to the description will be omitted in the drawings in order to clearly explain the present disclosure. Similar parts are denoted by similar reference numerals throughout the specification. Throughout the specification, when one part is referred to as being “connected” to another part, it should be understood that the former may be “directly connected” to the latter or “electrically connected” to the latter via an intervening part. Further, when a part is referred to as “including” a component, this means that the part may include another component, and does not exclude another component unless specifically stated otherwise. FIG.1is a diagram illustrating a configuration of a motor control system including a motor control apparatus according to the present disclosure. A control apparatus100ofFIG.1is an apparatus for controlling an operation of a motor200in a motor control system1000configured to control the driving of the motor200. The motor200whose operation is controlled by the control apparatus100refers to a three-phase motor having a stator and a rotor, and alternating current (AC) power of a predetermined frequency is applied to a coil of the stator of each phase of three phases so that the rotor rotates. For example, the motor may be one of a surface-mounted permanent magnet synchronous motor (SMPMSM), an interior permanent magnet synchronous motor (IPMSM), and a synchronous reluctance motor (Synrm). The motor control apparatus100may be an apparatus for controlling the driving of the motor200by supplying driving power to the motor200. In addition, the control apparatus100may be an apparatus for controlling the operation of the motor200to control the driving of a compressor including the motor200. The control apparatus100may be an apparatus for controlling the motor200using an inverter method. That is, the control apparatus100may be an inverter for controlling the driving of the motor200, or an apparatus including the inverter. The control apparatus100may control the operation of the motor200by controlling the inverter in relation to the driving of the motor200. Meanwhile, the control apparatus100may be an apparatus for controlling the operation of the motor200using a sensorless method. As shown inFIG.1, the control apparatus100includes an inverter part110configured to convert direct current (DC) power into AC power and output the AC power to the motor200, and a control part120configured to control the inverter part110in relation to the driving of the motor200. In the control apparatus100including the inverter part110and the control part120, the control part120controls the inverter part110such that a first input voltage for estimating resistance of the stator of the motor200is applied before a second input voltage for estimating a position of the rotor of the motor200, in response to a start-up of the driving of the motor200. Further, the control part120controls the driving of the motor200on the basis of the resistance of the stator estimated according to the result of applying the first input voltage. Here, the first input voltage may include a third input voltage and a fourth input voltage. In this case, a maximum value of the third input voltage and a maximum value of the fourth input voltage may be different, and angles of composite magnetic fluxes of the motor generated due to the third input voltage and the fourth input voltage may be the same. Further, the control part120may control the inverter part110to apply the fourth input voltage when a time which exceeds a first threshold time has passed from a point at which a response current for the third input voltage is 0 [A]. Meanwhile, a resistance value of the stator may be estimated in a section in which the response current for each of the third input voltage and the fourth input voltage is less than 0 [A]. Meanwhile, the third input voltage and the fourth input voltage may each have a sine wave form. Further, the control part120may control the inverter part110not to apply the first input voltage before a time which exceeds a second threshold time has passed after receiving a stop command from the motor200. Meanwhile, the resistance value of the stator may be newly estimated when the time which exceeds the second threshold time has passed after receiving the stop command of the motor200. Further, the resistance value of the stator may not be updated even when the first input voltage is applied before the time which exceeds the second threshold time has passed after receiving the stop command of the motor200. Meanwhile, the first input voltage may be a voltage in which a compensation voltage is added to a command voltage in order to correct an error between the command voltage and a voltage output to the motor200. At this time, when the inverter part110includes a plurality of inverters, the compensation voltage may be changed according to a sum of voltage errors when power is applied to the plurality of inverters. That is, when the motor200starts operating, the control apparatus100estimates the position of the rotor after estimating the resistance of the stator without aligning the position of the rotor of the motor200. As such, since the position of the rotor is not aligned before the motor200starts operating, a time taken to align the position of the rotor may be reduced. In addition, a signal for estimating the resistance of the stator and a signal for estimating the position of the rotor are separated, and the signal for estimating the resistance is applied before the signal for estimating the position, so that the accuracy of the resistance estimation may be improved. That is, since the resistance of the stator is estimated before estimating the position of the rotor, an influence due to the signal for estimating the position may be minimized. FIGS.2A and2Bare diagrams illustrating an example of a current applied in response to a start-up of driving of a motor in the related art. In the related art sensorless control of a motor of a washing machine, an initial position alignment may be carried out to stabilize initial starting characteristics and detection performance. In the initial position alignment operation, as shown inFIG.2, a DC is applied to align a position of the motor to a specific position, and then the motor is driven, and by using voltage and current information at this time, a stator resistance RS, which is an essential parameter for sensorless control, is detected. Referring toFIGS.2A and2B, section A may be a section in which the position alignment of the rotor is carried out, and section B may be a section in which swing start-up of the motor is carried out. In addition, the starting of the motor may begin from section C. Referring toFIG.2A, it can be seen that, in section A, a motor speed is close to zero because it is before starting the motor, but when it reaches section C after passing section B, the motor speed increases. Meanwhile, the current applied to the motor control apparatus in section A may be the same as that shown inFIG.2B. In the related art, the resistance of the stator was simultaneously estimated while aligning the position of the rotor. However, there is a problem in that a time (0to Ti) taken to align the position is relatively long, and the greater the inertia of an object to be driven by the motor, the longer the time taken to align the position. FIGS.3A and3Bare diagrams illustrating an example of a current applied in response to a start-up of driving of a motor according to the present disclosure. The motor control apparatus and method according to an example embodiment of the present disclosure may estimate the resistance of the stator and estimate the position of the rotor instead of performing a process of aligning the position of the rotor. Thus, the motor control apparatus may estimate the resistance of the stator and the position of the rotor in section A′ shown inFIG.3Ainstead of section A shown inFIG.2A. At this time, a current applied to the motor control apparatus and system from section A′ to section C may be the same as that shown inFIG.3B. Referring toFIGS.2A through3B, it can be seen that a time (section A′) taken to estimate the resistance of the stator and the position of the rotor is shorter than a time (section A) taken to align the position of the rotor. Accordingly, the motor control apparatus and method according to an example embodiment of the present disclosure may reduce noise and/or vibration of the motor. FIG.4is a diagram illustrating an example of a resistance estimation section and an initial position estimation section according to an example embodiment of the present disclosure. Referring toFIG.4, in a motor control apparatus and method according to an example embodiment of the present disclosure, an inverter part may be controlled such that a first input voltage410for estimating resistance and a second pattern voltage420for estimating an initial position are applied in response to a start-up of the driving of the motor. Meanwhile, when a signal for estimating the resistance and a signal for estimating the initial position are applied, rotational torque is generated by the signals so that the motor may be moved. This movement of the motor may affect the performance of estimating the resistance. When this is taken into consideration, the first input voltage410of the present disclosure may be applied before the second pattern voltage420is applied. Accordingly, the motor control apparatus and method according to an example embodiment of the present disclosure may prevent the performance of estimating the resistance from being degraded due to the second pattern voltage420. Further, in order to offset a voltage error due to non-linearity of the inverter, the first input voltage410may be applied to the same position of the rotor. Whether the first input voltage is applied to the same position of the rotor is detected by detecting an angle of a composite magnetic flux of the motor due to the first input voltage. For example, the first input voltage410may be applied at 0°, but the position at which the first input voltage410is applied is not limited thereto. FIG.5is a diagram illustrating an example of a resistance estimation section according to another example embodiment of the present disclosure. Referring toFIG.5, a first input voltage510for estimating resistance may include a third input voltage520having a maximum value of V1and a fourth input voltage730having a maximum value of V2. For example, V1may be 0.75*V2, but the relationship between V1and V2is not limited thereto. In other words, the first input voltage510may include the third input voltage520and the fourth input voltage530having different amplitudes. However, the shape of the first input voltage510is not limited thereto, and the first input voltage510may be at least one of a square wave, a half wave, a pulse, and a sine wave. FIG.6is a diagram illustrating an example of a resistance detection section included in the resistance estimation section according to an example embodiment of the present disclosure. When the first input voltage includes a third input voltage and a fourth input voltage, a first detection section610ofFIG.6indicates a section in which a response current for the third input voltage is less than 0 [A]. In addition, a second detection section620indicates a section in which a response current for the fourth input voltage is less than 0 [A]. The motor control apparatus and method according to an example embodiment may estimate the resistance of the stator in the first detection section610and the second detection section620. Meanwhile, a stator resistance RSmay be calculated on the basis of Equations 1 to 4 below. First, a motor voltage equation based on a stationary coordinate system is given by Equation 1. Vα=RsIα+LsdIαdt-ωre∅fsinθre[Equation1] Here, LSdenotes a stator inductance. Meanwhile, assuming that ωre=0, integral values for voltages detected in the first detection section610and the second detection section620may be respectively expressed as Equations 2 and 3. ∫Vα1dt=Rs∫Iα1dt+Ls∫dIα1dtdt[Equation2] Here, Vα1denotes a voltage detected in the first detection section610, and Iα1denotes a current detected in the first detection section610. ∫Vα2dt=Rs∫Iα2dt+Ls∫dIα2dtdt[Equation3] Here, Vα2denotes a voltage detected in the second detection section620, and Iα2denotes a current detected in the second detection section620. Accordingly, using Equations 2 and 3, the stator resistance RSmay be calculated as Equation 4. Rs=((∫Vα2-∫Vα1)-Ls(Iα2-Iα1))(∫Iα2-∫Iα1)[Equation4] FIG.7is a diagram for describing a method of estimating a stator resistance according to an example embodiment of the present disclosure. Referring toFIG.7, the motor control apparatus and method according to an example embodiment of the present disclosure may control a switching operation such that a fourth input voltage is applied when a time which exceeds a first threshold time has passed from a point at which a response current for a third input voltage is 0 [A] after the third input voltage has applied. Referring toFIG.7, when a third input voltage710is applied, a response current for the third input voltage710may be detected. In the motor control apparatus and method according to an example embodiment, a fourth input voltage730may be applied when a time which exceeds the first threshold time has passed from a point720at which the response current for the third input voltage710is 0 [A]. For example, the first threshold time may be 15 ms, but the first threshold time is not limited thereto. When the motor moves while the resistance is estimated, a voltage due to a counter electromotive force may be generated so that a resistance estimation error may occur. Thus, when starting the motor in a state in which the motor is not completely stopped, a signal for estimating resistance may not be applied, or a resistance value may not be newly detected even when the signal for estimating the resistance is applied. Here, the expression “the resistance value is not newly detected” may include that a previously estimated resistance value is not updated by a new resistance value even when the resistance value is newly estimated. For example, although it depends on a load amount and quantity, when a time period from a motor stop command to a motor start command is within 3.5 seconds, the resistance value may not be updated. In addition, the resistance value of the stator may not be detected or updated in a motion in which the motor starts immediately after stopping, such as a short circuit after detecting eccentricity, or cloth wetting pattern, in addition to a motion that controls the operation of the motor with a net acting ratio, such as washing or tumbling. Accordingly, the motor control apparatus and method according to an example embodiment may reduce a resistance error. FIGS.8A and8Bare diagrams for describing non-linear characteristics of the inverter and a compensation voltage according to the non-linear characteristics of the inverter. In the motor control apparatus and method according to an example embodiment of the present disclosure, a first input voltage obtained by adding a compensation voltage to a command voltage may be applied to compensate for non-linearity of the inverter included in the inverter part. When a voltage for estimating resistance is applied, non-linear characteristics of the inverter due to dead time or the like in a low voltage region may be present. Accordingly, in order to reduce an influence of the non-linear characteristics of the inverter, a voltage error occurring at the dead time may be calculated and may be added to the command voltage. As such, the voltage added to the command voltage may be defined as a compensation voltage. Referring toFIG.8A, a first voltage error represents a voltage error when power is applied to a lower inverter of the motor, and a second voltage error represents a voltage error when power is applied to an upper inverter of the motor. It can be seen that the voltage error occurs in a section except for a point at which a magnitude of a phase current is 0 [A] when the first voltage error and the second voltage error are summed. FIG.8Billustrates a compensation voltage determined in consideration of the sum of the voltage errors shown inFIG.8A. Referring toFIGS.8A and8B, it can be seen that the compensation voltage has a sign opposite to that of the voltage error. FIGS.9A and9Bare diagrams for describing a difference between a command voltage according to the compensation voltage and a voltage output to the motor, according to an example embodiment. Meanwhile, the motor control apparatus and method according to an example embodiment of the present disclosure may estimate the resistance of the stator in a detection section in which the response current for the first input voltage is less than 0 [A]. Meanwhile, the first detection section is highly likely to be similar to a first section910ofFIGS.9A and9B, and thus in the description ofFIGS.9A and9B, it is assumed that the motor control apparatus and method of the present disclosure estimate the resistance of the stator in the first section910. Meanwhile,FIG.9Ais a diagram illustrating the command voltage and an output voltage, which is a voltage output to the motor before the non-linearity of the inverter is compensated for. It can be seen that a difference occurs between the command voltage and the output voltage in the first section910because the compensation voltage is not considered. Meanwhile, after the compensation voltage is added to the command voltage to compensate for the non-linearity of the inverter, the output voltage may be the same as that inFIG.9B. Referring toFIG.9B, it can be seen that the command voltage and the output voltage are substantially similar to each other in the first section910. FIG.10is a flowchart illustrating a motor control method according to another example embodiment of the present disclosure. In operation S1010, the present disclosure may control the inverter part such that a first input voltage for estimating resistance of the stator of the motor is applied before a second input voltage for estimating a position of the rotor of the motor in response to a start-up of the driving of the motor. Here, the first input voltage may include a third input voltage and a fourth input voltage. In this case, a maximum value of the third input voltage and a maximum value of the fourth input voltage may be different, and angles of composite magnetic fluxes of the motor generated due to the third input voltage and the fourth input voltage may be the same. Meanwhile, the third input voltage and the fourth input voltage may each have a sine wave form. Operation S1010may include controlling the inverter part to apply the fourth input voltage when a time which exceeds a first threshold time has passed from a point at which a response current for the third input voltage is 0 [A]. Further, operation S1010may include controlling the inverter part not to apply the first input voltage before a time which exceeds a second threshold time has passed after receiving a stop command from the motor. In operation S1020, the present disclosure may control the driving of the motor on the basis of the resistance of the stator estimated according to the result of applying the first input voltage. Meanwhile, a resistance value of the stator may be estimated in a section in which the response current for each of the third input voltage and the fourth input voltage is less than 0 [A]. Further, the resistance value of the stator may be newly estimated when the time which exceeds the second threshold time has passed after receiving the stop command of the motor. Further, the resistance value of the stator may not be updated even when the first input voltage is applied before the time which exceeds the second threshold time has passed after receiving the stop command of the motor. Meanwhile, the first input voltage may be a voltage in which a compensation voltage is added to a command voltage in order to correct an error between the command voltage and a voltage output to the motor. Here, when the inverter part includes a plurality of inverters, the compensation voltage may be a voltage that is changed according to a sum of voltage errors when power is applied to the plurality of inverters. The above-described example embodiments of the motor control apparatus, the motor control system, and the motor control method according to the present disclosure may be applied to and implemented on a motor control apparatus provided in a motor, for example, an inverter apparatus for controlling the motor, a motor including the same, a control method for the motor, or the like. In particular, the example embodiments may be effectively applied to and implemented on a control apparatus, a control system, and a control method for controlling an initial operation of a motor, a control apparatus, a control system, and a control method for aligning a position of a motor, a control apparatus, a control system, a control method for detecting a position of a motor, or the like. In addition, the example embodiments may also be effectively applied to and implemented on a compressor control apparatus provided in a compressor including a motor, for example, an inverter apparatus for controlling a motor of a compressor, a compressor including the same, a control method for the compressor, or the like. However, techniques disclosed herein are not limited thereto, and may also be applied to and implemented on all motor control apparatuses, motor control systems and motor control methods, home appliances including the motor, control apparatuses for home appliances including the motor, and control systems and control methods for home appliances including the motor to which the technical concept of the present disclosure is applicable. Further, in the present specification, the terms “er (or) etc.” may be a hardware component, such as a processor or circuit, and/or a software component executed by the hardware configuration, such as a processor. An aspect provides a motor control apparatus, a motor control system, and a motor control method for estimating resistance of a stator before starting a motor. The technical goals to be achieved by the present example embodiments are not limited to the above-described technical aspects, and other technical aspects which are not described may be inferred from the following example embodiments. According to an aspect, there is provided a motor control apparatus including an inverter part configured to convert direct current (DC) power into alternating current (AC) power and output the AC power to a motor, and a control part configured to control the inverter part in relation to driving of the motor, and the control part may control the inverter part such that a first input voltage for estimating resistance of a stator of the motor is applied before a second input voltage for estimating a position of a rotor of the motor in response to a start-up of the driving of the motor, and control the driving of the motor on the basis of the resistance of the stator estimated according to a result of applying the first input voltage. According to another aspect, there is also provided a motor control method of a motor control apparatus including an inverter part configured to convert direct current (DC) power into alternating current (AC) power and output the AC power to a motor and a control part configured to control the inverter part in relation to driving of the motor, the method including controlling the inverter part such that a first input voltage for estimating resistance of a stator of the motor is applied before a second input voltage for estimating a position of a rotor of the motor in response to a start-up of the driving of the motor, and controlling the driving of the motor on the basis of the resistance of the stator estimated according to a result of applying the first input voltage. Specific details of other example embodiments are included in the detailed descriptions and drawings. According to an example embodiment of the present disclosure, one or more of the following effects can be achieved. First, a time for aligning an initial position of a motor can be omitted so that it is possible to drive the motor more rapidly. Second, the accuracy of detecting resistance of a stator of the motor can be improved so that it is possible to control the operation of the motor accurately and stably. Effects of the present disclosure will not be limited to the above-mentioned effects and other unmentioned effects will be clearly understood by those skilled in the art from the following claims. The above description of the present disclosure is only exemplary, and it will be understood by those skilled in the art that various modifications may be made without departing from the scope of the present disclosure and without changing essential features. Therefore, the example embodiments described above should be understood as being illustrative in all aspects instead of limiting. For example, each component described as a single entity may be distributed and implemented, and components described as being distributed may also be implemented in a combined form. The scope of the present disclosure will be defined by the following claims rather than the above detailed description, and all changes and modifications derived from the meaning and the scope of the claims and equivalents thereof should be understood as being included in the scope of the present disclosure. It will be understood that when an element or layer is referred to as being “on” another element or layer, the element or layer can be directly on another element or layer or intervening elements or layers. In contrast, when an element is referred to as being “directly on” another element or layer, there are no intervening elements or layers present. As used herein, the term “and/or” includes any and all combinations of one or more of the associated listed items. It will be understood that, although the terms first, second, third, etc., may be used herein to describe various elements, components, regions, layers and/or sections, these elements, components, regions, layers and/or sections should not be limited by these terms. These terms are only used to distinguish one element, component, region, layer or section from another region, layer or section. Thus, a first element, component, region, layer or section could be termed a second element, component, region, layer or section without departing from the teachings of the present invention. Spatially relative terms, such as “lower”, “upper” and the like, may be used herein for ease of description to describe the relationship of one element or feature to another element(s) or feature(s) as illustrated in the figures. It will be understood that the spatially relative terms are intended to encompass different orientations of the device in use or operation, in addition to the orientation depicted in the figures. For example, if the device in the figures is turned over, elements described as “lower” relative to other elements or features would then be oriented “upper” relative to the other elements or features. Thus, the exemplary term “lower” can encompass both an orientation of above and below. The device may be otherwise oriented (rotated 90 degrees or at other orientations) and the spatially relative descriptors used herein interpreted accordingly. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention. As used herein, the singular forms “a”, “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprises” and/or “comprising,” when used in this specification, specify the presence of stated features, integers, steps, operations, elements, and/or components, but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or groups thereof. Embodiments of the disclosure are described herein with reference to cross-section illustrations that are schematic illustrations of idealized embodiments (and intermediate structures) of the disclosure. As such, variations from the shapes of the illustrations as a result, for example, of manufacturing techniques and/or tolerances, are to be expected. Thus, embodiments of the disclosure should not be construed as limited to the particular shapes of regions illustrated herein but are to include deviations in shapes that result, for example, from manufacturing. Unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. It will be further understood that terms, such as those defined in commonly used dictionaries, should be interpreted as having a meaning that is consistent with their meaning in the context of the relevant art and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein. Any reference in this specification to “one embodiment,” “an embodiment,” “example embodiment,” etc., means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment of the invention. The appearances of such phrases in various places in the specification are not necessarily all referring to the same embodiment. Further, when a particular feature, structure, or characteristic is described in connection with any embodiment, it is submitted that it is within the purview of one skilled in the art to effect such feature, structure, or characteristic in connection with other ones of the embodiments. Although embodiments have been described with reference to a number of illustrative embodiments thereof, it should be understood that numerous other modifications and embodiments can be devised by those skilled in the art that will fall within the spirit and scope of the principles of this disclosure. More particularly, various variations and modifications are possible in the component parts and/or arrangements of the subject combination arrangement within the scope of the disclosure, the drawings and the appended claims. In addition to variations and modifications in the component parts and/or arrangements, alternative uses will also be apparent to those skilled in the art. | 31,294 |
11863095 | DETAILED DESCRIPTION OF THE EXEMPLARY EMBODIMENTS It is understood that the term “vehicle” or “vehicular” or other similar term as used herein is inclusive of motor vehicles in general such as passenger automobiles including sports utility vehicles (SUV), buses, trucks, various commercial vehicles, watercraft including a variety of boats and ships, aircraft, and the like, and includes hybrid vehicles, electric vehicles, plug-in hybrid electric vehicles, hydrogen-powered vehicles and other alternative fuel vehicles (e.g. fuels derived from resources other than petroleum). As referred to herein, a hybrid vehicle is a vehicle that has two or more sources of power, for example both gasoline-powered and electric-powered vehicles. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the disclosure. As used herein, the singular forms “a,” “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprises” and/or “comprising,” when used in this specification, specify the presence of stated features, integers, steps, operations, elements, and/or components, but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or groups thereof. As used herein, the term “and/or” includes any and all combinations of one or more of the associated listed items. Throughout the specification, unless explicitly described to the contrary, the word “comprise” and variations such as “comprises” or “comprising” will be understood to imply the inclusion of stated elements but not the exclusion of any other elements. In addition, the terms “unit”, “-er”, “-or”, and “module” described in the specification mean units for processing at least one function and operation, and can be implemented by hardware components or software components and combinations thereof. Further, the control logic of the present disclosure may be embodied as non-transitory computer readable media on a computer readable medium containing executable program instructions executed by a processor, controller or the like. Examples of computer readable media include, but are not limited to, ROM, RAM, compact disc (CD)-ROMs, magnetic tapes, floppy disks, flash drives, smart cards and optical data storage devices. The computer readable medium can also be distributed in network coupled computer systems so that the computer readable media is stored and executed in a distributed fashion, e.g., by a telematics server or a Controller Area Network (CAN). Hereinafter, a motor driving device according to various embodiments of the present disclosure will be described in detail with reference to the attached drawings. FIG.1is a circuit diagram illustrating a motor driving device according to an embodiment of the present disclosure. Referring toFIG.1, a motor driving device according to an embodiment of the present disclosure is a motor driving device that supplies driving power to a motor100having a plurality of windings C1to C3corresponding to a plurality of phases, and may be configured to include a first inverter10that includes a plurality of first switching elements S11to S16and is connected to first stages of the windings of the motor100, a second inverter20that includes a plurality of second switching elements S21to S26and is connected to second stages of the windings of the motor100, third switching elements N31S31to N33S33that selectively interconnect/disconnect points that divide the number of turns of the windings of the motor100by a preset ratio of N1:N2, and a controller40that controls ON/OFF states of the first switching elements S11to S16, the second switching elements S21to S26, and the third switching elements N31S31to N33S33on the basis of a required output of the motor100. The first inverter10and the second inverter20may convert direct current (DC) power stored in a battery200into three-phase alternating current (AC) power and provide the three-phase AC power to the motor100, or convert regenerative braking energy generated due to occurrence of regenerative braking torque of the motor100during regenerative braking into a direct current and provide the direct current to the battery200. The conversion between the DC power and the AC power may be performed by pulse width modulation control of the plurality of first switching elements S11to S16and the plurality of second switching elements S21to S26that are provided to the first inverter10and the second inverter20, respectively. The first inverter10may include a plurality of legs11to13to which a DC voltage formed at a DC link capacitor300connected between opposite ends of the battery200is applied. The legs11to13may correspond to the plurality of phases of the motor100to obtain electrical connection. In particular, a first leg11may include two switching elements S11and S12connected in series between opposite ends of the DC link capacitor300, and a connected node between the two switching elements S11and S12may be connected to one end of the winding C1for one phase in the motor100such that AC power corresponding to one of the plurality of phases is input/output. Likewise, a second leg12may include two switching elements S13and S14connected in series between the opposite ends of the DC link capacitor300, and a connected node between the two switching elements S13and S14may be connected to one end of the winding C2for one phase in the motor100such that AC power corresponding to one of the plurality of phases is input/output. In addition, a third leg13may include two switching elements S15and S16connected in series between the opposite ends of the DC link capacitor300, and a connected node between the two switching elements S15and S16may be connected to one end of the winding C3for one phase in the motor100such that AC power corresponding to one of the plurality of phases is input/output. The second inverter20may also have a configuration similar to that of the first inverter10. The second inverter20may include a plurality of legs21to23to which the DC voltage formed at the DC link capacitor300connected between the opposite ends of the battery200is applied. The legs21to23may correspond to the plurality of phases of the motor100to obtain electrical connection. In particular, a first leg21may include two switching elements S21and S22connected in series between the opposite ends of the DC link capacitor300, and a connected node between the two switching elements S21and S22may be connected to the second end of the winding C1for one phase in the motor100such that AC power corresponding to one of the plurality of phases is input/output. Likewise, a second leg22may include two switching elements S23and S24connected in series between the opposite ends of the DC link capacitor300, and a connected node between the two switching elements S23and S24may be connected to the second end of the winding C2for one phase in the motor100such that AC power corresponding to one of the plurality of phases is input/output. In addition, a third leg23may include two switching elements S25and S26connected in series between the opposite ends of the DC link capacitor300, and a connected node between the two switching elements S25and S26may be connected to the second end of the winding C3for one phase in the motor100such that AC power corresponding to one of the plurality of phases is input/output. The first inverter10is connected to first ends of the windings C1to C3of the motor100, and the second inverter20is connected to second ends of the windings C1to C3of the motor100. That is, open end winding type electrical connection may be formed in which the opposite ends of the windings C1to C3of the motor100are respectively connected to the first inverter10and the second inverter20. In the embodiment of the present disclosure, the third switching element30is configured to selectively interconnect/disconnect points that divide the number of turns of the plurality of windings C1to C3included in the motor100by a preset ratio of N1:N2. For example, the third switching element30may be made up of a total of three switching elements S31to S33. First ends of the switching elements S31to S33may be connected to a point at which the number of turns of the plurality of windings C1to C3is divided by a preset ratio of N1:N2, and the second ends of the switching elements S31to S33may be interconnected (here, N1 and N2 are the real number of turns). In this connection structure, in the case where the third switching element30is turned off, the motor100may be operated with the windings having the number of turns of N1+N2. In the case where the third switching element30is turned on, the windings C1to C3of the motor100form a Y-connection at a position at which the third switching element30is connected. For example, in the case where the third switching element30is turned on, all the plurality of switching elements S21to S26inside the second inverter20are turned off and are not operated, and the first inverter10is operated to drive the motor100, so that the motor100may be driven as a motor having the number of turns of N1. As the third switching elements S31to S33, various switching devices known in the art such as a metal oxide semiconductor field effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), a thyristor, and a relay may be employed. The controller40is an element that basically has pulse width modulation control over the switching elements S11to S16and S21to S26included in the first inverter10and the second inverter20such that the motor100can be driven on the basis of the required output of the motor100. Especially, in various embodiments of the present disclosure, the controller40may determine an inverter used to drive the motor on the basis of the required output of the motor100, thus determine an ON/OFF state of the third switching element30, and have pulse width modulation control over the switching elements of the inverter determined to drive the motor. In particular, if the required output of the motor100is smaller than a preset reference value, the controller40can have pulse width modulation control over the switching elements S11to S16of the first inverter10without operating the second inverter20and drive the motor100(for convenience of description, referred to as “first driving mode”). In this case, the controller40may control the third switching elements S31to S33to be in an ON state. Accordingly, the windings C1to C3of the motor100form a Y-connection for which the points at which the number of turns from first ends thereof connected to the first inverter20is N1 are mutually interconnected, and thus the motor100is operated as a motor whose windings have the number of turns of N1. In this way, in the first driving mode, the third switching element30is turned on by operating the first inverter10, and thereby control of driving the motor100whose windings have the number of turns of N1 and are Y-connected can be performed. Driving the motor in the first driving mode can be accomplished in such a way that a DC voltage of the first inverter10, a phase current detected and provided to the motor100by a current sensor50, and a motor angle detected by a motor rotor sensor (not illustrated) installed on the motor100are input into the controller40, and then the controller40has pulse width modulation control over the first switching elements S11to S16of the first inverter10. Since various techniques for having pulse width modulation control over one inverter and driving the motor100are already well-known in the art, a more detailed description of a technique for the pulse width modulation control of the inverter performed in the first driving mode will be omitted. If the required output of the motor100is greater than a preset reference value, the controller40can operate both the first inverter10and the second inverter20to drive the motor100(for convenience of description, referred to as “second driving mode”). In this case, the controller40may control the third switching elements S31to S33to be in an OFF state. Accordingly, the windings C1to C3of the motor100have the number of turns of N1+N2, first ends thereof are connected to the first inverter10, and the second ends thereof are connected to the second inverter20. That is, the motor100in the second driving mode becomes an open end winding motor in which opposite ends of the windings C1to C3are in an open state, and may be driven by having pulse width modulation control over the two inverters10and20connected to the opposite ends of the windings C1to C3. Driving the motor in the second driving mode can be accomplished in such a way that a DC voltage of the first inverter10, a DC voltage of the second inverter20, a phase current detected and provided to the motor100by the current sensor50, and a motor angle detected by the motor rotor sensor (not illustrated) installed on the motor100are input into the controller40, and then the controller40has pulse width modulation control over the first switching elements S11to S16of the first inverter10and the second switching elements S21to S26of the second inverter20. Since various techniques for having pulse width modulation control over two inverters connected to the opposite ends of the windings of the open end winding motor and driving the motor are already well-known in the art, a more detailed description of a technique for the pulse width modulation control of the inverters performed in the second driving mode will be omitted. FIG.2is a graph illustrating a motor rotational speed (RPM)-motor torque curve and a high-efficiency region related to each motor driving mode of the motor driving device according to the embodiment of the present disclosure. As described above, the motor driving device according to the embodiment of the present disclosure drives the Y-connected motor having the number of turns of N1 in the first driving mode by controlling the first inverter10, and drives the open end winding motor having the number of turns of N1+N2 in the second driving mode by controlling the first inverter10and the second inverter20. As illustrated inFIG.2, in the case where the motor100is applied to the driving of a vehicle, a major drive point of the vehicle is classified into a drive point Y1during city driving (e.g., stop and go conditions) and a drive point Y2during highway driving, and the drive points Y1and Y2are preferably included in a high-efficiency region of a motor-inverter system. In the embodiment of the present disclosure, in the case where the motor100is applied to a vehicle, the number of turns of N1 is preferably determined such that the high-efficiency region of the motor-inverter system includes the major drive points Y1and Y2in the first driving mode. Since efficiency of the motor-inverter system is determined by a voltage utilization factor of the inverter, the number of turns of N1 of the windings C1to C3of the motor100in the first driving mode is preferably determined such that a region R1where the voltage utilization factor of the inverter is more than or equal to a preset reference value by which the voltage utilization factor is excellent can be determined to be included in the major drive points Y1and Y2of the vehicle. The second driving mode is a mode for driving the open end winding motor. It is known that, in comparison with the case where a motor for Y-connected windings having the same number of turns is simply driven by one inverter, output of the inverter can be set to be high by about √{square root over (3)} times in an open end winding driving type. That is, in the case where the open end winding driving type that is for the second driving mode is applied, the number of turns of the motor can be increased by √{square root over (3)} times, and thus current output for the motor producing the same output can be reduced by √{square root over (3)} times. In this way, in the case where the open end winding driving type is applied, a current of the inverter can be reduced to increase efficiency compared to the driving mode of the Y-connected motor in order to produce the same output, and material costs can be reduced due to a reduction in amount of use of power semiconductors applied as switching elements. As illustrated inFIG.2, in the case where the second driving mode is applied, a region R2that is more than or equal to a preset reference value by which, as the output of the motor increases, a voltage utilization factor of the motor-inverter system can be determined to be excellent hardly includes the major drive points Y1and Y2. Therefore, as described above, it is preferred that the motor100is preferably operated to relatively reduce an output current of the inverter and reduce the amount of use of the power semiconductors by driving the motor100to improve efficiency in the first driving mode in a low torque region within which the major drive points Y1and Y2fall and by driving the motor100in the second driving mode in a region in which high output is required. In addition, to further improve efficiency of the first driving mode operated in the low torque region within which the major drive points Y1and Y2fall, MOSFETs formed of SiC that is a material having a relatively small switching loss are more preferably employed as the switching elements S11to S16applied to the first inverter10. In contrast, IGBTs formed of Si that is an inexpensive material are preferably employed as the switching elements S21to S26applied to the second inverter20operated in a high output region. As described above, the motor driving device according to various embodiments of the present disclosure can determine whether to divide a turn ratio of the windings of the motor on the basis of the required output of the motor, divide the number of turns of the windings in a low output region such that the major drive points of the vehicle are included in a high efficiency region of the motor-inverter system in order to improve efficiency of the motor-inverter system, and realize high torque with a low current using the whole number of turns of the windings in a high output region. Accordingly, the motor driving device according to various embodiments of the present disclosure can improve efficiency in an entire torque region to contribute to improving fuel economy of the vehicle, compared to the case where the conventional Y-connected motor is driven by one inverter. Meanwhile, in the second driving mode, or a mode in which the open end winding motor is driven by opening all the third switching elements to simultaneously operate the first inverter10and the second inverter20, the second switching elements S21to S26included in the second inverter20can be formed of, for instance, Si having a great switching loss, and thus entire motor driving efficiency is reduced. In the second driving mode, sizes of the second switching elements S21to S26should be increased to be driven with the same currents as the first switching elements S11to S16of the first inverter10which are formed of SiC, and thus an increase in material costs may be caused. Therefore, a control technique capable of reducing a switching loss caused by the second switching elements S21to S26to increase efficiency and implementing the second driving mode without increasing the sizes of the second switching elements S21to S26is required. Hereinafter, the mode in which the motor100is driven as the open end winding motor by opening all the third switching elements N31to N33to simultaneously operate the first inverter10and the second inverter20will be described in greater detail. FIG.3is a diagram illustrating voltage vectors composed by two inverters applied to an open end winding motor, andFIG.4is a diagram illustrating voltage vectors composed by two inverters when an open end winding type motor is driven. In the circuit structure illustrated inFIG.1, the first inverter10and the second inverter20may include the plurality of pairs of switching elements S11and S12, S13and S14, S15and S16, and S21and S22, S23and S24, and S25and S26of the legs corresponding to the phases of the motor, and the switching elements of each leg may be operated complementarily to each other. In the vector diagram illustrated inFIG.3, the vertices and origin of a hexagon represent voltage vectors according to states of the switching elements of each inverter. For example, a point indicated by an index A represents a (100) voltage vector of the first inverter10. Here, the (100) voltage vector means that the upper switching elements S11of the switching elements S11and S12included in the leg corresponding to a phase A of the first inverter is in an ON state and the upper switching elements S13and S15of the switching elements included in the legs corresponding to the other phases are in an OFF state. Further, the center of the hexagon corresponds to a (000) or (111) voltage vector, which means that all the upper switching elements of the legs of the inverter are in an ON or OFF state. As illustrated inFIG.3, the voltage vectors that can be composed by each inverter may be a total of eight voltage vectors by composition of the ON and OFF states of the six switching elements. If the motor100is driven in an open end winding type after the third switching element30is turned off, the motor100has a structure in which the legs of the first inverter10and the second inverter20are connected to the opposites ends of the windings corresponding to the phases of the motor100, and thus a phase voltage of each phase applied to the motor is applied by a difference between the phase voltage of the first inverter and the phase voltage of the second inverter. Further, because the first inverter10and the second inverter20are independently controlled, the voltage vectors that are actually applied to the motor may form a total of 64 voltage vectors as illustrated inFIG.4by a combination of eight voltage vectors that can be composed by the first inverter10and eight voltage vectors that can be composed by the second inverter20. Meanwhile, a common mode voltage in each of the inverters10and20may be defined as a value obtained by dividing the sum of the phase voltages of the phases by 3, and a voltage vector in which a difference between the common mode voltages of the two inverters is not zero when the open end winding type motor is driven is not preferably used when the motor is driven. This is because a flow of current generated by the difference between the common mode voltages of the two inverters10and20gives rise to a loss of power in the motor. Voltage vectors indicated by reference symbols “51” to “56” inFIG.4are voltage vectors in which the difference between the common mode voltages of the two inverters is not zero. For example, since an OS vector (a 13′ vector) inFIG.4has a switching state of the first inverter which is (100) and a switching state of the second inverter which is (010), the common mode voltage of the first inverter may be calculated like “{(Vdc/2)+0+0}/3=Vdc/6”, and the common mode voltage of the second inverter may be calculated like “{0+(Vdc/2)+0}/3=Vdc/6”. Therefore, the difference between the two common mode voltages becomes zero. In contrast, since an OG vector (a 14′ vector) inFIG.4has a switching state of the first inverter which is (100) and a switching state of the second inverter which is (011), the common mode voltage of first inverter may be calculated like “{(Vdc/2)+0+0}/3=Vdc/6”, and the common mode voltage of second inverter may be calculated like “{0+(Vdc/2)+(Vdc/2)}/3=Vdc/3”. Therefore, the difference between the two common mode voltages becomes “−Vdc/6”. Here, Vdc may be a DC input voltage of the inverter, or a voltage of the battery200. If the difference between the two common mode voltages of the two inverters is calculated in the way given by way of example above, it can be found that the voltage vectors indicated by reference symbols “51” and “56” inFIG.4are the voltage vectors in which the difference between the common mode voltages of the two inverters is not zero. FIG.5is a vector diagram excluding voltage vectors generating the difference between two common mode voltages of two inverters from the vector diagram illustrated inFIG.4. As illustrated inFIG.5, if usable composite voltage vectors excluding the voltage vectors generating the difference between the two common mode voltages of the two inverters are interconnected, they are represented by a hexagon. Six regions divided by diagonals of the hexagon may control switching states of the second inverter in the same way. For example, a first region ofFIG.5has a configuration in which a switching state of the second inverter can be implemented by 4′ or (011), and a second region has a configuration in which a switching state of the second inverter can be implemented by 5′ or (001). Further, a third region has a configuration in which a switching state of the second inverter can be implemented by 6′ or (101), and a fourth region has a configuration in which a switching state of the second inverter can be implemented by 1′ or (100). In addition, a fifth region has a configuration in which a switching state of the second inverter can be implemented by 2′ or (110), and a sixth region has a configuration in which a switching state of the second inverter can be implemented by 3′ or (010). Each of the first to sixth regions is a region corresponding to an electrical angle of 60°, and all the first to sixth regions may correspond to an electrical angle of 360°. In view of this point, the embodiment of the present disclosure is configured such that, to minimize the switching loss of the second inverter20using the switching elements formed of Si having a great switching loss, the switching elements of the second inverter according to each region are controlled to continuously maintain the switching state corresponding to the relevant region, and is configured to switch the switching elements of the first inverter10to compose desired voltage vectors. As a result, the embodiment of the present disclosure curbs the switching of the switching elements having a great switching loss to the utmost to minimize the switching loss, and thereby can improve the efficiency of the system. FIG.6is a diagram illustrating switching examples of the region-specific first and second inverters illustrated inFIG.5. Referring toFIG.6, in the case where the voltage vector is intended to be composed in the first region ofFIG.5during space vector pulse width modulation for driving the motor100, the switching elements in the second inverter INV2are switched to maintain a (011) state that is a switching state corresponding to 4′, and the switching elements in the first inverter INV1are switched to output switching states corresponding to 2, 4, and 6, and thereby a desired voltage vector may be composed. A switching duty of each phase in the first inverter may be properly adjusted according to a position of the voltage vector to be composed. Further, in the case where the voltage vector is intended to be composed in the second region ofFIG.5during the space vector pulse width modulation for driving the motor100, the switching elements in the second inverter INV2are switched to maintain a (001) state that is a switching state corresponding to 5′, and the switching elements in the first inverter INV1are switched to output switching states corresponding to 1, 3, and 5. Thereby, a desired voltage vector may be composed. In addition, in the case where the voltage vector is intended to be composed in the third region ofFIG.5during the space vector pulse width modulation for driving the motor100, the switching elements in the second inverter INV2are switched to maintain a (101) state that is a switching state corresponding to 6′, and the switching elements in the first inverter INV1are switched to output switching states corresponding to 2, 4, and 6. Thereby, a desired voltage vector may be composed. In the case where the voltage vector is intended to be composed in the fourth region ofFIG.5during the space vector pulse width modulation for driving the motor100, the switching elements in the second inverter INV2are switched to maintain a (100) state that is a switching state corresponding to 1′, and the switching elements in the first inverter INV1are switched to output switching states corresponding to 1, 3, and 5. Thereby, a desired voltage vector may be composed. In the case where the voltage vector is intended to be composed in the fifth region ofFIG.5during the space vector pulse width modulation for driving the motor100, the switching elements in the second inverter INV2are switched to maintain a (101) state that is a switching state corresponding to 2′, and the switching elements in the first inverter INV1are switched to output switching states corresponding to 2, 4, and 6. Thereby, a desired voltage vector may be composed. In the case where the voltage vector is intended to be composed in the sixth region ofFIG.5during the space vector pulse width modulation for driving the motor100, the switching elements in the second inverter INV2are switched to maintain a (101) state that is a switching state corresponding to 3′, and the switching elements in the first inverter INV1are switched to output switching states corresponding to 1, 3, and 5. Thereby, a desired voltage vector may be composed. FIG.7is a diagram illustrating switching states of switching elements in a first inverter and switching states of switching elements in a second inverter according to the embodiment of the present disclosure.FIG.8is a diagram illustrating switching states of switching elements in a first inverter and switching states of switching elements in a second inverter when a conventional open end winding type motor is driven. As illustrated inFIG.7, it can be found that switching of the second inverter is remarkably reduced in the embodiment of the present disclosure. This can be more clearly found from the switching states when the conventional open end winding type motor ofFIG.8is driven. As illustrated inFIG.8, when the conventional open end winding type motor is driven, the switching elements in the second inverter also have a high-speed switching region b2. In the case where the second inverter is implemented by switching elements having a great switching loss like the switching elements formed of Si, a switching loss caused by switching the second inverter is greatly increased, which may lead to reducing efficiency of the entire system. In contrast, the embodiment of the present disclosure can improve efficiency of the entire system by minimizing the switching of the second inverter having the switching elements having a great switching loss as represented by “b1” that is a region corresponding to “b2” ofFIG.8and driving the first inverter having the switching elements formed of, for instance, SiC having a relatively small switching loss. Although specific embodiments of the present disclosure have been described and illustrated, it will be apparent to those skilled in the art that the present disclosure car be variously modified and changed within the scope of the claims. | 31,470 |
11863096 | In the drawings, like reference numerals are sometimes used to designate like structural elements. It should also be appreciated that the depictions in the figures are diagrammatic and not to scale. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The present application relates generally to pulsed control of a wide variety of electric machines (e.g., electric motors and generators) that would otherwise be operated in a continuous manner By pulsed control, the machine is intelligently and intermittently pulsed on and off to both (1) meet operational demands while (2) improving overall efficiency. More specifically, under selected operating conditions, an electric machine is intermittently pulse-driven at more efficient energy conversion operating levels to deliver the desired average output more efficiently than would be attained by conventional continuous machine operation. Pulsed operation results in deliberate modulation of the electric machine torque; however, the modulation is managed in such a manner such that levels of noise or vibration are minimized for the intended application. For the sake of brevity, the pulsed control of a wide variety of electric machines as provided herein is described in the context of a three-phase induction electric motor in a vehicle. This explanation, however, should not be construed as limiting in any regard. On the contrary, the pulse control as described herein can be used for many types of electric machine, meaning both electric motors and generators. For instance, the machine pulsed control as described herein may be used with any type of machine regardless if AC (e.g., induction, synchronous, any number of poles, etc.) or DC (e.g., brushless, electrically excited, permanent magnet, series wound, shunt brushed, compound, etc.). In addition, pulsed control of such electric machines may be used in any application, not just limited to electric vehicles. In particular, pulsed control may be used in systems that require lower acceleration and deceleration rates than vehicle applications, such as electric motors for heating, cooling, and ventilating systems. Pulsed engine control is described in U.S. patent application Ser. No. 16/353,159 filed on Mar. 14, 2019, and U.S. Provisional Patent Application Nos. 62/644,912, filed on Mar. 19, 2018; 62/658,739, filed on Apr. 17, 2018; and 62/810,861 filed on Feb. 26, 2019. Each of the foregoing applications is incorporated herein by reference in their entirety. Three-Phase Induction Machine An induction machine includes two main components, a stationary stator and a rotating rotor. In a three-phase machine, the stator may include a three-coil winding that is excited by a three-phase AC input. When the three-phase AC input is passed through the three-phase winding, a rotating magnetic field (RMF) is generated. The rotational rate of the RMF is known as the synchronous speed (Ns) of the electric machine. The rotor is typically either a “squirrel cage” or a “wound” type rotor, both having a plurality of electrically conductive elements that are electrically shorted at their ends. In accordance with Faraday's law, the RMF induces a current within the conductive elements of the rotor. The induced current establishes an induced magnetic field, which interacts with the magnetic field produced in the stator coils. The interaction of the rotor and stator magnetic fields generates an electromagnetic force (EMF) causing the rotor rotation. This type of motor is called an induction motor because electrical current is induced on the rotor conductive elements by electromagnetic induction, as opposed to a direct electrically conductive path. Three-phase induction motors provide a number of advantages. First, they are inherently self-starting. Second, the rotational speed of the rotor is easy to control. The rotational speed of the rotor (Nr) is always slightly less than the synchronous speed (Ns). This difference is known as slip, which may be expressed in terms of a percentage: Slip %=(Ns−Nr)/NsEq. (1) The frequency of the three-phase AC power energizing the stator windings controls the RMF rotational rate and thus the synchronous frequency. In turn, the rotational speed of the rotor can be controlled based on Eq. (1) defined above. While the frequency provided to the three-phase winding controls the synchronous speed (Ns), the amplitude of the applied AC controls the output torque of the electric machine. When the amplitude is higher or lower, the output of the machine is higher or lower, respectively. Vehicle Motor Efficiency Map Referring toFIG.1, an exemplary vehicle motor efficiency map10under different load and speed conditions is illustrated. The map10plots torque (N*m) along the vertical axis as a function of motor speed (RPM) along the horizontal axis. The maximum steady-state output power is given by curve12. The area under the peak-torque/speed curve12is mapped into a plurality of regions, each labeled by an operational efficiency percentage. For the particular motor shown, the following characteristics are evident:The most efficient or “sweet-spot” region of its operating range is the operating region labeled14, which is generally in the range of 4,500-6,000 RPM with a torque output in the range of about 40-70 N*m. In region14, the energy conversion efficiency is on the order of 96%, making it the “sweet spot”, where the motor is operating in its most efficient operating range.As the motor speed increases beyond approximately 6,000+ RPM, the efficiency tends to decrease, regardless of the output torque.As the output torque increases beyond 70 N*m or falls below 40 N*m, the efficiency percentage tends to decrease from its peak, in some situations rather significantly. For example, when the motor is operating at approximately 2,000 RPM and an output torque of 100 N*m, the efficiency is approximately 86%. When torque output falls below about 30 N*m, regardless of the motor speed, the efficiency drops, approaching zero at zero load.At any particular motor speed, there will be a corresponding most efficient output torque, which is diagrammatically illustrated by a maximum efficiency curve16. The map10as illustrated was derived from an electric motor used in a 2010 Toyota Prius. Map10is for an internal permanent magnet synchronous motor. It should be understood that this map10is merely illustrative and should not be construed as limiting in any regard. A similar map can be generated for just about any electric motor, for example a 3-phase induction motor, regardless if used in a vehicle or in some other application. As can be seen from the map10, the motor is generally most efficient when operating within the speed and torque ranges of the sweet spot14. If the operating conditions can be controlled so that the motor operates a greater proportion of time at or near its sweet spot14, the overall energy conversion efficiency of the motor can be significantly improved. From a practical point of view, however, many driving situations dictate that the motor operate outside of the speed and torque ranges of the sweet spot14. In electric vehicles it is common to have no transmission and as such have a fixed ratio of the electric motor rotation rate to the wheel rotation rate. In this case, the motor speed may vary between zero, when the vehicle is stopped, to a relatively high RPM when cruising at highway speeds. The torque requirements may also vary widely based on factors such as whether the vehicle is accelerating or decelerating, going uphill, going downhill, traveling on a level surface, braking, etc. As can be seen inFIG.1, at any particular motor speed, there will be a corresponding most efficient output torque which is diagrammatically illustrated by maximum efficiency curve16. From a conceptual standpoint, when the desired motor torque is below the most efficient output torque for the current motor speed, the overall efficiency of the motor can be improved by pulsing the motor, so as to operate the motor a proportion of time at or near its sweet spot and the remainder of the time at a low or zero torque output level. The average torque thus generated is controlled by controlling the duty cycle of sweet spot operation. Referring toFIG.2, a graph20plotting total applied current to an electric motor on the vertical axis versus time on the horizontal axis is illustrated. The applied current may be the sum of the current on all three-phases for a three-phase motor. For illustrative purposes, it will be assumed that each ampere of applied current will produce 1 N*m of output torque. In this particular example, a desired motor output torque is 10 N*m, which will require 10 amps of current as represented by the dashed line22. In this example, the most efficient torque output for the motor is 50 N*m corresponding to 50 amps of applied current. During conventional operation, the motor would continuously generate 10 N*m so long as the desired torque remained at this value. With pulsed-control operation, the motor is pulsed, as represented by pulses24, to deliver 50 N*m of torque for 20% of the time. The remaining 80% of the time, the motor is off. The net output of the motor therefore meets the operational demand of 10 N*m. Since the motor operates more efficiently when it is delivering 50 N*m than when it delivers 10 N*m, the motor's overall efficiency can thus be improved by pulsing the motor using a 20% duty cycle while still meeting the average torque demand. In the above example, the duty cycle is not necessarily limited to 20%. As long as the desired motor output, does not exceed 50 N*m, the desired motor output can be met by changing the duty cycle. For instance, if the desired motor output changes to 20 N*m, the duty cycle of the motor operating at 50 N*m can be increased to 40%; if the desired motor output changes to 40 N*m, the duty cycle can be increase to 80%; if the desired motor output changes to 5 N*m, the duty cycle can be reduced to 10% and so on. Generally, pulsed motor control can potentially be used advantageously any time that the desired motor torque falls below the maximum efficiency curve16ofFIG.1. On the other hand, when the desired motor torque is at or above the maximum efficiency curve16, the motor may be operated in a conventional (continuous or non-pulsed) manner to deliver the desired torque. Pulsed operation offers opportunity for efficiency gains when the motor is required to deliver an average torque below the average torque corresponding to its maximum operating efficiency point. It should be noted that current and torque values and time scale provided inFIG.2are merely illustrative and are not intended to be limiting in any manner. In actual motor pulsing embodiments, the pulse duration used may widely vary based on the design needs of any particular system. In generally, however, the scale of the periods for each on/off cycle is expected to be on the order of 10 μsec to 0.10 seconds (i.e. pulsing at a frequency in the range of 10 to 100,000 Hz), as for example between 0.2 and 20 milliseconds (50 to 5000 Hz) as will be discussed in more detail below. Furthermore, there are a wide variety of different motors and each motor has its own unique efficiency characteristics. Further, at different motor speeds, a given motor will have a different efficiency curve. The nature of the curve may vary depending on the particular motor or a particular application. For example, the current pulses need not be flat topped as depicted inFIG.2and/or the current need not go to zero during the off periods but may be some non-zero value. Regardless of the particular curve used, however, at some proportion of the time the motor is operating is preferably at or near its highest efficiency region for a given motor speed. Efficiency Improvements with Improved Rate of Torque Rise The vast majority of current motor converters are typically designed for continuous, not pulsed operation. Such motors generally transition from the unenergized to an energized state relatively infrequently. As a result, little design effort is made in managing such transitions. To the extent any design effort is made in managing the transition, it is typically directed to achieving a smooth transition as opposed to a fast transition. The transition from the energized to energized states for most motors is therefore often rate limited (i.e., relatively not fast). The Applicant has discovered that for a motor system that regularly transitions from an unenergized motor state to peak efficiency state such as with pulsed operation, even further efficiency improvements can be realized when the transitions occur as fast as possible. With fast transitions, for example from zero torque to the peak efficiency torque, the overall average motor efficiency is improved because the motor spends less time in transition where efficiency is less than the peak. This relationship is depicted inFIG.3AandFIG.3B Referring toFIG.3A, a torque versus efficiency map for an exemplary motor operating at a fixed speed (e.g. 6000 rpms) is illustrated. In the exemplary map, a range of torque outputs from 0.0 Nm to 250 Nm is plotted along the horizontal axis, while the efficiency of the motor from 0.0 percent to 100 percent is plotted along the vertical axis. The curve26depicts the transition of the motor from zero to peak efficiency torque. During this transition, as depicted by the shaded region27, the peak efficiency torque has a much lower efficiency at the peak efficiency torque28. Referring toFIG.3B, a map is provided illustrating torque versus work lost for an exemplary motor operating at a fixed speed during a transition from zero to peak efficiency torque. In this map, the work losses (W) are plotted along the vertical axis, while the torque output of the motor is plotted along the horizontal axis. As demonstrated by the curve29, the work losses of the motor increase as the torque output increases during the transition from zero to peak efficiency torque. Therefore, the faster that transition time from zero to peak efficiency torque, the less work is performed and the less energy is consumed by the motor. By substituting time in place of torque along the horizontal axis and then integrating the area under the curve29, the energy consumed by the motor can be calculated for a given transition time. For instance, the Applicant found that with an exemplary motor, 7234.5 Joules of energy was used with a transition time of 0.5 seconds, while only 723.4 Joules of energy were used a transition time of 0.05 second. This comparison demonstrates that the faster the transition time from zero to peak efficiency torque, the lower the energy consumed in losses. It should be noted that with this example, it is assumed that no acceleration of the load has taken place, so no energy has been added to the load inertia. For different motors, the transition of the motor from zero to peak efficiency torque, the peak efficiency torque and the work losses will all vary. The maps ofFIG.3AandFIG.3Bshould therefore be viewed as merely exemplary and should not be construed as limiting in any regard. Power Converter Power inventers are known devices that are used with electric motors for converting a DC power supply, such as that produced by a battery or capacitor, into three-phase AC input power applied to motor stator windings. In response, the stator windings generate the RMF as described above. Referring toFIG.4, a diagram of a power controller30for pulsed operation of an electric machine is illustrated. The power controller30includes a power converter32, a DC power supply34, an electric machine36, and a pulse controller38. The power converter32may be operated as a power inverter or power rectifier depending on the direction of energy flow through the system. When the electric machine is operated as a motor, the power converter32is responsible for generating three-phased AC power from the DC power supply34to drive the induction machine36. The three-phased input power, denoted as phase A37a, phase B37b, and phase C37c, is applied to the windings of the stator of the electric machine36for generating the RMF as described above. The lines depicting the various phases,37a,37b, and37care depicted with arrows on both ends indicating that current can flow both from the power converter32to the electric machine36when the machine is used as a motor and that current can flow from the electric machine36to the power converter32when the machine is used as a generator. When the electric machine is operating as a generator, the power converter32operates as a power rectifier and the AC power coming from the electric machine36is converted to DC power being stored in the DC power supply. The pulse controller38is responsible for selectively pulsing the three-phased input power. During conventional (i.e., continuous) operation, the three-phased input power is continuous or not pulsed. On the other hand, during pulsed operation, the three-phased input power is pulsed. Pulsed operation may be implemented, in non-exclusive embodiments, using any of the approaches described herein, such as but not limited to the approaches described with regard toFIG.5B,FIG.5CandFIGS.8through14. Referring toFIG.5A-5C, plots are provided for illustrating the difference between continuous and pulsed three-phased input power provided to the induction motor36. In each plot, current is plotted on the vertical axis and time is plotted along the horizontal axis. FIG.5Aillustrates conventional sinusoidal three-phased input current42a,42b, and42cdelivered to the induction machine36. Phase B, denoted by curve42blags phase A, denoted by42aby 120 degrees. Phase C, denoted by curve42c, lags phase B by 120 degrees. The sine wave period is T. The three-phased input current42a,42b, and42cis continuous (not pulsed) and has a designated maximum amplitude of approximately 50 amps. It should be appreciated that 50 amps is only a representative maximum current and the maximum current may have any value. FIG.5BandFIG.5Cillustrate two examples of different pulsed three-phased current waveforms44a,44b, and44cand46a,46b, and46cthat each has a 50% duty cycle and peak amplitude of approximately 100 amps. As inFIG.5Athe period of the base sine wave is τ, however, now the sine wave is modulated on and off. The delivered current inFIG.5BandFIG.5Cdelivers the same average torque as the continuously applied three-phased input current ofFIG.5A(assuming torque is proportional to current, which is often the case). The difference between pulsed currents44a-cand46a-cis the duration of their respective current pulses and the interleaved “off” periods. InFIG.5B, the current pulses44a-care interleaved with “off” periods of equal length. The length of each on and off period is 2τ. InFIG.5C, the current pulses46a-cand the interleaved “off” periods again have equal duration. In this case the duration is τ/2. In both examples, the duty cycle is 50%. However, the duration of the “on” and “off” time durations is different, i.e. the frequency of the pulsed modulation is different. The frequency of the pulsed modulation may vary based on the type of electrical machine used, noise and vibration considerations, current operating rotor speed, and other factors. FIG.5BandFIG.5Cillustrate applications in which the “on” motor drive pulses are evenly spaced while the motor is operated at a steady state desired output level. Such an approach works well in many circumstances but is not a requirement. The duty cycle need not be 50% but can be adjusted to match the desired average output torque. InFIG.5BandFIG.5Cthe phase of the on/off pulses is synchronized with the applied AC power; however, the phase of the on/off pulses need not be synchronized with the phase of the applied AC power in some embodiments. Thus, the relative sizes and/or timing of the motor drive pulses can be varied as long as they average out to deliver the desired average torque. Motor Physics and Constraints With any given motor, physics ultimately limits how fast a zero to peak efficiency torque transition can be. In general, the transition speed is based on the physics of how fast the electric fields can be built up in the motor, which in turn, are limited by the applied voltage, electric motor back emf (“BEMF”) and the inductance of the motor windings. If we assume that the set point of the power converter32is incremented at time zero and the feedback is zero, then the control to the output stages of each phase will be saturated. As a result, either the low or high output power devices for each motor phase will be turned on hard. This results in six possible combinations, including:1. Phase A and B positive with phase C negative,2. Phase A positive with phase B and C negative,3. Phase B and C positive with phase A negative,4. Phase B positive with phase A and C negative,5. Phase C and A positive with phase B negative, and6. Phase C positive with phase A and B negative. With each of these six possible combinations, the current flow in the motor36at time zero will be (a) the full current in one phase and (b) while the other two phases split the current. The ratio of these currents will depend, as further described below, by the rotor position at time zero. Referring toFIG.6A, a representative circuit modeling the current flows through the three phases A, B and C is shown. Each phase A, B and C is represented by its self inductance (“LS”), it mutual inductance (“LM”), its resistance (“R”) and its BEMF. In the case shown, Ic=Ia+Ib. The sum of the currents flowing in the mutual inductance is zero, and therefore, the mutual inductance has no effect on the current flow. The resulting reduced equivalent circuit assuming the BEMF of the motor is zero as illustrated inFIG.6B. This circuit takes time to build the current to a given value: i=VR(1-e-RtL) If the BEMF is not zero, then the applied voltage to each phase will differ. Because the phase impedances and phase currents are balanced, the neutral point of the winding is for this case=Vbus*2/3. If winding B was connected to the negative rail then the neutral voltage would be=Vbus/3. This defines the currents Ia, Ib, and Ic for phases A, B and C as: Ia=[Vbus×13-bemfa]×(1-e-RtL)RIb=[Vbus×13-bemfb]×(1-e-RtL)RIc=[Vbus×23-bemfc]×(1-e-RtL)R As all the values above are the instantaneous, the values at time zero are dependent upon the instantaneous value of the BEMF of each phase, which in turn is dependent upon the location of the rotor within one electrical cycle or pole pair pitch. It must also be noted that as time progresses, so does the instantaneous BEMF voltage per phase, the voltage applied to the motor inductance and the rate of rise of the motor phase current. The intent is for the current to reach its desired value and phase to provide the demanded torque. The current is normally controlled using Field Oriented Control or “FOC”, and hence, the phase currents are transposed to the rotating frame values of “iq” (quadrature current) and “id” (direct current) where the vector sum of id and iq equals the peak magnitude of the phase current and Arc Tan id/iq is the angle. The cosine of the angle is the power factor. So deducing the values of id and iq using the Direct Quadrature Zero transform gives: id=Vbus*[Cos∅3+sin∅3]×(1-e-RtL)Riq={Vbus*[Cos∅3-sin∅3]+Vpk}×(1-e-RtL)R When inspecting the above formulae, the BEMF waveform Vpk only influences iq (quadrature current), both are influenced by the bus voltage, Vbus, and the angular position of the rotor, θ. Neither the angle nor the motor BEMF can be changed without a change of motor so the only parameter that can be controlled to affect the rate of rise of the phase currents, and hence the motor torque, is the applied bus voltage, Vbus. One aspect of this invention, therefore, proposes that the bus voltage be temporarily increased or “boosted” to a higher value than the normal operating bus voltage for the duration of transit time from zero to the peak efficiency torque during pulsing, thereby reducing that transit time. It should be noted that when the converter is turned off the energy stored in the electric motor windings is returned to the bus voltage supply. If the supply cannot absorb this energy, then the bus voltage will rise as the bus capacitance absorbs this energy. Due to the amount of capacitance across the bus supply, this normal process will typically only increase the bus voltage by a small percentage, generally not enough to be considered as boosting the bus voltage. However, if this energy is captured independently, for example captured and stored in a storage device such as a capacitor or battery, then it could be recycled back to the motor in the form of a boost voltage. Alternatively, during the “OFF” period the bus voltage could be augmented by a separate boost voltage source using a charge pump or separate voltage source. This boost supply should not be designed to charge the main bus capacitance but a separate capacitance that can be discharged into the motor over the on-transition time from zero to the demanded torque. Conventional Power Converter Circuit The inherent inductance of the motor can thus transitorily delay/slow the voltage/power steps between the on and off motor states. During continuous (non-pulsed) operation, these transitory effects tend to have a relatively minimal impact on overall motor operation. However, when rapid pulsing is used as contemplated herein, the transitory effects can have a larger net impact, and therefore, there is an incentive to reduce the leading and falling edge pulse transition times. Referring toFIG.7A, a circuit diagram of a representative prior art power converter32is illustrated. The power converter circuit32includes three pairs of switches, denoted as S1thru S6. Each pair of switches S1-S2, S3-S4, and S5-S6are connected in series between two voltage buses (+VBUS) and (−VBUS). The electrical potential between two voltage buses (+VBUS) and (−VBUS) is the available potential to operate the electric machine36. Each of the switches, S1-S6, may have a bypass diode (D1-D6) connected electrically in parallel to the switch. These diodes help to prevent switch damaging voltage spikes that may be generated during switch operation. The diodes also provide a path for recycling current which the switch may block. This is especially important when the electric machine36is used as a generator. The switches S1-S6may be each be a MOSFET (metal-oxide semiconductor field-effect transistor) switch with integrated diodes. Alternatively, other types of transistors, such as, but not limited to, insulated gate bipolar transistors (IGBT) may be used. A connection to a stator coil winding of the electric machine36is made between each switch pair. For phase A, the connection is between switch pair S1-S2and is designated as37a. For phase B, the connection is between switch pair S3-S4and is designated as37b. For phase C, the connection is between switch pair S5-S6and is designated as37c. Within the electric machine36, each phase stator winding may be modeled as an inductor31, a resistor33, and a mutual inductance35. These elements are only labeled inFIG.7Afor phase C, but analogous elements are present in the phase A and phase B windings. The switches S1-S6may be collectively referred to as a switching network that controls power to and from the electric machine36. When the electric machine36is operated as a motor, the switches S1-S6operate in a conventional manner to apply current to each of the stator windings. For example, the switches may be operated as a six-step inverter, which provides AC power to the electric machine36. FIG.7Bshows the switching sequence to obtain a six-step output from the power converter32. Each switch is open for ½ of a cycle period in a staggered manner. For each winding, current can flow through one switch on the top row and one or two switches on the bottom row. The switch pairs, S1-S2, S3-S4, and S5-S6are never simultaneously turned on as this would short out the DC power supply34. FIG.7Cshows the voltage between points A and B as voltage Vab. Similarly,FIG.7DandFIG.7Eshow the voltage between points B and C and C and A, respectively. Summing these voltages allows the voltage between each phase and neutral to be determined. FIG.7Fshows the resultant phase voltage for phase A. The resultant 6-step waveform approximates a sine wave with a frequency, co, and is commonly referred to as the modulation signal. The phase voltage for phase B and C are shifted in phase by 120° and 240°, respectively, relative to the phase A voltage. It should be appreciated that electric machine36may be operated as a generator as well as a motor. When operating as a generator, the energy flow is from the electric machine36to the DC supply34. The power converter32acts as a 3-phase rectifier rather than an inverter. In typical prior art systems, the switching network is used to control the power flowing to the electric motor by pulse width modulation (PWM) control. PWM control reduces the time that the switching network is in an active configuration of the switches S1-S6where power can flow to the electric motor. That is, the fraction of time that the switches S1-S6are in an inactive configuration, either S1, S3, and S5or S2, S4, and S6are all turned off, increases as the desired electric motor torque output decreases. Power Converter with Boost FIG.8shows a power converter circuit132including a boost circuit according to a non-exclusive embodiment of the current invention. As compared to the prior art power converter circuit32shown inFIG.7A, power converter circuit132also includes additional switches SA and SB, which are each controlled by a pulse controller38. They may be controlled by a common signal line41(as shown inFIG.8) or they may have independent control lines (not shown inFIG.8). When switch SA is turned on, the positive power supply voltage (+VDC) is coupled to (+VBUS). When switch SB is turned on, the negative power supply voltage (−VDC) is coupled to (−VBUS). During operation, the pulse controller38operates to selectively turn switches SA or SB on and off by applying a pulsed waveform to signal line41, which electrically connects pulsed controller38to switches SA and SB. When switch SA and SB are turned on, current may be delivered to the electric machine36. Conversely, when SA and/or SB are turned off, no current, or only a transient current, is delivered to the electric machine36. The power converter circuit132also includes a capacitor C1, which has one conductive plate coupled to (+VBUS) and the other conductive plate coupled to (−VBUS). Collectively the switches SA and SB and the capacitor C1may be referred to as a boost circuit, since their purpose is to increase the initial voltage on the +VBUSand −VBUSbuses at the beginning of an “on” pulse as described below. In various embodiments, boost circuit may be incorporated into the switching network or may include elements distinct from the switching network. As previously noted, the goal of pulsed motor control is to operate the electric machine36at substantially its most efficient level for the current machine speed during “on” periods and to cut-off power (provide zero or negligible power) during the “off” periods. For example, the power supplied during the off periods may be less than 10%, 5%, 1%, 0.5%, or 0.1% of the power supplied during the “on” period. The operating point while operating during the “on” period may have an efficiency within 5%, 2%, or 1% of a maximum operating efficiency point of the motor at the current motor speed. The transitions thru the low efficiency operating region between the “off” and “on” periods should be as fast as possible to maximize efficiency. Thus, the power transitions between the machine power “on” and “off” states ideally have a leading edge that transitions vertically straight up and a following edge that vertically transitions straight down. Such “perfect” pulses60are diagrammatically illustrated inFIG.9A, which illustrates the ideal motor drive current versus time for pulsed control having a duty cycle of 50%. In this figure, the current pulse represents the sum of the current in all the phases. While the current pulse is shown as flat topped, this will not necessarily be the case. In the real-world, a number of practical limitations make generation of such perfect pulses difficult to achieve. For instance, inductive aspects of both the electric machine36and the power converter32circuitry slow down the current rise and fall times. The actual response of a particular machine will vary with the electrical characteristics of the electric machine36, the rotational speed of the electric machine and the available bus voltages. In general, the actual rise and fall of pulses occur more gradually, meaning the transitions occur over time. The nature of the rise and fall in the real-world is diagrammatically illustrated inFIG.9B. As seen therein, there is a ramp-up period (rise time)62required for the current to actually rise from zero to the desired “on” power level and a ramp-down period (fall time)64required for the current to actually fall from the “on” power level down to zero. During the power ramp-up and ramp-down periods, the electric machine36continues to consume or generate power. However, the machine operates less efficiently during these transition periods. In general, the machine efficiency will drop as the operating current drops from its maximum efficiency condition (curve16FIG.1) towards zero, with the energy conversion efficiency getting noticeably worse as the current level approaches zero. Thus, the pulse distortion represented by the current ramp-up and ramp-down periods detract from efficiency gains resulting from pulsed operation. In general, the smaller the ratio of the rise/fall times to the pulse length, the less the transitory switching effects impact the machine's energy conversion efficiency during pulsing. It should be appreciated that the transitory effects shown inFIG.9Bare illustrative in nature and do not necessarily reflect actual rise/fall times associated with operation of any particular electric machine. The relative scale of the rise time to the pulse length ratio can vary widely based on the characteristics of the machine used (which primarily dictates the rise and fall times), the frequency of the pulsing (which is primarily dictated by the control scheme used) and the pulse width (which is dictated by the control scheme and machine load). The voltage available to power the electric machine and machine rotation speed will also impact the pulse rise and fall times. If the pulsing is slow compared to the machine response, the rise/fall times may be a small fraction of the pulse width and the transitory switching effects may have a minimal impact on machine performance Conversely, if the pulsing is very rapid and/or the machine response is low, the rise/fall times may be a significant fraction of the pulse width and can even exceed the pulse width in some situations. If not managed carefully, the transitory efficiency losses associated with switching can significantly reduce or even eliminate any theoretical gains that can be attained by pulsed operation. Thus, it is important to consider the transitory switching effects associated with pulsed operation when determining the pulsing frequency and control schemes that are appropriate for any particular application. The capacitor C1included in the power converter circuit132ofFIG.8is provided to improve the current rise and fall times. The capacitor C1may store energy from the electric machine36during the ramp-down period and supply energy to the electric machine36during the ramp-up period. This results in faster turn-on and turn-off transitions than would occur without the capacitor C1. To better understand operation of the power converter132, assume the power converter132is initially in an “on” state and the electric machine36is operating as a motor. This implies that, the switches SA and SB are turned on so that current can flow from the positive terminal of the DC supply34thru the power converter132to the electric machine36and return to the negative terminal of the DC supply34. The switches S1thru S6will oscillate in the configurations shown inFIG.7Bto apply AC power to the electric machine36. To terminate motor operation the switches SA and SB may be turned off, allowing the +VBUSand −VBUSbuses to have a different potential than their respective terminals of the DC power source34. Since the circuit is now open, current must cease to flow thru the circuit; however, there may be significant energy associated with current generated magnetic fields in the electric machine36. At least some of this energy may be extracted from the electric machine36and is captured in capacitor C1where it is stored. This will increase the electrical potential difference between the positive voltage bus and negative voltage bus. For example, the potential on line +VBUSmay increase above that of the positive terminal of the DC power source, +VDC, and the potential on line −VBUSmay decrease below that of the negative terminal of the DC power source, −VDC. Note that the switches S1-S6all have bypass diodes, which allow unidirectional current to flow from the electric machine36to the +VBUSline and from the −VBUSline to the electric machine36independent of the switch position. Coincident with, or nearly coincident with, the turning off of switches SA and SB any of the switches S1-S6that may have been turned on when switches SA and SB opened are turned off, so that current does not flow through any of these switches between the +VBUSand −VBUSlines to the electric machine36. When motor operation is once again desired, the switches S1-S6may be turned on in one of the patterns shown inFIG.7B. The switching pattern must correspond to the rotor rotation angle so that the phase of the applied current matches the correct phasing to once again supply power to the electric machine36. Switches SA and SB are closed when the voltage on the +VBUSdrops to +VDCand −VBUSrises to −VDC, respectively. This circuit configuration and control method is arranged to provide a higher initial voltage to be applied to the electric machine36at the start of the “on” phase, which advantageously reduces the pulse rise time. FIG.10illustrates exemplary +VBUSand −VBUSwaveforms versus time for the circuit shown inFIG.8. The pulse generator38generates a digital waveform43consisting of a string of digital “0's” and “1's”. The 1's may correspond to the electric machine36being turned “on” and the 0's may correspond to the electric machine being turned “off”. InFIG.10, the electric machine36is being pulsed at a 40% duty cycle; however, this is exemplary only and any duty cycle may be used. The voltage45on the +VBUSrail increases to +VBOOSTduring the ramp-down period and the voltage47on the −VBUSrail drops to −VBOOSTduring the ramp-down period. The magnitude of the voltage change on the +VBUSand −VBUSbuses may be equal or different. The +VBUSvoltage45and −VBUSvoltage47remain relatively constant during the motor off time as the energy from the motor ramp-down is stored in capacitor C1. When the pulse generator waveform43returns to a digital “1”, the energy stored in capacitor C1is supplied to the electric machine36thru the switch array S1-S6. This causes the voltage45on the +VBUSrail to return to +VDCand the voltage47on the −VBUSrail to return to −VDCas the charge in capacitor C1is dissipated and the energy stored in capacitor C1is used to drive the motor. Motor operation during the “on” period is sustained by turning on switches SA and SB, so that energy from the DC power supply34may be used to drive the motor. Effectively the boost circuit increases an available electric potential between the positive voltage bus and the negative voltage bus to drive the electric machine at the beginning of at least one pulse in the series of pulses. It should be appreciated that positive voltage bus and negative voltage bus are relative terms and the electric potential on each of these buses relative to ground potential may be either positive or negative. The boost circuit may be used to increase the available electric potential to drive the electric machine for all pulses in the series of pulses. While the exemplary power converter with boost circuit is shown inFIG.8as having a switch adjacent both the positive and negative terminal of the DC power supply, this is not a requirement. In some embodiments, only a single switch may be required The switches SA and SB in conjunction with the capacitor C1can thus be used to reduce the power rise and fall times, in some cases by factors of 2, 5, 10 or more. The voltage across capacitor C1can be increased above that of the power supply by storing energy recovered from the motor during its ramp down. The magnitude of the voltage increases with the amount of magnetic energy that can be extracted and captured. This can significantly reduce potential deleterious transitory switching effects associated with pulsed operation. Examples of improved rise and fall times are schematically shown inFIG.9C. As evident in the figure, the ramp-up rise time66on the pulse leading edge is faster/shorter as compared to the corresponding ramp-up time62shown inFIG.9B. Similarly, the ramp-down time68of the pulse trailing edge is faster/shorter as compared to the corresponding ramp-down time64shown inFIG.9B. Therefore, it should be appreciated that electric machines designed with pulsed control in mind or modified to improve the transient response of the machine to power pulses, can benefit even more from pulsed operation than existing machines. It should be appreciated that the appropriate pulsing frequency implemented by the pulse controller38for different machines may be very different based on the machine's construction, operating environment and operational range. For some electric machines, switching frequencies on the order of 10-50 kHz may be appropriate—whereas for other machines much lower switching frequencies, as for example 10-500 Hz range may be more appropriate. The most appropriate pulsing frequency for any particular machine will depend on a wide variety of circumstances, such as the type of machine, the load of the machine, and/or the application of the machine. It should be appreciated that the details of the boost circuit used to shorten the rise and fall times of the power to or from an electric machine may vary depending on the type of electric machine and its operating regime. For example, in some cases the one of the switches SA or SB can be deleted from the power converter circuit132. Other types of power converter circuits and control strategies may be used. For example, a Z-source inverter, where a diode, two inductors, and two capacitors are situated between the power supply and switching network may be used in some situations. The voltage boost level and size of capacitor C1can be chosen appropriately for the electric machine and its inductive and resistive characteristics to shorten the transient rise/fall times associated with pulsing the machine on and off. Preferably, the respective capacitance and boost voltage levels are also selected to maximize overall machine efficiency during pulsing, including inefficiencies associated with the transients themselves and the effects of any overshoot that may occur due to use of the capacitor C1. Since the capacitor C1is used to improve transient response, it may be opportunistically recharged in the periods when the motor is not being supplied power—as for example during the electric machine off periods. This mode of operation is explained in more detail in the description below regardingFIG.11. Depending on the motor speed and load there may be insufficient energy stored in the magnetic fields of the motor to adequately boost the +VBUSand −VBUSvoltages for sharp rise and fall times. In such cases it may be desirable to boost the potential difference across the electric machine during the off periods between pulses. An exemplary voltage waveform showing two boost cycles73aand73bis shown inFIG.11. It should be appreciated that more or less than two boost cycles may be used depending on the operating conditions of the electric machine. An appropriate switching network and control strategy is required to implement this type of control. Referring toFIG.12, another power converter200including a boost circuit202in accordance with another embodiment of the invention is illustrated. The power converter200includes switches S1and S2for phase A, switches S3and S4for phase B and switches S5and S6for phase C. Each pair of switches S1-S2, S3-S4, and S5-S6are connected in series between two voltage buses (+VBUS) and (−VBUS). The electrical potential between two voltage buses (+VBUS) and (−VBUS) is the available potential to operate the electric machine36. The switches S1-S6are collectively referred to as a switching network that controls the power to and from the machine36. When operating as a motor, power from a DC supply is provided via the switching network of switches S1-S6. In turn, the switching network provides phased energy to the three phases of stator windings of the machine36as previously described. Similarly, when operating as a generator, the energy flow is from the machine36to a storage device, such as a battery. The boost circuit202includes a boost supply204, switch206, a capacitor C1, a battery and a control signal208generated by the pulse controller38. As the pulse controller38was previously described, a detailed explanation is not repeated herein for the sake of brevity. In various embodiments, the boost supply204can be a dedicated circuit (e.g., charge pump or separate voltage source) capable of generating a boost voltage and/or a storage device such as another capacitor and/or battery. With the later embodiments, at least some of the energy stored by the storage device may be derived from the motor38itself. For example, when the machine36is operating as a generator, or when the machine36is acting as a motor and transitions from on to off states, such as during pulsing, the produced energy can be diverted to and saved certain components in the boost circuit202, such the capacitor C1and/or the battery. The saved energy can then be used to “boost” the positive rail (+VBUS) during positive transitions as described below. The switch206can be any type of switch that is capable of switching between the positive (+) and negative (−) electrodes of the boost supply204. It is anticipated that this switch will be constructed using semiconductor devices. In a specific but non-exclusive embodiment, the switch206is a single pull double throw switch. During continuous motor operation, phased power is provided to the stator windings of the machine36via the switches S1and S2for phase A, switches S3and S4for phase B and switches S5and S6for phase C as is well known in the art. The net result is a continuous torque output of the motor as previously described. During pulsed operation, the pulsed controller38controls the switch206via the control signal208to control the boost circuit202. With a positive pulse transition, the switch206is activated to connect the positive rail (+VBUS) to the positive (+) terminal of the boost supply204. As a result, the boost supply204, operating in cooperation with the capacitor C1and battery, act to boost the voltage on the positive rail (+VBUS). With the Increased or Boosted Voltage on the Positive Rail, the transition time is reduced. Once the stored energy in the boost circuit has diminished or the peak torque level has been achieved the control signal208directs the switch to connect the positive rail (+VBUS) to the negative (−) terminal of the boost supply204. As a result, the boost voltage is effectively removed from the positive rail (+VBUS). The effect of the boost circuit202is also illustrated inFIGS.9A-9C. In particular,FIG.9Ashows an ideal pulse with no transition time,FIG.9Bshows a “real world” pulse with a transition time designated by reference numeral62. As previously noted, inductive aspects of both the electric machine36and the circuitry of the power converter200slow down the current rise and fall times.FIG.9Cshows a transition aided by the boost circuit202. As can be readily understood by a comparison, the “boosted” transition time66as shown inFIG.9Cis significantly smaller (i.e. faster) than the transition time62as shown inFIG.9B. With theFIG.12embodiment, the capacitor C1is arranged in parallel with each of the switch pairs S1-S2, S3-S4and S5-S6between the positive rail (+VBUS) and negative rail (−VBUS). In a non-exclusive embodiment, the size of C1is derived from the ripple current of the power converter200when acting as an inverter. With this arrangement, the ability of the boost circuit202to reduce the rise and fall times of pulses is improved. Operational Flow Diagrams FIG.13is a flow diagram70illustrating steps for pulsed control operation of an electric motor with characteristics such as those depicted inFIG.1. In the initial step72, the current motor output and current motor speed are ascertained. In decision step74, a determination is made based on the current motor output and current motor speed if the motor should be operated in a continuous mode or a pulsed mode. In other words, a determination is made if the desired motor torque is above or below the most efficient output torque for the current motor speed (i.e., the maximum efficiency curve16of the motor map illustrated inFIG.1). If above, the motor is operated in the continuous mode. If below, the motor may advantageously be operated in the pulsed mode. In step76, the motor is operated in the continuous mode76if the current motor torque is above the most efficient output torque for the current motor speed. In step78, the power output or magnitude of the “on” pulses that provide for substantially maximum efficiency operation at the current motor speed is determined. In step80, the desired pulse duty cycle for operation in the pulsed mode is determined so that the average output power or torque matches the desired output. In step82, the motor is operated in the pulsed mode using the determined pulse duty cycle and pulsed power output. The use of the power controller30with the boosted power converter circuits132or some other power converter circuit capable of storing and releasing magnet energy from the electric machine reduces, often significantly, the rise and fall times of the pulses, further improving motor efficiency. The above steps72-82are continuously performed while the motor is in operation. At any particular motor speed, there will be a corresponding most efficient output torque which is diagrammatically illustrated by maximum efficiency curve16inFIG.1. As the instantaneous motor output request and/or current motor speed change, a decision is made to operate the motor in either the continuous or pulsed mode as appropriate. From a conceptual standpoint, when the desired motor torque is below the most efficient output torque for the current motor speed, the overall efficiency of the motor can be improved by pulsing the motor. As a result, for electric motor-powered vehicles the overall efficiency of the vehicle is improved, meaning the vehicle range between battery recharging is extended. FIG.14is a diagram illustrating a system300for modulating the energy supplied to a machine36in accordance with another non-exclusive embodiment of the invention. The system300includes the machine36, a power converter32, a torque control decision module302, a feedback sensor304for generating a feedback signal306indicative of the angular position of the rotor of the machine36and a torque and speed estimator308. During operation of the system300, the torque modulation decision module302receives a torque demand. In response, the torque modulation decision module302makes a determination if the requested torque is less than the peak efficiency torque of the machine36when operating as a motor. If not, meaning the torque demand is larger than the peak efficiency torque, the machine36is operated as a motor in the continuous mode. In which case, the torque demand waveform310provided to the power converter32is indicative of continuous operation of the machine36operating as a motor. On the other hand if the torque demand is less than the peak efficiency torque of the machine36, then the machine36is operated as a motor in the pulsed mode. In which case, the torque modulation decision module302produces a modulated waveform310for the power converter32, causing the machine36operating as a motor to switch or pulse between the peak efficiency torque of the motor and a lower torque, the average of which is substantially equal to the demanded torque. In various embodiments, the lower torque can be zero, but it is not necessarily zero. The lower torque can be some other torque value above zero, provided the average of the lower and peak efficiency torque is substantially equal to the demanded torque. The power converter32includes a switching network including pairs of switches S1-S2for phase A, switches S3-S4for phase B and switches S5-S6for phase C, all of which are not shown in the figure for the sake of simplicity. As previously noted, the switches S1-S6are controlled by the power converter32to operate the machine36either (1) continuously as a motor resulting in generating a continuous torque output when the torque demand is greater than the peak efficiency torque or (2) in the pulse mode when the torque demand is less than the peak efficiency torque. The power converter32can control the energy supplied to the machine36using any of a number of different protocols, such as Pulse Width Modulation (PWM), Direct Torque Control (DTC), hysteresis, or “dead beat” control, which is a form of current modulation. In alternative embodiments, a boosted power converter such as132ofFIG.8or200ofFIG.12may be used. With boosted versions of the power converter32, efficiency and performance of motor operation of the machine36is improved due to the faster rise and/or fall times of pulses during pulsed operation. The feedback sensor304generates the feedback signal306, which is indicative of the angular position of the rotor of the machine36. The feedback signal is provided to each of the power converter32and the torque and speed estimator308. With the angular position of the rotor known, the torque and speed estimator308can provide accurate estimates of the torque and speed of the motor to the torque modulation decision module302. In response, the waveform310can be adjusted as necessary so that the timing of switching network (i.e., the timing of turning the switches S1-S6on/off) within the power converter32can be precisely controlled so that each of the phases A, B and C of energy are timed to coincide with the current position of the rotor. As a result, the operation of the machine36as a motor is both smooth and efficient. It should be noted that the use of a feedback sensor304is not mandatory and that other techniques can be used for measuring or estimating the angular position of the rotor of the machine36. For instance, any of a number of sensorless approaches may be used as well. Other Motor and Generator Types There are a wide variety of machines, both electric motor and generator, that are known and commercially available, including both DC and AC motors/generators. Although the structure, control and energy conversion efficiency of the various types of electric motors and generators vary significantly, most electric motors and generators are designed to operate over a range of operating conditions and their energy conversion efficiency will vary over that operating range, often significantly. In general, the control principles described herein can be applied to any type of machine to improve efficiency, provided the operating range includes regions below the equivalent of the maximum efficiency curve16illustrated inFIG.1. Some prior art motors are currently operated using pulse width modulation (PWM) control. However, such motors are driven without consideration of what might be their most efficient energy conversion level. As such, the described approach can also be used to improve the energy conversion efficiency of such motors as well. Many types of motors, including brushless DC motors, induction motors, synchronous AC motors, switched reluctance motors, etc. are traditionally driven by a continuous, albeit potentially varying, drive current to deliver the desired torque output. Frequently, the drive current is controlled by controlling the output voltage of an inverter and/or converter (which serves as the voltage input to the motor). Generally, by changing the relative phasing between the rotor and stator magnetic fields a motor can be operated as a generator. Thus, circuits and control methods described in terms of a motor are equally applicable to using an electric machine as a generator. The described pulsed control is particularly beneficial when such motors and generators are operated in regions below their respective maximum energy conversion efficiency points. Therefore, the present embodiments should be considered illustrative and not restrictive and the invention is not to be limited to the details given herein but may be modified within the scope and equivalents of the appended claims. | 57,816 |
11863097 | DETAILED DESCRIPTION Preferred embodiments according to the present invention will be described in detail with reference to the drawings. FIG.1is a schematic diagram showing a motor controller10according to one embodiment of the present invention. The motor controller10is used for driving a three-phase motor M, where the three-phase motor M has a first coil L1, a second coil L2, and a third coil L3. The motor controller10comprises a switch circuit100, a driving circuit110, and a pulse width modulation circuit120. The switch circuit100includes a first transistor101, a second transistor102, a third transistor103, a fourth transistor104, a fifth transistor105, a sixth transistor106, a first terminal V, a second terminal U, a third terminal W, a fourth terminal VCC, and a fifth terminal GND, where the switch circuit100is coupled to the three-phase motor M for driving the three-phase motor M. The first terminal V has a first voltage signal VO. The second terminal U has a second voltage signal UO. The third terminal W has a third voltage signal WO. The first transistor101is coupled to the fourth terminal VCC and the first terminal V while the second transistor102is coupled to the first terminal V and the fifth terminal GND. The third transistor103is coupled to the fourth terminal VCC and the second terminal U while the fourth transistor104is coupled to the second terminal U and the fifth terminal GND. The fifth transistor105is coupled to the fourth terminal VCC and the third terminal W while the sixth transistor106is coupled to the third terminal W and the fifth terminal GND. Each of the first transistor101, the third transistor103, and the fifth transistor105may be a p-type MOSFET. Each of the second transistor102, the fourth transistor104, and the sixth transistor106may be an n-type MOSFET. Moreover, the fourth terminal VCC has an input voltage, where the input voltage may be a power supply voltage. The fifth terminal GND has a ground voltage. The system may provide the input voltage for the motor controller10via the fourth terminal VCC, thereby enabling the motor controller10to work normally. For example, the input voltage may be 12 volts and the ground voltage may be 0 volt. Therefore, the motor controller10may be applied to a high voltage configuration. The first coil L1 is coupled to the first terminal V and a sixth terminal COM. The second coil L2 is coupled to the second terminal U and the sixth terminal COM. The third coil L3 is coupled to the third terminal W and the sixth terminal COM. That is to say, the first coil L1, the second coil L2, and the third coil L3 form a Y-shaped configuration. The driving circuit110generates a first control signal C1, a second control signal C2, a third control signal C3, a fourth control signal C4, a fifth control signal C5, and a sixth control signal C6 for respectively controlling the ON/OFF states of the first transistor101, the second transistor102, the third transistor103, the fourth transistor104, the fifth transistor105, and the sixth transistor106. The pulse width modulation circuit120generates a pulse width modulation signal Vp to the driving circuit110, where the pulse width modulation signal Vp has a duty cycle. The motor controller10may control the speed of the three-phase motor M by adjusting the duty cycle. FIG.2is a timing chart according to one embodiment of the present invention, where the current ILW indicates the current flowing through the third coil L3. Please refer toFIG.1andFIG.2simultaneously. According to one embodiment of the present invention, when the motor controller10starts a floating phase for detecting a phase switching time point, the motor controller10may enable that at least one transistor within the switch circuit100is operated in a linear region, where the floating phase is formed in the third coil L3. At this moment the motor controller10enables that the fifth transistor105and the sixth transistor106are turned off to form the floating phase. In order to avoid generating switching noise, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may enable that the voltage of the first terminal V is greater than the ground voltage and the voltage of the first terminal V is less than the input voltage. More specifically, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may enable that the voltage of the first terminal V is locked at a specific voltage or the current flowing through the first terminal V is locked at a specific current, according to the duty cycle of the pulse width modulation signal Vp. The specific voltage may be relevant to the duty cycle and the specific current may be relevant to the duty cycle. For instance, when the duty cycle is 50%, the input voltage is 12 volts and the ground voltage is 0 volt, the motor controller10may enable that the voltage of the first terminal V is locked at 6 volts. That is to say, if the motor controller10adopts a voltage lock mode, when the duty cycle increases, the specific voltage may increase. Similarly, if the motor controller10adopts a current lock mode, when the duty cycle increases, the specific current may increase. Thus, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may reduce switching noise of the three-phase motor M and increase the success rate of switching phases by the voltage lock mode or the current lock mode. Furthermore, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may not need to start an ON time detecting mode or an OFF time detecting mode for detecting aback electromotive force, resulting that the motor controller10may be applied to a high frequency configuration. When the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may avoid generating switching noise and increase the detecting accuracy by the embodiments as follows:1. When the motor controller10enables that the fifth transistor105and the sixth transistor106are turned off to form the floating phase, the floating phase is formed in the third coil L3. When the motor controller10enables that the floating phase is formed in the third coil L3, the motor controller10may enable that the first transistor101and the second transistor102are partially turned on. That is, both the first transistor101and the second transistor102are operated in the linear region. At this moment the motor controller10may enable that the third transistor103is turned off and the fourth transistor104is partially turned on or fully turned on. The motor controller10may modulate the ON resistance of the first transistor101and the ON resistance of the second transistor102, such that the motor controller10enters the voltage lock mode or the current lock mode to avoid generating switching noise. As shown inFIG.2, the motor controller10may detect the zero point of the back electromotive force by comparing the voltage of the third terminal W with the voltage of the sixth terminal COM during the floating phase time interval. Therefore, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may not need to start the ON time detecting mode or the OFF time detecting mode for detecting the back electromotive force.2. When the motor controller10enables that the fifth transistor105and the sixth transistor106are turned off to form the floating phase, the floating phase is formed in the third coil L3. When the motor controller10enables that the floating phase is formed in the third coil L3, the motor controller10may enable that the first transistor101and the second transistor102are partially turned on. That is, both the first transistor101and the second transistor102are operated in the linear region. At this moment the motor controller10may enable that the third transistor103is partially turned on and the fourth transistor104is partially turned on or fully turned on. The motor controller10may modulate the ON resistance of the first transistor101and the ON resistance of the second transistor102, such that the motor controller10enters the voltage lock mode or the current lock mode to avoid generating switching noise. As shown inFIG.2, the motor controller10may detect the zero point of the back electromotive force by comparing the voltage of the third terminal W with the voltage of the sixth terminal COM during the floating phase time interval. Therefore, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may not need to start the ON time detecting mode or the OFF time detecting mode for detecting the back electromotive force.3. When the motor controller10enables that the fifth transistor105and the sixth transistor106are turned off to form the floating phase, the floating phase is formed in the third coil L3. If the three-phase motor M is operated at a full speed, the motor controller10may enable that the first transistor is fully turned on and the second transistor is turned off. That is, the first transistor is operated in a saturation region. At this moment the motor controller10may enable that the third transistor103is turned off and the fourth transistor104is partially turned on. As shown inFIG.2, the motor controller10may detect the zero point of the back electromotive force by comparing the voltage of the third terminal W with the voltage of the sixth terminal COM during the floating phase time interval. Therefore, when the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may not need to start the ON time detecting mode or the OFF time detecting mode for detecting the back electromotive force. According to one embodiment of the present invention, the motor controller10may be applied to a brushless direct current motor system. Moreover, the motor controller10may be applied to the high voltage configuration and the high frequency configuration. When the motor controller10starts the floating phase for detecting the phase switching time point, the motor controller10may be operated in the voltage lock mode or the current lock mode, such that an voltage of an output terminal within the switch circuit100is greater than the ground voltage and the voltage of the output terminal is less than the input voltage. Based on the above disclosed technology, the motor controller10may reduce switching noise of the three-phase motor M and increase the success rate of switching phases. While the present invention has been described by the preferred embodiments, it is to be understood that the invention is not limited to the disclosed embodiments. On the contrary, it is intended to cover various modifications. Therefore, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications. Those skilled in the art will readily observe that numerous modifications and alterations of the device and method may be made while retaining the teachings of the invention. Accordingly, the above disclosure should be construed as limited only by the metes and bounds of the appended claims. | 11,420 |
11863098 | Referring to theFIGS.1aand1b, two different 3-level inverter topologies for one phase leg are illustrated. Both of these 3-level inverter topologies are per se known.FIG.1aillustrates a neutral point clamped (NPC) inverter circuit topology (for one phase leg) whereasFIG.1billustrates a T-type three-level inverter circuit topology (for one phase leg). In the illustrated embodiments, the inverters2are diode clamped inverters, however it will be appreciated that other inverter types such as flying capacitor inverters may be employed without departing from the scope of the invention. Moreover,FIGS.1aand1billustrate only one phase of a three-level inverter, but it will be appreciated that the inverter according to embodiments of the invention may be a multiphase inverter having two, three or more phases depending on the application. For instance, as illustrated inFIGS.5and6, many electrical motors are three phase U, V, W and thus the inverter according to an embodiment of the invention for such application is a three phase inverter. The principles described herein for one phase leg apply to each phase leg of the multiphase inverter. Referring toFIGS.5and6, according to an aspect of the invention, an electrical motor system comprises a multiphase U, V, W electrical motor3connected to a drive1and a DC power supply4. The drive1comprises a power electronics module5connected to a gate drive unit6. The gate drive unit includes or is connected to a control module7that receives voltage VDC, VNPand phase current iu, iw, ivmeasurement inputs8,9from the power supply4and motor3to control the gate drive and power electronics for control of the motor3. The power electronics module5comprise a multi-level multi-phase inverter2according to embodiments of the invention, as illustrated inFIGS.5and6. Although the illustrated embodiments show a three-level inverter for each phase leg, the invention also applies to higher level inverters, such as five level or seven level inverters per phase leg that enable the generation of smoother stepped output waveforms with lower harmonic distortions, but that also face the problem of neutral point control under certain operating conditions similarly to three-level inverters. The present invention overcomes the problem of neutral point deviation by operating the multi-level inverter with a standard modulation scheme having three or more levels of modulation patterns, depending on the number of modulation levels of the inverter, when neutral point stability is high, and operating the multi-level inverter with a two-level modulation pattern when neutral point stability is low. The operation of the multi-level inverter may switch between three (or higher) level modulation and two-level modulation patterns depending on one or more operating parameters of the inverter and the load connected to the inverter. The load may in particular comprise a multiphase, for instance three phase U, V, W, electrical motor3connected to a multiphase, for instance three phase inverter2according to embodiments of the invention as schematically illustrated inFIGS.5and6. Using the two level modulation pattern when starting a motor without sensor feedback also improves stability and control of the sensorless start. Using the two level modulation pattern in a multi-level inverter according to embodiments of the invention also advantageously allows to reduce the size of the DC link capacitors C1, C2compared to conventional multi-level inverters. The inverter according to embodiments of the invention is connected to a gate drive unit6comprising or connected to a controller7that, inter alia, is connected to the inverter transistors Q1-Q4to control the switching of the transistors for the generation of the modulation patterns, in particular the pulse width modulation (PWM) patterns of the standard multi-level modulation scheme and the two-level modulation scheme. The switching states of the transistors Q1, Q2, Q3and Q4of three-level inverters of both embodiments ofFIGS.1aand1bfor three-level modulation is illustrated in the table below: Voltage between the neutral pointSwitching stateNP and the output Vout-NPQ1Q2Q3Q4Positive (P)+VDC/2OnOnOffOffNeutral (0)0OffOnOnOffNegative (N)−VDC/2OffOffOnOn The corresponding three-level PWM signal is illustrated inFIG.4awhereby the PWM signal has an inverter phase leg output voltage VOut-Gwith values +VDC, +VDC/2 and 0 for the positive portion of voltage signal, and values 0, −VDC/2, and −VDCfor the negative portion of voltage signal. The switching states of the transistors Q1, Q2, Q3and Q4of the three-level inverters of both embodiments ofFIGS.1aand1bfor operation in two-level modulation mode is illustrated in the table below: Voltage between the neutral pointSwitching stateNP and the output VOut-NPQ1Q2Q3Q4Positive (P)+VDC/2OnOnOffOffNegative (N)−VDC/2OffOffOnOn The corresponding two-level PWM signal is illustrated inFIG.4bwhereby the PWM signal has an inverter phase leg output voltage VOut-Gwith values +VDC, and 0 for the positive portion of voltage signal, and values 0, and −VDCfor the negative portion of voltage signal. In both inverter circuit designs ofFIGS.1aand1b, a two-level inverter mode of operation can easily be achieved by avoiding the switching state “Neutral (0)”. It may be noted that in two-level mode the gate drive unit is configured to drive the transistor pair Q1-Q3and transistor pair Q2-Q4always in complementary manner such that they are in an opposite switching states. The gate drive unit controller receives measurement signals of the operating parameters from the load and the inverter that are used to determine the operating mode, in particular the level of the PWM modulation pattern level, and to control the switching between standard multi-level modulation, two-level modulation, or no modulation. An illustration of these modulation modes or states is schematically illustrated inFIG.2, whereby the transition between states (no modulation, 3 level modulation, 2 level modulation) in this example is briefly explained in the table below. TransitionDescription1Initial and automatic transition2Modulation of inverter is requested to switch off. No pulses are generated any more.3Modulation of inverter is requested to be switched on.4The 2-level modulation strategy is selected depending on operating parameters selectedfrom any one or more of:1)High neutral point potential deviation2)Low stator frequency of the motor connected to the inverter3)Low power factor cos(ϕ) of the motor connected to the inverter4)Low phase current magnitude5)Low PWM frequency5The 3-level modulation strategy is selected for all other cases than listed under transition 4.6, 7, 8Automatic transitions Operating parameters that may be measured or computed and used by the gate drive unit6to determine whether to operate the inverter in two-level or multi-level modulation mode, where the electrical load is an electrical motor3, are presented below: ParameterDescriptionTypical operating range of valuesVnpNeutral point voltageA predetermined value dependingon the characteristics of thesystem (motor and supply) [V]εnpDeviation of neutral point voltageThreshold value having a value( | VDC/2 − VNP| ) in %;between 0 to 50 [%];ωStator frequency0 to 40% of nominal speedcos(φ)Motor power factor = Real power/0 to 1 [—]Apparent powerIphMotor phase currentA predetermined value depending onthe characteristics of the motor [A]fpwmModulation (PWM) frequency0 to 100 [kHz] According to embodiments of the invention, the threshold values of the above mentioned parameters are set in the following ranges: Threshold values for determiningParameterDescriptiontransition to two-level modulationωminMinimum motor stator frequency0 to 40% of nominal speedεnp_maxAllowed deviation of neutral0 to 50 [%]point voltage in %PFminMotor minimum power factor0 to 0.7 [—]Iph_minMotor minimum phase currentA predetermined value depending onthe characteristics of the motor [A]fpwm_minMinimum PWM frequency0 to 50 [kHz] In advantageous embodiments, the threshold values may in particular be set in the following ranges: Threshold values for determiningParameterDescriptiontransition to two-level modulationωminMinimum motor stator frequency5 to 30% of nominal speedεnp_maxAllowed deviation of neutral5 to 30 [%]point voltage in %PFminMotor minimum power factor0.1 to 0.5 [—]Iph_minMotor minimum phase current5% to 40% of Nominal current [A]fpwm_minMinimum PWM frequency1 to 30 [kHz] Preferably, according to advantageous embodiments of the invention, the threshold values may be set in the following ranges: Threshold values for determiningParameterDescriptiontransition to two-level modulationωminMinimum motor stator frequency5% to 15% of nominal speedεnp_maxAllowed deviation of neutral10 to 20 [%]point voltage in %PFminMotor minimum power factor0.1 to 0.3 [—]Iph_minMotor minimum phase current10% to 30% of Nominal current [A]fpwm_minMinimum PWM frequency5 to 15 [kHz] An example of threshold values of the above mentioned parameters in an advantageous embodiment is illustrated below: Threshold values for determiningParameterDescriptiontransition to two-level modulationωminMinimum motor stator frequency10% of nominal speedεnp_maxAllowed deviation of neutral15 [%]point voltage in %PFminMotor minimum power factor0.2 [—]Iph_minMotor minimum phase current20% of Nominal current [A]fpwm_minMinimum PWM frequency8 [kHz] A range of values, or a threshold value (e.g. a minimum or maximum value), or a plurality of threshold values (lower bound and upper bound), may be set for any one or more of the above parameters and stored for instance in a lookup table in a memory, preferably in a memory of the gate drive unit. The gate drive unit receives measurement values of said any one or more of the above parameters and compares them with the stored values to determine the control mode of the inverter. For instance, if frequency ω of motor stator coils drops below a threshold value of 40% of the nominal speed (frequency) of the motor in the above example, the gate drive unit transitions operation of the inverter from standard multi-level (three-level in the illustrated embodiments) operation to two-level modulation, and when frequency ω of the motor stator coils increases above a threshold value of 40% of the nominal speed (frequency) the gate drive unit transitions operation of the inverter from two-level modulation back to standard multi-level (three-level in the illustrated embodiments) operation. Thus, when the operating parameter is in a first range on one side of a threshold value, in which the neutral point is stable (either below or above the threshold value depending on the parameter), the inverter operates in standard multi-level modulation mode, and if the operating parameter value crosses the threshold value to the other side constituting a second range in which the neutral point is potentially unstable, the inverter mode of operation switches to the two level modulation mode. It may be noted that, in embodiments, the transition threshold values of the parameters used to control operation of the inverter between two-level and multi-level modulation patterns, may be different depending on the direction of the transition to ensure a partial overlap of ranges of values determining the transition. In particular, the transition from two-level to multi-level modulation may have a different threshold value than the threshold value for the transition from multi-level modulation to two-level modulation such that the range of operation in two-level mode overlaps partially the range of operation in three or high level operation mode in order to avoid instability around the transition value. For instance, the threshold value of the stator frequency ω may be 40% of the nominal motor frequency in the direction of decreasing frequency for transition from three to two-level modulation, whereas the threshold value of the stator frequency ω may be 45% of the nominal motor frequency in the direction of increasing frequency for transition from two to three-level modulation. The aforegoing feature may apply to any one or more of the values having a threshold to control the transition from one modulation pattern to the other modulation pattern Any parameter or combination of parameters that are indicative of a deviation or an instability of the neutral point voltage beyond desired amplitudes for accurate control of the system (e.g. electrical motor) may be used to set transition values for changing the modulation pattern of the inverter. In embodiments of an electrical motor connected to an inverter, the preferred parameters for determining the modulation pattern may include: neutral point potential deviation Vnp, stator frequency ω of the motor, power factor cos(ϕ) of the motor, phase current magnitude Iphand PWM frequency fpwm. All of these preferred parameters may be used to control the operation mode of the inverter between three-level (multi-level) or two-level as illustrated in the flow diagram ofFIG.3. It may however be noted that only one, or only a subset of the preferred parameters may be used to control the operation mode of the inverter between three-level (multi-level) or two-level. Moreover, as mentioned above, other parameters that are indicative of a deviation or an instability of the neutral point voltage beyond desired amplitudes may be employed to control the operation mode of the inverter between multi-level and two-level operation. Furthermore, in variants, two or more of the control parameters may also be combined, for instance a ratio or a multiplication of parameters to form a composite value, for instance a ratio of phase current magnitude with respect to PWM frequency Iph/fpwmor a product of the phase current magnitude and the PWM frequency Iph×fpwmAnother example would be a weighted sum between phase current magnitude, PWM frequency and stator frequency, the factors kxare scaling factors k1*Iph+k2*fpwm+k3*ω. | 14,051 |
11863099 | DETAILED DESCRIPTION Embodiment 1 FIG.1is a block diagram schematically illustrating a configuration of an air conditioner100using a refrigeration cycle, according to Embodiment 1. The air conditioner100includes an outdoor unit110installed outdoors, an indoor unit150installed indoors, and a remote controller160. FIG.2is a diagram illustrating a schematic configuration of the outdoor unit110. The outdoor unit110includes a motor111, an outdoor air conditioning unit114, and a driving device120. The motor111uses a plurality of coils to which three-phase AC voltages from the driving device120is applied, to produce a driving force for driving a specific portion included in the outdoor air conditioning unit114. The motor111receives the three-phase AC voltages applied from the driving device120, via a U-phase power line113U, a V-phase power line113V, and a W-phase power line113W. The motor111includes a U-phase coil112U connected to the U-phase power line113U, a V-phase coil112V connected to the V-phase power line113V, and a W-phase coil112W connected to the W-phase power line113W. The outdoor air conditioning unit114performs an outdoor operation in the refrigeration cycle. The outdoor air conditioning unit114includes devices such as a compressor114a, an outdoor heat exchanger114b, and an outdoor fan114c. The compressor114aobtains the driving force from the motor111to compress a refrigerant used in the refrigeration cycle. The outdoor heat exchanger114bperforms a heat exchange of the refrigerant. The outdoor fan114cserves as a fan which blows air to the outdoor heat exchanger114bfor the outdoor unit110. The driving device120is a device for controlling each part of the outdoor unit110and driving the motor111. The driving device120includes a power supply121, a reactor122, a converter123, an inverter126, a connection switching unit128, a current detection circuit134, a controller135, and a communication unit136. The power supply121serves as an AC power supply which outputs an AC voltage. The converter123receives the AC voltage from the power supply121via the reactor122, performs rectifying, smoothing and the like with respect to the AC voltage and thus generates a DC voltage. The converter123includes bridge diodes124A to124D which rectify the AC voltage, and a smoothing capacitor125which smooths an output voltage. The inverter126receives the DC voltage input from the converter123, generates three-phase AC voltages from the DC voltage and outputs the generated three-phase AC voltages to the motor111. The inverter126includes a first U-phase switching element126Ua, a second U-phase switching element126Ub, a first V-phase switching element126Va, a second V-phase switching element126Vb, a first W-phase switching element126Wa, and a second W-phase switching element126Wb which are connected in three phase bridge of the U, V, and W phases. The first U-phase switching element126Ua corresponds to a U-phase upper arm, and the second U-phase switching element126Ub corresponds to a U-phase lower arm. The first U-phase switching element126Ua and the second U-phase switching element126Ub are connected to the U-phase power line113U. A first U-phase diode127Ua is connected in parallel with the first U-phase switching element126Ua, and a second U-phase diode127Ub is connected in parallel with the second U-phase switching element126Ub. The first V-phase switching element126Va corresponds to a V-phase upper arm, and the second V-phase switching element126Vb corresponds to a V-phase lower arm. The first V-phase switching element126Va and the second V-phase switching element126Vb are connected to the V-phase power line113V. A first V-phase diode127Va is connected in parallel with the first V-phase switching element126Va, and a second V-phase diode127Vb is connected in parallel with the second V-phase switching element126Vb. The first W-phase switching element126Wa corresponds to a W-phase upper arm, and the second W-phase switching element126Wb corresponds to a W-phase lower arm. The first W-phase switching element126Wa and the second W-phase switching element126Wb are connected to the W-phase power line113W. A first W-phase diode127Wa is connected in parallel with the first W-phase switching element126Wa, and a second W-phase diode127Wb is connected in parallel with the second W-phase switching element126Wb. Each of the switching elements126Ua to126Wb can be implemented as a transistor such as IGBT (Insulated Gate Bipolar Transistor). ON or OFF of each of the switching elements126Ua to126Wb is controlled in accordance with driving signals DS from the controller135. The connection switching unit128switches the connection states of the plurality of coils of the motor111between a first connection state and a second connection state. The second connection state is defined herein as a state in which the line voltage of the inverter126is lower than that in the first connection state. For example, the first connection state is defined as the Y-connection state, and the second connection state is defined as the Δ-connection state. The connection switching unit128includes a U-phase switch129U, a V-phase switch129V, and a W-phase switch129W. The U-phase switch129U serves as a switching unit which switches the connection destination of the U-phase coil112U. The V-phase switch129V serves as a switching unit which switches the connection destination of the V-phase coil112V. The W-phase switch129W serves as a switching unit which switches the connection destination of the W-phase coil112W. FIG.3is a schematic diagram illustrating the connection state between the motor111and the connection switching unit128. The U-phase coil112U has one end112Ua connected to the U-phase power line113U and the other end112Ub connected to a common contact130U of the U-phase switch129U. A first switching contact131U of the U-phase switch129U is connected to a first switching contact131V of the V-phase switch129V and a first switching contact131W of the W-phase switch129W. A second switching contact132U of the U-phase switch129U is connected to the V-phase power line113V. The U-phase switch129U can switch connections to the common contact130U between the first switching contact131U and the second switching contact132U. The V-phase coil112V has one end112Va connected to the V-phase power line113V and the other end112Vb connected to a common contact130V of the V-phase switch129V. The first switching contact131V of the V-phase switch129V is connected to the first switching contact131U of the U-phase switch129U and the first switching contact131W of the W-phase switch129W. A second switching contact132V of the V-phase switch129V is connected to the W-phase power line113W. The V-phase switch129V can switch connections to the common contact130V between the first switching contact131V and the second switching contact132V. The W-phase coil112W has one end112Wa connected to the W-phase power line113W and the other end112Wb connected to a common contact130W of the W-phase switch129W. The first switching contact131W of the W-phase switch129W is connected to the first switching contact131U of the U-phase switch129U and the first switching contact131V of the V-phase switch129V. A second switching contact132W of the W-phase switch129W is connected to the U-phase power line113U. The W-phase switch129W can switch connections to the common contact130W between the first switching contact131W and the second switching contact132W. Since the connection switching unit128is configured as above, the motor111can be set in the Y-connection state in which the other end112Ub of the U-phase coil112U, the other end112Vb of the V-phase coil112V, and the other end112Wb of the W-phase coil112W are connected to each other, by connecting the common contact130U to the first switching contact131U in the U-phase switch129U, connecting the common contact130V to the first switching contact131V in the V-phase switch129V, and connecting the common contact130W to the first switching contact131W in the W-phase switch129W. The motor111can be set in the Δ-connection state in which the end112Ua of the U-phase coil112U and the other end112Wb of the W-phase coil112W are connected to the U-phase power line113U, the end112Va of the V-phase coil112V and the other end112Ub of the U-phase coil112U are connected to the V-phase power line113V, and the end112Wa of the W-phase coil112W and the other end112Vb of the V-phase coil112V are connected to the W-phase power line113W, by connecting the common contact130U to the second switching contact132U in the U-phase switch129U, connecting the common contact130V to the second switching contact132V in the V-phase switch129V, and connecting the common contact130W to the second switching contact132W in the W-phase switch129W. The U-phase switch129U, the V-phase switch129V, and the W-phase switch129W can individually switch the connection destinations of the common contacts130U to130W between the first switching contacts131U to131W and the second switching contacts132U to132W in accordance with switching signals CSU, CSV, and CSW from the controller135. The U-phase switch129U, the V-phase switch129V, and the W-phase switch129W are specified as c-contact switches, but they are not limited to such an example. The U-phase switch129U, the V-phase switch129V, and the W-phase switch129W need only be implemented as bidirectionally openable and closable switches. For example, each of the U-phase switch129U, the V-phase switch129V, and the W-phase switch129W may be implemented as a combination of a-contact switches or a combination of b-contact switches, or may be implemented as a semiconductor switch. For the U-phase switch129U, the V-phase switch129V, and the W-phase switch129W, a switch of a low conduction loss in turn-on is preferable and mechanical switches such as relays or contactors can be used. However, the use of switching elements employing a WBG (Wide Band Gap) semiconductor such as SiC or GaN as these switches allows low ON resistances, low losses, and less element heat generation. Especially when the connection states are switched during driving, these switches are preferably made up of semiconductors. A structure in which the connection state is the Y-connection state when a normally-on element of the semiconductor is ON-state can reduce the loss on the light load (Y-connection), and the structure suitable for the air conditioner100which has a high contribution rate in the light load. To return toFIG.2, when the inverter126is switched, a shunt resistor133converts a current flowing through buses L1and L2into a voltage proportional to the current and transmits the voltage to the current detection circuit134. The current detection circuit134serves as a current detection unit which detects the value of the current on the input side of the inverter126. In Embodiment 1, the current detection circuit134detects the value of the bus current (input current) of the inverter126, but the detection of the current value is not limited to such an example. For example, the value of the current on the input side of the inverter126may be calculated by using the phase current of the inverter126. The controller135serves as a control unit which controls each part of the outdoor unit110. The controller135controls, in particular, the outdoor air conditioning unit114, the inverter126, and the connection switching unit128. The controller135detects an abnormality of the connection switching unit128. For example, the controller135controls the inverter126and the connection switching unit128, and detects an abnormality of the connection switching unit128, based on the current value detected by the current detection circuit134. For example, the controller135. A method for detecting an abnormality by the controller135will be described later. When the controller135detects an abnormality of the connection switching unit128, it notifies the indoor unit150that the abnormality has been detected, via the communication unit136. The controller135can individually control the U-phase switch129U, the V-phase switch129V, and the W-phase switch129W. For example, the controller135is connected to the connection switching unit128with three control lines which are a U-phase control line135U for controlling the U-phase switch129U, a V-phase control line135V for controlling the V-phase switch129V, and a W-phase control line135W for controlling the W-phase switch129W. The controller135can control switching of the U-phase switch129U by transmitting a U-phase switching signal CSU for controlling the U-phase switch129U to the U-phase switch129U through the U-phase control line135U. Similarly, the controller135can control switching of each of the V-phase switch129V and the W-phase switch129W by transmitting a V-phase switching signal CSV to the V-phase switch129V through the V-phase control line135V and transmitting a W-phase switching signal CSW to the W-phase switch129W through the W-phase control line135W. The communication unit136communicates with the indoor unit150. For example, the communication unit136transmits a notification signal indicating that an abnormality of the connection switching unit128has been detected, to the indoor unit150in accordance with an instruction from the controller135. When the outdoor unit110is connected to the indoor unit150with three lines, and no communication line is included, the communication unit136performs communication with power lines. When the outdoor unit110is connected to the indoor unit150with four lines, and a communication line is included, the communication unit136performs communication with the communication line. The connection between the outdoor unit110and the indoor unit150may be wireless connection or connection with a dedicated wire and the like, and the communication unit136may perform communication using such connection. FIG.4is a block diagram schematically illustrating a configuration of the indoor unit150in Embodiment 1. The indoor unit150includes an indoor air conditioning unit151, a first communication unit152, a second communication unit153, a display unit154, and a control unit155. The indoor air conditioning unit151performs an indoor operation in the refrigeration cycle. The indoor air conditioning unit151includes devices such as an indoor heat exchanger151aand an indoor fan151b. The indoor heat exchanger151aperforms heat exchange of the refrigerant. The indoor fan151bserves as a fan which blows air to the indoor heat exchanger151afor the indoor unit150. The first communication unit152communicates with the outdoor unit110. The second communication unit153communicates with the remote controller160or a smartphone161serving as a user terminal. The display unit154displays the content of instructions issued from the control unit155. The control unit155controls each part of the indoor unit150. The control unit155notifies a user that an abnormality of the connection switching unit128has been detected, when the first communication unit152receives the notification signal from the outdoor unit110. For example, the control unit155notifies the user by performing at least one of display on the display unit154, display on the remote controller160, and display on the smartphone161. More specifically, the control unit155instructs the display unit154to display information indicating that the abnormality of the connection switching unit128has been detected. The control unit155can also cause the remote controller160to display information indicating that the abnormality of the connection switching unit128has been detected, by causing the second communication unit153to transmit, to the remote controller160, a specific signal indicating that the abnormality of the connection switching unit128has been detected. In this case, the second communication unit153can be implemented by, for example, a communication interface using infrared rays. The control unit155can even cause the smartphone161to display information indicating that the abnormality of the connection switching unit128has been detected, by causing the second communication unit153to transmit, to the smartphone161, notification data indicating that the abnormality of the connection switching unit128has been detected. In this case, the second communication unit153can be implemented by a wireless LAN (Local Area Network) communication interface. The remote controller160functions as an input reception unit which receives input of various instructions. For example, the remote controller160receives input to start the operation of the air conditioner100. The smartphone161may function as the input reception unit. Part or the whole of the controller135of the outdoor unit110and the control unit155of the indoor unit150described above can be implemented by, for example, a memory10, and a processor11such as a CPU (Central Processing Unit) which executes a program stored in the memory10, as illustrated inFIG.5(A). The program may be provided via a network or with a recording medium recording the program. Part or the whole of the controller135and the control unit155can also be implemented by, for example, a processing circuit12such as a single circuit, a complex circuit, a programmed processor, a parallel-programmed processor, ASICs (Application Specific Integrated Circuits), or an FPGA (Field Programmable Gate Array), as illustrated inFIG.5(B). An operation for detecting an abnormality of the connection switching unit128by the controller135of the outdoor unit110will be described below. FIG.6is a flowchart illustrating an abnormality detection sequence by the controller135in Embodiment 1. Assume herein that the connection switching unit128is set in a Y-connection state or a Δ-connection state. First, the controller135obtains, as a first current value, a current value detected by the current detection circuit134when a voltage is applied to only the U-phase power line113U and the V-phase power line113V (S10). For example, the controller135applies a voltage to only the U-phase power line113U and the V-phase power line113V by transmitting driving signals DS to the inverter126to turn on only the first U-phase switching element126Ua and the second V-phase switching element126Vb. In this case, for example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, if the connection switching unit128has no abnormality, a current flows through the U-phase coil112U and the V-phase coil112V. The value of the bus current in this case is detected as the first current value. Then, the controller135obtains, as a second current value, a current value detected by the current detection circuit134when a voltage is applied to only the V-phase power line113V and the W-phase power line113W (S11). For example, the controller135applies a voltage to only the V-phase power line113V and the W-phase power line113W by transmitting driving signals DS to the inverter126to turn on only the first V-phase switching element126Va and the second W-phase switching element126Wb. In this case, for example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, if the connection switching unit128has no abnormality, a current flows through the V-phase coil112V and the W-phase coil112W. The value of the bus current in this case is detected as the second current value. The controller135obtains as a third current value, a current value detected by the current detection circuit134when a voltage is applied to only the W-phase power line113W and the U-phase power line113U (S12). For example, the controller135applies a voltage to only the W-phase power line113W and the U-phase power line113U by transmitting driving signals DS to the inverter126to turn on only the first W-phase switching element126Wa and the second U-phase switching element126Ub. In this case, for example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, if the connection switching unit128has no abnormality, a current flows through the W-phase coil112W and the U-phase coil112U. The value of the bus current in this case is detected as the third current value. The controller135detects the presence or absence of an abnormality of the connection switching unit128by comparing the first current value, the second current value, and the third current value (S13). A method for detecting the presence or absence of an abnormality of the connection switching unit128by the controller135will be described below with reference toFIGS.7to10. The case where the connection switching unit128is in the Y-connection state, as illustrated inFIG.2, and has no abnormality will be described below with reference toFIG.7. Symbols inFIG.7will be described here. Symbol UP denotes a driving signal DS for the first U-phase switching element126Ua; symbol VP denotes a driving signal DS for the first V-phase switching element126Va; and symbol WP denotes a driving signal DS for the first W-phase switching element126Wa. Symbol UN denotes a driving signal DS for the second U-phase switching element126Ub; symbol VN denotes a driving signal DS for the second V-phase switching element126Vb; and symbol WN denotes a driving signal DS for the second W-phase switching element126Wb. Symbol Iu denotes a current value of the U-phase power line113U assuming that the direction from the inverter126to the motor111is positive; symbol Iv denotes a current value of the V-phase power line113V assuming that the direction from the inverter126to the motor111is positive; and symbol Iw denotes a current value of the W-phase power line113W assuming that the direction from the inverter126to the motor111is positive. Symbol I denotes the value of the bus current detected by the current detection circuit134. As illustrated inFIG.7, when the first U-phase switching element126Ua and the second V-phase switching element126Vb are turned on at time t1, a current flows through the U-phase coil112U and the V-phase coil112V, and the current detection circuit134detects the current as the first current value. When the first V-phase switching element126Va and the second W-phase switching element126Wb are turned on at time t2, a current flows through the V-phase coil112V and the W-phase coil112W, and the current detection circuit134detects the current as the second current value. When the first W-phase switching element126Wa and the second U-phase switching element126Ub are turned on at time t3, a current flows through the W-phase coil112W and the U-phase coil112U, and the current detection circuit134detects the current as the third current value. The first current value, the second current value, and the third current value may be peak or average values. As illustrated inFIG.7, when the connection switching unit128has no abnormality, the first current value, the second current value, and the third current value are nearly equal values. A first case where the connection switching unit128is in the Y-connection state, as illustrated inFIG.2, and has an abnormality will be described below with reference toFIG.8.FIG.8illustrates that the U-phase switch129U has opening failure. Symbols inFIG.8are the same as those inFIG.7. As illustrated inFIG.8, even when the first U-phase switching element126Ua and the second V-phase switching element126Vb are turned on at time t4, if the U-phase switch129U is open, no current flows through the U-phase coil112U and the V-phase coil112V. Accordingly, the current detection circuit134does not detect the first current value. When the first V-phase switching element126Va and the second W-phase switching element126Wb are turned on at time t5, a current flows through the V-phase coil112V and the W-phase coil112W, and the current detection circuit134detects the current as the second current value. Even when the first W-phase switching element126Wa and the second U-phase switching element126Ub are turned on at time t6, if the U-phase switch129U is open, no current flows through the W-phase coil112W and the U-phase coil112U. Accordingly, the current detection circuit134does not detect the third current value. Hence, as illustrated inFIG.8, when the connection switching unit128has opening failure, two of the first current value, the second current value, and the third current value are not detected. A second case where the connection switching unit128is in the Y-connection state, as illustrated inFIG.2, and has an abnormality will be described below with reference toFIG.9.FIG.9illustrates that the U-phase switch129U has short-circuit failure on the side of the second switching contact132U, as illustrated inFIG.10. Symbols inFIG.9are the same as those inFIG.7. As illustrated inFIG.9, when the first U-phase switching element126Ua and the second V-phase switching element126Vb are turned on at time t7, if the U-phase switch129U has short-circuited on the side of the second switching contact132U, a current flows through the U-phase coil112U while no current flows through the V-phase coil112V. Accordingly, the current detection circuit134detects the current value obtained when a current flows through only the U-phase coil112U as the first current value. When the first V-phase switching element126Va and the second W-phase switching element126Wb are turned on at time t8, a current flows through the V-phase coil112V and the W-phase coil112W, and the current detection circuit134detects the current as the second current value. When the first W-phase switching element126Wa and the second U-phase switching element126Ub are turned on at time t9, if the U-phase switch129U has short-circuited on the side of the second switching contact132U, a current flows through the W-phase coil112W and the U-phase coil112U and also flows through the V-phase coil112V. Accordingly, the current detection circuit134detects the current value obtained when a current flows through the U-phase coil112U, the V-phase coil112V, and the W-phase coil112W as the third current value. Hence, as illustrated inFIG.9, when the connection switching unit128has short-circuit failure, at least one of the first current value, the second current value, and the third current value is detected as a value different from another one. As described above, the controller135compares the first current value, the second current value, and the third current value with each other. Therefore, for example, it is possible to determine that the connection switching unit128has no abnormality if the absolute values of the differences between these values are equal to or smaller than a predetermined threshold, and determine that the connection switching unit128has an abnormality if the absolute values of the differences between these values are larger than the predetermined threshold or if there is any current value that cannot be detected in these values. Although pulse signals are used as the driving signals DS for the inverter126inFIGS.7to9, the use of, for example, PWM signals having a fixed duty ratio as the driving signals DS allows the current detection circuit134to detect current values which depend not on the inductances of the U-phase coil112U, the V-phase coil112V, and the W-phase coil112W, but on the resistances. In this case, the first current value, the second current value, and the third current value are desirably average values. In the flowchart illustrated inFIG.6, the first current value is detected by applying a voltage to the U-phase power line113U and the V-phase power line113V, the second current value is detected by applying a voltage to the V-phase power line113V and the W-phase power line113W, and the third current value is detected by applying a voltage to the W-phase power line113W and the U-phase power line113U, but this Embodiment is not limited to such an example. For example, one of the U-phase power line113U, the V-phase power line113V, and the W-phase power line113W is set as a first line, another is set as a second line, and the other is set as a third line. the first current value may be detected by applying a voltage to only the first line and the second line by the inverter126, the second current value may be detected by applying a voltage to only the second line and the third line by the inverter126, and the third current value may be detected by applying a voltage to only the third line and the first line by the inverter126. In such a case, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the first line is set as a first coil, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the second line is set as a second coil, and a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the third line is set as a third coil. A switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the first coil is set as a first switching unit, a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the second coil is set as a second switching unit, and a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the third coil is set as a third switching unit. When the controller135detects an abnormality of the connection switching unit128, it can notify a user and take measures according to the failure mode. When, for example, the connection switching unit128has opening failure, the controller135stops (halts) the operation of the air conditioner100. When the connection switching unit128has short-circuit failure at a switching contact on the Y-connection side, the controller135drives the air conditioner100in the Y-connection state. When the connection switching unit128has short-circuit failure at a switching contact on the Δ-connection side, the controller135drives the air conditioner100in the Δ-connection state. By such driving, the air conditioner100can even wait for repair by support, while continuing its operation. Next, the timing of performing an abnormality detection sequence by the controller135will be described. When the connection switching unit128is broken, this results in an abnormal connection state, and other circuits may also be broken due to over current, excessive heat generation, or the like. When one continues the operation without being aware of step-out, the compressor114amay stall. Hence, the controller135desirably executes the abnormality detection sequence before the compressor114ais started. For example, as illustrated inFIG.11, when an operation instruction signal from the remote controller160is input to the indoor unit150(T00), the indoor fan151bstarts to be driven (T01), the outdoor fan114cof the outdoor unit110starts to be driven (T02), and the compressor114aof the outdoor unit110is driven (T04). In this case, the controller135desirably executes an abnormality detection sequence at time T03between time T02and time104. The remote controller160receives from a user input to start the operation of the air conditioner100and then transmits an operation instruction signal to the indoor unit150. In this way, lengthening the start-up time of the compressor114acan be suppressed. An abnormality detection sequence can be executed at any timing with no problem as long as this is done before the start-up of the compressor114a, but in order not to hear much noise, this is done desirably after the start of driving the indoor fan151band more desirably after the start of driving the outdoor fan114c. This can keep the volume of a sound generated by the abnormality detection sequence relatively low among unit driving sounds. As illustrated inFIG.12, in the case where an operation instruction signal from the remote controller160is input to the indoor unit150(T10), the indoor fan151bstarts to be driven (T12), the outdoor fan114cof the outdoor unit110starts to be driven (T13), and the compressor114aof the outdoor unit110is driven (T14), the abnormality detection sequence may be executed immediately after the operation instruction signal from the remote controller160is input to the indoor unit150. More specifically, the abnormality detection sequence may be executed in response to the reception, by the remote controller160, of input to start the operation of the air conditioner100from the user. This makes it possible to detect an abnormal state and notify a user while the user focuses on the remote controller160or the indoor unit150. As described above, according to Embodiment 1, an abnormality of the connection switching unit128can be easily detected. Embodiment 2 An air conditioner200according to Embodiment 2 includes an outdoor unit210, an indoor unit150, and a remote controller160, as illustrated inFIG.1. The indoor unit150and the remote controller160of the air conditioner200according to Embodiment 2 are the same as the indoor unit150and the remote controller160according to Embodiment 1. The outdoor unit210according to Embodiment 2 includes a motor111, an outdoor air conditioning unit114, and a driving device220, as illustrated inFIG.2. The motor111and the outdoor air conditioning unit114of the outdoor unit210according to Embodiment 2 are the same as the motor111and the outdoor air conditioning unit114according to Embodiment 1. The driving device220in Embodiment 2 includes a power supply121, a reactor122, a converter123, an inverter126, a connection switching unit128, a shunt resistor133, a current detection circuit134, a controller235, and a communication unit136. The driving device220in Embodiment 2 is the same as the driving device120in Embodiment 1, except for the controller235. The controller235in Embodiment 2 serves as a control unit which controls each part of the outdoor unit210. The controller235controls, in particular, the outdoor air conditioning unit114, the inverter126, and the connection switching unit128. The controller235in Embodiment 2 is different from the controller135in Embodiment 1 in terms of the sequence of detecting an abnormality of the connection switching unit128by controlling the inverter126and the connection switching unit128. FIG.13is a flowchart illustrating an abnormality detection sequence by the controller235in Embodiment 2. First, the controller235obtains, as a first current value, a current value detected by the current detection circuit134when the connection switching unit128is set in the Y-connection state and a voltage is applied to only the U-phase power line113U and the V-phase power line113V (S20). Then, the controller235obtains, as a second current value, a current value detected by the current detection circuit134when the U-phase switch129U of the connection switching unit128is switched to the second switching contact132U and a voltage is applied to only the U-phase power line113U and the V-phase power line113V (S21). For example, the controller235switches the U-phase switch129U to the second switching contact132U by transmitting a U-phase switching signal CSU to the connection switching unit128through the U-phase control line135U. The controller235applies a voltage to only the U-phase power line113U and the V-phase power line113V by transmitting driving signals DS to the inverter126to turn on only the first U-phase switching element126Ua and the second V-phase switching element126Vb. The controller235determines the presence or absence of an abnormality of the U-phase switch129U by comparing the first current value and the second current value (S22). In step S20, if the U-phase switch129U has no abnormality, a current flows through the U-phase coil112U and the V-phase coil112V. In step S21, if the U-phase switch129U has no abnormality, a current flows through the U-phase coil112U. This means that the first current value is expected to be about half the second current value. Therefore, when the first current value is different from the second current value, the controller235can determine that the U-phase switch129U has no abnormality. Note that the controller235can determine that the first current value is different from the second current value, based on whether the absolute value of the difference of between the values is larger than a predetermined threshold. The controller235obtains, as a third current value, a current value detected by the current detection circuit134when the connection switching unit128is set in the Y-connection state and a voltage is applied to only the V-phase power line113V and the W-phase power line113W (S23). The controller235obtains, as a fourth current value, a current value detected by the current detection circuit134when the V-phase switch129V of the connection switching unit128is switched to the second switching contact132V and a voltage is applied to only the V-phase power line113V and the W-phase power line113W (S24). For example, the controller235switches the V-phase switch129V to the second switching contact132V by transmitting a V-phase switching signal CSV to the connection switching unit128through the V-phase control line135V. The controller235applies a voltage to only the V-phase power line113V and the W-phase power line113W by transmitting driving signals DS to the inverter126to turn on only the first V-phase switching element126Va and the second W-phase switching element126Wb. The controller235determines the presence or absence of an abnormality of the V-phase switch129V by comparing the third current value and the fourth current value (S25). In step S23, if the V-phase switch129V has no abnormality, a current flows through the V-phase coil112V and the W-phase coil112W. In step S25, if the V-phase switch129V has no abnormality, a current flows through the V-phase coil112V. Therefore, when the third current value is different from the fourth current value, the controller235can determine that the V-phase switch129V has no abnormality. A method for this determination is the same as that in step S22. The controller235obtains, as a fifth current value, a current value detected by the current detection circuit134when the connection switching unit128is set in the Y-connection state and a voltage is applied to only the W-phase power line113W and the U-phase power line113U (S26). The controller235obtains, as a sixth current value, a current value detected by the current detection circuit134by switching the W-phase switch129W of the connection switching unit128to the second switching contact132W and applying a voltage to only the W-phase power line113W and the U-phase power line113U (S27). For example, the controller235switches the W-phase switch129W to the second switching contact132W by transmitting a W-phase switching signal CSW to the connection switching unit128through the W-phase control line135W. The controller235applies a voltage to only the W-phase power line113W and the U-phase power line113U by transmitting driving signals DS to the inverter126to turn on only the first W-phase switching element126Wa and the second U-phase switching element126Ub. The controller235determines the presence or absence of an abnormality of the W-phase switch129W by comparing the fifth current value and the sixth current value (S28). In step S26, if the W-phase switch129W has no abnormality, a current flows through the W-phase coil112W and the U-phase coil112U. In step S27, if the W-phase switch129W has no abnormality, a current flows through the W-phase coil112W. Therefore, when the fifth current value is different from the sixth current value, the controller235can determine that the W-phase switch129W has no abnormality. A method for this determination is the same as that in step S22. In the flowchart illustrated inFIG.13, the first current value and the second current value are detected by applying a voltage to the U-phase power line113U and the V-phase power line113V, the third current value and the fourth current value are detected by applying a voltage to the V-phase power line113V and the W-phase power line113W, and the fifth current value and the sixth current value are detected by applying a voltage to the W-phase power line113W and the U-phase power line113U, but this Embodiment is not limited to such an example. For example, one of the U-phase power line113U, the V-phase power line113V, and the W-phase power line113W is set as a first line, another is set as a second line, and the other is set as a third line. The first current value and the second current value may be detected by applying a voltage to only the first line and the second line by the inverter126, the third current value and the fourth current value may be detected by applying a voltage to only the second line and the third line by the inverter126, and the fifth current value and the sixth current value may be detected by applying a voltage to only the third line and the first line by the inverter126. In this case, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the first line is set as a first coil, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the second line is set as a second coil, and a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the third line is set as a third coil. A switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the first coil is set as a first switching unit, a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the second coil is set as a second switching unit, and a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the third coil is set as a third switching unit. As described above, according to Embodiment 2, an abnormality of the connection switching unit128can be more accurately detected. The present invention is not limited to above-described Embodiments 1 and 2. For example, the sequence illustrated inFIG.6is executed in one of the Y-connection state and the Δ-connection state in Embodiment 1, but an abnormality of the connection switching unit128can be more accurately detected by executing, for example, the sequence illustrated inFIG.6in one of the Y-connection state and the Δ-connection state, and executing the sequence illustrated inFIG.6in the other state when an abnormality is detected. When the sequence illustrated inFIG.6is executed in the other state, the current value detected in step S10is obtained, as a fourth current value, the current value detected in step S11is obtained, as a fifth current value, and the current value detected in step S12is obtained, as a sixth current value. In this case, a first threshold and a second threshold (first threshold<second threshold) can also be prepared in advance as thresholds used in step S13ofFIG.6, so that the controller135determines that the connection switching unit128is normal when the absolute value of the difference is equal to or smaller than the first threshold, determines that the connection switching unit128has an abnormality when the absolute value of the difference is larger than the second threshold, and executes the sequence illustrated inFIG.6in the other state when the absolute value of the difference is larger than the first threshold and equal to or smaller than the second threshold. The sequence illustrated inFIG.13may even be executed when an abnormality is detected in the sequence illustrated inFIG.6. Embodiment 3 An air conditioner300according to Embodiment 3 includes an outdoor unit310, an indoor unit150, and a remote controller160, as illustrated inFIG.1. The indoor unit150and the remote controller160of the air conditioner300according to Embodiment 3 are the same as the indoor unit150and the remote controller160according to Embodiment 1. The outdoor unit310according to Embodiment 3 includes a motor111, an outdoor air conditioning unit114, and a driving device320, as illustrated inFIG.2. The motor111and the outdoor air conditioning unit114of the outdoor unit310according to Embodiment 3 are the same as the motor111and the outdoor air conditioning unit114according to Embodiment 1. The driving device320in Embodiment 3 includes a power supply121, a reactor122, a converter123, an inverter126, a connection switching unit128, a shunt resistor133, a current detection circuit134, a controller335, and a communication unit136. The driving device320in Embodiment 3 is the same as the driving device120in Embodiment 1, except for the controller335. The controller335in Embodiment 3 serves as a control unit which controls each part of the outdoor unit310. The controller335controls, in particular, the outdoor air conditioning unit114, the inverter126, and the connection switching unit128. The controller335in Embodiment 3 is different from the controller135in Embodiment 1 in terms of the sequence of detecting an abnormality of the connection switching unit128by controlling the inverter126and the connection switching unit128. FIG.14is a flowchart illustrating an abnormality detection sequence by the controller335in Embodiment 3. Assume here that the connection switching unit128is set in the Y-connection state or the Δ-connection state. First, the controller335obtains, as a first current value, a current value detected by the current detection circuit134when a voltage is applied to supply a current in only the direction from the U-phase power line113U to the V-phase power line113V and the W-phase power line113W (S30). For example, the controller335applies a voltage to supply a current in only the direction from the U-phase power line113U to the V-phase power line113V and the W-phase power line113W, by transmitting driving signals DS to the inverter126to turn on only the first U-phase switching element126Ua, the second V-phase switching element126Vb, and the second W-phase switching element126Wb. In this case, for example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, as long as the connection switching unit128has no abnormality, a current flows through the U-phase coil112U, the V-phase coil112V, and the W-phase coil112W. The value of the bus current in this case is detected as the first current value. Then, the controller335obtains as a second current value, the current value detected by the current detection circuit134when a voltage is applied to supply a current in only the direction from the V-phase power line113V to the U-phase power line113U and the W-phase power line113W (S31). For example, the controller335applies a voltage to supply a current in only the direction from the V-phase power line113V to the U-phase power line113U and the W-phase power line113W, by transmitting driving signals DS to the inverter126to turn on only the first V-phase switching element126Va, the second U-phase switching element126Ub, and the second W-phase switching element126Wb. In this case, for example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, as long as the connection switching unit128has no abnormality, a current flows through the U-phase coil112U, the V-phase coil112V, and the W-phase coil112W. The value of the bus current in this case is detected as the second current value. The controller335obtains as a third current value, the current value detected by the current detection circuit134when a voltage is applied to supply a current in only the direction from the W-phase power line113W to the U-phase power line113U and the V-phase power line113V (S32). For example, the controller335applies a voltage to supply a current in only the direction from the W-phase power line113W to the U-phase power line113U and the V-phase power line113V, by transmitting driving signals DS to the inverter126to turn on only the first W-phase switching element126Wa, the second U-phase switching element126Ub, and the second V-phase switching element126Vb. In this case, for example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, if the connection switching unit128has no abnormality, a current flows through the U-phase coil112U, the V-phase coil112V, and the W-phase coil112W. The value of the bus current in this case is detected as the third current value. The controller335detects the presence or absence of an abnormality of the connection switching unit128by comparing the first current value, the second current value, and the third current value with each other (S33). A method for this comparison is the same as that in Embodiment 1. For example, as illustrated inFIG.2, when the connection switching unit128is in the Y-connection state, if the connection switching unit128has no abnormality, the first current value, the second current value, and the third current value are equal values (normal values). When, for example, the U-phase switch129U has opening failure, no current flows through the U-phase coil112U, the V-phase coil112V, and the W-phase coil112W in step S30. Accordingly, the current detection circuit134detects no first current value. When the U-phase switch129U has opening failure, a current flows through the V-phase coil112V and the W-phase coil112W while no current flows through the U-phase coil112U in step S31. The second current value is tree-fourths times the normal value. When the U-phase switch129U has opening failure, a current flows through the V-phase coil112V and the W-phase coil112W while no current flows through the U-phase coil112U in step S32. The third current value is tree-fourths times the normal value. When, for example, the U-phase switch129U has short-circuit failure on the side of the second switching contact132U, a current flows through the U-phase coil112U while little current flows through the V-phase coil112V and the W-phase coil112W in step S30. The first current value is three-seconds times the normal value. When the U-phase switch129U has short-circuit failure on the side of the second switching contact132U, only the U-phase coil112U is present between the V-phase power line113V and the U-phase power line113U and only the V-phase coil112V and the W-phase coil112W are present between the V-phase power line113V and the W-phase power line113W in step S31. Therefore, the second current value is nine-fourths times the normal value. When the U-phase switch129U has short-circuit failure on the side of the second switching contact132U, a current flows through only the V-phase coil112V and the W-phase coil112W in step S32. The third current value is three-fourths times the normal value. As described above, by comparing the first current value, the second current value, and the third current value, for example, the controller335can determine that the connection switching unit128has no abnormality when the absolute values of the difference between these values are equal to or smaller than a predetermined threshold, and determine that the connection switching unit128has an abnormality when the absolute values of the difference between these values are larger than the predetermined threshold or there is any current value that cannot be detected in these values. In the flowchart illustrated inFIG.14, the first current value is detected by applying a voltage to supply a current in only the direction from the U-phase power line113U to the V-phase power line113V and the W-phase power line113W, the second current value is detected by applying a voltage to supply a current in only the direction from the V-phase power line113V to the U-phase power line113U and the W-phase power line113W, and the third current value is detected by applying a voltage to supply a current in only the direction from the W-phase power line113W to the U-phase power line113U and the V-phase power line113V, but this Embodiment is not limited to such an example. For example, one of the U-phase power line113U, the V-phase power line113V, and the W-phase power line113W is set as a first line, another one of them is set as a second line, and the remaining one of them is set as a third line. the first current value may be detected by applying a voltage to supply a current in only the first direction from the first line to the second line and the third line by the inverter126, the second current value may be detected by applying a voltage to supply a current in only the second direction from the second line to the first line and the third line by the inverter126, and the third current value may be detected by applying a voltage to supply a current in only the third direction from the third line to the first line and the second line by the inverter126. As another example, the first current value may be detected by applying a voltage to supply a current in only the first direction from the first line and the second line to the third line by the inverter126, the second current value may be detected by applying a voltage to supply a current in only the second direction from the second line and the third line to the first line by the inverter126, and the third current value may be detected by applying a voltage to supply a current in only the third direction from the first line and the third line to the second line by the inverter126. In these cases, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the first line is set as a first coil, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the second line is set as a second coil, and a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the third line is set as a third coil. A switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the first coil is set as a first switching unit, a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the second coil is set as a second switching unit, and a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the third coil is set as a third switching unit. Embodiment 4 An air conditioner400according to Embodiment 4 includes an outdoor unit410, an indoor unit150, and a remote controller160, as illustrated inFIG.1. The indoor unit150and the remote controller160of the air conditioner400according to Embodiment 4 are the same as the indoor unit150and the remote controller160according to Embodiment 1. The outdoor unit410according to Embodiment 4 includes a motor111, an outdoor air conditioning unit114, and a driving device420, as illustrated inFIG.2. The motor111and the outdoor air conditioning unit114of the outdoor unit410according to Embodiment 4 are the same as the motor111and the outdoor air conditioning unit114according to Embodiment 1. The driving device420in Embodiment 4 includes a power supply121, a reactor122, a converter123, an inverter126, a connection switching unit128, a shunt resistor133, a current detection circuit134, a controller435, and a communication unit136. The driving device420in Embodiment 4 is the same as the driving device120in Embodiment 1, except for the controller435. The controller435in Embodiment 4 serves as a control unit which controls each part of the outdoor unit410. The controller435controls, in particular, the outdoor air conditioning unit114, the inverter126, and the connection switching unit128. The controller435in Embodiment 4 is different from the controller135in Embodiment 1 in terms of the sequence of detecting an abnormality of the connection switching unit128by controlling the inverter126and the connection switching unit128. FIG.15is a flowchart illustrating an abnormality detection sequence by the controller435in Embodiment 4. First, the controller435obtains as, a first current value, a current value detected by the current detection circuit134when the connection switching unit128is set in the Y-connection state and a voltage is applied to supply a current in only the direction from the U-phase power line113U to the V-phase power line113V and the W-phase power line113W (S40). Then, the controller435obtains, as a second current value, a current value detected by the current detection circuit134when the U-phase switch129U of the connection switching unit128is switched to the second switching contact132U and a voltage is applied to supply a current in only the direction from the U-phase power line113U to the V-phase power line113V and the W-phase power line113W (S41). The controller435determines the presence or absence of an abnormality of the U-phase switch129U by comparing the first current value and the second current value (S42). For example, when the first current value is different from the second current value, the controller435can determine that the U-phase switch129U has no abnormality. Note that the controller435can determine that the first current value is different from the second current value, based on whether the absolute value of the difference between these values is larger than a predetermined threshold. The controller435obtains, as a third current value, a current value detected by the current detection circuit134when the connection switching unit128is set in the Y-connection state and a voltage is applied to supply a current in only the direction from the V-phase power line113V to the U-phase power line113U and the W-phase power line113W (S43). The controller435obtains, as a fourth current value, a current value detected by the current detection circuit134when the V-phase switch129V of the connection switching unit128is switched to the second switching contact132V and a voltage is applied to supply a current in only the direction from the V-phase power line113V to the U-phase power line113U and the W-phase power line113W (S44). The controller435determines the presence or absence of an abnormality of the V-phase switch129V by comparing the third current value and the fourth current value (S45). For example, when the third current value is different from the fourth current value, the controller435can determine that the V-phase switch129V has no abnormality. A method for this determination is the same as that in step S42. The controller435obtains, as a fifth current value, a current value detected by the current detection circuit134when the connection switching unit128is set in the Y-connection state and a voltage is applied to supply a current in only the direction from the W-phase power line113W to the U-phase power line113U and the V-phase power line113V (S46). The controller435obtains, as a sixth current value, a current value detected by the current detection circuit134by switching the W-phase switch129W of the connection switching unit128to the second switching contact132W and applying a voltage to supply a current in only the direction from the W-phase power line113W to the U-phase power line113U and the V-phase power line113V (S47). The controller435determines the presence or absence of an abnormality of the W-phase switch129W by comparing the fifth current value and the sixth current value (S48). For example, when the fifth current value is different from the sixth current value, the controller435can determine that the W-phase switch129W has no abnormality. A method for this determination is the same as that in step S42. In the flowchart illustrated inFIG.15, the first current value and the second current value are detected by applying a voltage to supply a current in only the direction from the U-phase power line113U to the V-phase power line113V and the W-phase power line113W, the third current value and the fourth current value are detected by applying a voltage to supply a current in only the direction from the V-phase power line113V to the U-phase power line113U and the W-phase power line113W, and the fifth current value and the sixth current value are detected by applying a voltage to supply a current in only the direction from the W-phase power line113W to the U-phase power line113U and the V-phase power line113V, but this Embodiment is not limited to such an example. For example, one of the U-phase power line113U, the V-phase power line113V, and the W-phase power line113W is set as a first line, another one of them is set as a second line, and the remaining one of them is set as a third line. The first current value and the second current value may be detected by applying a voltage to supply a current in only the first direction from the first line to the second line and the third line by the inverter126, the third current value and the fourth current value may be detected by applying a voltage to supply a current in only the second direction from the second line to the first line and the third line by the inverter126, and the fifth current value and the sixth current value may be detected by applying a voltage to supply a current in only the third direction from the third line to the first line and the second line by the inverter126. As another example, the first current value and the second current value may be detected by applying a voltage to supply a current in only the first direction from the first line and the second line to the third line by the inverter126, the third current value and the fourth current value may be detected by applying a voltage to supply a current in only the second direction from the second line and the third line to the first line by the inverter126, and the fifth current value and the sixth current value may be detected by applying a voltage to supply a current in only the third direction from the first line and the third line to the second line by the inverter126. In these cases, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the first line is set as a first coil, a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the second line is set as a second coil, and a coil (112U,112V, or112W) having one end (112Ua,112Va, or112Wa) connected to the third line is set as a third coil. A switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the first coil is set as a first switching unit, a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the second coil is set as a second switching unit, and a switch (129U,129V, or129W) which switches the connection destination of the other end (112Ub,112Vb, or112Wb) of the third coil is set as a third switching unit. The present invention is not limited to above-described Embodiments 3 and 4. For example, the sequence illustrated inFIG.14is executed in one of the Y-connection state and the Δ-connection state in Embodiment 3, but an abnormality of the connection switching unit128can be more accurately detected by executing, for example, the sequence illustrated inFIG.14in one of the Y-connection state and the Δ-connection state, and executing the sequence illustrated inFIG.14in the other state when an abnormality is detected. When the sequence illustrated inFIG.14is executed in the other state, the current value detected in step S30is obtained as a fourth current value, the current value detected in step S31is obtained as a fifth current value, and the current value detected in step S32is obtained as a sixth current value. In this case, a first threshold and a second threshold (first threshold<second threshold) can also be prepared in advance as thresholds used in step S33ofFIG.14, so that the controller335determines that the connection switching unit128is normal when the absolute value of the difference is equal to or smaller than the first threshold, determines that the connection switching unit128has an abnormality when the absolute value of the difference is larger than the second threshold, and executes the sequence illustrated inFIG.14in the other state when the absolute value of the difference is larger than the first threshold and equal to or smaller than the second threshold. The sequence illustrated inFIG.15may even be executed when an abnormality is detected in the sequence illustrated inFIG.14. As described above, according to Embodiments 1 to 4, an abnormality of the connection switching unit128can be easily detected based on the current value detected by the current detection circuit134. Executing the abnormality detection sequence before driving the compressor114acan prevent failure of the air conditioner100or200such as a stall of the compressor114a. Executing the abnormality detection sequence after driving the indoor fan151ballows the sound of the indoor fan151bto drown out a sound occurring in the abnormality detection sequence, so that a user in the room does not feel discomfort. Executing the abnormality detection sequence after driving the outdoor fan114callows the sound of the outdoor fan114clocated outdoors to drown out a sound occurring outdoors in the abnormality detection sequence, so that a user in the room does not feel discomfort. Executing the abnormality detection sequence in response to reception, by the remote controller160, of input to start the operation can notify a user of an abnormality immediately after the operation of the remote controller160, so that it is possible to provide the user with such a notification reliably. In such a case, executing the abnormality detection sequence before driving the indoor fan151bmakes it possible to provide the user with the notification more reliably, when an abnormality occurs. According to Embodiment 1, since it is only necessary to compare a plurality of current values detected in the first connection state or the second connection state, an abnormality of the connection switching unit128can be easily detected. Furthermore, for example, when, in the power-OFF of the air conditioner100, connection switching unit128is in one of the first connection state and the second connection state, an abnormality of the connection switching unit128can be detected without operating the connection switching unit128by executing the abnormality detection sequence in this state. Therefore, a waste of power due to switching operation can be reduced and shortening the life due to an increase in the number of operation of the connection switching unit128can also be reduced. When an abnormality is detected in the abnormality detection sequence executed in one of the first connection state and the second connection state, an abnormality of the connection switching unit128can be more reliably detected by executing the abnormality detection sequence in the other state. Since the U-phase switch129U, the V-phase switch129V, and the W-phase switch129W can be operated individually, the power supply capacity required for the switching operation can be reduced compared to the case where all the switches are operated simultaneously, and a low-capacity power supply circuit can be used. This can curb the raise in cost due to increase in the capacity of the power supply circuit. A place where a failure has occurred can be more accurately detected by individually operating these switches. The presence or absence of an abnormality can be more reliably detected by abnormality detection based on the current values detected before and after switching of the U-phase switch129U, the V-phase switch129V, or the W-phase switch129W. The use of semiconductor switches, especially switching elements made by using a WBG semiconductor, as the U-phase switch129U, the V-phase switch129V, and the W-phase switch129W allows low ON resistances, low losses, and reduction in element heat generation. | 67,951 |
11863100 | The diagrams depicted herein are illustrative. There can be many variations to the diagram or the operations described therein without departing from the spirit of the disclosure. For instance, the actions can be performed in a differing order or actions can be added, deleted or modified. In the accompanying figures and following detailed description of the described embodiments, the various elements illustrated in the figures are provided with two or three-digit reference numbers. With minor exceptions, the leftmost digit(s) of each reference number corresponds to the figure in which its element is first illustrated. DETAILED DESCRIPTION Disclosed herein is an induction machine (it should be noted that the terms “machine,” “motor,” and “induction machine” may be used interchangeably herein) with localized voltage unbalance compensation (VUC). In particular, the present disclosure may relate to a VUC device that may be used in conjunction with the induction machine and may correct for a voltage unbalance in the induction machine resulting from a voltage source feeding the induction machine. For example, the VUC may correct for the voltage unbalance in the induction machine by providing a supplemental voltage to compensate for a voltage unbalance. The VUC may be connected to the machine in parallel. This may allow the VUC to receive any voltages being applied to the induction machine by the voltage source in parallel with the induction machine so that the VUC may identify voltage unbalances in voltages being received by the induction machine. The VUC may improve the functionality of the machine by maintaining proper working conditions of the machine during intervals of voltage unbalance. Also disclosed herein is a control method for operating the VUC as well as simulation results that confirm the validity of the control method. The results show that the VUC may allow a machine to maintain performance with balanced, rated voltage even during most voltage-unbalanced conditions, as well as conditions of symmetrical voltage sags or surges. To ensure that the VUC is able to compensate for an unbalance when a source voltage drops to a low value, the VUC may be provided with additional power so that a thyristor bridge (described below) may have enough power to allow the VUC to function. In some embodiments, the VUC may also provide additional benefits beyond the benefits described above. For example, the VUC may eliminate harmonic current flow within the machine. This may be accomplished by modification of control signals fed to the VUC, and may be nearly a zero cost addition, which may add significant value to the overall concept. The VUC may also improve a power factor as viewed at the terminals of the machine (e.g., by supplying a leading voltage in series with the machine). The power factor of squirrel cage induction machines operating without a VUC may normally be poor, typically 0.8-0.9. This, in turn, may cause the current required to power the machine to rise to values between 10 and 20% beyond is optimal value at unity power factor. While operation at these lagging power factors are inherent to the machine torque production, the 10-20% increase in current may cause the power utility to encounter a corresponding loss in their distribution, transformers, and distribution lines as a result. Consequently, industries are typically charged a penalty for a power factor poorer than unity. However, by use of the VUC, the machine can be controlled to present itself as a unity power factor load to the utility regardless of the loading condition of the motor. Again, this feature can be incorporated into the VUC circuit simply by modification of the VUC digital controller at nearly zero additional cost. In addition, the presence of a VUC can be used to ease the shock of starting the machine by reducing inrush current. The VUC may also only need to supply the voltage difference between the desired voltage and the actual unbalanced voltage. Hence, the compensation range for the unbalance voltages may be wide compared against typical unbalance correcting systems. Furthermore, because the VUC is in parallel with the induction machine, the induction machine may essentially function the same as if the VUC were not present, which results in ease in implementing the VUC into induction machines. In some embodiments, the VUC may provide even further benefits in that it may only need to be rated only at a fraction of the rating of the induction machine. For example, in the case of a 5% voltage unbalance, the volt-ampere rating of the VUC needs to be only rated at roughly 5% that of the induction machine rating that it is protecting. Thus, the VUC may be a low-cost means of maintaining 100% motor load capability even in the presence of a 5% voltage unbalance. The circuit principle is not limited to a 5% unbalance but can correct for as much as a 25% voltage unbalance or more. Embodiments of the disclosure are described more fully hereinafter with reference to the accompanying drawings, in which example embodiments of the disclosure are shown. This disclosure may, however, be embodied in many different forms and should not be construed as limited to the example embodiments set forth herein; rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the disclosure to those skilled in the art. Like numbers refer to like, but not necessarily the same or identical, elements throughout. The following embodiments are described in sufficient detail to enable at least those skilled in the art to understand and use the disclosure. It is to be understood that other embodiments would be evident based on the present disclosure and that process, mechanical, material, dimensional, process equipment, and parametric changes may be made without departing from the scope of the present disclosure. In the following description, numerous specific details are given to provide a thorough understanding of various embodiments of the disclosure. However, it will be apparent that the disclosure may be practiced without these specific details. Moreover, to avoid obscuring the present disclosure, some well-known system configurations and process steps may not be disclosed in full detail. Likewise, the drawings showing embodiments of the disclosure are semi-diagrammatic and not to scale, and, particularly, some of the dimensions may be exaggerated in the drawings for the clarity of presentation. In addition, where multiple embodiments are disclosed and described as having some features in common, for clarity and ease of illustration, description, and comprehension thereof, similar and like features will ordinarily be described with like reference numerals even if the features are not identical. “An embodiment,” “various embodiments,” and the like indicate embodiment(s) so described may include particular features, structures, or characteristics, but not every embodiment necessarily includes the particular features, structures, or characteristics. Some embodiments may have some, all, or none of the features described for other embodiments. “First,” “second,” “third,” and the like describe a common object and indicate different instances of like objects are being referred to. Such adjectives do not imply objects so described must be in a given sequence, either temporally, spatially, in ranking, or in any other manner. “Connected” may indicate elements are in direct physical or electrical contact with each other, and “coupled” may indicate elements co-operate or interact with each other, but they may or may not be in direct physical or electrical contact. Also, while similar or same numbers may be used to designate same or similar parts in different figures, doing so does not mean all figures, including similar or same numbers, constitute a single or same embodiment. The terms “perpendicular,” “orthogonal,” “coplanar,” and/or “parallel” may mean substantially perpendicular, orthogonal, coplanar, or parallel, respectively. For example, “perpendicular” can mean perpendicular within ±20, 15, 10, or 5 degrees. Further, the figures shown herein may not have precisely vertical or horizontal edges, but rather may have some finite slope and have surface roughness, as is to be expected for fabricated devices. The terms “about,” “substantially,” “approximately,” and variations thereof, are intended to include a degree of error associated with a measurement of the particular quantity using equipment available at the time of filing the application. For example, “about” can include a range of ±8% or 5%, or 2% of a given value. Example Systems and Associated Architecture Turning to the figures,FIG.1depicts a circuit diagram for a machine102equipped with a Voltage Unbalance Compensator (VUC)101in accordance with one or more example embodiments of the disclosure. In some instances, the machine102may be an induction machine. The VUC101may include a Pulse Width Modulation (PWM) inverter106, thyristor bridge110, and DC link capacitor and discharge circuit108. The VUC101may be connected in parallel with stator windings of the machine102. The VUC101may also be connected to a control system (e.g., control system200, shown inFIG.2). In some embodiments, the VUC101may serve to correct for voltage unbalances originating from a voltage source112. The voltage source112may provide a three-phase voltage to a first input114of the machine102. In some instances, a voltage unbalance may mean that the voltages on each line of the three-phase voltage applied to the first input114of the machine102may not be the same. Voltage unbalances at the first input114of the machine102may result in suboptimal functionality of the machine102. For example, a voltage unbalance may result in a significant increase in an amount of current applied to the machine102, which may potentially be damaging to the machine102. The VUC101may mitigate the impact of such voltage unbalances from the voltage source112. To accomplish this, the VUC101may be connected to the machine102in parallel such that the voltage received from the voltage source112by the machine102may also be received by the VUC101. Based on this received voltage from the voltage source112, the VUC101may determine the amount of voltage unbalance from the voltage source112(e.g., voltage unbalance across any of the three phases of the three-phase voltage) and produce a compensating output voltage to a second input116of the machine102to partially or fully correct for the unbalanced voltage (e.g., three-phase voltage) provided to the first input114of the machine102by the voltage source112. At least some of the circuit elements of the VUC101may be described in further detail with reference toFIG.3below. FIGS.2A-2Bshows a control system200that may be used to maintain balanced voltages (e.g., remove voltage unbalanced as described above) across the windings of the machine102. The control block diagram for the VUC101may be comprised of three parts: line voltage control, direct current (DC) link voltage control, and discharge control, however, in some embodiments, any other number or combination of controller portions may be used as well. The line voltage control may operate the compensating voltage for unbalanced voltages. The DC link voltage control may maintain the DC link voltage for driving the inverter. The discharge controller may work to discharge the DC link capacitor to suppress an increase of the DC link voltage. FIG.3Adepicts a system300(which may be a simplified version of system100) to illustrate the basic principle of the control system200as presented inFIGS.2A-2B. For example, system300may include a voltage source312, a machine302, and/or a VUC301, which may be the same as voltage source112, machine102, and/or VUC101. The system300may also include a thyristor bridge306and/or a PWM inverter308. InFIG.2, it may be shown that the PWM inverter308may be used to supply a voltage set on one side of the stator phases of the machine302. As a result, the PWM inverter308may only need supply the a voltage that is equivalent to the difference between a desired voltage (balanced voltage) and an identified unbalanced voltage. Thus, the voltage rating of the DC link and PWM inverter switches may only be a small fraction of the voltage rating of the machine and need only be equal, roughly, to the maximum value of voltage unbalance to be corrected. In some embodiments, a thyristor bridge306may be used to feed the PWM inverter308and may convert the input AC voltage to a controlled DC link voltage. A DC link capacitor discharge circuit may be used to discharge the DC link capacitor when a voltage unbalance occurs, which exceeds the rated voltage of the thyristor bridge306. If desired, a second PWM inverter also can be used instead of the thyristor bridge306, thereby eliminating the need for the discharge circuit. In addition, the switching loss also would increase compared to a thyristor bridge306. Because the discharge circuit may only functions when an unbalance voltage occurs significantly beyond the rated voltage of the machine302, its size may remain relatively small. FIG.3Bmay depict a system350, which may be another simplified version of the system100depicted inFIG.1. As withFIG.3A,FIG.3Bmay depict a voltage source312, a machine302, and/or a VUC301.FIG.3Bmay serve as a reference point for how various variables with respect to system100may be determined. For example, iu, iv, and iwmay represent currents being provided to the induction machine, and may be determined by Equations 1, 2, and 3. E0may be the neutral point voltage described as Equation 4. Quantities va′, vb′ and vc′ may be the output voltages of the PWM inverter (e.g., PWM inverter308described with respect toFIG.3Aas well as any other PWM inverter described herein). Quantities va′, vb′and vc′may be voltages provided by the voltage source312. iu={va−(va′+E0)}/Zm(Equation 1) iv={vb−(vb′+E0)}/Zm(Equation 2) iw={vc−(vc′+E0)}/Zm(Equation 3) E0=(va/Zm+vb/Zm+vc/Zm)/(1/Zm+1/Zm+1/Zm)=⅓(va+vb+vc) (Equation 4) Ideal phase currents iui, iviand iwimay be those that exist when the induction machine is driven directly by balanced voltage sources. They may be derived as Equation 5, Equation 6, and Equation 7. Note that vai, vbiand vcimay be ideal phase voltages. iui=vai/Zm(Equation 5) ivi=vbi/Zm(Equation 6) iwi=vci/Zm(Equation 7) From the above equations, output phase voltages of the PWM inverter may be calculated as: va′=va−E0−vai(Equation 8) vb′=vb−E0−vbi(Equation 9) vc′=vc−E0−vci(Equation 10) If these output phase voltages are used as command values by the control system200, and are inserted into Equations 1, 2, and 3, ideal motor current values may be obtained as Equations 5, 6, and 7. However, these command values may not be able to be utilized because the neutral voltage E0may not be known. The neutral voltage E0may often be measured by accessing the neutral point voltage of machine. However, one may not be able to access the neutral point of the voltage sources. Hence, phase voltage control may not be able to be employed, and line voltage control may have to be used in order to avoid needing to know the neutral voltage. The command output line voltage of inverter vab′, vbc40and vca′ can be derived as Equations 11, 12, and 13 from the subtracting Equations 8−9, 9−10, and 10−8, respectively. vab′=vab−vabi(Equation 11) vbc′=vbc−vbci(Equation 12) vca′=vca−vcai(Equation 13) The vab, vbcand vcamay now the source line voltages. The quantities vabi, vbciand vcaimay be ideal line voltages assuming line voltages are balanced. Note that the neutral voltage E0may not be required in Equations 11, 12, and 13. Though the PWM inverter may only be able to supply the line voltage, the command phase voltage va′, vb′ and vc′ may still be required to control the output line voltage of the inverter. Therefore, the command output phase voltage of the inverter va′ may be calculated from the results of PI control of the output line voltage about vab′ and vabout. The command output line voltage of the inverter vab′ and measured output line voltage of the inverter vaboutmay be used for line voltage PI control. Then the output of PI control may be named as vacom, the vacommay be used as the command output phase voltage of inverter va′. From the equations that relate the line voltage and phase voltage, the command output phase voltage of inverter may be calculated, so that finally: va′=vacom(Equation 14) vc′=vca′+vacom(Equation 15) vb′=vbc′+vc′=vbc′+vca′+vacom(Equation 16) The command output phase voltages may be made by the command output line voltage of inverter and output of the PI control. Note that a coordinate transform may be used to accomplish the PI control because a dq-axes representation is utilized, as shown theFIG.3A. The control may require that the two input waveforms have a 90-degree phase difference. The ideal line voltages vabi, vbciand vcaimay be obtained from the measured line voltage of tri-phase sinusoidal voltage sources during balance voltage. The line voltages may be measured and stored as ideal dq-axes values during periods of balanced voltage. Consequently, the ideal line voltages vabi, vbciand vcaimay be calculated based on these stored values. Four voltage sensors may be needed to control the VUC301. Two voltage sensors may be used for measuring the line voltages of the polyphase voltage sources. Note that summation of the line voltages may be equal to zero even during unbalance voltage, so that if two line voltages are measured by voltage sensor, the remaining one can be calculated based on these measured values. One voltage sensor may be used for measuring the output line voltage of the inverter vabout. Lastly, one voltage sensor may be used for measuring the DC link voltage. It may be used for both line voltage control and DC link voltage control. Next, DC link voltage control is described. The DC link voltage may be controlled by a thyristor bridge. The command of DC link voltage may be expressed by: vdc_c=3/π×√{square root over (2)}×vab(Equation 17) The command value of the ideal DC link voltage may be calculated from the output of the diode bridge during conditions of balanced voltage. The command of phase angle of the thyristor bridge may be determined by the output of the DC link voltage PI control, as shown theFIG.2. When the DC link voltage control fails because the input line voltage drops due to an unbalanced voltage condition, the command phase angle may be set to zero so that the thyristor bridge then simply operates as a three-phase diode bridge. The discharge circuit may only operate when the DC link voltage increases beyond the DC link voltage limitation Vdclimit_c. Here, the limitation value may be set from Equation 17 to be the rated voltage of the DC link capacitor Vdcrated. The limitation value may be expressed in Equation 18. The coefficient of kdcmay represent the voltage range used in the VUC. Vdclimit_c=kdc×Vdcrated(Equation 18) Simulation Results FIGS.4-9Bpertain to simulations used to validate the effectiveness of the VUC as described above.FIG.4illustrates the basic simulation strategy. The total simulation time may be set to 12 seconds, and a balanced voltage period may occur from 0s to 5s (for example a period of the simulation during which a voltage provided to an induction machine are balanced). The unbalanced voltage period may start from 5s and may end at 12s (for example, this may represent a period of time during which a voltage imbalance is reduced to test the effectiveness of the VUC). The induction machine with the VUC may be operated from 1s to 12s. The rated operation for the induction machine starts from 3.5s to 12s. Before it starts to drive, initial charging for the DC link capacitor may be finished charging during the first second. The command ideal line voltages may also be computed and stored during one second to drive the VUC. Two simulation conditions were conducted to investigate the operation of the induction machine with the VUC. Simulation I shows a case where only one phase voltage is changed and the voltage unbalance ratio is changed. Simulation II is a condition in which two or three-phase source voltages are changed, and corresponding unbalanced voltages occur. For the purpose of simulation, a 15 kW class induction machine model was used and was selected as a middle-capacity size induction machine. The induction machine parameters were derived based on a NEMA B model. Table 1 lists the induction machine parameters used in the study. TABLE IINDUCTION MACHINE PARAMETERSParametersValuesRated power Pout[kW]15Rated voltage Va[V]240Rated current Ia[A]44Pole number—4Rated frequency fe[Hz]60Stator resistance Rs[Ω]0.115Rotor resistance Rr[Ω]0.092Magnetizing inductance Lm[mH]25Leakage inductance of stator L1s[mH]0.885Leakage inductance of rotor L1s[mH]0.885Leakage impedance of stator X1s[Ω]0.334Leakage impedance of rotor X1r[Ω]0.334Magnetizing impedance Xm[Ω]9.427 Voltage sources were chosen to also consider the influence of the impedance of the power grid. The impedances of the power grid were assumed balanced, and that changing the phase voltages only causes unbalanced voltages in the simulation and not the impedances. Table 2 lists the detail of the simulation conditions for simulation case I and simulation case II. FIG.5shows the simulation results when only the source phase voltage a is dropped to 70%. Note that the motor currents, iu, ivand iwremain balanced even during the voltage unbalance. After examining a multitude of cases, it was clear that the VUC can compensate for any type of single-phase unbalanced voltage. From the results, the inverter of compensator is able to supply the desired line voltages following the commanded line voltages of the inverter. In addition, the DC link voltage was also controlled constantly by using the thyristor bridge. TABLE IISIMULATION CONDITIONSParametersValuesAnalysis total time12 sTime step6.7 usSimulation I: InputTwo phase voltage: 138.6 V,phase voltage va, vb, vconly one phasevoltage va: unbalanced(i) Only Phase a: 207.8 V(150%)(ii) Only Phase a: 180.1 V(130%)(iii) Only Phase a: 152.4 V(110%)(iv) Only Phase a: 124.7 V(90%)(v) Only Phase a: 97.0 V(70%)(vi) Only Phase a: 83.1 V(60%)(vii) Only Phase a: 69.3 V(50%)(viii) Only Phase a: 55.4 V(40%)(ix) Only Phase a: 41.6 V(30%)Note: phase angle no change.Simulation II: Input phase(i) va= 124.7 V(90%),voltage va, vb, vcvb= 117.8 V(85%),Note: two or three phasevc= 138.6 V(100%)voltages are changed.(i) va= 124.7 V(90%),vb= 124.7 V(90%),vc= 117.8 V(85%)Note: phase angle no changeFrequency of input voltage60 HzPower grid impedance10 uH, 10 mΩLgand RgResistances for initial5 Ωcharge of DC linkcapacitator RinReactors connected to10 mH, 10 mΩthe Thyristor LTand RTReactor of discharge10 mH, 10 mΩcircuit Ldisand RdisDC link capacitator3000 uF, 10 mΩCdcand RdcInduction machine15 kW (parameters arelisted in Table 1)Switching frequency60 Hzof ThyristorSwitching frequency2 kHzof inverter (IGBT)Switching frequency2 kHzof discharge circuit(IGBT)Control method ofInverter: Line voltage PI controlthe subcircuitThyristor: DC linkvoltage PI controlDischarge circuit: Dischargecontrol of DClink capacitatorPeriod of voltage5 s to 12 sunbalancedPeriod of rated operation3.5 s to 12 sof the inductionmachineCommand of the DC link324 Vvoltage vdc_cLimitation of the DC link350 Vvoltage vdelimit_c FIG.6shows the simulation results for a voltage surge when phase a voltage is increased to about 110% of the rated voltage. As a result, voltage unbalance occurs again. However, the motor currents, iu, ivand iwremain balanced and the VUC is again able to compensate the unbalance voltages. The command output of line voltages of inverter and actual voltages across the motor phases match well. The DC link voltage was also controlled constantly by using the thyristor bridge and discharge circuit. FIG.7shows the motor currents (e.g., simulation I) when the voltage unbalance ratio was changed. From this result, the motor currents were again completely balanced during all of the simulation conditions. Furthermore, amplitude of motor currents was completely same as rated motor current. It means that the rated motor operation was not affected by the operation of the VUC. From the result, it was evident the validity of the control method of the VUC. However, when one of the phase voltages drops to about the 30% of the rated voltage, balance of the motor currents could not be maintained. This was caused by the fact that the VUC cannot supply the correct value of compensating voltages since the thyristor bridge is unable to receive enough power to drive the inverter to its desired value. FIGS.9aand9bshow the torque waveforms when the one of the phase voltages was changed to 70% and to 110% of rated voltage, respectively. Normal torque is also illustrated in the figure when the voltages maintain balance. From the results, solid-state compensator enables one to keep torque constant to within 4% rated torque. On the other hand, the torque fluctuates 23% and 9% rated torque, respectively, without the solid-state compensator. The effectiveness of the proposed approach is evident from these results. Various illustrative embodiments have been discussed above. These and other example embodiments of the disclosure will be described in more detail hereinafter through reference to the accompanying drawings. The drawings and the corresponding description are provided merely for illustration and are not intended to limit the disclosure in any way. It should be appreciated that numerous other embodiments, variations, and so forth are within the scope of this disclosure. Example Methodology FIG.10illustrates a flow diagram of an example process1000in accordance with one or more embodiments of the disclosure. In various embodiments, block1002, may comprise receiving an input voltage. In some embodiments, the input voltage may be received from a voltage source. The voltage source may provide a three-phrase voltage, for example, for use with a three-phrase machine (e.g., an induction machine, which may be the same as machine102described with respect toFIG.1above). The input voltage may be received by a machine, such as an induction machine. The input voltage may also be received by a voltage compensation device, which may be connected in parallel with the machine. The input voltage may be received by a first set of three-phase inputs on the machine. The machine may have a total of six inputs, which may allow the machine to receive two three-phase voltage inputs. In various embodiments, block1004may comprise determining a voltage unbalance of the input voltage, wherein the voltage unbalance comprises a difference between an actual voltage and a desired voltage. In some embodiments, the unbalance determination may be made by the voltage compensation device. For example, the voltage compensation device may comprise a control system capable of making such determinations (for example, the control system depicted inFIG.2). The voltage unbalance may exist across all three of the three-phase voltage lines, or may only exist on one of the voltage phase lines. In various embodiments, block1006may comprise providing an output voltage, wherein the output voltage is equivalent, or comparable to, the difference between the actual voltage and the desired voltage. In some embodiments, the output voltage may be provided by the voltage compensation device to the machine. The output voltage may be provided to a second set of three-phase inputs included on the machine. The output voltage may serve to compensate for the unbalances in the input voltage received by the machine in the first set of three-phase inputs. Thus, the combination of the input voltage and the output voltage from the voltage compensation device may result in a combined voltage that is equivalent to or close to the desired voltage of the machine. Many modifications and other implementations of the disclosure set forth herein will be apparent, having the benefit of the teachings presented in the foregoing descriptions and the associated drawings. Therefore, it is to be understood that the disclosure is not to be limited to the specific implementations disclosed and that modifications and other implementations are intended to be included within the scope of the appended claims. Although specific terms are employed herein, they are used in a generic and descriptive sense only and not for purposes of limitation. The disclosure is described above with reference to block and flow diagrams of systems, methods, apparatuses, and/or computer program products according to example embodiments of the disclosure. It will be understood that one or more blocks of the block diagrams and flow diagrams, and combinations of blocks in the block diagrams and the flow diagrams, respectively, can be implemented by computer-readable program instructions. Likewise, some blocks of the block diagrams and flow diagrams may not necessarily need to be performed in the order presented, or may not necessarily need to be performed at all, according to some embodiments of the disclosure. Various block and/or flow diagrams of systems, methods, apparatus, and/or computer program products according to example embodiments of the disclosure are described above. It will be understood that one or more blocks of the block diagrams and flow diagrams, and combinations of blocks in the block diagrams and flow diagrams, respectively, can be implemented by computer-readable program instructions. Likewise, some blocks of the block diagrams and flow diagrams may not necessarily need to be performed in the order presented, or may not necessarily need to be performed at all, according to some embodiments of the disclosure. Accordingly, blocks of the block diagrams and flow diagrams support combinations of means for performing the specified functions, combinations of elements or operations for performing the specified functions, and program instruction means for performing the specified functions. It will also be understood that each block of the block diagrams and flow diagrams, and combinations of blocks in the block diagrams and flow diagrams, can be implemented by special purpose, hardware-based computer systems that perform the specified functions, elements or operations, or combinations of special purpose hardware and computer instructions. Many modifications and other embodiments of the disclosure set forth herein will be apparent having the benefit of the teachings presented in the foregoing descriptions and the associated drawings. Therefore, it is to be understood that the disclosure is not to be limited to the specific embodiments disclosed and that modifications and other embodiments are intended to be included within the scope of the appended claims. Although specific terms are employed herein, they are used in a generic and descriptive sense only and not for purposes of limitation. | 31,181 |
11863101 | DETAILED DESCRIPTION Hereinafter, a driving apparatus and an air-conditioning apparatus according to each embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to the embodiments. First Embodiment FIG.1is a diagram illustrating an example configuration of a driving apparatus100according to a first embodiment of the present invention. The driving apparatus100is connected to a power source1and a permanent-magnet motor2. The driving apparatus100includes an inverter unit3that drives the permanent-magnet motor2, a control unit4that controls the inverter unit3, and a current detection unit5that detects a motor current flowing through the winding of the permanent-magnet motor2. The power source1may be a direct-current power supply including a cell, a battery, and the like, or may be an alternating-current-to-direct-current power converter including a known converter that converts an alternating-current voltage into a direct-current voltage.FIG.2is a diagram illustrating an example of the power source1connected to the driving apparatus100according to the first embodiment. As illustrated inFIG.2, the power source1may be an alternating-current-to-direct-current power converter1aincluding a known converter that converts an alternating-current voltage supplied from a three-phase alternating-current power supply11a, into a direct-current voltage.FIG.3is a diagram illustrating another example of the power source1connected to the driving apparatus100according to the first embodiment. As illustrated inFIG.3, the power source1may be an alternating-current-to-direct-current power converter1bincluding a known converter that converts an alternating-current voltage supplied from a single-phase alternating-current power supply11b, into a direct-current voltage. Although not illustrated, a booster circuit such as a well-known direct current (DC)-to-DC converter may be inserted into a direct-current bus between each of the alternating-current-to-direct-current power converters1aand1band the inverter unit3to boost a direct-current voltage. The permanent-magnet motor2is a motor including a permanent magnet. The permanent-magnet motor2is connected to the inverter unit3of the driving apparatus100. The inverter unit3converts a direct-current voltage supplied from the power source1into a three-phase alternating-current voltage for driving the permanent-magnet motor2at a desired frequency, that is, rotational speed, in accordance with a PWM signal for each phase. The PWM signal is a drive signal generated on the basis of a voltage command in the control unit4. The inverter unit3applies the three-phase alternating-current voltage to the permanent-magnet motor2. That is, the inverter unit3generates the three-phase alternating-current voltage from the direct-current voltage supplied from the power source1. The inverter unit3drives the permanent-magnet motor2in accordance with the drive signal from the control unit4. The inverter unit3includes six switching elements and six feedback diodes. The inverter unit3includes two switching elements connected in series to each other on a phase-by-phase basis, and a feedback diode connected in anti-parallel to each switching element. The inverter unit3is a known three-phase inverter that drives a general three-phase motor, and each switching element thereof performs a switching operation in accordance with a PWM signal for the corresponding phase. The switching element and the feedback diode may be an element including silicon, or an element including silicon carbide, gallium nitride, diamond, or the like, which is a wide bandgap semiconductor with high withstand voltage and capable of high-temperature operation. The current detection unit5detects a motor current flowing through the permanent-magnet motor2. The current detection unit5may be a known current sensor such as an alternating-current current transformer (ACCT) or a direct-current current transformer (DCCT) provided on wiring connected to the winding of the permanent-magnet motor2. Alternatively, the current detection unit5may be a known shunt resistor for phase current detection inserted in a bus connected to the power source1or switching elements on a negative side of the inverter unit3. The control unit4generates a voltage command to control an operation of the inverter unit3. The control unit4includes an overcurrent protection unit41and a magnet temperature estimation unit42. The overcurrent protection unit41performs a protection operation on the inverter unit3on the basis of a motor current flowing through the winding of the permanent-magnet motor2detected by the current detection unit5and an overcurrent protection threshold Ilim set by a method as described later. The magnet temperature estimation unit42estimates a magnet temperature Tmag of the permanent magnet of the permanent-magnet motor2. Although the present embodiment employs a configuration in which the overcurrent protection unit41and the magnet temperature estimation unit42are included in the control unit4, this is merely an example, and there is no limitation thereto. A configuration may be employed in which each component individually performs processing independently with a processor such as a microcomputer, and the control unit4, the overcurrent protection unit41, and the magnet temperature estimation unit42cooperate with one another only for signals necessary to perform the protection operation. Next, an operation of the driving apparatus100will be described. The control unit4generates a three-phase alternating-current voltage command necessary for driving the permanent-magnet motor2at a desired frequency, that is, at a desired rotational speed. Specifically, the control unit4executes well-known control computation such as feedforward control computation and vector control computation with a processor such as a microcomputer for each control computation period Δt in the control unit4, and generates a three-phase alternating-current voltage command of a desired frequency. Hereinafter, the desired frequency is defined as an output voltage frequency finv of the inverter unit3. Although not illustrated, in these control computations, the control unit4may use a motor current flowing through the winding of the permanent-magnet motor2detected by the current detection unit5, or may use information on a position or a speed detected by a position sensor or a speed sensor attached to the permanent-magnet motor2. Furthermore, the control unit4generates a carrier wave having a carrier frequency fc. The carrier frequency is a predetermined frequency or a frequency calculated on the basis of the output voltage frequency finv of the inverter unit3. That is, the control unit4sets the carrier frequency fc to a constant value or to a positive integral multiple of the output voltage frequency finv of the inverter unit3. The control unit4converts the three-phase alternating-current voltage command into a modulated wave, generates a PWM signal for each phase, which is a drive signal for the inverter unit3, on the basis of a result of magnitude comparison between the carrier wave and the modulated wave, and outputs the PWM signal to the inverter unit3. There are modes called an asynchronous PWM mode and a synchronous PWM mode for generating the carrier wave in the control unit4. The former is a mode in which the carrier frequency fc is set independently of the output voltage frequency finv, and the latter is a mode in which the carrier frequency fc is set to M times the absolute value of the output voltage frequency finv. Note that M as a parameter is a positive integer, and a multiple of 3 is mainly used as M. FIG.4is a diagram illustrating examples of a carrier wave generation mode in the control unit4according to the first embodiment. Specifically,FIG.4is a diagram illustrating an example of a relationship among a voltage phase of a voltage command, a carrier wave in the asynchronous PWM mode, a carrier wave of M=9, that is, a synchronous nine-pulse carrier wave in the synchronous PWM mode, and a carrier wave of M=6, that is, a synchronous six-pulse carrier wave in the synchronous PWM mode. Illustrated in a first stage ofFIG.4is the voltage command. Illustrated in second to fourth stages ofFIG.4are the carrier waves. For the asynchronous PWM mode, generally, a value of fc/finv is empirically set to approximately 9 or more, i.e., fc/finv≥9 in order to maintain the symmetry of the waveform of the three-phase alternating-current voltage output from the inverter unit3to reduce the deterioration of the driving performance due to distortion in the waveform of the voltage. The ratio fc/finv is a ratio between the carrier frequency fc and the output voltage frequency finv. In addition, since there is also an upper limit of the carrier frequency fc due to constraints such as loss and heat generation accompanying switching operations of the switching elements of the inverter unit3, there is also an upper limit of the output voltage frequency finv in order to maintain the relationship of fc/finv≥9. For the synchronous PWM mode, on the other hand, the value of fc/finv is a positive integer M, and even if fc/finv≤9 is satisfied, the symmetry of the waveform of the three-phase alternating-current voltage output from the inverter unit3is maintained. The synchronous PWM mode makes it possible to set a higher maximum value of the output voltage frequency finv than that in the asynchronous PWM mode even under the constraint on the upper limit of the carrier frequency fc, and achieve the driving even in a high speed zone where the driving is difficult in the asynchronous PWM mode. It is noted that the carrier frequency fc constantly changes depending on set values of the output voltage frequency finv and the positive integer M. In order to stably output the three-phase alternating-current voltage from the inverter unit3, there is also a lower limit of the carrier frequency fc. In an actual operation, thus, generally, the asynchronous PWM mode is switched to the synchronous PWM mode or vice versa, and the carrier frequency fc in the asynchronous PWM mode and the positive integer M in the synchronous PWM mode are set to appropriate values, for example, in correspondence to the output voltage frequency finv. The present embodiment is based on the assumption that the control unit4performs such operations as switching the PWM mode, and the positive integer M. FIG.5is a diagram illustrating an example of a relationship between the carrier wave and the control computation period Δt of the control unit4according to the first embodiment. The current detection unit5detects the motor current flowing through the winding of the permanent-magnet motor2at each of positive and negative maximum amplitude time points of the carrier wave, and outputs a detection value to the control unit4. The control unit4performs control computation for driving the permanent-magnet motor2on the basis of a current value detected by the current detection unit5, and compares the carrier wave with an instantaneous value of the voltage command for each phase obtained as a result of the control computation. The control unit4generates a PWM signal for each phase, which is a drive signal for the inverter unit3, on the basis of a result of the comparison, and performs a protection operation on the inverter unit3. The positive and negative maximum amplitude time points of the carrier wave illustrated inFIG.5correspond to a half period of the carrier wave. In addition, the positive and negative maximum amplitude time points are timings at which the switching operations of the switching elements of the inverter unit3are performed and the motor current flowing through the winding of the permanent-magnet motor2pulsates. Such timings are constantly away from intersections of the carrier wave and the modulated wave. The current detection unit5can detect the motor current without being affected by the pulsation of the motor current at each of the positive and negative maximum amplitude time points of the carrier wave illustrated inFIG.5. It is desirable for the control unit4to execute the control computation synchronizing the control computation period with a period of the carrier wave or a period which is an integral multiple of the period of the carrier wave. Alternatively, the control unit4desirably executes the control computation synchronizing the control computation period with a half period of the carrier wave or a period which is an integral multiple of the half period of the carrier wave. This makes it possible for the control unit4to use the current value detected by the current detection unit5, without being affected by the pulsation of the motor current. Consequently, the control unit4sets, on the basis of the carrier frequency fc, the control computation period Δt so that Δt=N/(2×fc) holds true in order to execute the control computation synchronizing the control computation period Δt with a period that is N/2 times the period of the carrier wave, that is, a period that is N times the half period of the carrier wave. That is, the control unit4sets the control computation period Δt to a positive integral multiple of a half of the carrier frequency fc. Note that N as a parameter is a positive integer. The control unit4executes the control computation every control computation period Δt. Thus, the control computation period Δt is a function of the carrier frequency fc, i.e., Δt=f(fc). That is, the control unit4sets the control computation period Δt in correspondence to the change in the carrier frequency fc.FIG.4also illustrates the relationship between the carrier wave and the voltage command to the inverter unit3in each of the PWM modes, and the length of the control computation period Δt varies depending on the PWM modes. FIG.5illustrates a relationship between the period of the carrier wave and the control computation period Δt in the case of N=1. It is noted that N is not necessarily limited to 1 (N=1), and a positive integer other than 1 may be selected as N, for example, N=2, 3, . . . may be selected. When the carrier frequency fc of the carrier wave is very high and the positive integer N is set to 1, the control unit4can fail to complete a process of the control computation within the set control computation period Δt, and the overflow can occur. When the carrier frequency fc is low and the value of the positive integer N is set to be large, the control computation period Δt may be so long that a desired control response may not be obtained. For these reasons, the control computation period Δt, that is, the positive integer N, is set on the basis of the carrier frequency fc in view of, for example, a processing load on the control computation, performance of the control unit4, and a control response required as a system. Next, an operation of the overcurrent protection unit41, which is a feature of the present embodiment, will be described. The overcurrent protection unit41determines whether there is an anomaly in an output current on the basis of the motor current flowing through the winding of the permanent-magnet motor2detected by the current detection unit5, and performs a protection operation on the inverter unit3. The motor current flowing through the winding of the permanent-magnet motor2is detected by the current detection unit5every control computation period Δt in synchronization with one or both of the positive and negative maximum amplitude time points of the carrier wave. The overcurrent protection unit41compares an absolute value of the current value detected by the current detection unit5with the overcurrent protection threshold Ilim set by a method described later. When the absolute value of the detected current value exceeds the overcurrent protection threshold Ilim, the overcurrent protection unit41determines that there is an anomaly. Desirably, the overcurrent protection unit41performs a series of operations from the current detection operation to the anomaly determination operation described above on a phase-by-phase basis, and performs the following protection operation on the inverter unit3when it is determined that there is an anomaly in any phase. As an example of the protection operation for the inverter unit3, the overcurrent protection unit41stops all the outputs of the PWM signals for the individual phases, which are drive signals for the inverter unit3. Alternatively, the overcurrent protection unit41outputs, to the inverter unit3, a signal that directly stops the switching operations of all the switching elements of the inverter unit3, thereby stopping the output of the three-phase alternating-current voltage from the inverter unit3to the permanent-magnet motor2. The overcurrent protection requires the overcurrent protection unit41to perform a protection operation on the inverter unit3immediately before an instantaneous value of the motor current exceeds a limit value, in order to prevent the motor current from exceeding the limit value when the motor current flowing through the winding of the permanent-magnet motor2increases due to a factor such as load fluctuation or control disturbance. The overcurrent protection threshold Ilim is set for this purpose. The above-described limit value needs to be set to a value that can prevent all phenomena which the inverter unit3or the permanent-magnet motor2may experience due to an overcurrent. These phenomena include a deterioration phenomenon such as dielectric breakdown due to heat generation when an overcurrent flows through the permanent-magnet motor2, and element destruction of the switching elements of the inverter unit3due to an overcurrent. The limit value is typically, a selected value which prevents occurrence of a phenomenon of possible overcurrent-caused phenomena, which phenomenon may occur due to a motor current having the smallest absolute value. The overcurrent protection threshold Ilim is set taking account of a margin relative to the limit value. In recent years, in many cases, a constraint due to irreversible demagnetization of the motor is strictest in considering the limit value because of, for example, an increase in the withstand voltage of a switching element, and critical design of a motor. FIG.6is a diagram illustrating an example of a relationship among the magnet temperature Tmag, a demagnetizing current Imag, and the overcurrent protection threshold Ilim according to the first embodiment.FIG.6is the diagram illustrating the example of the relationship among the magnet temperature Tmag, the demagnetizing current Imag, and the overcurrent protection threshold Ilim particularly in a case where an overcurrent protection function is designed on the basis of a constraint on a magnet of the permanent-magnet motor2in which the constraint due to the irreversible demagnetization of the motor is strictest. The diagram ofFIG.6is for high-temperature demagnetization in which demagnetization occurs as temperature increases, but the following idea can similarly apply also to low-temperature demagnetization differing in that the gradient of the demagnetizing current Imag with respect to the magnet temperature Tmag tends to increase. In the case where the magnet temperature Tmag of the permanent-magnet motor2is unknown, the overcurrent protection threshold Ilim is set as a constant value having a margin relative to a reference, or a demagnetizing current Imag that is minimum within an expected temperature range during operation of the permanent-magnet motor2. In this case, the margin is excessive for the protection of the inverter in the temperature zone, specifically, in a low-temperature zone ofFIG.6for the high-temperature demagnetization ofFIG.6. Such an excessive margin will limit the output performance of the permanent-magnet motor2more than necessary. In view of this, the magnet temperature estimation unit42estimates the magnet temperature Tmag through a method as described later, and the overcurrent protection unit41sets the overcurrent protection threshold Ilim to a value having a margin relative to the demagnetizing current Imag at the estimated magnet temperature Tmag. By changing the overcurrent protection threshold Ilim in correspondence to the magnet temperature Tmag, the overcurrent protection unit41can reduce an excess margin and extend the output performance of the permanent-magnet motor2. The overcurrent protection unit41intermittently compares the current detection value with the overcurrent protection threshold Ilim every control computation period Δt to determine necessity of the protection operation. This means that the longer the control computation period Δt is, the longer an interval between the determinations made by the overcurrent protection unit41is, which results in failure to perform the protection operation in time when a sudden change in the motor current occurs. FIG.7is a diagram illustrating an influence of a difference in the control computation period Δt of the overcurrent protection unit41of the control unit4according to the first embodiment on whether an overcurrent can be detected. InFIG.7, on condition that the positive integer N, which is a parameter for setting the control computation period Δt, is constant, the control computation period Δt is prolonged from a control computation period Δt1to a control computation period Δt2as the carrier frequency fc becomes lower, that is, a period of a carrier wave is prolonged. Note that the control computation period Δt1<the control computation period Δt2holds true. The carrier frequency fc becoming lower means the output voltage frequency finv of the inverter unit3becoming lower in the synchronous PWM mode. In the case ofFIG.7, when the permanent-magnet motor2changes as indicated by a thick solid line, a period from a certain current detection timing to the next current detection timing changes depending on the control computation period Δt. Therefore, when the control computation period Δt is the control computation period Δt1, the overcurrent protection unit41can detect where a motor current exceeds the overcurrent protection threshold Ilim, and perform a protection operation on the inverter unit3before the motor current reaches the demagnetizing current Imag. On the other hand, when the control computation period Δt is the lengthened control computation period Δt2, the motor current may exceed the demagnetizing current Imag at the next current detection timing of the overcurrent protection unit41. In summary, the different control computation periods are set in the case of the same current change in the permanent-magnet motor2, but there is a difference between when the control computation period Δt is the control computation period Δt1and when the control computation period Δt is the control computation period Δt2as to whether the overcurrent protection unit41can detect the overcurrent before the motor current reaches the demagnetizing current Imag. As a result of performing the overcurrent protection operation on the basis of the same overcurrent protection threshold Ilim all the time, the overcurrent protection unit41can fail to operate when protection for the inverter unit3is necessary. For this reason, the overcurrent protection unit41desirably sets a margin of the overcurrent protection threshold Ilim relative to and the demagnetizing current Imag, the margin being in proportion to the length of the control computation period Δt. A value of the overcurrent protection threshold Ilim is desirably set to be smaller as the control computation period Δt is prolonged. In addition, since the control computation period Δt and the carrier frequency fc are desirably in an inversely proportional relationship as described above, the overcurrent protection unit41may set the value of the overcurrent protection threshold Ilim to be smaller as the carrier frequency fc decreases. Furthermore, in a case where the synchronous PWM mode applies, it is desired that the output voltage frequency finv and the carrier frequency fc be in a proportional relationship and the control computation period Δt and the carrier frequency fc be in an inversely proportional relationship. The overcurrent protection unit41may thus set the value of the overcurrent protection threshold Ilim to be smaller as the output voltage frequency finv decreases. FIG.8is a conceptual diagram illustrating an example of a relationship between the control computation period Δt and the overcurrent protection threshold Ilim according to the first embodiment. Specifically,FIG.8illustrates the relationship between the control computation period Δt and the overcurrent protection threshold Ilim at various magnet temperatures Tmag=T1, T2, and T3in a case of high-temperature demagnetization. Assume that T1>T2>T3holds true. The overcurrent protection unit41sets the relationship among the overcurrent protection threshold Ilim, the control computation period Δt, and the magnet temperature Tmag so that the overcurrent protection threshold Ilim relative to the control computation period Δt decreases as the magnet temperature Tmag increases, as illustrated inFIG.8. In practice, the overcurrent protection unit41stores the relationship between the control computation period Δt and the overcurrent protection threshold Ilim under a plurality of magnet temperature Tmag conditions in the form of a table or a mathematical formula. FIG.9is a conceptual diagram illustrating an example of a relationship between the carrier frequency fc and the overcurrent protection threshold Ilim according to the first embodiment. The overcurrent protection unit41may set the relationship among the overcurrent protection threshold Ilim, the carrier frequency fc, and the magnet temperature Tmag so that the overcurrent protection threshold Ilim relative to the carrier frequency fc decreases as the magnet temperature Tmag increases, as illustrated inFIG.9. FIG.10is a conceptual diagram illustrating an example of a relationship between the output voltage frequency finv and the overcurrent protection threshold Ilim according to the first embodiment.FIG.10illustrates the relationship between the output voltage frequency finv and the overcurrent protection threshold Ilim in the case where the synchronous PWM mode applies. The overcurrent protection unit41may set the relationship among the overcurrent protection threshold Ilim, the output voltage frequency finv and the magnet temperature Tmag so that the overcurrent protection threshold Ilim relative to the output voltage frequency finv decreases as the magnet temperature Tmag increases, as illustrated inFIG.10. The overcurrent protection unit41may store the relationship between the magnet temperature Tmag and the overcurrent protection threshold Ilim as illustrated inFIG.6as a table of correction coefficients for performing correction by multiplication by a coefficient on the basis of the control computation period Δt, instead of tables or mathematical formulae indicating the relationships illustrated inFIGS.8to10. The overcurrent protection unit41may store a relationship between the overcurrent protection threshold Ilim and the carrier frequency fc, which is a parameter related to setting the control computation period Δt, instead of the relationship between the overcurrent protection threshold Ilim and the control computation period Δt. Alternatively, the overcurrent protection unit41may store a relationship between the overcurrent protection threshold Ilim and the output voltage frequency finv of the inverter unit3in the synchronous PWM mode. Consequently, in order to perform the protection operation on the inverter unit3appropriately regardless of the length of the control computation period Δt, the overcurrent protection unit41needs to change the overcurrent protection threshold Ilim in correspondence to the control computation period Δt, the carrier frequency fc that can be a parameter related to setting the control computation period Δt, or the output voltage frequency finv of the inverter unit3in the synchronous PWM mode. The overcurrent protection unit41sets the overcurrent protection threshold Ilim on the basis of the magnet temperature Tmag and any one of the control computation period Δt of the control unit4, the output voltage frequency finv of the inverter unit3, and the carrier frequency fc based on the output voltage frequency finv of the inverter unit3. The magnet temperature Tmag is a magnet temperature estimated value of the permanent magnet of the permanent-magnet motor2. Next, an operation of the magnet temperature estimation unit42will be described. The magnet temperature estimation unit42estimates the magnet temperature Tmag of the permanent magnet of the permanent-magnet motor2on the basis of the three-phase alternating-current voltage output from the inverter unit3and the motor current flowing through the permanent-magnet motor2. The magnet temperature estimation unit42can use a current value detected by the current detection unit5as the motor current necessary for estimating the magnet temperature Tmag. While the driving apparatus100requires a voltage sensor for detecting the three-phase alternating-current voltage, the three-phase alternating-current voltage output from the inverter unit3includes many harmonic components generated by the switching operations of the switching elements. For this reason, instead of using a detection value of the three-phase alternating-current voltage, the magnet temperature estimation unit42may use the voltage command of the control unit4on the basis of which the inverter unit3generates the three-phase alternating-current voltage. That is, the magnet temperature estimation unit42may estimate the magnet temperature Tmag on the basis of the voltage command and the motor current. The magnet temperature estimation unit42may convert coordinates for the voltage command and the motor current into known rotating orthogonal two-axis coordinates, that is, d-q axis coordinates, and use values having been subjected to the coordinate conversion. By performing the coordinate conversion, the magnet temperature estimation unit42can handle an alternating-current amount including voltage, current, etc. as a direct-current amount, and can simplify an operation process. A voltage output from the inverter unit3and a voltage generated in the permanent-magnet motor2are balanced with each other, under which condition the permanent-magnet motor2is driven. The voltage generated in the permanent-magnet motor2is a winding resistance voltage and an induced voltage. The winding resistance voltage is a voltage generated mainly by a flow of the motor current through the winding of the permanent-magnet motor2. The induced voltage is a voltage generated in the winding of the permanent-magnet motor2under an action of the magnetic flux of the permanent-magnet motor2and the rotation of the permanent-magnet motor2itself. The magnetic flux of the permanent-magnet motor2is constituted by magnetic flux generated by a flow of the motor current through the winding of the permanent-magnet motor2and magnetic flux caused by the permanent magnet of the permanent-magnet motor2. Since the magnetic flux caused by the permanent magnet has a correlation with the magnet temperature Tmag, the correlation between the magnet temperature Tmag and the magnetic flux caused by the permanent magnet is obtained in advance from actual measurement, a data sheet of permanent magnet characteristics, or the like. The magnet temperature estimation unit42can estimate the magnet temperature Tmag of the permanent-magnet motor2by extracting a component related to the magnetic flux caused by the permanent magnet from the voltage generated in the permanent-magnet motor2, that is, the voltage output from the inverter unit3. Among the winding resistance voltage and the induced voltage are components caused by the magnetic flux generated by a flow of the motor current through the winding of the permanent-magnet motor2. These components can be calculated from the motor current, the rotational speed, and the like if a winding resistance value, inductance, and the like, which are parameters indicating the characteristics of the permanent-magnet motor2, are known. The control unit4obtains the parameters indicating the characteristics of the permanent-magnet motor2through, for example, actual measurement, and subtracts those components from the voltage generated in the permanent-magnet motor2, thereby extracting the component related to the magnetic flux caused by the permanent magnet. Note that a more detailed configuration of the magnet temperature estimation unit42can be implemented by, for example, a configuration disclosed in Japanese Patent No. 5788057, and thus, will be omitted herein. In the light of the above description, a series of operations of the driving apparatus100from startup thereof and start of driving of the permanent-magnet motor2to the protection operation performed on the inverter unit3will be described. FIG.11is a flowchart illustrating operations of the driving apparatus100according to the first embodiment from startup thereof and start of driving of the permanent-magnet motor2to a protection operation performed on the inverter unit3. These operations are mainly controlled by the control unit4. First, the control unit4starts up the driving apparatus100(step S1) and performs initial setting (step S2). Specifically, the control unit4sets, as the initial setting, the control computation period Δt, the carrier frequency fc, the overcurrent protection threshold Ilim, and the like necessary for driving the permanent-magnet motor2. The control unit4sets an appropriate value of the overcurrent protection threshold Ilim corresponding to the control computation period Δt as described above. After the initial setting performed by the control unit4, the inverter unit3controls an operation related to output of a voltage on the basis of the output voltage frequency finv set by the control unit4, and starts driving the permanent-magnet motor2(step S3). The control unit4determines whether the carrier frequency fc has been changed during driving of the permanent-magnet motor2(step S4). If the carrier frequency fc has been changed (step S4: Yes), the control unit4sets the carrier frequency fc to a desired value (step S5), and also sets the control computation period Δt to an appropriate value corresponding to the carrier frequency fc (step S6). If the carrier frequency fc has not been changed (step S4: No), the control unit4skips the operations of steps S5and S6. In the control unit4, the magnet temperature estimation unit42appropriately estimates the magnet temperature Tmag of the permanent-magnet motor2(step S7), and sets the overcurrent protection threshold Ilim in correspondence to the magnet temperature Tmag (step S8). Furthermore, the magnet temperature estimation unit42corrects the overcurrent protection threshold Ilim in correspondence to the control computation period Δt (step S9). The temperature change in the magnet temperature Tmag of the permanent-magnet motor2is moderate relative to the change in the motor current. For this reason, the magnet temperature estimation unit42need not estimate the magnet temperature as frequently as current anomaly determination performed by the overcurrent protection unit41. That is, the magnet temperature estimation unit42need not perform the magnet temperature estimation operation in the same period as the control computation period Δt. The magnet temperature estimation unit42may perform the magnet temperature estimation operation and the setting operation of the overcurrent protection threshold Ilim corresponding to the magnet temperature Tmag, at timings extracted from the timings of the control computation performed every control computation period Δt. When the magnet temperature estimation unit42sets and corrects the overcurrent protection threshold Ilim, the overcurrent protection unit41compares the motor current flowing through the winding of the permanent-magnet motor2detected by the current detection unit5, with the overcurrent protection threshold Ilim (step S10), and determines whether there is an anomaly in the output current. If the overcurrent has not been detected (step S11: No), the overcurrent protection unit41returns to the operation of step S4. If the overcurrent has been detected (step S11: Yes), the overcurrent protection unit41performs a protection operation on the inverter unit3(step S12). By performing the protection operation on the inverter unit3in accordance with the flowchart illustrated inFIG.11as described above, the driving apparatus100can achieve the protection operation on the inverter unit3regardless of the magnitude of the carrier frequency fc and the length of the control computation period Δt, and can prevent demagnetization of the permanent magnet due to an overcurrent. Next, a hardware configuration of the control unit4included in the driving apparatus100will be described.FIG.12is a diagram illustrating an example of a hardware configuration that realizes the control unit4included in the driving apparatus100according to the first embodiment. The control unit4is realized by a processor201and a memory202. The processor201is a central processing unit (CPU, also referred to as a processing device, an arithmetic device, a microprocessor, a microcomputer, a processor, or a digital signal processor (DSP)), or system large scale integration (LSI). As the memory202, a nonvolatile or volatile semiconductor memory such as a random access memory (RAM), a read only memory (ROM), a flash memory, an erasable programmable read only memory (EPROM), or an electrically erasable programmable read only memory (EEPROM (registered trademark)) can be exemplified. The memory202is not limited thereto, and may be a magnetic disk, an optical disk, a compact disc, a mini disk, or a digital versatile disc (DVD). As described above, according to the first embodiment, the control unit4of the driving apparatus100sets the overcurrent protection threshold Ilim for the inverter unit3in correspondence to the control computation period Δt, or a parameter related to the control computation period Δt, for example, the carrier frequency fc, and the output voltage frequency finv of the inverter unit3in the synchronous PWM mode. Accordingly, the driving apparatus100achieves overcurrent protection of the inverter unit3regardless of the length of the control computation period Δt set on the basis of the carrier frequency fc. As a result, the driving apparatus100provides an effect that it is possible to prevent element destruction of the switching elements of the inverter unit3and demagnetization of the permanent-magnet motor2connected to the inverter unit3due to an overcurrent. In addition, since the driving apparatus100makes it possible to design a reduced margin relative to the demagnetizing current Imag, the output performance of the permanent-magnet motor2can be extended, and a permanent magnet having a lower demagnetization resistance can be used as a permanent magnet for the permanent-magnet motor2. As a result, there is an effect that manufacturing cost of the permanent-magnet motor2can be reduced. Furthermore, the driving apparatus100has a mode for setting the carrier frequency fc to M times the output voltage frequency finv of the inverter unit3, that is, the synchronous PWM mode applies to the driving apparatus100. Applying the synchronous PWM mode the driving apparatus100enables a maximum value of the output voltage frequency finv to be set higher than that in the asynchronous PWM mode. The driving apparatus100can perform driving even in a high speed zone where driving is difficult in the asynchronous PWM mode, and thus has an effect that appropriate overcurrent protection of the inverter unit3can be achieved even in the high speed zone. Second Embodiment In the first embodiment, the driving apparatus100performs the protection operation on the inverter unit3, taking account of the magnet temperature Tmag of the permanent-magnet motor2on the basis of the control computation period Δt. A second embodiment shortens the control computation period Δt, that is, an overcurrent determination period in correspondence to the magnet temperature Tmag to thereby improve responsiveness to overcurrent protection under the condition that requires a more accurate protection operation because of, for example, high-temperature demagnetization, or the demagnetizing current Imag decreases with an increase in the temperature of the permanent-magnet motor2. Differences from the first embodiment will be described. FIG.13is a flowchart illustrating operations of the driving apparatus100according to the second embodiment from startup thereof and start of driving of the permanent-magnet motor2to a protection operation performed on the inverter unit3. These operations are mainly controlled by the control unit4. InFIG.13, the operations of steps S1to S7are similar to those in the flowchart of the first embodiment illustrated inFIG.11. The magnet temperature estimation unit42compares the estimated magnet temperature Tmag with a predetermined temperature threshold Th (step S21). If the estimated magnet temperature Tmag is equal to or higher than the temperature threshold Th (step S21: Yes), the magnet temperature estimation unit42resets the control computation period Δt, that is, the overcurrent determination period to be shorter in correspondence to the magnet temperature Tmag in order to improve the responsiveness to the overcurrent protection (step S22). That is, the magnet temperature estimation unit42sets the control computation period Δt on the basis of the magnet temperature Tmag which is a magnet temperature estimated value. A constraint on the shortest time of the control computation period Δt arises from the above-described upper limit of the setting of the carrier frequency fc and a constraint on a process of the control computation performed by the processor. Under such a constraint, the magnet temperature estimation unit42resets the carrier frequency fc (step S23). That is, the magnet temperature estimation unit42sets the carrier frequency fc on the basis of the output voltage frequency finv of the inverter unit3and the control computation period Δt set on the basis of the magnet temperature Tmag which is a magnet temperature estimated value. To reset the carrier frequency fc, specifically, the magnet temperature estimation unit42selects the PWM mode from among the asynchronous PWM mode and the synchronous PWM mode on the basis of the magnitude of the output voltage frequency finv and selects the above-described parameters, or positive integers M and N, provided that the constraint of fc/finv≥9 is substantially satisfied when the asynchronous PWM mode applies, as the control computation period Δt and the carrier frequency fc have a correlation as described above. If the estimated magnet temperature Tmag is less than the temperature threshold Th (step S21: No), the magnet temperature estimation unit42skips the operations of steps S22and S23. The operations of subsequent steps S8to S12are similar to those in the flowchart of the first embodiment illustrated inFIG.11. As described above, according to the second embodiment, the driving apparatus100performs the protection operation on the inverter unit3in accordance with the flowchart illustrated inFIG.13. Accordingly, the driving apparatus100shortens the control computation period Δt, that is, the overcurrent determination period in correspondence to the magnet temperature Tmag to thereby improve responsiveness to overcurrent protection under the condition that requires a more accurate protection operation because the demagnetizing current Imag decreases with the increase in the temperature of the permanent-magnet motor2. As a result, the driving apparatus100can more reliably prevent demagnetization of the permanent magnet due to an overcurrent. Third Embodiment To perform the protection operation for the inverter unit3, in the first embodiment, the overcurrent protection unit41outputs, to the inverter unit3, a signal that stops all the outputs of the PWM signals of the individual phases, which are drive signals for the inverter unit3, or a signal that directly stops the switching operations of all the switching elements of the inverter unit3, thereby stopping the output of the three-phase alternating-current voltage from the inverter unit3to the permanent-magnet motor2. To perform the protection operation on the inverter unit3, in a third embodiment, the overcurrent protection unit41reduces the output voltage frequency finv of the three-phase alternating-current voltage output from the inverter unit3in order to continue the operation of the permanent-magnet motor2. The overcurrent protection unit41reduces power output from the inverter unit3to the permanent-magnet motor2, thereby performing the protection operation on the inverter unit3. The protection operation of the third embodiment is the same as that in the first embodiment in the operation involved in the overcurrent protection unit41determining the anomaly of the output current. It is noted that a drive constraint under which to perform the protection operation in the third embodiment is more moderate than a drive constraint under which to stop the supply of the three-phase alternating-current voltage from the inverter unit3to the permanent-magnet motor2in the first embodiment. It is therefore desirable to set a condition under which the output current is more likely to be determined as anomaly than under a condition in the first embodiment. For example, the third embodiment sets the overcurrent protection threshold Ilim to a value smaller than a set value in the first embodiment. To operate the protection operation on the inverter unit3, for example, the overcurrent protection unit41controls the control unit4such that an absolute value of the output voltage frequency finv of the inverter unit3is reduced by a frequency reduction amount Δf. That is, assuming that the output voltage frequency finv, which is set as a target value, before a reducing action based on the frequency reduction amount Δf is denoted by finv0, a relationship of finv=finv0+Δf holds true, where Δf<0 if finv0>0, and Δf>0 if finv0<0. It is noted that |finv0| is set to be larger than |Δf| so that signs of finv and finv0are not inverted between before and after the reducing action. The overcurrent protection unit41obtains the frequency reduction amount Δf on the basis of a proportional value corresponding to the excess of the output current with respect to the overcurrent protection threshold Ilim compared in the determination of the anomaly of the output current, and an integration value of the excesses for each control computation period Δt. While, in the third embodiment, a deviation occurs in the output voltage frequency finv of the inverter unit3with respect to a desired frequency, that is, a target value, and a speed deviation occurs in the permanent-magnet motor2, the operating state of the permanent-magnet motor2is continued unlike the stop of the supply of the three-phase alternating-current voltage from the inverter unit3. As described above, according to the third embodiment, the driving apparatus100has an effect that it is possible to achieve protection of the permanent-magnet motor2, while continuing driving the permanent-magnet motor2. Fourth Embodiment In a fourth embodiment, an example will be described in which the driving apparatus100described in the first to third embodiments is applied to an air-conditioning apparatus. By applying the driving apparatus100to an air-conditioning apparatus, it is possible to constitute an air-conditioning apparatus using a compressor that uses a rotational force of the permanent-magnet motor2as a drive source, and cooling and heating capacity can be improved with the extension of an operation limit. FIG.14is a diagram illustrating an example configuration of an air-conditioning apparatus200according to the fourth embodiment. The air-conditioning apparatus200includes the driving apparatus100and a compressor60using the permanent-magnet motor2as a drive source. In the air-conditioning apparatus200, a refrigerant circuit is constituted in which a refrigerant pipe70connects the compressor60, a four-way valve62, a heat source-side heat exchanger63, a heat source-side expansion valve64, a load-side expansion valve65, a load-side heat exchanger66, the four-way valve62, and the compressor60in this order. In the air-conditioning apparatus200, a refrigeration cycle is established by refrigerant flowing in the refrigerant circuit. Although not illustrated inFIG.14, an accumulator that stores excess refrigerant may be provided on a suction side of the compressor60. In controlling the refrigerant circuit, an air-conditioning control unit69controls the four-way valve62, the heat source-side expansion valve64, and the load-side expansion valve65. The configuration of the refrigeration cycle of the air-conditioning apparatus200illustrated inFIG.14is merely an example, and the refrigeration cycle does not necessarily have the same configuration. Next, an operation of the air-conditioning apparatus200will be described using a cooling operation as an example. Although a detailed description of a heating operation will be omitted, the air-conditioning apparatus200can also realize the heating operation by performing switching of flow paths in the four-way valve62. In the cooling operation, the air-conditioning apparatus200performs switching of the flow paths to adjust a direction of the four-way valve62so that refrigerant discharged in advance from the compressor60is directed to the heat source-side heat exchanger63and refrigerant flowing out of the load-side heat exchanger66is directed to the compressor60. By the driving apparatus100driving the permanent-magnet motor2, a compression element61coupled to the permanent-magnet motor2compresses the refrigerant into high-temperature and high-pressure refrigerant, and the compressor60discharges the high-temperature and high-pressure refrigerant. That is, the compressor60compresses the refrigerant in the refrigeration cycle. The high-temperature and high-pressure refrigerant discharged from the compressor60flows into the heat source-side heat exchanger63via the four-way valve62, and exchanges heat with external air to radiate heat in the heat source-side heat exchanger63. The refrigerant flowing out of the heat source-side heat exchanger63is subjected to expansion and pressure reduction in the heat source-side expansion valve64to change into low-temperature and low-pressure gas-liquid two-phase refrigerant. The refrigerant in such a state is subjected to expansion and pressure reduction in the load-side expansion valve65, flows into the load-side heat exchanger66, exchanges heat with air in a space to be air-conditioned to be evaporated, changes into low-temperature and low-pressure refrigerant, and flows out of the load-side heat exchanger66. The refrigerant flowing out of the load-side heat exchanger66is sucked into the compressor60via the four-way valve62, and is compressed again. In the air-conditioning apparatus200, the above operations are repeated. For the purpose of mainly cooling the inverter unit3of the driving apparatus100, a cooling plate may be brought into contact with a power module which is a component of the inverter unit3, and the refrigerant pipe70may be further brought into contact with the cooling plate to cause the refrigerant flowing in the refrigerant pipe70to absorb heat generated in the inverter unit3. With the above arrangement, a temperature increase in the inverter unit3can be efficiently reduced. In the configuration ofFIG.14, both of the heat source-side expansion valve64on the outdoor unit67side and the load-side expansion valve65on the indoor unit68side are included. The configuration is employed in order to make it possible to control the cooling capacity of the driving apparatus100independently by each of the two expansion valves, i.e., the heat source-side expansion valve64and the load-side expansion valve65. With such a configuration, the air-conditioning apparatus200is suitable for finely controlling the refrigerant so as to adjust the temperature of the inverter unit3to a desired temperature while detecting the temperature, and the temperature of the driving apparatus100, in particular, the temperature of the power module which is a component of the inverter unit3is not lowered unnecessarily, so that the generation of dew condensation can be reduced, and the temperature can be controlled so as not to increase. The configuration ofFIG.14is merely an example of finely controlling the temperature of the driving apparatus100, and both of the two expansion valves, i.e., the heat source-side expansion valve64and the load-side expansion valve65may not be necessarily included. Either the heat source-side expansion valve64on the outdoor unit67side or the load-side expansion valve65on the indoor unit68side may be included. In the fourth embodiment, the example has been described in which the driving apparatus100is applied to the air-conditioning apparatus200, but there is no limitation thereto, and the driving apparatus100can be applied to a device including a refrigeration cycle such as a heat pump device and a refrigeration device, in addition to the air-conditioning apparatus200. Under a situation that requires the heating operation of the air-conditioning apparatus200at the low ambient temperature with the low temperature of the permanent magnet of the permanent-magnet motor2in the case of high-temperature demagnetization, the design margin with respect to the demagnetizing current Imag is reduced and the overcurrent protection threshold Ilim is increased to increase a current that can flow through the permanent-magnet motor2, thereby improving the heating capacity and improving comfortability. The driving apparatus100of the air-conditioning apparatus200performs, particularly during heating operation, an operation of estimating the magnet temperature Tmag of the permanent magnet of the permanent-magnet motor2, and an operation of setting the overcurrent protection threshold Ilim on the basis of the magnet temperature estimated value of the permanent-magnet motor2and any one of the control computation period Δt of the control unit4, the output voltage frequency finv of the inverter unit3, and the carrier frequency fc based on the output voltage frequency finv of the inverter unit3. Accordingly, the air-conditioning apparatus200can further enhance the effect of the present invention. As described above, according to the fourth embodiment, the application of the driving apparatus100to the air-conditioning apparatus200can extend the operation limit of the permanent-magnet motor2which is a drive source of the compressor60. In particular, when a permanent magnet having a characteristic of the high-temperature demagnetization is used as the permanent magnet of the permanent-magnet motor2, the demagnetizing current Imag of the permanent magnet increases at a low temperature requiring a large heating capacity, and thus the current that can flow through the permanent-magnet motor2increases, so that there is an effect of improving the heating capacity at a low temperature. Examples of utilization of the driving apparatus100include a refrigerator, a dehumidifier, a heat pump water heater, a heat pump drying/washing machine, and a refrigeration device in addition to the air-conditioning apparatus200equipped with the compressor60using the rotational force of the permanent-magnet motor2as a driving force. Furthermore, the driving apparatus100is also applicable to a product not equipped with a compressor, such as a dryer, a washing machine, or a vacuum cleaner, which obtains a driving force by the rotational force of the permanent-magnet motor2, and is also applicable to a fan motor and the like. The configurations described in the embodiments above are merely examples of the content of the present invention and can be combined with other known technology and part thereof can be omitted or modified without departing from the gist of the present invention. | 56,677 |
11863102 | FIG.1shows a patient lifter1comprising stabilizer legs2equipped with drive wheels, a mast3connected to the stabilizer legs2and a cantilever4with one end pivotably connected to the mast3. To the other end of the cantilever4, a lifting hook5for lifting a patient is secured. The patient lifter1comprises an electric linear actuator system having an electric linear actuator6, a control box7with a controller, a power supply8, an emergency stop9and an operating unit10. The cantilever4may be raised and lowered by means of a linear actuator6, which with one end is secured to the frame and with the other end is secured to the cantilever4. The control box7is fixed to the mast3with a bracket16. FIG.2shows a patient lifter1which carries a load. As it can be seen, the geometry of the patient lifter1is changed due to the lifting of the cantilever4. More specifically the length of the cantilever arm in the horizontal and vertical orientation changes which influences the power needed for the linear electrical actuator in order to push the load. As it appears inFIG.3, showing a diagram with the linear electric actuator pushing a load, the patient lifter requires a big difference in push force from the actuator (and thereby a big difference in current draw) at different height positions. If the patient lifter is of the type where the controller is having a power limiting circuit limiting the power to be supplied to the at least one linear electric actuator to one programmed power limit value stored with the controller there could potentially be a safety problem. In the diagram, the maximum current recorded is 3.5 A to lift the load. Because the current draw is only 0.5 A to lift the load at low positions, the patient lifter will in theory be able to lift more than seven times the rated load as the current is set to cut off at 3.5 A. FIG.4shows an overall diagram of a linear electric actuator system featuring one linear electric actuator6and a controller11. For the sake of simplicity is only shown the components relevant for the explanation. The linear electric actuator6is equipped with a system for position determination12of the spindle nut on its travel on the spindle which gives input to the controller11. The controller11features a power limiting circuit13that is configured to cut the power to the linear electric actuator6in case the current draw exceeds a programmed value. Further the controller11is equipped with a table14of power limit values which is a reference to a specific position of the spindle nut during its travel on the spindle and a corresponding power limit value to be used by the controller when the spindle nut reaches the specific position. The controller has an algorithm that regularly updates the power limiting circuit13with the relevant up to date value such that the power limit circuit is configured to steadily live up to the requirements of the standard and to safeguard the mechanics of the application, here the patient lifter. The table14of power limit values can be provided by programming the controller with the specific values that suits the application. A learn mode can be implemented where the application over the full movement carries a max load and data pairs of position and current are recorded at suitable intervals to be used for the power limiting circuit. What is achieved by the invention is that the actuator system is arranged such that the threshold value of the maximum permissible power in the power limiting circuit may be changed, and that this change may be performed via reference to the position of the spindle nut on the spindle as a look up in a table showing a corresponding value for the power limit to be utilized by the power limiting circuit. | 3,738 |
11863103 | DETAILED DESCRIPTION FIG.1is a simplified block diagram of an example motorized window treatment system100. The motorized window treatment system100may comprise a plurality of motorized window treatments, for example, motorized roller shades110as shown inFIG.1. Each motorized roller shade110may comprise a respective flexible shade fabric112rotatably supported by a respective roller tube114. Each motorized window shade110may comprise a respective motor drive unit120that may be located inside of the roller tube114. The motor drive units120may each rotate the respective roller tube114to adjust a present position PPRESof the shade fabric112between a fully-open position POPENand a fully-closed position PCLOSED. The motor drive units120may be coupled to a communication link122and may receive commands from a keypad124and/or other control device across the communication link. The communication link122may comprise a wired communication link or a wireless communication link, such as, for example, a radio-frequency (RF) communication link or an infrared (IR) communication link. FIG.2is a simplified block diagram of an example motor drive unit200of a motorized window treatment. The motor drive unit200may comprise a motor210(e.g., a direct-current (DC) motor) that may be coupled to a roller tube of the motorized window treatment for rotating the roller tube for raising and lowering a flexible material (e.g., a shade fabric). The motor drive unit200may comprise a motor drive circuit220(e.g., an H-bridge drive circuit) that may receive a bus voltage VBUSand generate a pulse-width modulated (PWM) voltage VPWMfor driving the motor210. The bus voltage VBUSmay be produced across a bus capacitor CBUS. The motor drive unit200may comprise a power supply212that may receive the bus voltage VBUSand generate a supply voltage VCC(e.g., approximately 3.3 V) for powering the low-voltage circuitry of the motor drive unit. The motor drive unit200may be configured to receive an input voltage VINfrom, for example, an external power supply, such as a direct-current (DC) supply and/or an alternating-current (AC) supply. Additionally or alternatively, the motor drive unit200may be powered by one or more batteries and/or a photovoltaic power source, such as a solar cell. While not shown inFIG.2, the motor drive unit200may also comprise a rectifier circuit and/or a power converter circuit for receiving the input voltage VINand generating the bus voltage VBUSacross the bus capacitor CBUS. The motor drive unit200may comprise a unit control circuit230(e.g., a primary control circuit) for controlling the operation of the motor210. The unit control circuit230may comprise, for example, a microprocessor, a programmable logic device (PLD), a microcontroller, an application specific integrated circuit (ASIC), a field-programmable gate array (FPGA), or any suitable processing device or control circuit. The unit control circuit230may be configured to generate a drive signal VDRVfor controlling the motor drive circuit220to control the rotational speed of the motor210. For example, the drive signal VDRVmay comprise a pulse-width modulated signal, and the rotational speed of the motor210may be dependent upon a duty cycle of the pulse-width modulated signal. In addition, the unit control circuit230may be configured to generate a direction signal Vow for controlling the motor drive circuit220to control the direction of rotation of the motor210and an enable signal VENABLEfor enabling and disabling the motor drive circuit220. The unit control circuit230may be configured to control the motor210to adjust a present position PPRESof the shade fabric of the motorized window treatment between a fully-open position POPENand a fully-closed position PCLOSED. The motor drive unit200may include a rotational position sensor, e.g., a Hall effect sensor (HES) circuit240, which may be configured to generate two Hall effect sensor (HES) signals VHES1, VHES2that may indicate the rotational position and direction of rotation of the motor210. The HES circuit240may comprise two internal sensing circuits for generating the respective HES signals VHES1, VHES2in response to a magnet that may be attached to a drive shaft of the motor. The magnet may be a circular magnet having alternating north and south pole regions, for example. For example, the magnet may have two opposing north poles and two opposing south poles, such that each sensing circuit of the HES circuit240is passed by two north poles and two south poles during a full rotation of the drive shaft of the motor. Each sensing circuit of the HES circuit240may drive the respective HES signal VHES1, VHES2to a high state when the sensing circuit is near a north pole of the magnet and to a low state when the sensing circuit is near a south pole. The unit control circuit230may be configured to determine that the motor210is rotating in response to the HES signals VHES1, VHES2generated by the HES circuit240. In addition, the unit control circuit230may be configured to determine the rotational position and direction of rotation of the motor210in response to the HES signals VHES1, VHES2. The motor drive unit200may include a communication circuit242that allows the unit control circuit230to transmit and receive communication signals, e.g., wired communication signals and/or wireless communication signals, such as radio-frequency (RF) signals. The motor drive unit200may further comprise a user interface244having one or more buttons that allow a user to provide inputs to the control circuit230during setup and configuration of the motorized window treatment. The unit control circuit230may be configured to control the motor210to control the movement of the covering material in response to a shade movement command received from the communication signals received via the communication circuit242or the user inputs from the buttons of the user interface244. The user interface244may also comprise a visual display, e.g., one one or more light-emitting diodes (LEDs), which may be illuminated by the unit control circuit230to provide feedback to the user of the motorized window treatment system. The unit control circuit230may be coupled to a memory246(e.g., a non-volatile memory) for storage of the present position PPRESof the shade fabric and/or the limits (e.g., the fully-open position POPENand the fully-closed position PCLOSED). While controlling the motor drive circuit220to drive the motor210, the unit control circuit230may be configured to reduce the power delivered to the motor210(e.g., by stopping the motor) in the event of a stall condition. When a stall condition occurs, the motor210may stop rotating and the HES circuit240may stop generating the HES signals VHES1, VHES2even though the unit control circuit230is actively attempting to rotate the motor by continuing to generate the drive signal VDRV. The unit control circuit230may be configured to monitor one or both of the HES signals VHES1, VHES2to detect a stall condition. For example, the unit control circuit230may be configured to detect that the motor210may have stalled if the HES circuit240is not generating one or both of the HES signals VHES1, VHES2while the unit control circuit230is generating the drive signal VDRVfor controlling the motor drive circuit220to drive the motor210. The unit control circuit230may be configured to reduce the power delivered to the motor210(e.g., by stopping the motor) after a first amount of time (e.g., one second) from first detecting a stall condition. For example, the unit control circuit230may be configured to stop the motor in response to detecting a stall condition by disabling the motor drive circuit220(e.g., by driving the magnitude of the enable signal VENABLElow towards circuit common). The unit control circuit230may provide a software-based implementation of a process for detecting and resolving a stall condition in the motor210. The motor drive unit200may further comprise a stall protection circuit250(e.g., a hardware stall protection circuit) that may be configured to turn off the motor210in the event of a stall condition if the unit control circuit230is unable to stop the motor in response to the stall condition. The stall protection circuit250may receive the drive signal VDRVfrom the unit control circuit230and at least one of the HES signals VHES1, VHES2from the HES circuit240(e.g., the first HES signal VHES1as shown inFIG.2). The stall protection circuit250may be coupled to the enable signal VENABLEthat is generated by the unit control circuit230and received by the motor drive circuit220for disabling the motor drive circuit and thus stopping the motor in response to detecting a stall condition. For example, the stall protection circuit250may be configured to override the unit control circuit230and pull the enable signal VENABLEdown towards circuit common if the stall protection circuit is receiving the drive signal VDRV, but not receiving the first HES signal VHES1. The stall protection circuit250may be configured to drive the enable signal VENABLElow after a second amount of time (e.g., approximately 2-4 seconds) from the time when the first HES signal VHES1is not being generated while the drive signal VDRVis being generated. The stall protection circuit250may operate slower than the unit control circuit230when detecting a stall condition, such that the unit control circuit230may typically stop the motor210in the event of a stall condition after the first predetermined amount of time. However, if the unit control circuit230is unable to stop the motor in the event of a stall condition, the stall protection circuit250may operate to stop the motor after the second predetermined amount of time. The stall protection circuit250may provide a hardware-based implementation of a process for detecting and resolving a stall condition in the motor210. The stall protection circuit250may be configured to latch the enable signal VENABLEin the low state in response to detecting a stall condition. When the unit control circuit230stops driving the motor210after detecting a stall condition, the motor may relax and rotate a small amount in a rotational direction that is opposite the rotational direction in which the motor was being driven. This relaxing of the motor210may cause the HES circuit240to generate either or both of the HES signals VHES1, VHES2, which could potentially cause the stall protection circuit250to cease pulling the enable signal VENABLEdown towards circuit common to disable the motor drive circuit220. Therefore, the stall protection circuit250may latch the enable signal VENABLEin the low state after detecting a stall condition, such that the stall protection circuit may not stop pulling the enable signal VENABLElow when the motor210relaxes. To unlatch the stall protection circuit250, the unit control circuit230may stop generating the drive signal VDR(e.g., in response to receiving a stop command via the communication circuit242and/or an actuation of one of the buttons of the user interface244). FIG.3is a simplified partial schematic diagram of an example motor drive unit300of a motorized window treatment (e.g., the motor drive unit200shown inFIG.2). The motor drive unit300may comprise a motor310and an H-bridge drive circuit320configured to receive a bus voltage VBUSand control the motor310. The motor drive unit300may comprise a unit control circuit330configured to control the operation of the H-bridge drive circuit320to control the rotational speed and direction of the motor310. The unit control circuit330may be configured to generate a drive signal VDRVfor controlling the rotational speed of the motor310, a direction signal VDIRfor controlling the direction of rotation of the motor, and an enable signal VENABLEfor enabling and disabling the H-bridge drive circuit320. The unit control circuit330may receive two HES signals VHES1, VHES2from a HES circuit (e.g., the HES circuit140) and may be configured to determine the state of the motor310(e.g., if the motor is rotating), the rotational speed of the motor, and/or the direction of rotation of the motor in response to the HES signals VHES1, VHES2. The H-bridge drive circuit320may comprise four switching transistors, such as field-effect transistors (FETs) Q321, Q322, Q323, Q324, and an H-bridge control circuit326. For example, the H-bridge control circuit326may comprise an integrated circuit (IC). The H-bridge control circuit326may generate gate signals VG1, VG2, VG3, VG4that are received by gates of the respective FETs Q321, Q322, Q323, Q324 for rendering the FETs conductive and non-conductive. The motor310may be coupled between the junction of the FETs Q321, Q323 and the junction of the FETs Q322, Q324. The H-bridge control circuit326may render two of the FETs Q321, Q322, Q323, Q324 conductive and may pulse-width modulate (PWM) at least one of the gate signals VG1, VG2, VG3, VG4(e.g., one of the FETs that is conductive) to generate a pulse-width modulated (PWM) signal VPWMacross the motor310and conduct a motor current IMthrough the motor310as shown inFIG.3. When the FETs Q321, Q324 are conductive, a positive voltage having a magnitude approximately equal to the bus voltage VBUSmay be coupled across the motor310and the motor may rotate in a first direction. When the FETs Q322, Q323 are conductive, a negative voltage having a magnitude approximately equal to the bus voltage VBUSmay be coupled across the motor310and the motor may rotate in a second direction. The H-bridge control circuit326may adjust a duty cycle of the PWM signal VPWMto adjust the rotational speed of the motor310(e.g., by pulse-width modulating at least one of the FETs that are being controlled to be conductive as described above). The H-bridge control circuit326may determine which FETs Q321, Q322, Q323, Q324 to control to set the direction of the motor310in response to the direction signal VDIRgenerated by the unit control circuit330. The H-bridge control circuit326may determine the duty cycle for the PWM signal VPWMin response to the drive signal VDRVgenerated by the unit control circuit330. The H-bridge drive circuit320may comprise a feedback resistor R328 (e.g., having a resistance of approximately 50 mΩ) that may be coupled between the junction of the FETs Q323, Q324 and circuit common. The feedback resistor328may conduct a half-bridge current IHB(e.g., which may indicate the magnitude of the motor current IMthrough the motor310). The feedback resistor328may generate a feedback signal VFBthat may be representative of the magnitude of the motor current IMand may be received by the H-bridge control circuit326. The H-bridge control circuit326may provide an overcurrent protection (OCP) feature in response to the magnitude of the feedback signal VFBto prevent overcurrent conditions in the FETs Q321, Q322, Q323, Q324. For example, the H-bridge control circuit326may render all of the FETs Q321, Q322, Q323, Q324 non-conductive if the magnitude of the feedback signal VFBexceeds a first overcurrent threshold VOCP1(e.g., approximately 1 volt). The H-bridge control circuit326may disable the operation of the FETs Q321, Q322, Q323, Q324 for a retry time period TRETRY(e.g., approximately 3 milliseconds) after detecting the overcurrent condition. While not shown inFIG.3, the H-bridge control circuit326may also receive signals that indicate the magnitudes of voltages developed across the respective FETs Q321, Q322, Q323, Q324. The H-bridge control circuit326may configured to detect an overcurrent condition in the FETs Q321, Q322, Q323, Q324 if the magnitude of any of the voltages across the FETs exceeds a second overcurrent threshold VOCP2(e.g., approximately 1 volt). The unit control circuit330may be configured to detect a stall condition by monitoring one or both of the HES signals VHES1, VHES2For example, the unit control circuit330may be configured to detect that the motor310may have stalled if the HES circuit is not generating one or both of the HES signals VHES1, VHES2while the unit control circuit330is generating the drive signal VDRVfor controlling the motor drive circuit320to drive the motor310. In response to detecting a stall condition, the unit control circuit330may be configured to reduce the power delivered to the motor310(e.g., by stopping the motor) after a first amount of time (e.g., one second) from first detecting the stall condition. For example, the unit control circuit330may be configured to stop the motor in response to detecting a stall condition by driving the magnitude of the enable signal VENABLElow towards circuit common to by disable the motor drive circuit320. The overcurrent protection feature of the H-bridge control circuit326may prevent the motor310and the FETs Q321, Q322, Q323, Q324 from being damaged during the stall condition before the unit control circuit330stops the motor at the end of the first amount of time. The motor drive unit300may comprise a stall protection circuit350that may comprise two comparators U360, U370. The stall protection circuit350may include an edge detect circuit361that may receive one of the HES signals VHES1, VHES2(e.g., the first HES signal VHES1as shown inFIG.3) and may generate an edge detect signal VEDthat indicates the edges (e.g., the rising edges) of the first HES signal VHES1. The edge detect circuit361may comprise a capacitor C362 (e.g., having a capacitance of capacitance of approximately 0.1 μF, a diode D364 (e.g., a Schottky diode), and a resistor R365 (e.g., having a resistance of approximately 100 kΩ). When the magnitude of the first HES signal VHES1transitions from low to high, the capacitor C362 may conduct current through the resistor R365, thus generating a pulse (e.g., a positive-polarity pulse) in the edge detect signal VED. When the magnitude of the first HES signal VHES1transitions from high to low, the capacitor C362 may conduct a pulse of current in the opposite direction through the diode D364 (e.g., without generating a pulse in the edge detect signal VED). The negative input of the first comparator U360 may receive the edge detect signal VEDgenerated by the edge detect circuit361, and the positive input may receive a first reference voltage VREF1, which may be generated by a resistive divider circuit having resistors R366, R368. For example, the resistor R366 may have a resistance of approximately 100 kΩ and the resistor R368 may have a resistance of approximately 10 kΩ, such that the first reference voltage VREF1may have a magnitude of approximately 0.3 V. The output of the first comparator U360 may have an open collector configuration and may be coupled to a capacitor C371 (e.g., having a capacitance of approximately 2 μF) through a resistor R372 (e.g., having a resistance of approximately 100 kΩ). The capacitor C371 may also be coupled to the drive signal VDRVthrough a resistor R374 (e.g., having a resistance of approximately 1 Me). When the magnitude of the edge detect signal VEDis less than the magnitude of the first reference voltage VREF1, the capacitor C371 may charge from the drive signal VDRv through the resistor R374, such that a DC voltage VDCis generated across the capacitor C371. When the magnitude of the edge detect signal VEDis greater than the magnitude of the first reference voltage VREF1, the first comparator U360 may drive the output low to discharge the capacitor C371 through the resistor R372 (e.g., to approximately zero volts). The negative input of the second comparator U370 may receive the DC voltage VDCacross the capacitor C371, and the positive input may receive a second reference voltage VREF2, which may be generated by a resistive divider circuit having resistors R376, R378. For example, the resistor R376 may have a resistance of approximately 10 kΩ and the resistor R378 may have a resistance of approximately 90.9 kΩ, such that the second reference voltage VREF1may have a magnitude of approximately 3 V (e.g., equal to approximately 90% of the magnitude of the supply voltage VCC). The output of the second comparator U370 may have an open collector configuration and may be coupled to the enable signal VENABLEthat is generated by the unit control circuit330and received by the H-bridge control circuit326. The junction of the resistors R376, R378 (e.g., that generates the second reference voltage VREF2) may be coupled to the output of the second comparator U370 through a diode D379. When the comparator U370 drives the output low, the magnitude of the second reference voltage VREF2may be decreased to approximately the forward voltage drop of the diode D379 (e.g., approximately 0.7 V). When the unit control circuit330is generating the drive signal VDRv and the motor is rotating correctly, the HES circuit will generate the HES signals VHES1, VHES2When the first HES signal VHES1is driven high towards the supply voltage VCC, the edge detect circuit361may generate a pulse in the edge detect signal VED, which may exceed the first reference voltage VREF1. As a result, the first comparator U360 may pull the output low to discharge the capacitor C371 and drive the DC voltage Vic down to approximately zero volts. When the magnitude of the edge detect signal VEDis low (e.g., at approximately circuit common), the capacitor C371 may charge from the drive signal VDRv. Since the first HES signal VHES1is a periodic signal, the capacitor C371 may periodically discharge to approximately zero volts when the first HES signal VHES1is driven from low to high. As a result, the magnitude of the DC voltage Vic across the capacitor C371 may not be able to increase above the second reference voltage VREF2, which allows the unit control circuit330to have full control of the enable signal VENABLEand the H-bridge drive circuit320. If the motor stalls, the HES circuit may stop generating the HES signals VHES1, VHES2. When the first HES signal VHES1is not present at the negative input of the first comparator U360, the capacitor C371 may charge from the drive signal VDRVthrough the resistor R374 (e.g., without periodically discharging). When the magnitude of the DC voltage VDCacross the capacitor C371 rises above the second reference voltage VREF2, the second comparator U370 may drive the output low, thus controlling the magnitude of the enable signal VENABLEto approximately zero volts and disabling the H-bridge control circuit326. The capacitance of the capacitor C371 and the resistance of the resistor R374 may be sized such that the magnitude of the DC voltage across the capacitor C371 may exceed the second reference voltage VREF2after a second amount of time (e.g., approximately 2-4 seconds) after the HES circuit stops generating the first HES signal VHES1. The overcurrent protection feature of the H-bridge control circuit326may prevent the motor310and the FETs Q321, Q322, Q323, Q324 from being damaged during the stall condition before the second comparator U370 disables the motor drive circuit320at the end of the second amount of time. Since the drive signal VDRVmay have a magnitude approximately equal to the supply voltage VCCwhen driven high and the magnitude of the second reference voltage may be equal to approximately 90% of the supply voltage, the H-bridge control circuit326may be disabled when first HES signal VHES1is not being generated and the duty cycle of the drive signal VDRVexceeds 90%. After the second comparator U370 drives the output low to disable the H-bridge control circuit326, the magnitude of the second reference voltage VREF2may be pulled down to approximately 0.7 volts through the diode D379, which may latch the H-bridge control circuit326in the disabled state by preventing the second comparator circuit U370 letting go of the enable signal VEN. The H-bridge control circuit may be unlatched from the disabled state if the unit control circuit330stops the motor310(e.g., in response to receiving a stop command and/or a user input) or if the unit control circuit330is reset (e.g., in response to a power cycle of the motor drive unit300). FIG.4is a simplified flowchart of a stall protection procedure400that may be executed by a control circuit of a motor drive unit (e.g., the unit control circuit230of the motor drive unit200and/or the unit control circuit330of the motor drive unit300). For example, the stall protection procedure400may be executed periodically at410. If the control circuit is not presently driving the motor at412, the stall protection procedure400may simply exit at434. If the control circuit is presently driving the motor at412, but a HES signal is not present at414, the control circuit may detect a potential stall condition. If a stall flag is not presently set at416, the control circuit may set the stall flag at418and then reset and start a timer at420, before the stall protection procedure400exits at434. The timer may be used to delay shutting down the motor for a period of time (e.g., approximately one second) after first detecting that the HES signal is not present while driving the motor. For example, the period of time may be sufficiently longer than the rate at which the stall protection procedure400is executed, such that the stall protection procedure400may be executed a number of times before the time expires. If the stall flag is set at416, but the timer has not expired at422, the stall protection procedure400simply exits at434. When the timer has expired at422, the control circuit may stop the motor at424and log a stall event in memory at426, before the stall protection circuit exits at434. When the HES signal is present at414and the error flag is set at428, the control circuit may clear the stall flag at430and stop the timer at432, before the stall protection procedure400exits. | 25,790 |
11863104 | DETAILED DESCRIPTION Hereinafter, a motor drive device, an electric blower, an electric vacuum cleaner, and a hand dryer according to an embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to the embodiment. Embodiment FIG.1is a diagram illustrating a configuration of a motor drive system including a motor drive device according to an embodiment of the present invention. A motor drive system1according to the embodiment of the present invention includes a power supply10, a motor drive device2, and a single-phase motor12. The power supply10is a direct-current power supply that supplies direct-current power to the motor drive device2. The power supply10is a converter, a battery, or the like. The power supply10may be any power supply that outputs direct-current power, and is not limited to the converter, the battery, or the like. The single-phase motor12is a brushless motor including a rotor12aof a permanent magnet type and a stator12b. The single-phase motor12may be any permanent magnet type motor that generates an induced voltage, and is not limited to the brushless motor. Four permanent magnets are arranged on the rotor12ain a circumferential direction. These permanent magnets are arranged such that directions of magnetic poles thereof are alternately inverted in the circumferential direction, and form a plurality of magnetic poles of the rotor12a. The number of permanent magnets is not limited to four, and is only required to be four or more. A winding (not illustrated) is wound around the stator12b. A motor current flows through the winding. The motor current is equal to an alternating current supplied from a single-phase inverter11to the single-phase motor12. The motor drive device2is a device that supplies alternating-current power to the single-phase motor12to drive the single-phase motor12. The motor drive device2includes a voltage sensor20, a position sensor21, the single-phase inverter11, a control unit25, and a drive signal generation unit32. The voltage sensor20detects a direct-current voltage Vdcoutput from the power supply10. The voltage sensor20may detect a voltage applied to an input end of the motor drive device2, or may detect a direct-current voltage applied to a wiring connected to an output end of the power supply10. The position sensor21detects a rotor rotational position, which is a rotational position of the rotor12a, and outputs information on the detected rotational position as a position sensor signal21a. The position sensor signal21ais a signal that has a potential of one of two values, i.e., a high level or a low level depending on a direction of a magnetic flux generated from the rotor12a. The single-phase inverter11is a power converter having a direct-current/alternating-current conversion function of converting a direct-current voltage output from the power supply10, into an alternating-current voltage and applying the alternating-current voltage to a motor. The control unit25generates PWM signals Q1, Q2, Q3, and Q4on the basis of the direct-current voltage Vdcand the position sensor signal21aoutput from the position sensor21. Hereinafter, the PWM signals Q1, Q2, Q3, and Q4may be simply referred to as PWM signals. The drive signal generation unit32amplifies the PWM signals output from the control unit25and outputs the amplified signals as drive signals S1, S2, S3, and S4for driving switching elements in the single-phase inverter11. The drive signals S1, S2, S3, and S4are signals obtained by amplifying the PWM signals Q1, Q2, Q3, and Q4, respectively. The control unit25includes a processor31, a carrier generation unit33, and a memory34. The processor31is a processing unit that performs various calculations regarding PWM control and advance angle control. Details of the PWM control and the advance angle control will be described later. As the processor31, a central processing unit (CPU, also referred to as a central processing device, a processing device, an arithmetic device, a microprocessor, a microcomputer, a processor, or a digital signal processor (DSP)), or system large scale integration (LSI) can be exemplified. As the memory34, a nonvolatile or volatile semiconductor memory such as a random access memory (RAM), a read only memory (ROM), a flash memory, an erasable programmable read only memory (EPROM), or an electrically erasable programmable read-only memory (EEPROM (registered trademark)) can be exemplified. The memory34is not limited thereto, and may be a magnetic disk, an optical disk, a compact disc, a mini disk, or a digital versatile disc (DVD). The memory34stores a program read by the processor31. The memory34is used as a work area when the processor31performs an arithmetic process. A function of the carrier generation unit33illustrated inFIG.1may be implemented by a processor that executes a dedicated program stored in the memory34, or may be implemented by dedicated hardware. Details of a configuration of the carrier generation unit33will be described later. FIG.2is a diagram illustrating a circuit configuration of the single-phase inverter illustrated inFIG.1. The single-phase inverter11includes a plurality of switching elements51,52,53, and54bridge-connected to one another. In addition to the plurality of switching elements51,52,53, and54of the single-phase inverter11,FIG.2illustrates the single-phase motor12connected to the single-phase inverter11. Each of the two switching elements51and53located on a high-potential side is referred to as an upper-arm switching element. Each of the two switching elements52and54located on a low-potential side is referred to as a lower-arm switching element. The switching element51is a first upper-arm switching element, and the switching element52is a first lower-arm switching element. The switching elements51and52, which are connected in series with each other, define a first arm50A. The switching element53is a second upper-arm switching element, and the switching element54is a second lower-arm switching element. The switching elements53and54, which are connected in series with each other, define a second arm50B. The second arm50B is connected in parallel with the first arm50A. The switching element51has a connection end11-1connected to the switching element52. The switching element53has a connection end11-2connected to the switching element54. The connection ends11-1and11-2define alternating-current ends in a bridge circuit. The single-phase motor12is connected to the connection ends11-1and11-2. Each of the plurality of switching elements51,52,53, and54is a MOSFET which is a metal-oxide-semiconductor field-effect transistor. The MOSFET is an example of a field-effect transistor (FET). A body diode51aconnected in parallel between a drain and a source of the switching element51is formed in the switching element51. A body diode52aconnected in parallel between a drain and a source of the switching element52is formed in the switching element52. A body diode53aconnected in parallel between a drain and a source of the switching element53is formed in the switching element53. A body diode54aconnected in parallel between a drain and a source of the switching element54is formed in the switching element54. Each of the body diodes51a,52a,53a, and54ais a parasitic diode formed inside a MOSFET and is used as a freewheeling diode. Each of the plurality of switching elements51,52,53, and54is, for example, a MOSFET formed of a silicon-based material. However, each of the plurality of switching elements51,52,53, and54is not limited to the MOSFET formed of a silicon-based material, and at least one of the plurality of switching elements51,52,53, and54may be a MOSFET formed of a wide band gap semiconductor such as silicon carbide, a gallium nitride-based material, or diamond. In general, wide band gap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Thus, using a wide band gap semiconductor in at least one of the plurality of switching elements51,52,53, and54increases the withstand voltage and the allowable current density of the switching elements51,52,53, and54, which makes it possible to reduce the size of a semiconductor module incorporating the switching elements51,52,53, and54therein. Since wide band gap semiconductors also have high heat resistance, it is possible to reduce the size of a heat dissipation unit for dissipating heat generated in a semiconductor module, and also to simplify a heat dissipation structure for dissipating the heat generated in the semiconductor module. FIG.3is a diagram illustrating a functional configuration for generating the PWM signals illustrated inFIG.1.FIG.4is a diagram illustrating in detail the carrier comparison unit and the carrier generation unit illustrated inFIG.3. A function of generating the PWM signals Q1, Q2, Q3, and Q4can be implemented by the carrier generation unit33and the carrier comparison unit38illustrated inFIG.3. The function of the carrier comparison unit38is implemented by the processor31illustrated inFIG.1. The carrier comparison unit38receives inputs of an advance phase θv, a reference phase ee, a carrier generated by the carrier generation unit33, the direct-current voltage Vdc, and a voltage amplitude command V* which is an amplitude value of a voltage command Vm. The carrier comparison unit38generates PWM signals on the basis of the advance phase θv, the reference phase ee, the carrier, the direct-current voltage Vdc, and the voltage amplitude command V*. The advance phase θvand the reference phase θeare used to generate voltage commands Vm1and Vm2illustrated inFIG.4. The advance phase θvis calculated by an advance phase calculation unit described later. The “advance phase” is a phase that represents an advance angle θvvwhich is an advanced angle of a voltage command. The “advanced angle” is a phase difference between a motor applied voltage and a motor induced voltage. The motor applied voltage is a voltage applied to a stator winding (not illustrated) by the single-phase inverter11. The motor induced voltage is a voltage induced in the stator winding. The motor applied voltage is synonymous with an inverter output voltage which is an output voltage of the single-phase inverter11. When the motor applied voltage advances relative to the motor induced voltage, the “advanced angle” takes a positive value. The reference phase θeis calculated by a rotation speed calculation unit described later. The reference phase ∴eis a phase obtained by converting a rotor mechanical angle, which is an angle of the rotor12afrom a reference position, into an electrical angle. As illustrated inFIG.4, the carrier generation unit33includes a carrier frequency setting unit33a. A carrier frequency fc[Hz], which is a frequency of a carrier, is set in the carrier frequency setting unit33a. The carrier frequency setting unit33agenerates a carrier synchronized with a cycle of the advance phase θv. The generated carrier is output to the carrier comparison unit38.FIG.4illustrates a waveform of a triangular wave which is an example of the carrier. The triangular wave is a signal wave whose peak value is “1” and whose valley value is “0”. The PWM control on the single-phase inverter11includes synchronous PWM control and asynchronous PWM control. In the case of the asynchronous PWM control, it is not necessary to synchronize the carrier with the advance phase θv. The carrier comparison unit38includes an absolute value calculation unit38a, a division unit38b, a multiplication unit38c, a multiplication unit38d, an addition unit38e, an addition unit38f, a comparison unit38g, a comparison unit38h, an output inversion unit38i, and an output inversion unit38j. The absolute value calculation unit38acalculates an absolute value |V*| of the voltage amplitude command V*. The division unit38bdivides the absolute value |V*| by the direct-current voltage Vdc. The power supply10is, for example, a battery in which case even when a battery voltage decreases, the division of the absolute value |V*| by the direct-current voltage Vdccan increase a modulation rate in such a manner as to prevent the motor applied voltage from decreasing due to that battery voltage decrease, as compared with a case where the battery voltage decreases and the division by the direct-current voltage Vdcis not performed. The battery voltage means an output voltage of the battery. When the power supply10is not a battery but a power conversion device that converts alternating-current power from a commercial power supply into direct-current power, a change in an output voltage of the power conversion device is smaller than a change in an output voltage of the battery because a change in a voltage of the commercial power supply is small. Thus, where the power supply10that outputs direct-current power, using a commercial power supply is connected to the single-phase inverter11, a voltage generated inside the motor drive device2, that is, a direct-current voltage whose voltage indicates a constant value may be input to the division unit38binstead of the direct-current voltage Vdcbeing input to the division unit38b. The multiplication unit38cadds the advance phase θvto the reference phase θe, thereby calculating a sine which is a result of the addition. The multiplication unit38ccalculates the voltage command Vmby multiplying the calculated sine by an output of the division unit38b. The addition unit38eadds 1 to the voltage command Vmwhich is an output of the multiplication unit38c. An output of the addition unit38eis input to the comparison unit38gas the voltage command Vm1for driving the two switching elements51and52illustrated inFIG.2. The voltage command Vm1and the carrier are input to the comparison unit38g. The comparison unit38gcompares the voltage command Vm1with the carrier, thereby providing the comparison result that is the PWM signal Q2. The output inversion unit38iinverts an output of the comparison unit38g. An output of the output inversion unit38iis the PWM signal Q1. The output inversion unit38iprevents the switching elements51and52from being turned on at the same time. The multiplication unit38dmultiplies, by −1, the voltage command Vmwhich is the output of the multiplication unit38c. The addition unit38fadds 1 to an output of the multiplication unit38d. An output of the addition unit38fis input to the comparison unit38has the voltage command Vm2for driving the two switching elements53and54illustrated inFIG.2. The voltage command Vm2and the carrier are input to the comparison unit38h. The comparison unit38hcompares the voltage command Vm2with the carrier, thereby providing the comparison result that is the PWM signal Q4. The output inversion unit38jinverts an output of the comparison unit38h. An output of the output inversion unit38jis the PWM signal Q3. The output inversion unit38jprevents the switching elements53and54from being turned on at the same time. FIG.5is a time chart illustrating waveforms of the voltage commands and the PWM signals illustrated inFIG.4, and the motor applied voltage.FIG.5illustrates waveforms of the position sensor signal, a rotor mechanical angle θm, the reference phase ee, the advance phase θv, the voltage commands Vm1and Vm2, the carrier, the PWM signals Q1, Q2, Q3, and Q4, and the motor applied voltage. The waveform of the voltage command Vm1is indicated by a broken line. The waveform of the voltage command Vm2is indicated by a dot-and-dash line. These waveforms are waveforms detected when the rotor12aincluding four permanent magnets makes one rotation, for example. A, B, C, D, and E indicated by arrows inFIG.5represent timing of commutation of a current flowing through each coil wound around the stator12bof the single-phase motor12. The carrier comparison unit38illustrated inFIG.4can generate the PWM signals Q1, Q2, Q3and Q4, using the voltage commands Vm1and Vm2having the waveforms illustrated inFIG.5. As a result of using such PWM signals Q1, Q2, Q3, and Q4to control the switching elements51,52,53, and54in the single-phase inverter11, a PWM-controlled motor applied voltage is applied to the single-phase motor12. The motor applied voltage is a signal that takes a high level, low level, or zero level potential. Known modulation methods used when generating the PWM signals Q1, Q2, Q3, and Q4include a bipolar modulation method and a unipolar modulation method. The bipolar modulation method is a modulation method that outputs a voltage pulse that changes between positive and negative potentials. The unipolar modulation method is a modulation method that outputs a voltage pulse that changes among three potentials every half cycle of a power supply, that is, a voltage pulse that changes among a positive potential, a negative potential, and a zero potential. The waveforms of the PWM signals Q1, Q2, Q3, and Q4illustrated inFIG.5are obtained by the unipolar modulation. Any modulation method may be used for the motor drive device2according to the present embodiment. Where it is necessary to bring the waveform of the motor applied voltage and the waveform of a current flowing through each coil of the single-phase motor12closer to a sinusoidal wave, the unipolar modulation having a smaller harmonic content is more preferably employed than the bipolar modulation. As described above, the motor applied voltage is determined by comparing the carrier with the voltage commands. As the number of motor rotations increases, the frequency of each voltage command increases, so that the number of voltage pulses included in the motor applied voltage output in one cycle of the electrical angle decreases. As a result, an influence of the number of voltage pulses on distortion of a current waveform increases. Generally, when the number of voltage pulses is an even number, even-order harmonics are superimposed on the motor applied voltage, and the symmetry between a positive-side waveform and a negative-side waveform disappears. Accordingly, in order to bring the waveform of the current flowing through each coil of the single-phase motor12closer to a sinusoidal wave in which the harmonic content is reduced, the number of voltage pulses in one cycle of the electrical angle is preferably controlled so as to be an odd number. Controlling the number of voltage pulses in one cycle of the electrical angle so as to be an odd number makes it possible to bring the waveform of the current flowing through each coil of the single-phase motor12closer to a sinusoidal wave. FIG.6is a diagram illustrating a path of a current flowing through the single-phase inverter when the motor applied voltage illustrated inFIG.5is 0 [V]. The motor applied voltage illustrated inFIG.5is 0 [V] during brake control. That is, the inverter output voltage during brake control is 0 [V]. When the inverter output voltage is 0 [V], the switching elements51and53are controlled such that the switching elements51and53are turned off, and the switching elements52and54are controlled such that the switching elements52and54are turned on. At that time, the single-phase inverter11is in a freewheeling mode. In the freewheeling mode, no current flows between the power supply and the single-phase inverter11, but a current as indicated by a solid line40flows between the single-phase inverter11and the single-phase motor12. This current is a braking current generated by the motor induced voltage. A direction in which the braking current flows is determined depending on a direction of a current flowing in the single-phase motor immediately before the freewheeling, that is, immediately before the brake control is started. The current, which has flowed out of the single-phase motor12, returns to the single-phase motor12by passing through the switching element54and the switching element52. FIG.7is a diagram illustrating an equivalent circuit of the switching elements52and54, and the single-phase motor illustrated inFIG.6. InFIG.7, Rm represents motor winding resistance of the single-phase motor12, Lm represents motor winding inductance of the single-phase motor12, and Ron represents ON-resistance of each of the switching elements52and54. Rp represents the resistance of a wiring line that connects the single-phase motor12, the switching element52, and the switching element54to one another, and Lp represents the inductance of the wiring line. Since the single-phase motor12is rotating at a rotation speed ω, an induced voltage Em is represented by the product of the rotation speed co and an induced voltage constant φ (Em=ωφ). A braking current I flowing when the induced voltage Em is generated can be expressed as I=ωφ/(Rm+2Ron+Rp+jω(Lm+Lp)). Some conventional technique short-circuits coils to thereby decrease a braking current. Unfortunately, the mere short-circuiting of the coils increases di/dt, which is a change component of a motor current. To reduce the change component of the motor current, it is necessary to use a braking resistor or a highly accurate and highly responsive current sensor. The motor drive device2of the present embodiment, which need not use a braking resistor, a current sensor, or the like, can simplify the configuration of the motor drive system1and improve reliability. Next, the advance angle control in the present embodiment will be described.FIG.8is a diagram illustrating a functional configuration for calculating an advance phase input to the carrier generation unit and the carrier comparison unit illustrated inFIGS.3and4. A function of each of a rotation speed calculation unit42, an advance phase calculation unit44, and a voltage amplitude command control unit45illustrated inFIG.8is implemented by the processor31and the memory34illustrated inFIG.1. That is, a computer program for executing processes of the rotation speed calculation unit42, the advance phase calculation unit44, and the voltage amplitude command control unit45is stored in the memory34, and then the processor31reads and executes the program, and thereby the functions of the rotation speed calculation unit42, the advance phase calculation unit44, and the voltage amplitude command control unit45are realized. The voltage amplitude command control unit45includes a comparator45aand a command adjustment unit45b. The rotation speed calculation unit42calculates the rotation speed co and the reference phase θeof the single-phase motor12on the basis of the position sensor signal21a. The reference phase θeis a phase obtained by converting the rotor mechanical angle θm, which is a rotation angle of the rotor12afrom the reference position, into an electrical angle. The advance phase calculation unit44calculates the advance phase θvon the basis of the rotation speed co and the reference phase θecalculated by the rotation speed calculation unit42. FIG.9is a diagram illustrating an example of a method of calculating the advance phase illustrated inFIG.8. The horizontal axis inFIG.9represents the number of motor rotations N, and the vertical axis inFIG.9represents the advance phase θv. The number of motor rotations N is the number of rotations per unit time and corresponds to a rotation speed. As illustrated inFIG.9, the advance phase θvcan be determined using a function in which the advance phase θvincreases as the number of motor rotations N increases. Although the example inFIG.9provides the advance phase θvdetermined by a first order linear function, the function determining the advance phase θvis not limited thereto. Any function other than the first order linear function may be used as long as the advance phase θvbecomes the same or large in correspondence to the increase in the number of motor rotations N. When the number of motor rotations N increases, the frequency of the position sensor signal21aincreases, so that the advance phase calculation unit44illustrated inFIG.8increases the advance phase θv. In a case where a load connected to the single-phase motor is reduced when the advance phase θvis increased in the manner as described above, the number of motor rotations N continues to increase. In the case of an electric vacuum cleaner having the single-phase motor installed therein, for example, a suction tool of the electric vacuum cleaner fails to suck in air due to contact with the floor, which is a surface to be cleaned. For this reason, an inside of the electric vacuum cleaner is brought into a reduced pressure state or a vacuum state. As a result, the air resistance of a fan that is a load connected to the single-phase motor is considerably reduced in which case the number of motor rotations N continues to increase. When the number of motor rotations N increases in this manner, the magnets provided on the rotor surface may scatter due to a centrifugal force. In addition, the centrifugal force may deform or destroy the fan. For this reason, generally, an upper limit value is imposed on the number of motor rotations N, and the number of motor rotations N is controlled such that the number of motor rotations N does not exceed the upper limit value. In the present embodiment, when the number of motor rotations N exceeds the upper limit value, a value of the voltage amplitude command V* is gradually decreased so that the amplitudes of the voltage commands Vm1and Vm2illustrated inFIG.5decrease to be smaller than the amplitudes of the voltage commands Vm1and Vm2before the number of motor rotations N exceeds the threshold. For example, the value of the voltage amplitude command V* is decreased by a fixed decrease amount per unit time. Due to the decrease in the voltage amplitude command V*, the width of a zero vector section of the motor applied voltage is gradually increased to be wider than the width of the zero vector section before the number of motor rotations N exceeds the upper limit value. The zero vector section is a section in which a potential of the motor applied voltage illustrated inFIG.5is zero level, that is, the motor applied voltage is 0 [V]. Next, a description will be described as to an operation of changing the width of the zero vector section of the motor applied voltage when the number of motor rotations exceeds the upper limit value.FIG.10is a flowchart explaining operations of the voltage amplitude command control unit illustrated inFIG.8and the carrier comparison unit illustrated inFIG.4. The comparator45aillustrated inFIG.8compares the rotation speed co calculated by the rotation speed calculation unit42with the rotation speed threshold ωth, thereby determining whether the rotation speed co has exceeded the rotation speed threshold ωth, that is, whether the number of motor rotations N has exceeded a specific threshold (step S1). If the rotation speed ω has not exceeded the rotation speed threshold ωth (step S1, No), the process in step S1is repeated until the rotation speed ω exceeds the rotation speed threshold ωth. At that time, the command adjustment unit45boutputs a voltage amplitude command V* that makes the amplitudes of the voltage commands Vm1and Vm2constant. If the rotation speed ω has exceeded the rotation speed threshold ωth (step S1, Yes), the comparator45aoutputs, to the command adjustment unit45b, speed excess information indicating that the rotation speed ω has exceeded the rotation speed threshold ωth (step S2). When the speed excess information is input to the command adjustment unit45b, the command adjustment unit45bgradually decreases the value of the voltage amplitude command V* (step S3). The voltage commands Vm1and Vm2generated on the basis of the voltage amplitude command V* are compared with the carrier, thereby generating the PWM signals. The voltage commands Vm1and Vm2are updated at timing when the carrier rises up to the peak or falls down to the valley. The command adjustment unit45bupdates the voltage amplitude command V* at the timing when the carrier rises up to the peak or falls down to the valley. To update the voltage amplitude command V*, for example, the command adjustment unit45bsubtracts a constant value from the previous voltage amplitude command V* to thereby update the latest voltage amplitude command V*. This operation is repeated to thereby gradually decrease the value of the voltage amplitude command V*. The decrease in the value of the voltage amplitude command V* results in a decrease in an amplitude value of the voltage command Vmin the carrier comparison unit38illustrated inFIG.4. The decrease in the amplitude value of the voltage command Vmresults in a decrease in an amplitude value of each of the voltage commands Vm1and Vm2that are to be compared with the carrier. Consequently, the width of an ON-interval of each PWM signal widens, so that the width of the zero vector section of the motor applied voltage illustrated inFIG.5is gradually increased to be wider than the width of the zero vector section before the rotation speed ω exceeds the rotation speed threshold ωth (step S4). As described above, even when the number of motor rotations N rapidly increases due to the reduced load, the motor drive device2according to the present embodiment performs control for widening the width of the zero vector section of the motor applied voltage, thereby making it possible to brake the motor while reducing demagnetization. After step S4, the processor31compares, for example, the direct-current voltage Vdcwith the induced voltage Em calculated by the product of the rotation speed ω and the induced voltage constant φ, and determines whether the induced voltage Em has decreased to be lower than the direct-current voltage Vdc(step S5). That is, the processor31uses the direct-current voltage Vdcto determine whether the induced voltage generated in the motor has decreased to be lower than the direct-current voltage. If the induced voltage Em is higher than the direct-current voltage Vdc(step S5, No), the processes from step S1onward are repeated until the induced voltage Em decreases to be lower than the direct-current voltage Vdc. If the induced voltage Em has decreased to be lower than the direct-current voltage Vdc(step S5, Yes), the processor31stops the carrier signal generation performed by the carrier comparison unit38(step S6). Consequently, the operation of the single-phase inverter11stops, thereby preventing the increase in the regenerative voltage from the single-phase inverter11, and thus preventing, for example, an increase in each of voltages applied to a smoothing capacitor3and the power supply10illustrated inFIG.6. As a result, the life of the smoothing capacitor3and the power supply10can be extended. FIG.11is a diagram illustrating a relationship between the motor current flowing through the winding of the stator illustrated inFIG.1and a brake torque generated in the motor.FIG.11illustrates a waveform of the position sensor signal, the rotor mechanical angle θm, a waveform of the induced voltage, a waveform of the motor current, and a waveform of the brake torque. In addition,FIG.11illustrates, for example, 0°, 45°, 90°, 135°, and 180° as the rotor mechanical angle θmwhen the rotor12arotates clockwise. When the rotor12arotates clockwise, the position sensor signal21acorresponding to the rotor mechanical angle θmis output. If the rotation speed ω is sufficiently high and ωL>>R holds, the motor current has a waveform having its phase lagging 90 degrees behind the induced voltage. Accordingly, the brake torque is determined by the product of the magnetic flux generated from the magnets provided on the rotor and the motor current, and changes as illustrated inFIG.11. The average value of the brake torque is 0. In a case where the PWM signals for turning off the switching elements51and53, and turning on the switching elements52and54, as illustrated inFIG.6, are continues while the induced voltage is generated, the motor current rapidly increases and a large braking current transiently flows. As a result, the magnets on the rotor may be demagnetized. In the present embodiment, when the PWM signals as illustrated inFIG.5are generated, the amplitudes of the voltage commands Vm1and Vm2are controlled such that the amplitudes of the voltage commands Vm1and Vm2are reduced to thereby gradually widen the width of the zero vector section. The gradual widening of the width of the zero vector section makes it possible to reduce an increase in the braking current when the number of motor rotations N rapidly increases. Since the increase in the braking current is reduced, the occurrence of demagnetization is reduced. In addition, since the width of the zero vector section is gradually widened, it is possible to decrease the braking current while decreasing the motor applied voltage. Therefore, stop time of the motor can be shortened as compared with a case where there is no zero vector section. In addition, since the increase in the braking current is reduced, the increase in the current flowing through each of the plurality of switching elements51,52,53, and54is also reduced. Therefore, it is possible to prevent each of the plurality of switching elements51,52,53, and54from breaking down by exceeding its own withstand current (maximum current). Furthermore, since it is possible to prevent a switching element from exceeding a withstand current, a switching element having a small-capacity switching element providing a small allowable amount of current can be used, and an increase in manufacturing cost of the single-phase inverter11can be reduced. Next, a configuration of the drive signal generation unit32will be described.FIG.12is a diagram illustrating an example configuration of a signal generation circuit included in the drive signal generation unit illustrated inFIG.1. A signal generation circuit32A illustrated inFIG.12is a circuit that generates the drive signals S1and S2. Note thatFIG.12does not illustrate a circuit that generates the drive signals S3and S4, but since the circuit is configured similarly to the signal generation circuit32A illustrated inFIG.12, a description of a configuration thereof will be omitted hereinafter. The signal generation circuit32A includes a control power supply300which is a direct-current voltage source, a bootstrap circuit200, a high-voltage drive circuit400, and a low-voltage drive circuit401. The bootstrap circuit200includes a boot diode201whose anode is connected to the control power supply300, and a boot capacitor202whose one end is connected to a cathode of the boot diode201. The other end of the boot capacitor202is connected to the connection end11-1of the switching element51to the switching element52. The boot capacitor202functions to increase a voltage for operating the high-voltage drive circuit400higher than a voltage output from the control power supply300. For the bootstrap circuit200configured as described above, the boot capacitor202is charged through a current flowing through a path defined by the control power supply300, the boot diode201, the boot capacitor202, and the switching element52when the switching element52is turned on. A capacitor voltage Vcgenerated across the charged boot capacitor202can be expressed as Vc=Vcc+VBD−Vfwhere Vccrepresents a voltage of the control power supply300, VBDrepresents a forward voltage of the body diode52a, and Vfrepresents a forward voltage of the boot diode201. The high-voltage drive circuit400uses, as a power supply voltage, a voltage output from the bootstrap circuit200to convert the PWM signal Q1into the drive signal S1and outputs the drive signal S1to a gate of the switching element51. The low-voltage drive circuit401uses, as a power supply voltage, the voltage output from the control power supply300to convert the PWM signal Q2into the drive signal S2and outputs the drive signal S2to a gate of the switching element52. Note that circuits similar to the high-voltage drive circuit400and the bootstrap circuit200illustrated inFIG.12are used to generate the drive signal S3for the switching element53illustrated inFIG.2. In addition, a circuit similar to the low-voltage drive circuit401illustrated inFIG.12is used to generate the drive signal S4for the switching element54illustrated inFIG.2. For the motor drive device2according to the present embodiment, the upper-arm switching elements are turned off and the lower-arm switching elements are turned on, thereby allowing the motor applied voltage to include a zero vector section in which a current circulates among the lower-arm switching elements and the single-phase motor12. Consequently, charge is stored in the boot capacitor202, which is a drive power supply for the upper-arm switching element, and a voltage required for an operation of the upper-arm switching element can be stabilized. A general electric blower is controlled such that the number of rotations is constant. For such a constant rotation number control, an overcurrent may flow through a motor. The reason why the overcurrent flows is that a current changes rapidly in an attempt to keep the number of motor rotations constant when a load changes. More specifically, when the control rotation number control is performed at a time of transition from a “light load state”, i.e., a “small load torque state” to a “heavy load state”, i.e., a “large load torque state”, a motor output torque needs to be increased in an attempt to keep the number of rotations unchanged, which results in an increased amount of change in a motor current. The present embodiment provides control performed so that the voltage amplitude command V* is constant during steady operation. Since the voltage amplitude command V* is constant, the voltage amplitude command V* is not changed when a load increases. As a result, the number of motor rotations decreases in correspondence to an increase in a load torque. This control prevents an abrupt change in the motor current and an overcurrent, thereby achieving an electric blower and an electric vacuum cleaner that rotate stably. In a case of the electric blower, the load torque increases due to an increase in the number of rotations of blades that are a load of the motor, and also increases due to an increase in the diameter of an air passage. The diameter of the air passage indicates, for example, a size of a suction port of an electric vacuum cleaner. For example, when the diameter of the air passage is large because nothing is in contact with the suction port, a force for sucking the wind is required. Accordingly, the load torque increases under the condition where the blades rotate at the same number of rotations. On the other hand, when the suction port is closed by something in contact with the suction port, the diameter of the air passage is narrowed and the force for sucking the wind is not necessary. Therefore, the load torque decreases under the condition where the blades rotate at the same number of rotations. Next, an effect of the advance angle control will be described. Increasing the advance phase θvin correspondence to an increase in the number of rotations can widen a range of the number of rotations. When the advance phase θvis set to “0”, the number of rotations is saturated at a point where the motor applied voltage is equal to the motor induced voltage. In order to further increase the number of rotations, the advance phase θvis advanced to weaken a magnetic flux generated in the stator due to an armature reaction, thereby reducing an increase in the motor induced voltage and thus increasing the number of rotations. Accordingly, a wide region of the number of rotations can be obtained by selecting the advance phase θvin correspondence to the number of rotations. In applying the advance angle control according to the present embodiment to the electric vacuum cleaner, a voltage command is kept constant regardless of a change in a closed state of the suction port, that is, regardless of the load torque, and the advance phase θvwhich is an advanced angle of the voltage command is increased in correspondence to an increase in the rotation speed. With such control, stable driving is possible in a wide rotation speed range. Next, a loss reduction method in the present embodiment will be described with reference toFIGS.13to16.FIG.13is a first diagram illustrating a path of a motor current depending on the polarity of the inverter output voltage.FIG.14is a second diagram illustrating the path of the motor current depending on the polarity of the inverter output voltage.FIG.15is a third diagram illustrating the path of the motor current depending on the polarity of the inverter output voltage.FIG.16is a schematic cross-sectional view illustrating a schematic structure of a MOSFET that can be used as the switching elements illustrated inFIG.2. First, the schematic structure of the MOSFET will be described with reference toFIG.16, and then the path of the motor current will be described with reference toFIGS.13to15. FIG.16illustrates an n-type MOSFET. In a case of the n-type MOSFET, a p-type semiconductor substrate600is used as illustrated inFIG.16. A source electrode S, a drain electrode D, and a gate electrode G are formed on the semiconductor substrate600. A high-concentration impurity is ion-implanted to form a region601of n-type at each of portions in contact with the source electrode S and the drain electrode D. In addition, the semiconductor substrate600has an oxide insulating film602formed thereon between the gate electrode G and a portion where the region601of n-type is not formed. That is, the oxide insulating film602is interposed between the gate electrode G and a region603of p-type in the semiconductor substrate600. When a positive voltage is applied to the gate electrode G, electrons are attracted to a boundary surface between the region603of p-type in the semiconductor substrate600and the oxide insulating film602, such that the boundary surface is negatively charged. In a portion where the electrons are gathered, the electron density becomes higher than the hole density, such that the portion is changed into an n-type portion. The n-type portion serves as a path for a current and is called a channel604. The channel604is an n-type channel in the example inFIG.16. The MOSFET is controlled such that the MOSFET is turned on, thereby allowing more current to flow through the channel604than through a body diode formed in the region603of p-type. When the polarity of the inverter output voltage is positive, as indicated by thick solid line (a) inFIG.13, the current flows into the single-phase motor12through a channel of the switching element51which is an upper arm of a first phase, and flows out of the single-phase motor12and through a channel of the switching element54which is a lower arm of a second phase. When the polarity of the inverter output voltage is negative, as indicated by thick broken line (b) inFIG.13, the current flows into the single-phase motor12through a channel of the switching element53which is an upper arm of the second phase, and flows out of the single-phase motor12and through a channel of the switching element52which is a lower arm of the first phase. Next, a current path when the inverter output voltage is zero, that is, when a zero voltage is output from the single-phase inverter11will be described. When the inverter output voltage becomes zero after the positive inverter output voltage is generated, a current flows in a freewheeling mode in which a current flows between the single-phase inverter11and the single-phase motor12without current flowing from a power supply side, as indicated by thick solid line (c) inFIG.14. More specifically, the direction of the current having flowed through the single-phase motor12immediately before that freewheeling mode is unchanged, such that the current flows out of the single-phase motor12and returns to the single-phase motor12through the channel of the switching element54which is the lower arm of the second phase and the body diode52aof the switching element52which is the lower arm of the first phase. When the inverter output voltage becomes zero after the negative inverter output voltage is generated, the direction of the current having flowed immediately therebefore is opposite to the above direction of flow of current immediately before the inverter output voltage changes from the positive inverter output voltage to zero. As a result, as indicated by thick broken line (d) inFIG.14, the direction of the freewheeling current is opposite to that indicated by thick solid line (c) inFIG.14. More specifically, the current flowing out of the single-phase motor12returns to single-phase motor12through the body diode51aof the switching element51which is the upper arm of the first phase and the channel of the switching element53which is the upper arm of the second phase. As described above, in the freewheeling mode in which the current freewheelingly flows between the single-phase motor12and the single-phase inverter11, the current flows through the body diode in either one of the first phase and the second phase. Generally, it is known that conduction loss is generally smaller when a current passes through a channel of a MOSFET than when a current passes through in a forward direction of a diode. Therefore, in the present embodiment, the MOSFET including a body diode which would allow a current to flow therethrough is controlled such that the MOSFET is turned on to thereby reduce a flow current flowing through that body diode in the freewheeling mode providing the freewheeling current flows. The switching element52is controlled such that the switching element52is turned on at a timing of the flow of the freewheeling current indicated by thick solid line (c) inFIG.14in the freewheeling mode. Such control on the switching element52allows most of the freewheeling current to flow through the channel of the switching element52having a small resistance value, as indicated by thick solid line (e) inFIG.15. Consequently, the conduction loss in the switching element52is reduced. In addition, the switching element51is controlled such that the switching element51is turned on at timing when the freewheeling current indicated by thick broken line (d) ofFIG.14flows. Such control on the switching element51allows most of the freewheeling current to flow through the channel of the switching element51having a small resistance value, as indicated by thick broken line (f) inFIG.15. Consequently, the conduction loss in the switching element51is reduced. As described above, the MOSFET including a body diode is controlled such that the MOSFET is turned on at the timing when the freewheeling current flows through that body diode. As a result, the loss in the switching element can be reduced. The MOSFET, which can be controlled in the above manner, can be surface-mounted on a substrate such that heat can be dissipated on the substrate. Part or all of the switching elements are formed of wide band gap semiconductors, such that heat generation in the MOSFET is reduced only by the substrate. Note that if heat can be dissipated only by the substrate, a heat sink is not required, which contributes to reduction in size of an inverter and can lead to reduction in size of a product. In addition to the above-described heat dissipation method, a further heat dissipation effect can be obtained by installing the substrate in the air passage. Here, the air passage is a space around a fan such as an electric blower generating an air flow, or a passage through which the wind generated by the electric blower flows. As a result of installation of the substrate in the air passage, heat in a semiconductor element on the substrate can be dissipated by the wind generated by the electric blower, so that heat generation in the semiconductor element can be significantly reduced. Next, an application example of the motor drive device according to the embodiment will be described.FIG.17is a configuration diagram of an electric vacuum cleaner including the motor drive device according to the embodiment of the present invention. An electric vacuum cleaner61includes a battery67which is a direct-current power supply, the motor drive device2illustrated inFIG.1, an electric blower64driven by the single-phase motor12illustrated inFIG.1, a dust collection chamber65, a sensor68, a suction port body63, an extension pipe62, and an operation portion66. The battery67corresponds to the power supply10illustrated inFIG.1. A user who uses the electric vacuum cleaner61holds the operation portion66and operates the electric vacuum cleaner61. The motor drive device2of the electric vacuum cleaner61drives the electric blower64by using the battery67as a power supply. By driving the electric blower64, dust is sucked from the suction port body63, and the sucked dust is collected in the dust collection chamber65via the extension pipe62. The electric vacuum cleaner61is a product whose number of motor rotations changes from 0 [rpm] to 100,000 [rpm]. The number of motor rotations may reach a value of 100,000 [rpm] or more. The control method according to the embodiment described above is suitable for driving such a product in which the single-phase motor12rotates at a high speed. As described above, the electric vacuum cleaner61is an application in which a load changes depending on a contact area between the suction port of the electric vacuum cleaner and the floor surface, and is also an application with a high acceleration rate. For this reason, the rotation speed of the motor mounted on the electric vacuum cleaner61may reach a limit value instantly. According to the motor drive device2of the present embodiment, since the brake control is performed in an application in which the rotation speed of the motor rapidly increases, the rotation speed of the motor can be kept within an operable range of the product, and thus, the operation quality of the electric vacuum cleaner61can be improved and the reliability of the electric vacuum cleaner61can be increased. In addition, according to the motor drive device2of the present embodiment, the brake torque is generated by turning on the lower-arm switching elements when the brake control is performed. At that time, the boot capacitor202is instantly charged via the lower-arm switching element. It is therefore possible to shorten or eliminate time for charging the boot capacitor202in restarting the single-phase inverter11immediately after the single-phase inverter11stops. Accordingly, the restart time of the single-phase inverter11can be shortened. FIG.18is a configuration view of a hand dryer including the motor drive device according to the embodiment of the present invention. A hand dryer90includes the motor drive device2, a casing91, a hand detection sensor92, a water receiving portion93, a drain container94, a cover96, a sensor97, an air inlet port98, and an electric blower95. The sensor97is either a gyro sensor or a motion sensor. When the hand is inserted into a hand insertion portion99located above the water receiving portion93of the hand dryer90, the electric blower95blows off water by air blow, and the blown-off water is collected in the water receiving portion93and then stored in the drain container94. The hand dryer90is a product whose number of motor rotations changes from 0 [rpm] to 100,000 [rpm] as in the electric vacuum cleaner61illustrated inFIG.17. Therefore, the control method according to the embodiment described above is suitable for the hand dryer90as well, and can provide an effect similar to that of the electric vacuum cleaner61. FIG.19is a diagram for explaining modulation control performed by the motor drive device according to the embodiment of the present invention. The left side ofFIG.18illustrates a relationship between the number of rotations and a modulation rate. The right side ofFIG.18illustrates a waveform of an inverter output voltage when the modulation rate is 1.0 or less, and a waveform of the inverter output voltage when the modulation rate exceeds 1.0. Generally, a load torque of a rotating body increases as the number of rotations increases. For this reason, it is necessary to increase a motor output torque as the number of rotations increases. In addition, generally, the motor output torque increases in proportion to a motor current, and an increase in the motor current requires an increase in the inverter output voltage. Accordingly, the number of rotations can be increased by increasing the modulation rate and increasing the inverter output voltage. Next, control on the number of rotations in the present embodiment will be described. The following description is based on the assumption that an electric blower is a load, and an operating range of the electric blower is divided as follows. (A) Low-speed rotation region (region of a low number of rotations): 0 [rpm] to 100,000 [rpm] (B) High-speed rotation region (region of a high number of rotations): 100,000 [rpm] or more A region sandwiched between (A) and (B) above is a gray area, and may be included in the low-speed rotation region or in the high-speed rotation region depending on applications. First, the control in the low-speed rotation region will be described. In the low-speed rotation region, PWM control is performed with a modulation rate of 1.0 or less. Setting the modulation rate to 1.0 or less allows the motor current to be controlled such that the motor current provides a sinusoidal wave, and higher efficiency of the motor can be achieved. When the motor is operated using the carrier frequency common to the low-speed rotation region and the high-speed rotation region, the carrier frequency conforms to the high-speed rotation region, and therefore PWM pulses in the low-speed rotation region tends to increase more than necessary. For this reason, a method of lowering the carrier frequency in the low-speed rotation region to reduce switching loss may be used. Alternatively, control may be performed in such a way that the carrier frequency is changed in synchronization with the number of rotations, thereby preventing the number of pulses from changing in correspondence to the number of rotations. Next, the control in the high-speed rotation region will be described. In the high-speed rotation region, the modulation rate is set to a value larger than 1.0. Setting the modulation rate to larger than 1.0 allows the inverter output voltage to increase and the number of times of switching performed by the switching elements in the inverter to decrease, thereby making it possible to reduce an increase in switching loss. The modulation rate exceeding 1.0 increases the motor output voltage, but decreases the number of times of switching, which results in concern about current distortion. However, the current distortion in the high-speed rotation region is smaller than that in the low-speed rotation region, thus lessening an effect on waveform distortion because a reactance component of the motor increases and di/dt which is a change component of the motor current decreases during high-speed rotation. Accordingly, in the high-speed rotation region, the modulation rate is set to a value larger than 1.0, and control is performed so that the number of switching pulses is reduced. By this control, an increase in switching loss can be reduced and higher efficiency can be achieved. As described above, the boundary between the low-speed rotation region and the high-speed rotation region is ambiguous. Therefore, a first rotation speed that determines the boundary between the low-speed rotation region and the high-speed rotation region is set in the control unit25. The control unit25performs control so that the modulation rate is set to 1.0 or less when the rotation speed of the motor or the load is equal to or lower than the first rotation speed, and the modulation rate is set to be exceeding 1 when the rotation speed of the motor or the load exceeds the first rotation speed. As described above, in the present embodiment, the example configuration has been described in which the motor drive device2is applied to the electric vacuum cleaner61and the hand dryer90, but the motor drive device2can be applied to an electric device on which a motor is mounted. Examples of the electric device on which a motor is mounted include an incinerator, a crusher, a dryer, a dust collector, a printing machine, a cleaning machine, a confectionery machine, a tea making machine, a woodworking machine, a plastic extruder, a cardboard machine, a packaging machine, a hot air generator, an office automation appliance, and an electric blower. The electric blower is a blowing means for object transportation, dust suction, or general blowing and exhausting. The configurations described in the embodiment above are merely examples of the content of the present invention and can be combined with other known technology and part thereof can be omitted or modified without departing from the gist of the present invention. | 57,180 |
11863105 | DETAILED DESCRIPTION In the following, a method for controlling an electric motor where a pulse-width modulation (PWM) switching frequency is determined based on a planned reference speed of the electric motor, a control system for controlling an electric motor, and an industrial robot comprising a control system, will be described. The same reference numerals will be used to denote the same or similar structural features. FIG.1schematically represents a block diagram of one example of an industrial robot10comprising a control system12and a manipulator14having a plurality of electric motors16. The industrial robot10according toFIG.1constitutes one of many possible implementations of a method for controlling an electric motor16according to the present disclosure. InFIG.1, the electric motors16of the manipulator14are used to control movements (e.g. rotational or translational) of a plurality of links (not shown) relative to each other. Each electric motor16is arranged to drive a joint (not shown) between two adjacent links. In the non-limiting example inFIG.1, each electric motor16is constituted by a rotary electric servomotor having 16 poles. The manipulator14is illustrated as comprising six electric motors16, but the number of electric motors16may be increased or reduced. The manipulator14further comprises a plurality of position sensors18, e.g. resolvers, associated with the electric motors16. Each position sensor18is arranged for real-time detection of the rotational position of an associated electric motor16. Signals representing the measured position20of each electric motor16are sent to the control system12. Optionally, the manipulator14further comprises one or more speed detection sensors (not shown) for real-time detection of the rotational speed of each electric motor16. The control system12comprises a plurality of drive units22. Each drive unit22is configured to produce an alternating current24with a certain electric frequency produced by PWM technique. In the example inFIG.1, the control system12comprises one drive unit22associated with each electric motor16. One drive unit22may however alternatively drive a plurality of electric motors16. Each drive unit22may comprise a rectifier for converting AC into DC, a frequency inverter, and a DC bus connected between the rectifier and the inverter. The inverter converts the DC current to a variable alternating current24. Each drive unit22may further comprise an inverter control unit that controls switching of one or more switching elements of the inverter according to a commanded switching frequency. The variable alternating current24from the inverter of the drive unit22is supplied to an associated electric motor16. The inverter of the drive unit22may further comprise an IGBT (insulated-gate bipolar transistor) module. The IGBT module has a lifetime depending on the power cycling. If high PWM switching frequencies are used also at lower speeds of the associated electric motor16, the lifetime of the IGBT module is reduced. The control system12of this example further comprises a main computer26. The main computer26comprises a data processing device28(e.g. a central processing unit, CPU) and a memory30. A computer program is stored in the memory30. A manipulator program, a model of the manipulator14and a path planner is implemented in the main computer26, e.g. in the memory30. The path planner plans the path of the manipulator14. For each electric motor16of the manipulator14, the path planner generates a signal representing planned reference speed32based on movement instructions from the manipulator program and the model of the manipulator14. The control system12is thereby configured to determine a planned reference speed32of the electric motor16. The path planner may further generate a signal representing a reference position34based on movement instructions from the manipulator program and the model of the manipulator14for each electric motor16. The planned reference speed32, and optionally the reference position34for each electric motor16, are sent to the associated drive unit22. The planned reference speed32for each electric motor16and the measured position20for each electric motor16are used by the associated drive unit22for close loop PID control of the electric motor16. The control parameters of the PID control (Kp, Ki, Kd) may be regulated with a function of the planned reference speed32and the PWM switching frequency. The alternating current24may for example be generated by means of SVPWM (Space Vector PWM) and current loop PI control, implemented in each drive unit22. The control parameters of the PI control (Kp, Ki) may be regulated with a function of the planned reference speed32and the PWM switching frequency. The control system12is further configured to determine a PWM switching frequency based on the planned reference speed32of the associated electric motor16. A high switching frequency may be set when the planned reference speed32is high and a low switching frequency may be set when the planned reference speed32is low. The alternating current24, produced by PWM switching with the determined PWM switching frequency, is output to the electric motor16when driving the electric motor16at the planned reference speed32. The PWM switching frequency may be set to a predetermined low value (e.g. 2 kHz) when the planned reference speed32is below a low speed threshold value (e.g. 250 rpm). The PWM switching frequency may be set to a predetermined high value (e.g., 10 kHz) when the planned reference speed32is above a high speed threshold value (e.g. 6000 rpm). The PWM switching frequency may be set to an intermediate value (between the predetermined low value and the predetermined high value) proportional to the planned reference speed32when the planned reference speed32is between the low speed threshold value and the high speed threshold value. While the present disclosure has been described with reference to exemplary embodiments, it will be appreciated that the present invention is not limited to what has been described above. For example, it will be appreciated that the dimensions of the parts may be varied as needed. | 6,201 |
11863106 | DESCRIPTION OF EXEMPLARY EMBODIMENTS For a better understanding the above description and for the sole purpose of providing an example, some non-limiting drawings are included that schematically depict exemplary embodiments. FIG.2shows a power supply control circuit200for a BLDC130having a “Y” formation. In another example, the BLDC130can have a “Delta” formation. The advantages of the configuration of the power supply control circuit200are cost reduction, the PCB area for placement of the control circuit is smaller compared to other configurations. Furthermore, one single ADC converters is needed. The control circuit200comprises a microcontroller/PWM control block210that generates a PWM signal for a power inverter120which produces an AC electric current that feeds the BLDC motor130. The microcontroller/PWM block210controls current in the brushless DC motor130based on a control algorithm implemented by the microcontroller. The control circuit200comprises means for calculating a DC current consumption ISbased on a phase current signal shunt of the BLDC motor, the means comprises a Rshunt215for measuring low-side current sensing and an input AC filter103. In order to obtain the DC current consumption IS, the voltage Vshuntcorresponding to the current Ishuntacross Rshunt215is measured, amplified, offset corrected and filtered as shown inFIG.4. The DC current consumption ISis calculated according to equation 1. FIG.3shows a power supply control circuit300for the BLDC130. The advantages of the configuration of the power supply control circuit300are that this configuration can obtain more precise phase currents readings and involve less acoustic noise and less total harmonic distortion (THD). Hence, the control algorithm can be improved when calculated DC current consumption ISis used to implement the control algorithm. The control circuit300comprises a microcontroller/PWM generator control block310that generates six PWM signals for the power inverter120that feeds the DC motor130. The microcontroller/PWM generator block310controls current in the brushless DC motor130based on the control algorithm. In some examples, the DC current consumption ISof the BLDC motor130can be used as input to the control algorithm. The control circuit300comprises means for calculating the DC current consumption ISbased on a phase current signal IShTIshunt) the means comprises a three shunt resistors RSh1, RSh2, and RSh3for measuring each phase current and an input AC filter103. In order to obtain the DC current consumption IS, three voltages VSh1, VSh2, and VSh3are measured that correspond to the currents ISh1, ISh2, and ISh3across the three shunt resistors RSh1, RSh2and RSh3, respectively. The total phase current signal can be calculated as IShT=ISh1ISh2ISh3. Furthermore, the three voltages VSh1, VSh2, and VSh3are amplified, offset corrected, summed, and filtered as shown inFIG.5. The DC current consumption ISis calculated according to equation 2. FIG.4shows the control circuit200previously shown inFIG.2comprising means for calculating a DC current consumption IS, the means comprises signal processing elements to obtain the DC current consumption ISof the BLDC motor130. In this example, the DC current consumption IScan be used as input for the microcontroller/PWM block210as shown in the figure. These signal processing elements comprise an amplifier405to amplify the voltage Vshuntand perform an offset correction and a low pass filter410to filter the voltage signal Vshuntfrom the amplifier405. Hence, an output voltage VOis obtained from the low pass filter410and the DC current consumption of the BLDC motor ISis obtained based on said voltage VOand according to the equation 1. FIG.5shows the control circuit300previously shown inFIG.3further comprising means for calculating a DC current consumption IS, the means comprises signal processing elements to obtain the DC current consumption ISof the BLDC motor130. In this example, the DC current consumption IScan also be used as input for the microcontroller/PWM block310as inFIG.4. As previously mentioned, the means for calculating a DC current consumption IScomprises three shunt resistors RSh1, RSh2, and RSh3. The signal processing elements comprise three amplifiers505,510, and515to amplify three measured voltage levels VSh1, VSh2, and VSh3corresponding to the shunt resistors RSh1, RSh2, and RSh3, respectively, and to perform an offset correction of the voltage levels. Furthermore, the signal processing elements comprise a summation module as, e.g., a summing amplifier (not drawn in the figure) but represented by reference520to obtain a total voltage VShT=VSh1VSh2VSh3. A low pass filter is also included as part of the control circuit300to filter the total voltage VShTin order to obtain an output voltage VO. Finally, the DC current consumption of the BLDC motor ISis obtained based on said voltage VOand according to the equation 2. Even though reference has been made to exemplary embodiments of the disclosure, it is obvious for a person skilled in the art that the BLDC power supply control circuit architectures described herein are susceptible to numerous variations and modifications, and that all the details mentioned can be substituted for other technically equivalent ones without departing from the scope of protection defined by the attached claims. | 5,370 |
11863107 | DETAILED DESCRIPTION OF THE INVENTION The present invention is described with reference to the attached figures, where like reference numerals are used throughout the figures to designate similar or equivalent elements. The figures are not drawn to scale and are provided merely to illustrate the instant invention. Several aspects of the invention are described below with reference to example applications for illustration. It should be understood that numerous specific details, relationships, and methods are set forth to provide a full understanding of the invention. One having ordinary skill in the relevant art, however, will readily recognize that the invention can be practiced without one or more of the specific details, or with other methods. In other instances, well-known structures or operations are not shown in detail to avoid obscuring the invention. The present invention is not limited by the illustrated ordering of acts or events, as some acts may occur in different orders and/or concurrently with other acts or events. Furthermore, not all illustrated acts or events are required to implement a methodology in accordance with the present invention. The following description is an embodiment of the present invention. The purpose of the present invention is to exemplify the general principles of the invention and should not be construed as limiting the scope of the invention, which is defined by the scope of the claims. FIG.1is a relationship diagram of position estimation error and the d-q axis current of a rotary electric machine control device, in accordance with an embodiment of the conventional art. InFIG.1, the vertical axis depicts position estimation errors estimated by the control device of a rotary electric machine, and the horizontal axis depicts the values of q-axis current (iq). Regarding experiment conditions, the magnitude of the d-axis current (id) can be adjusted between 0%-100% of a maximum rated current, and the magnitude of the q-axis current can be adjusted between 0%-100% of the maximum rated current. The unit of the position estimation error on the vertical axis is degree. In this experiment, according toFIG.1, the d-axis current is individually within 0%-100% of different six sectional curves. When the d-axis current remains a constant value, no matter the q-axis current is a forward or reverse current, the value of q-axis current becomes larger, and the position estimation error generated by the rotary electric machine control device becomes larger. For example, under the case where the d-axis current is maintained at 100% of the maximum rated current, the position estimation error increases as the q-axis current value increases. The position estimation error is a DC offset current caused by the cross-coupling effect between d axis and q axis. The DC offset current can directly or indirectly influences the accuracy of the rotary electric machine's rotor positions estimated by the rotary electric machine control device. For solving the problems mentioned above, the present invention proposes a control device of the electric machine for promoting the accuracy of estimating the rotary electric machine's rotor positions. The control device provided by the present invention can be operated in a more simplified manner to promote the accuracy of estimating the rotary electric machine's rotor positions. The operation principle and procedure of the present invention will be described in detail below. FIG.2is a block diagram of the control device100of the rotary electric machine200, in accordance with an embodiment of the present invention. In the present invention, the control device100, used for controlling the rotary electric machine200, includes a current command unit110, a controller130, a current conversion device134, a voltage conversion device132, a signal demodulation device140, an error compensation unit190, an adding device145, a position estimation device170, an encoder device150, a subtraction device160, an error controller180and a plurality of switches (182,184,195). When the rotary electric machine200operates normally, the switches182,184are turned off, and the switch195is turned on. As a result, when the rotary electric machine200operates normally, the signals output from the error controller180cannot be provided to the adding device145and error compensation unit190. In the present invention, the voltage conversion device132is electrically coupled to the current command unit110and the rotary electric machine200. At first, the current command unit110in the control device100is configured to provide the d-axis current command Id and the q-axis current command Iq. The controller130receives the d-axis current command Id, the q-axis current command Iq and a high-frequency signal generated and inputted by a high-frequency signal generator120, and then the controller130correspondingly outputs the d-axis voltage Vd and the q-axis voltage Vq on the synchronous coordinate. Then, by using the voltage conversion device132, the d-axis voltage Vd and the q-axis voltage Vq are converted to three-phase voltages (Va, Vb, Vc) on the stationary coordinate for rotating the rotary electric machine200. In this embodiment, the voltage conversion device132in the present invention includes synchronous/stationary axis converter, stationary/three-phase axis converter and inverter . . . etc. However, the present invention is not limited to this. In some embodiments of the present invention, the rotary electric machine200is a three-phase permanent magnet synchronous motor (three-phase PMSM), and the control of this kind of motors is usually based on the synchronous coordinate. As a result, if the voltage conversion device132mentioned above makes use of a synchronous/stationary axis converter, the d-axis voltage Vd and the q-axis voltage Vq output from the controller130could be converted to the d-axis voltage and the q-axis voltage on the stationary coordinate. By using a stationary/three-phase axis converter, the d-axis voltage and the q-axis voltage on the stationary coordinate could be further converted to three-phase voltages Va, Vb, Vc. In this embodiment, the voltage conversion device132can also include an inverter for adjusting the amplitude and frequency of the three-phase voltages Va, Vb, Vc to the rotary electric machine200. One having ordinary skill in the art will comprehend the operation principles of the synchronous/stationary axis converter and the stationary/three-phase axis converter and inverter, and the present disclosure does not repeatedly recite the description herein and does not show them inFIG.2. In the present invention, the current conversion device134retrieves the rotary electric machine currents Ia, Ib, Ic passing through the rotary electric machine200, and then converts the rotary electric machine currents Ia, Ib, Ic to a synchronous reference coordinate current. The rotary electric machine currents Ia, Ib, Ic can be individually defined as a magnetic flux current idand a torque current iqfor the components of d-axis and q-axis current on the synchronous coordinate. As a result, through a coordinate transformation, the rotary electric machine currents Ia, Ib, Ic can be converted to the synchronous reference coordinate current. The synchronous reference coordinate current further can be resolved into the magnetic flux current idand the torque current iq. The three-phase rotary electric machine currents Ia, Ib, Ic are the stator current of the rotary electric machine200, and the magnetic flux current idand the torque current iqare the currents on the synchronous coordinate. Because of the different references of coordinate systems, the current conversion device134includes a three-phase/stationary axis converter and a stationary/synchronous axis converter. The current conversion device134can convert the rotary electric machine currents Ia, Ib, Ic on the stationary coordinate to the magnetic flux current idand the torque current iqon the synchronous coordinate. After the magnetic flux current idand the torque current iqare processed by a high-pass filter (not shown) in the signal demodulation device140, the high-frequency magnetic flux current idhand the high-frequency torque current iqhcan be generated. One having ordinary skill in the art will comprehend that the operation principles of the three-phase/stationary axis converter and the stationary/synchronous axis converter, the present disclosure does not depict the same inFIG.2. It is particularly noted that in the present invention, the high-frequency magnetic flux current idhis a partial current signal of the d-axis current flowing through a rotor of the rotary electric machine200, and the high-frequency torque current iqhis a partial current signal of the q-axis current flowing through the rotor of the rotary electric machine200. In the vector control concepts of rotary electric machines, controlling the d-axis current or voltage can adjust the stator flux of the rotary electric machine200, and controlling the q-axis current or voltage can adjust the output torque of the rotary electric machine200. In this embodiment, the signal demodulation device140receives the magnetic flux current idand the torque current iqoutput from the current conversion device134, and then generates the high-frequency magnetic flux current idhand the high-frequency torque current iqhby using a motor mathematical model (or a high-frequency current equation) and the high-pass filter. The high-frequency current equation is shown below: iqh=vhp[(Lq+Ld2)2-(Lq-Ld2)2-Ldq2](Lq-Ld2sin(2Δθ)+Ldqcos(2Δθ))idh=vhp[(Lq+Ld2)2-(Lq-Ld2)2-Ldq2](Lq+Ld2+Lq-Ld2cos(2Δθ)+Ldqsin(2Δθ)), wherein the iqhis the high-frequency torque current, idhis the high-frequency magnetic flux current, p is a differential operator, Ldqis a cross-coupling inductance, Ldis a d-axis inductance, Lqis a q-axis inductance, and wherein Ldand Lqare measured values. Additionally, the cross-coupling inductance Ldqis the main factor causing the reduction of the accuracy of rotary electric machine's rotor position estimated by the control device100of rotary electric machine200. The value of cross-coupling inductance Ldqis proportional to the high-frequency torque current value iqh. The high-frequency torque current value iqhbecomes larger, and the cross-coupling inductance Ldqis increased. The operation of the control device100to reduce the effect of the cross-coupling inductance Ldqis described in detail below. In this embodiment, the signal demodulation device140of the control device100is electrically coupled to the current conversion device134, and includes a high-pass filter (not shown). The signal demodulation device140receives the magnetic flux current idand the torque current iqoutput from the current conversion device134. The signal demodulation device140further computes a current variation Δiqhof the high-frequency torque current iqhand a current variation Δidhof the high-frequency magnetic flux current idhaccording to a demodulation equation shown below: Δiqh=vhΔT[(Lq+Ld2)2-(Lq-Ld2)2-Ldq2](Lq-Ld2sin(2Δθ)+Ldqcos(2Δθ))Δidh=vhΔT[(Lq+Ld2)2-(Lq-Ld2)2-Ldq2](Lq+Ld2+Lq-Ld2cos(2Δθ)+Ldqsin(2Δθ)) Wherein vhis a high-frequency signal, and value of vhcan be positive or negative. In this embodiment, the high-frequency signal can be a square wave signal, but the invention is not limited thereto. It should be noted that the cross-coupling inductance Ldqcan be discovered in the current variation Δiqhof the high-frequency torque current iqhand the current variation Δidhof the high-frequency magnetic flux current idhso that the cross-coupling inductance Ldqcan be considered as a DC offset current of the position estimation error. The main technique of the present invention is to eliminate the effects that the cross-coupling inductance Ldqinfluences the current variation Δiqhof the high-frequency torque current iqh, the current variation Δidhof the high-frequency magnetic flux current idhand the control device100of the rotary electric machine200. In addition, the current variation Δi output by the signal demodulation device140is the high-frequency current variation on the synchronous coordinate mentioned above. The current variation Δiqhof the high-frequency torque current iqh, the current variation Δidhof the high-frequency magnetic flux current idh, or a combination of both can represent as the high-frequency current variation Δi on the synchronous coordinate. In order to briefly illustrate the following embodiments, the current variation Δi is only used to represent the current variation Δiqhof the high-frequency torque current iqh, the current variation Δidhof the high-frequency magnetic flux current idh, or a combination of both, but the present invention is not limited to this. In this embodiment, the error compensation unit190is electrically coupled to the current command unit110, and receives the d-axis current command Id and the q-axis current command Iq from the current command unit110. The error compensation unit190outputs a first correction value C1 based on the d-axis current command Id and the q-axis current command Iq. In this embodiment, there is at least one table to be configured and built in the error compensation unit190, and the error compensation unit190searches, look-up, or indexes the table to acquire a first correction value C1 that corresponds to the d-axis current command Id and the q-axis current command Iq at present. Then, the error compensation unit190outputs the searched first correction value C1 corresponding to the d-axis current command Id and the q-axis current command Iq at present to the adding device145. The Table 1 shown below is an example that configured and built in the error compensation unit190, and it only shows part of the first correction values C1. The d-axis current command Id and the q-axis current command Iq corresponding to each of the related first correction value C1 are not fully identical. For example, as shown in Table 1, C1(Id1,Iq1) means that the first correction value C1 corresponds the d-axis current command Id1 and the q-axis current command Iq1; while C1(Id2,Iq2) means that the first correction value C1 corresponds the d-axis current command Id2 and the q-axis current command Iq2; . . . and so on. However, the present invention is not limited to this. TABLE 1q-axis current commandd-axis current commandIq1Iq2Iq3Id1C1 (Id1, Iq1)C1 (Id1, Iq2)C1 (Id1, Iq3)Id2C1 (Id2, Iq1)C1 (Id2, Iq2)C1 (Id2, Iq3)Id3C1 (Id3, Iq1)C1 (Id3, Iq2)C1 (Id3, Iq3)Id4C1 (Id4, Iq1)C1 (Id4, Iq2)C1 (Id4, Iq3) In this embodiment, the adding device145is electrically coupled to the error compensation unit190, the signal demodulation device140and the position estimation device170. The adding device145is used for adding the current variation Δi and the first correction value C1 to generate a second correction value C2. The adding device145then outputs the generated second correction value C2 to the position estimation device170. Eventually, according to the second correction value C2, the position estimation device170adjusts a phase estimation value θato the current conversion device134and the voltage conversion device132for adjusting the magnetic flux current idand the torque current iqrespectively. The phase estimation value θais the estimated position of the rotor. According to the phase estimation value θa, the current conversion device134performs a coordinate transformation calculation to acquire the magnetic flux current idand the torque current iq; afterwards, the adjusted magnetic flux current idand torque current iqcan be delivered to the controller130for modifying the values of d-axis voltage Vd and q-axis voltage Vq. The voltage conversion device132then receives the signals mentioned above to indirectly adjust the three-phase voltages Va, Vb, Vc. As a result, the rotor position of rotary electric machine200can be adjusted, oscillation when the rotary electric machine200rotates at low speed can be reduced effectively, and the efficiency of operation of the rotary electric machine200can be increased. In summary, the error compensation unit190outputs the first correction value C1 corresponding to the d-axis current command and the q-axis current command by searching the table. Then, the adding device145adds the searched first correction value C1 and the current variation Δi to acquire the second correction value C2. As a result, the position estimation device170can perform the calculation based on the acquired second correction value C2 for effectively controlling and increasing the accuracy of phase estimation value θa. The procedure of building the table in the error compensation unit190is described in detail as below. FIG.3depicts a block diagram of the control device100of the rotary electric machine200operates in a test mode, in accordance with an embodiment of the present invention. In this embodiment, before building a table in the error compensation unit190, the operators usually need to process a test mode to the rotary electric machine200. As a result, the procedures of building the table described below are finished when the rotary electric machine200is continuously processed in the test mode. In this embodiment, the switches182and184in the control device100are turned on, and the switch195is turned off. The encoder device150is used to measure the rotor position of the rotary electric machine200for outputting an actually measured phase measurement value θr. The phase measurement value θrthat is a really measured value which can be considered as the real position of the rotor. At the same time, the position estimation device170continuously outputs the phase estimation value θato the current conversion device134and the subtraction device160. In some embodiments, the subtraction device160is electrically coupled to the encoder device150, the position estimation device170and the error controller180. When the rotary electric machine200operates in the test mode, through the subtraction device160, the phase estimation value θais subtracted from the actual measured phase measurement value θrto generate a phase error Δθ for the error controller180. As described above, the phase error Δθ output by the subtraction device160is the difference between the actual rotor position and the estimated rotor position. When the rotary electric machine200operates in the test mode, because of the switches182and184are turned on, the error controller180can continuously generate a revised value R1 to the adding device145based on the phase error Δθ. It should be noted that when the rotary electric machine200operates in the test mode, the switch195between the error compensation unit190and the adding device145is turned off so that the adding device145cannot receive the first correction value C1 output by the error compensation unit190. When the phase error Δθ falls within a target range, the error controller180assigns the revised value R1 at present as the first correction value C1. Then, the error compensation unit190stores the revised value R1 which assigned as the first correction value C1, the d-axis current command Id and the q-axis current command corresponding to the first correction value C1. In some other embodiments, the error controller180can detect the phase error Δθ. In general conditions, if the error controller180detects that the phase error Δθ is not substantially equal to zero, the error controller180then outputs the revised value R1 to the adding device145. The adding device145then adds the revised value R1 and the current variation Δi at present to generate the second correction value C2 for the position estimation device170. Then, the position estimation device170then adjusts the phase estimation value θafor the subtraction device160so that the phase error Δθ output by the subtraction device160is adjusted. When the error controller180detects that the phase error Δθ is within the target range (Ex: the target range is 2%-4%), the error controller180takes the revised value R1 as the redefined and assigned new first correction value C1 at present, and the error compensation unit190then stores the revised value R1. At this time, the error compensation unit190records the revised value R1 assigned as the first correction value C1 and records the corresponding d-axis current command Id and q-axis current command Iq at present simultaneously. In addition; on contrary, when the error controller180detects that the phase error Δθ is out of the target range, the control device100performs the previous procedures for searching another phase error Δθ, and the error controller180identifies whether it is in the target range again. When the current command unit110continuously provides different d-axis current command Id and q-axis current command Iq each time, the current variation Δi, for example, output by the signal demodulation device140can be directly or indirectly modified so that the phase error Δθ output by the subtraction device160is correspondingly modified. The error controller180will continuously adjust the revised value R1. When the error controller180detects that the phase error Δθ falls within the target range (Ex: the target range is 2%-4%), the error controller180stops adjusting the revised value R1 and then stores the revised value R1 at present in a table of the error compensation unit190. The error compensation unit190stores each revised value R1 corresponding to the d-axis current command Id and q-axis current command Iq, and assigns the revised value R1 to be the redefined and assigned new first correction value C1 that corresponds to the present current commands continuously so as to build the table. As a result, the table includes a plurality of the tested first correction values C1, and each of the d-axis currents command Id and q-axis currents command Iq corresponding to the first correction values C1 is not fully identical. FIG.4depicts a flow chart of control method400for controlling rotary electric machine200, in accordance with an embodiment of the present invention. Please refer toFIG.2andFIG.4for illustrating the procedures of the control method400of the rotary electric machine200. The control method400starts in step410. In step410, when the current command unit110starts to output the d-axis current command Id and q-axis current command Iq, the error compensation unit190starts to receive the d-axis current command Id and the q-axis current command Iq, and then step420is executed. At the same time, according to the d-axis current command Id and the q-axis current command Iq output by the current command unit110, the controller130and the voltage conversion device132output the three-phase voltages Va, Vb, Vc to drive the rotary electric machine200. In step420, according to the d-axis current command Id and the q-axis current command Iq output by the current command unit110, the error compensation unit190outputs a first correction value C1 corresponding to the d-axis current command Id and the q-axis current command Iq at present. The error compensation unit190includes a table, as shown in Table 1 above. The error compensation unit190will finally find out a suitable first correction value C1 corresponding to the d-axis current command Id and the q-axis current command Iq at present in the Table 1, and then step430is executed. In step430, when the rotary electric machine200starts to rotate, the current conversion device134retrieves the rotary electric machine currents Ia, Ib, Ic passing through the rotary electric machine200, and then computes the magnetic flux current idand the torque current iq, and then step440is executed. In step440, the signal demodulation device140receives the magnetic flux current idand the torque current iq, and then computes the high-frequency torque current iqhand the high-frequency magnetic flux current idhbased on a motor mathematical model or a high-frequency current equation. The motor mathematical model or the high-frequency current equation are identical with the description above, so the same descriptions will not be elaborated upon. The signal demodulation device140in the control device100also computes the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idh, according to the high-frequency torque current iqhand/or the high-frequency magnetic flux current idh. The demodulation equation for computing the current variation Δiqhof the high-frequency torque current iqhand the current variation Δidhof the high-frequency magnetic flux current idhis mentioned above, so the same will not be elaborated upon. After steps420and440, the control device100starts to execute step450. The adding device145in the control device100adds the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idhwith the assigned first correction value C1 to generate the second correction value C2, and then the step460is executed. In step460, the position estimation device170in the control device100adjusts the phase estimation value θabased on the generated second correction value C2, and then delivers the adjusted phase estimation value θato the current conversion device134, and then step470is executed. In step470, according to the adjusted phase estimation value θa, the current conversion device134performs a coordinate transformation calculation to acquire the magnetic flux current idand the torque current iq, and then the current conversion device134delivers the magnetic flux current idand the torque current iqto the controller130. The controller130then can adjust the values of the d-axis voltage Vd and the q-axis voltage Vq based on the magnetic flux current idand the torque current iq, and then the controller130delivers the signals mentioned previously to the voltage conversion device132. Then, the voltage conversion device132performs the coordinate transformation calculation to indirectly adjust the three-phase voltages Va, Vb, Vc. As a result, the control device100can adjust the rotor position of the rotary electric machine200, and the control device100can effectively reduce the oscillation caused by the rotary electric machine200rotating at low speed. According to the control method400inFIG.4, the control device100in the present invention, by means of the error compensation unit190, outputs the first correction value C1 corresponding to the d-axis current command and the q-axis current command at present by means of searching a table. Then, the adding device145adds the assigned new first correction value C1 and the current variation Δi (the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idh) to acquire the second correction value C2. As a result, the position estimation device170can perform the calculation based on the acquired second correction value C2 for effectively controlling and increasing the accuracy of the phase estimation value θa. Therefore, building the table in the error compensation unit190is one of the main features of the present invention. The procedure of building the table in the error compensation unit190is described in detail below. FIG.5depicts a flow chart of control method500for controlling rotary electric machine200operates in a test mode, in accordance with an embodiment of the present invention. Please refer toFIG.3andFIG.5for illustrating the procedures of the control method500of the rotary electric machine200. The procedures of the control method500inFIG.5are mainly used for building the table stored or built in the error compensation unit190. In this embodiment, before building a table in the error compensation unit190, the operators usually need to make the rotary electric machine200operating in a test mode. As a result, the procedures of building a table described below are finished by the control device100when the rotary electric machine200is operated in a test mode continuously. The control method500starts at step510, and the rotary electric machine200remains a stationary status at this time. Also, by means of appropriate methods, the rotor is stopped from rotation. When the current command unit110starts to output the d-axis current command Id and the q-axis current command Iq, the controller130and the voltage conversion device132output the three-phase voltages Va, Vb, Vc based on the d-axis current command Id and the q-axis current command Iq output by the current command unit110for driving rotor of the rotary electric machine200. In step510, the current conversion device134retrieves the rotary electric machine currents Ia, Ib, Ic passing through the rotary electric machine200, and performs the coordinate transformation calculation to acquire the magnetic flux current idand the torque current iq, and then step520is executed. In step520, according to the magnetic flux current idand the torque current iqoutput from the current conversion device134, the signal demodulation device140in the control device100computes the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idhto the adding device145. The equations for computing the current variation Δiqhof the high-frequency torque current iqhand the current variation Δidhof the high-frequency magnetic flux current idhare identical with the description provided above, so they are not repeatedly recited herein. At the same time, although the adding device145only receives the current variation Δi (the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idh), the adding device145still executes step540: to add a revised value R1 of the error controller180and the current variation Δi (the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idh) to output a second correction value C2, and then step550is executed. It should be noted that when the control device100executes step540for the first time, the revised value R1 output by the error controller180is zero. As a result, at this time, the second correction value C2 output by the adding device145for the first time is in accordance with the current variation Δi (the current variation Δiqhof the high-frequency torque current iqhand the current variation Δidhof the high-frequency magnetic flux current idh). In step550, according to the second correction value C2, the position estimation device170outputs the phase estimation value θato the subtraction device160, and then step560is executed. When the rotary electric machine200is controlled by the three-phase voltages Va, Vb, Vc, the control device100also executes step530: the encoder device150in the control device100measures the rotor positions of the rotary electric machine200and outputs the actual measured phase measurement value θrto the subtraction device160, and then step560is executed. In step560, the subtraction device160receives the phase estimation value θaand the actual measured phase measurement value θrfrom step550and step530respectively to compute the phase error Δθ between the phase estimation value θaand the actual measured phase measurement value θr. After computing the phase error Δθ, the control device100starts to perform step570. In step570, the error controller180detects whether the phase error Δθ falls within a target range. As a result, when the error controller180detects that the phase error Δθ falls within the target range, the control device100executes step590and the error controller180controls the error compensation unit190to store the revised value R1 at present as the redefined and assigned new first correction value C1, and simultaneously records the d-axis current command Id and the q-axis current command Iq corresponding to the first correction value C1 at present. However, when the error controller180detects that the phase error Δθ is not within the target range, the control device100executes step580and outputs the revised value R1 based on the phase error Δθ at present. It should be noted that, due to the previous step540, the revised value R1 received by the adding device145in the initial step is zero, so when the control device100executes step570for the first time, the phase error Δθ received by the error controller180may possibly be outside the target range. As a result, the control device100will perform step570to step580. After the error controller180outputs the revised value R1, the control device100performs step540. When the control device100performs step540again, the adding device145adds the revised value R1 of the error controller180and current variation Δi (the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idh), and then adjusts the second correction value C2 to the position estimation device170. The control device100continuously performs step550. In step550, according to the adjusted second correction value C2, the position estimation device170adjusts the phase estimation value θato the subtraction device160. In step560, the subtraction device160computes the phase error Δθ between the phase estimation value θaand the actual measured phase measurement value θr, then Step570is executed. In step570, the error controller180detects whether the phase error Δθ falls within the target range again. If the error controller180detects that the phase error Δθ falls within the target range, the control device100performs step590, and then the error controller180controls the error compensation unit190to store the revised value R1 as the redefined and assigned new first correction value C1, and simultaneously records the d-axis current command Id and the q-axis current command Iq corresponding to the first correction value C1 at present. As mentioned above, if the error controller180detects that the phase error Δθ is not within the target range, the control device100performs step580, and the error controller180continuously adjusts the revised value R1 to the adding device145based on checking whether the phase error Δθ and the target range are matched. The control device100will uninterruptedly perform from the Steps540to580until the error controller180detects that the phase error Δθ falls within the target range in step570. When the current command unit110modifies the d-axis current command Id and the q-axis current command Iq each time, the control device100performs the procedures of the control method500mentioned above. As a result, when the current command unit110continuously modifies the d-axis current command Id and the q-axis current command Iq a plurality of times or loops, the error compensation unit190can build the table. The operators can determine how many times or loops the control method500is performed, depending on demand. If the control device100performs the control method500more times or loops, there would be more sets of current commands recorded in the table of the error compensation unit190. When the rotary electric machine200works normally, the rotor position of the rotary electric machine200estimated by the control device100becomes more accurate accordingly. As a result, it is much more effective to solve the phenomenon of the oscillation caused by the rotary electric machine200rotating at low speed, and increase the efficiency of the operation of the rotary electric machine200. FIG.6is a practical operation block diagram of the control device100of the rotary electric machine200, in accordance with an embodiment of the present invention. The speed control unit115is an embodiment of the current command unit110mentioned previously, but the present invention is not limited to. The speed control unit115can provide the d-axis current command Id, and a speed controller115ain the speed control unit115can provide the q-axis current command Iq. Then, the controller130and the voltage conversion device132generate the three-phase voltages Va, Vb, Vc to the rotary electric machine200. The voltage conversion device132includes a synchronous/stationary axis converter132a, a stationary/three-phase axis converter132band an inverter132c. The rotary electric machine200rotates based on the three-phase voltages Va, Vb, Vc. The synchronous/stationary axis converter132a, the stationary/three-phase axis converter132band the inverter132care prior arts, so the present disclosure does not recite them again. In one of the other embodiments, the speed control unit115further includes a speed controller115a. The processors, micro-processors or other computing devices can provide a rotating speed command ω1to the speed control unit115. One having ordinary skill in the art will comprehend the techniques that the processors, micro-processors and other computing devices can provide with regard to the rotating speed command ω1, and it is not drawn inFIG.6. The position estimation device170also can generate the estimated speed feedback ω2to the speed control unit115. The speed control unit115still can generate the rotating speed error ω3based on computing the rotating speed command ω1and the estimated speed feedback ω2. The rotating speed error ω3is a rotating speed error between the rotating speed commands ω1and estimated speed feedback ω2mentioned previously, so that the speed controller115agenerates the q-axis current command Iq based on the rotating speed error ω3. When the rotary electric machine200rotates based on the three-phase voltages Va, Vb, Vc, the current conversion device134receives the rotary electric machine currents Ia, Ib, Ic passing through the rotary electric machine200, and then computes the torque current iqand the magnetic flux current id. The current conversion device134includes a three-phase/stationary axis converter134a, a stationary/synchronous axis converter134band a low-pass filter134c. The three-phase/stationary axis converter134aand the stationary/synchronous axis converter134bin the current conversion device134can compute the torque current iq1and the magnetic flux current id1on the synchronous coordinate. It should be noted that in this embodiment, the present invention simply selectively makes use of the torque current iq1on the synchronous coordinate to perform the calculation for increasing the accuracy of position estimation, but the present invention is not limited to. The high-frequency current equations defined by the three-phase/stationary axis converter134aand the stationary/synchronous axis converter134bare as mentioned above, so is the same are not repeatedly recited herein. After the current conversion device134finished the computation of the torque current iq1on the synchronous coordinate, the signal demodulation device140computes the high-frequency torque current iqhand the current variation Δiqhof the high-frequency torque current iqh. The signal demodulation device140includes a high-pass filter140aand a high-frequency signal demodulation device140b. The high-pass filter140acomputes the high-frequency torque current iqhbased on the torque current iq1. The high-frequency signal demodulation device140bhas a demodulation equation therein for computing the current variation Δiqh. The demodulation equation is described above, so it will not be recited here again. When the rotary electric machine200is operated in the test mode, the method of building a table using the control device100has been described in detail previously, so the description will not be repeated. In conclusion, the present invention makes use of the error compensation unit190to receive the d-axis current command Id and the q-axis current command Iq for outputting the first correction value C1. Then, the present invention computes the current variation Δiqhof the high-frequency torque current iqhand/or the current variation Δidhof the high-frequency magnetic flux current idhfor effectively controlling and increasing the accuracy of the phase estimation value θaestimated by the position estimation device170. Compared to the current techniques, the present invention greatly reduces the amount of (signal) calculation in the processor, and more effectively improves the operation efficiency of the control device of the rotary electric machine200. In addition, the present invention can also solve the oscillating phenomenon generated when the rotary electric machine200is operated at a low speed, and improve the operating efficiency of the rotary electric machine200. While the invention has been described above in terms of a preferred embodiment, it is not intended to limit the scope of the invention, and it should be understood by those of ordinary skill in the art without departing from the spirit and scope of the invention. Instead, the scope of the invention should be determined by the scope of the appended claims. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to limit the invention. As used herein, the singular forms “a,” “an,” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. Furthermore, to the extent that the terms “including,” “includes,” “having,” “has,” “with,” or variants thereof are used in either the detailed description and/or the claims, such terms are intended to be inclusive in a manner similar to the term “comprising.” | 41,785 |
11863108 | FIG.1shows a polyphase rotary electric machine1, in particular for a motor vehicle, to which the invention can be applied. This rotary electric machine can form an alternator or a starter-alternator of the vehicle. This rotary electric machine can be powered, via a power electronics component9comprising an inverter/rectifier, by a battery, the nominal voltage of which is 12 V or 48 V or has a value above 300 V, for example. The rotary electric machine1comprises a casing2. Inside this casing2, it further comprises a shaft3, a rotor4which rotates as one with the shaft3and a stator5surrounding the rotor4. The rotational movement of the rotor4takes place about an axis X. In this example, the casing2comprises a front bearing6and a rear bearing7which are assembled together. These bearings6,7are hollow in form and each bear, centrally, a respective ball bearing10,11for the rotational mounting of the shaft3. A pulley12is, in the example under consideration, fixed to a front end of the shaft3, at the front bearing6, for example using a nut bearing on the bottom of the cavity of this pulley. This pulley12makes it possible to transmit the rotational movement to the shaft3and it can be connected, via a belt, to the crankshaft of the combustion engine of the vehicle. The rear end of the shaft3bears, in this instance, slip rings belonging to a commutator and connected by wire connections to the winding. Brushes belonging to a brush holder8are arranged so as to rub against the slip rings. The front bearing6and the rear bearing7can further comprise substantially lateral openings for the passage of air in order to make it possible for the rotary electric machine to be cooled by the circulation of air generated by the rotation of a front fan13on the front dorsal face of the rotor4, that is to say at the front bearing6, and of a rear fan14on the rear dorsal face of the rotor, that is to say at the rear bearing7. In this exemplary embodiment, the stator5comprises a body15in the form of a stack of laminations which is provided with notches, for example of the semi-closed or open type, equipped with notch insulator for the mounting of the polyphase electrical winding of the stator. Each phase comprises a winding16passing through the notches of the body15and forming, with all the phases, a front bundle and a rear bundle on either side of the body of the stator. The windings16are, for example, obtained from a continuous wire covered with enamel or from conductive elements in the form of a bar such as pins connected to one another. The electric winding of the stator is, for example, three-phase, then implementing a star or delta configuration, the outputs of which are connected to the power electronics component9. The rotor4ofFIG.1is a claw-pole rotor. It comprises two pole wheels17. The first pole wheel17faces the power electronics component9, while the second pole wheel17faces the pulley12. Each of the pole wheels17comprises a bottom18extending radially on either side of the axis X, the wheel defining a series of claws19of trapezoidal overall shape. Each claw of a pole wheel17extends axially in the direction of the other pole wheel from a base arranged on the radially outer periphery of the bottom18. The rotor4further comprises, between the radially inner portions20and the claws19, a coil wound on a coil insulator22. The rotor4can also comprise permanent magnets (which are not shown) interposed between two adjacent claws19at the outer periphery of the rotor. As a variant, the rotor4can be devoid of such permanent magnets. There can be any number of pairs of poles which is defined by the rotor4, for example six or eight. The machine also comprises sensors for measuring the position of the rotor4, for example three Hall effect sensors, grouped together in the same housing made from plastic. These sensors are, for example, positioned at the rear bearing7of the machine and they interact with a magnetic target30which rotates as one with the rotor. In the example which is going to be described, this magnetic target30defines eight pairs of poles. The measurements delivered by these sensors are used by the circuit100for determining the angular position of the rotor4, which is now going to be described with reference toFIGS.2to4. In a known manner, the circuit100comprises a block101performing discretization of the signals s1 to s3 acquired by each position sensor. At the output of this block101, the various signals originating from the sensors and which have been discretized attack a dynamic normalization circuit102which also receives at input an image of the signal representative of the position of the rotor. This image is in this instance a linear combination of the cosine and the sine of the angle θmeasured with respect to a reference position of this rotor, the time derivative of this angle corresponding to the rotational speed of this rotor4. In the example under consideration, each signal originating from a sensor then discretized and which is received at the input of the dynamic normalization circuit102is first of all corrected in105by subtracting the zero error. The signal thus corrected then undergoes, in106, synchronous and coherent demodulation by the linear combination of the cosine and the sine of the angle of the rotor. The signal xiresulting from this demodulation then enters a low-pass filter107which has, in the example under consideration, a variable cutoff frequency. In a first operating range, for example for a speed of the rotor of between 0 rpm and 2000 rpm, the cutoff frequency of this filter107is between 2 Hz and 50 Hz in the example under consideration. In a second operating range, for example for a speed of the rotor above 2000 rpm, the cutoff frequency of the filter is of the order of 0.5 Hz in the example under consideration. The low-pass filter107has another input corresponding to a predefined amplitude value x0ifor the first harmonic of the signal received at the input of the dynamic normalization circuit102. This value x0ican be used as an input datum by the low-pass filter107at the start of the first operating range. At the output of the low-pass filter107, the value of the first harmonic of the signal received at the input of the dynamic normalization circuit102can be extracted, from the demodulated signal, for each of the aforementioned operating modes. Phase-shift compensation and saturation are respectively applied to this amplitude of the first harmonic of the signal received at the input of the dynamic normalization circuit102by respective blocks110and111, and the resulting signal is received at the input of a selector115. This selector also receives a control signal via an input116, as well as, via another input117, the predefined value of the amplitude of the first harmonic also received at the input of the filter107. According to the circumstances, there can, via the control signal at the input116, be imposed at the output of the selector115:the first amplitude value for the signal received at the input of the circuit102resulting from the demodulation operation, orthe predefined first amplitude value x0ifor this first harmonic of said signal. This value at the output of the selector115is then used to normalize the signal received at the input of the circuit102, a block120dividing this signal received at the input of the circuit102by the value of the amplitude of the first harmonic for this signal present at the output of the selector115. What has just been described above is applied in parallel to each signal originating from a sensor for sensing the position of the rotor of the electric machine. The signals thus obtained at the output of the circuit102are received at the input of a block122performing a mathematical transformation for modeling the system, which is three-phase in the example, as a two-phase system. This transformation uses, for example, a Clarke or Concordia matrix. Other transformations can be used when the number of position sensors is different from 3. The signals at the output of this block122are received at the input of a circuit130producing a control loop for controlling the position of the rotor4, delivering at output a signal representative of the position of the rotor4, which is the aforementioned angle θ. This circuit130modulates, via a block131, the sine signal at the output of the block122by the cosine of the angle θ, and this circuit130also modulates, via a block132, the cosine signal at the output of the block122by the sine of the angle θ. The difference between the signals at the output of the blocks131and132is received at the input of a corrector135. The output of the corrector135delivers the rotational speed of the rotor4, which delivers another output of the circuit130, and this rotational speed enters an integrator136of the circuit130, in order to obtain the angle θ. The speed and angle values thus obtained can be used to control the rotary electric machine1. FIG.5shows, on the same graph, a plurality of responses originating from a position sensor during a speed hike210for the rotor4moving from 0 to 10,000 rpm. The curve200represents the signal delivered by the sensor according to the prior art in reaction to this speed hike, upstream of the determining device100. The curve201represents the actual value of this amplitude of the first harmonic in reaction to this speed hike. The curve202represents the amplitude of the first harmonic of this signal, as determined using the circuit102described above, in reaction to this speed hike. It can be observed that, while the curve200is neither quick nor stable nor precise, the curve202follows the curve201from the low speeds despite interference such as the offset and the other harmonics. The invention is not limited to the example which has just been described. | 9,862 |
11863109 | DETAILED DESCRIPTION In order to make the purpose, technical solutions, and advantages of the disclosure clearer, the technical solutions in the disclosure will be clearly and completely described below in combination with the drawings in the disclosure. It is apparent that the described embodiments are not all but part of embodiments of the disclosure. All other embodiments obtained by those of ordinary skill in the art based on the embodiments in the disclosure without paying any creative work shall fall within the scope of protection of the disclosure. It is to be noted that similar reference numerals and letters represent similar terms in the drawings and thus a term, once being defined in a drawing, is not required to be further defined and explained in subsequent drawings. In addition, terms “first”, “second” and the like in the descriptions of the disclosure are only adopted for distinguishing descriptions and may not be understood to indicate or imply relative importance. Rail transit has become an important means of transportation in the modern society. With the diversified development of a technology, more and more types of rail transits have been present, e.g., bullet train, locomotive, metro, light rail, monorail, etc. An existing rail transit vehicle is usually a train set consisting of multiple carriages. The train set is usually formed in a trailer-tractor combination manner and in fixed marshalling, and thus has a more stable power. The trailer-tractor combination manner refers to a manner of marshalling tractor carriages and trailer carriages according to a certain sequence. A motor is mounted in the tractor carriage to provide power for the train set. The motor may be a permanent magnet motor, a permanent magnet assisted reluctance motor, etc. The motor refers to an electromagnetic device converting or transmitting electric energy according to the law of electromagnetic induction, and includes components such as a stator, a rotor, a housing, etc. At present, the motor in the rail transit vehicle mostly adopts an alternating current (AC) motor. An AC asynchronous motor is applied extensively due to its advantages of simple structure, low production cost, simplicity for maintenance, quick dynamic response, etc. A permanent magnet motor has gradually been applied to the rail transit vehicle due to its advantages of high energy efficiency, as well as high performance, high-precision transmission, and quick dynamic response. When the permanent magnet motor operates, a three-phase current is input to a stator winding, and a rotating magnetic field is formed in the stator winding after the current is input. Since a permanent magnet with fixed magnetic poles is mounted on a rotor, the rotating magnetic field generated in a stator may drive the rotor to rotate according to the principle that like poles repel and unlike poles attract. When the rotor rotates, a magnetic field of a permanent magnet rotor performs a cutting motion on a coil wire to further generate a counter electromotive force (CEMF) in the coil wire. When an inter-turn short circuit occurs to a coil of the motor, the rail transit vehicle may keep running under the driving of other motors even though the motor is shut down. After the motor is shut down, a short-circuited coil in the stator may form a closed loop during the operation of the motor, and a permanent magnet on the rotor keeps operating with the vehicle to further generate a rotating magnetic field, i.e., a CEMF. The closed loop formed by the short-circuited coil generates an induced current in the magnetic field under the CEMF. Since a wire of the short-circuited coil has a resistance, electric energy in the short-circuited coil may be converted into heat energy under the CEMF. As the operational time of the vehicle extends, the heat energy on the short-circuited coil is continuously accumulated, and the temperature thereof keeps rising. In such case, when the vehicle does not timely stop or reduce its running speed, the conditions where the coil is fused, a fused matter enters an air gap between the stator and the rotor, and a bearing clearance, etc., are likely to occur to the motor, to further result in the collision of the stator and the rotor and a bearing failure, which may eventually cause secondary failures such as the separation of the motor, etc. In order to solve the foregoing problems, the disclosure provides a method and device for adjusting a permanent magnet motor, an equipment, and a storage medium. An electronic equipment may determine an operational time of any short-circuited coil of a permanent magnet motor to be adjusted in a production stage of the permanent magnet motor to be adjusted, according to an electromotive force (EMF) parameter and information of an electromagnetic structure of the permanent magnet motor to be adjusted and a minimum impedance value of the short-circuited coil. In the operational time, the permanent magnet motor to be adjusted operates at a maximum rotational speed, and the short-circuited coil may not be fused. When the operational time is shorter than a preset time, it is considered that the permanent magnet motor to be adjusted has a potential safety hazard, and an electromagnetic structure of the permanent magnet motor to be adjusted is required to be adjusted. In addition, the electronic equipment may also determine a target rotational speed of the permanent magnet motor to be adjusted when the rail transit vehicle runs, according to the information of the electromagnetic structure of the permanent magnet motor to be adjusted, the minimum impedance value of any short-circuited coil, and a target operational time. The running speed of the rail transit vehicle is further adjusted according to the target rotational speed to ensure that the rotational speed of the permanent magnet motor to be adjusted is lower than the target rotational speed when the rail transit vehicle runs. The effect of ensuring safe running of the rail transit vehicle is further achieved. An execution body of the disclosure is an electronic equipment, which may be a computer, a server, an on-board computer, a controller, or the like. No limitations are made thereto in the disclosure. FIG.1is a flowchart of a method for adjusting a permanent magnet motor according to an embodiment of the disclosure. As shown inFIG.1, the method of the embodiment may include the following operations. In operation S101, an electronic equipment acquires a CEMF parameter, information of an electromagnetic structure of a permanent magnet motor to be adjusted and a minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, here the CEMF parameter is a CEMF value of the permanent magnet motor at a maximum rotational speed. In the embodiment, the electronic equipment may acquire information such as the CEMF value of the permanent magnet motor to be adjusted at the maximum rotational speed, the minimum impedance value of any short-circuited coil, and the information of the electromagnetic structure, etc. in a manner of requesting from other equipments, reading from a memory, obtaining through an input, or the like. No limitations are made thereto in the disclosure. The electronic equipment further acquires an operational time of the short-circuited coil when an inter-turn short circuit occurs to the permanent magnet motor to be adjusted at the maximum rotational speed, according to the information. The CEMF parameter is a CEMF value generated by the coil by cutting a magnetic induction line of a magnetic pole during the operation of the permanent magnet motor to be adjusted. In an example, the electronic equipment may transmit an acquisition instruction to a test permanent magnet motor, and receive the CEMF parameter returned by the test permanent magnet motor according to the acquisition instruction. Or, the CEMF value of the permanent magnet motor at the maximum rotational speed is determined through a simulation technology. The test permanent magnet motor is a permanent magnet motor with the same information of an electromagnetic structure as the permanent magnet motor to be adjusted. The electronic equipment may acquire the CEMF parameter from the test permanent magnet motor as follows. The electronic equipment transmits an acquisition instruction to a voltage measurement device on the test permanent magnet motor, and acquires the CEMF value, measured by the voltage measurement device, of the test permanent magnet motor at the maximum rotational speed. The electronic equipment may also simulate, through the simulation technology and according to the information of the electromagnetic structure of the permanent magnet motor to be adjusted, the permanent magnet motor to be adjusted and an operational condition of the permanent magnet motor to be adjusted at the maximum rotational speed in an environment such as a simulation software, etc., to further determine the CEMF value of the permanent magnet motor to be adjusted at the maximum rotational speed. The permanent magnet motor to be adjusted may have at least one coil which may be arranged on the stator. The coil of the permanent magnet motor generates a magnetic field when a current flows through the coil, and the rotor rotates under the interaction of a magnetic field of a magnetic pole and the magnetic field of the coil to implement the electric energy conversion or transmission of the permanent magnet motor. There is an insulating material or insulating coating on a surface of a wire in the coil. During usage, the insulating material may be fractured, abraded, separated, etc., to further cause two turns of the wire in the coil contacting each other, resulting in a phenomenon of inter-turn short circuit of the coil. The minimum impedance value of any short-circuited coil is an impedance value of any turn of the wire in the coil. The coil in the permanent magnet motor may also be arranged on the rotor. In such case, a permanent magnet is arranged on the stator. In an example, the information of the electromagnetic structure of the permanent magnet motor to be adjusted includes one or more of: information of a cooling structure of the permanent magnet motor, a melting point of an insulating material, a permanent magnet parameter, a slot shape parameter of the motor, an electromagnetic wire parameter, etc. The cooling structure of the permanent magnet motor is configured to cool the permanent magnet motor. The information of the cooling structure of the permanent magnet motor may include cooling efficiency, a cooling manner, etc. The insulating material is used for insulation protection over the wire on the coil of the permanent magnet motor. A permanent magnet is used for the rotor of the permanent magnet motor. The permanent magnet parameter includes the magnetic performance, size, etc., of the permanent magnet. In operation S102, the electronic equipment determines, according to the CEMF parameter, the information of the electromagnetic structure and the minimum impedance value, an operational time of a smallest short-circuited coil. In the embodiment, after acquiring the CEMF parameter, the information of the electromagnetic structure and the minimum impedance value of the permanent magnet motor to be adjusted according to the operation S101, the electronic equipment imports the information to a parameter model, and obtains the operational time of the short-circuited coil through the parameter model. The parameter model may be a mapping relationship model, or an algorithm model. In operation S103, the electronic equipment transmits, to a production equipment, an adjustment instruction configured to instruct the production equipment to adjust the electromagnetic structure of the permanent magnet motor to be adjusted, when the operational time is inconsistent with a preset time, so that the production equipment adjusts the electromagnetic structure of the permanent magnet motor to be adjusted. In the embodiment, the electronic equipment acquires the operational time of the short-circuited coil of the permanent magnet motor to be adjusted from the operation S102. The electronic equipment compares the operational time and the preset time. When the operational time is more than or equal to the preset time, it is considered that the electromagnetic structure of the permanent magnet motor to be adjusted satisfies a condition and is not required to be adjusted. For example, the preset time is one hour, and the operational time of the short-circuited coil is three hours, then it is considered that the electromagnetic structure of the permanent magnet motor to be adjusted satisfies the condition and is not required to be adjusted. When the operational time is shorter than the preset time, it is considered that the electromagnetic structure of the permanent magnet motor to be adjusted does not satisfy the condition and is required to be adjusted. For example, the preset time is one hour, and the operational time of the short-circuited coil is half an hour, then it is considered that the electromagnetic structure of the permanent magnet motor to be adjusted does not satisfy the condition and is required to be adjusted. The electronic equipment transmits the adjustment instruction to the production equipment when determining that the electromagnetic structure of the permanent magnet motor to be adjusted is required to be adjusted. The production equipment adjusts the electromagnetic structure of the permanent magnet motor to be adjusted according to the adjustment instruction, after receiving the adjustment instruction transmitted by the electronic equipment. Preferably, the adjustment instruction is specifically configured to instruct the production equipment to perform one or more of: adjusting the permanent magnet parameter, replacing the insulating material, adjusting a cooling structure of the motor, adjusting the slot shape parameter, adjusting the electromagnetic wire parameter, etc., on the permanent magnet motor to be adjusted. The permanent magnet parameter and the electromagnetic structure determine an intensity of the magnetic field of the magnetic pole. When the same coil is used, the intensity of the magnetic field is higher at the same rotational speed, then the CEMF value generated by the coil by cutting the magnetic induction line is greater. When the rotational speed is kept unchanged, reduction of the CEMF value in the permanent magnet motor where the inter-turn short circuit occurs may effectively prolong the operational time of the short-circuited coil of the permanent magnet motor. Therefore, reduction of the permanent magnet parameter in the permanent magnet motor may effectively prolong the operational time of the short-circuited coil of the permanent magnet motor, optimize the permanent magnet motor, and improve the safety of the permanent magnet motor. The reduction of the permanent magnet parameter may affect the electromagnetic performance of the permanent magnet motor. A reluctance torque, e.g., a permanent magnet assisted reluctance motor, may further be increased on the basis of reducing the permanent magnet parameter, to ensure the electromagnetic performance The insulating material is used for the wire of the coil. After the inter-turn short circuit occurs to the permanent magnet motor, the short-circuited coil converts electric energy into heat energy under the CEMF and the resistance of the short-circuited coil, which causes the temperature of the coil rising continuously. The insulating material may be molten due to the over-temperature of the coil to further aggravate the short circuit failure of the coil. Therefore, replacing the insulating material of the permanent magnet motor to improve the heat resistance class and heat dissipation performance of the insulating material may ensure that the short-circuited coil of the permanent magnet motor operates for a longer time, optimize the permanent magnet motor, and improve the safety of the permanent magnet motor. The cooling structure of the permanent magnet motor is configured to cool the permanent magnet motor. After the inter-turn short circuit occurs to the permanent magnet motor, the short-circuited coil converts electric energy into heat energy under the CEMF and the resistance of the short-circuited coil, which causes the temperature of the coil rising continuously. The cooling structure of the permanent magnet motor may be configured to dissipate heat by ventilation, heat conduction, and other manners to control the temperature of the permanent magnet motor and avoid an over-temperature. Therefore, adjusting the cooling structure of the permanent magnet motor to enhance the heat dissipation effect of the cooling structure of the permanent magnet motor may ensure that the short-circuited coil of the permanent magnet motor operates for a longer time, optimize the permanent magnet motor, and improve the safety of the permanent magnet motor. In addition, adjusting the electromagnetic structure of the permanent magnet motor may further include adjusting the slot shape parameter, adjusting the electromagnetic wire parameter, adjusting the turns of the coil, changing a wire material, etc. According to the method for adjusting a permanent magnet motor provided in the disclosure, the electronic equipment acquires the CEMF parameter, the information of the electromagnetic structure of the permanent magnet motor to be adjusted and the minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, to determine the operational time of the short-circuited coil. The electronic equipment further judges, according to the operational time of the short-circuited coil, whether the adjustment instruction is required to be transmitted to the production equipment. When the operational time is inconsistent with the preset time, the electronic equipment transmits the adjustment instruction to the production equipment. The production equipment adjusts, according to the adjustment instruction, the electromagnetic structure of the permanent magnet motor to be adjusted. In the disclosure, the operational time of the short-circuited coil is acquired to judge whether the permanent magnet motor to be adjusted may operate for the preset time safely, and the adjustment instruction is transmitted, according to a judgment result, to the production equipment of the permanent magnet motor to be adjusted, to implement optimization and adjustment of the permanent magnet motor. As such, the short-circuited coil of the permanent magnet motor may operate for a longer time, the permanent magnet motor is optimized, and the safety of the permanent magnet motor is improved. A specific implementation of determining, according to the CEMF parameter, the information of the electromagnetic structure and the minimum impedance value, the operational time of the short-circuited coil in the operation S102inFIG.1will be described in detail based on the embodiment shown inFIG.1. FIG.2is a flowchart of another method for adjusting a permanent magnet motor according to an embodiment of the disclosure. As shown inFIG.2, the method of the embodiment may include the following operations. In operation S201, an electronic equipment acquires a CEMF parameter, information of an electromagnetic structure of a permanent magnet motor to be adjusted and a minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, here the CEMF parameter is a CEMF value of the permanent magnet motor at a maximum rotational speed. An implementation of the operation S201is similar to that of the operation S101in the embodiment shown inFIG.1, and will not be elaborated in the embodiment. In operation S202, the electronic equipment determines, according to the CEMF parameter and the minimum impedance value, a generated heat value of the short-circuited coil. In the embodiment, the electronic equipment acquires the CEMF parameter and the minimum impedance value from the operation S201. In the short-circuited coil of the permanent magnet motor, a CEMF acts on the short-circuited coil to convert electric energy into heat energy. A generated heat value of the short-circuited coil in a unit time is calculated through the following formula: Q=Vet/R(1). Here Q is the generated heat value in Joule. V is the CEMF value in volt. R is a resistance value in ohm. t is the unit time, i.e., one hour. In operation S203, the electronic equipment determines, according to the generated heat value of the short-circuited coil and the information of the electromagnetic structure, an operational time of the short-circuited coil. The information of the electromagnetic structure of the permanent magnet motor includes information of cooling electromagnetic structure of the permanent magnet motor, a melting point of an insulating material, a permanent magnet parameter, a slot shape parameter of the motor, an electromagnetic wire parameter, etc. The electronic equipment may acquire the generated heat value of the short-circuited coil in the unit time according to the operation S202, and acquire a heat dissipation value generated by a cooling electromagnetic structure of the permanent magnet motor in the unit time according to the information of cooling electromagnetic structure of the permanent magnet motor. The electronic equipment may calculate a practical generated heat value of the permanent magnet motor according to the generated heat value and the heat dissipation value. The practical generated heat value is a generated heat value that may still be accumulated after the interaction of heating and heat dissipation. With the accumulation of this part of generated heat value, the temperature of the permanent magnet motor may rise continuously. A high enough temperature of the permanent magnet motor may cause the insulating material melting and the coil fusing, to further bring the safety problem of the permanent magnet motor. Therefore, an operational time of the permanent magnet motor before reaching the high enough temperature is the operational time of the short-circuited coil of the permanent magnet motor. The temperature value may be an empirical value, and it is determined empirically that the permanent magnet motor is easily damaged after reaching the temperature. The temperature value may also be a calculated value, and it is calculated according to melting points of the insulating material, a wire material, etc. The temperature value may also be a trial value, and it is determined according to tests that the permanent magnet motor is easily damaged after reaching the temperature. In operation S204, the electronic equipment transmits, to a production equipment, an adjustment instruction configured to instruct the production equipment to adjust the electromagnetic structure of the permanent magnet motor to be adjusted, when the operational time is inconsistent with a preset time, so that the production equipment adjusts the electromagnetic structure of the permanent magnet motor to be adjusted. An implementation of the operation S204is similar to that of the operation S103in the embodiment shown inFIG.1, and will not be elaborated in the embodiment. According to the method for adjusting a permanent magnet motor provided in the disclosure, the electronic equipment acquires the CEMF parameter and the minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, to determine the generated heat value of the short-circuited coil. The electronic equipment determines, according to the generated heat value of the short-circuited coil and the information of the electromagnetic structure of the permanent magnet motor to be adjusted, the operational time during which no safety problem may be brought to the short-circuited coil. The electronic equipment further judges, according to the operational time of the short-circuited coil, whether the adjustment instruction is required to be transmitted to the production equipment. When the operational time is inconsistent with the preset time, the electronic equipment transmits the adjustment instruction to the production equipment. The production equipment adjusts, according to the adjustment instruction, the electromagnetic structure of the permanent magnet motor to be adjusted. In the disclosure, the electronic equipment calculates the generated heat value of the short-circuited coil according to the CEMF parameter and the minimum impedance value. The electronic equipment determines, according to the generated heat value and the electromagnetic structure parameter of the permanent magnet motor, the operational time of the short-circuited coil. The electronic equipment optimizes and adjusts, according to the operational time, the permanent magnet motor to be adjusted. As such, the short-circuited coil of the permanent magnet motor may operate for a longer time, the permanent magnet motor is optimized, and the safety of the permanent magnet motor is improved. A specific implementation of determining, according to the CEMF parameter and the minimum impedance value, the generated heat value of the short-circuited coil in the operation S202inFIG.2will be described in detail based on the embodiments shown inFIGS.1and2. FIG.3is a flowchart of another method for adjusting a permanent magnet motor according to an embodiment of the disclosure. As shown inFIG.3, the method of the embodiment may include the following operations. In operation S301, an electronic equipment acquires a CEMF parameter, information of an electromagnetic structure of a permanent magnet motor to be adjusted and a minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, here the CEMF parameter is a CEMF value of the permanent magnet motor at a maximum rotational speed. An implementation of the operation S301is similar to that of the operation S101in the embodiment shown inFIG.1, and will not be elaborated in the embodiment. In operation S302, the electronic equipment acquires a coil parameter of a coil of the permanent magnet motor to be adjusted. The electronic equipment may acquire the coil parameter of the coil corresponding to the permanent magnet motor to be adjusted, by reading from a memory. The electronic equipment may also acquire the coil parameter of the coil corresponding to the permanent magnet motor to be adjusted, according to input information. No limitations are made thereto in the disclosure. The coil parameter of the coil of the permanent magnet motor includes a cross-sectional area of a wire, resistivity of the wire, a length of the wire, turns of the coil, etc. In operation S303, the electronic equipment determines, according to the CEMF parameter and the minimum impedance value, a current value of the short-circuited coil. In the embodiment, the short-circuited coil generates the current value under the CEMF. The current value is a quotient of the CEMF value over the minimum impedance value. In operation S304, the electronic equipment determines, according to the coil parameter and the current value, a current density of the short-circuited coil. In the embodiment, the current density of the short-circuited coil is a value of a current flowing through a unit area. The current density is a ratio of the current of the short-circuited coil to the cross-sectional area of the wire. In operation S305, the electronic equipment determines, according to the current density and the minimum impedance value, a generated heat value of the short-circuited coil. In the embodiment, a generated heat value of the short-circuited coil in a unit time may be calculated according to the current and the minimum impedance value through the following formula: Q=I2Rt(1). Here Q is the generated heat value in Joule. I is the current value in Ampere. R is a resistance value in ohm. t is the unit time, i.e., one hour. In operation S306, the electronic equipment determines, according to the generated heat value of the short-circuited coil and the information of the electromagnetic structure, an operational time of the short-circuited coil. An implementation of the operation S306is similar to that of the operation S203in the embodiment shown inFIG.2, and will not be elaborated in the embodiment. In operation S307, the electronic equipment transmits, to a production equipment, an adjustment instruction configured to instruct the production equipment to adjust the electromagnetic structure of the permanent magnet motor to be adjusted, when the operational time is inconsistent with a preset time, so that the production equipment adjusts the electromagnetic structure of the permanent magnet motor to be adjusted. An implementation of the operation S307is similar to that of the operation S103in the embodiment shown inFIG.1, and will not be elaborated in the embodiment. According to the method for adjusting a permanent magnet motor provided in the disclosure, the electronic equipment acquires the CEMF parameter, the minimum impedance value and the coil parameter of any short-circuited coil of the permanent magnet motor to be adjusted, to calculate the current value and the current density. The electronic equipment determines, according to the current value, the current density and the minimum impedance value, the generated heat value of the short-circuited coil in the unit time. The electronic equipment determines, according to the generated heat value and the information of the electromagnetic structure of the permanent magnet motor to be adjusted, the operational time of the short-circuited coil. The electronic equipment further judges, according to the operational time, whether the adjustment instruction is required to be transmitted to the production equipment. When the operational time is inconsistent with the preset time, the electronic equipment transmits the adjustment instruction to the production equipment. The production equipment adjusts, according to the adjustment instruction, the electromagnetic structure of the permanent magnet motor to be adjusted. In the disclosure, the electronic equipment calculates the current value, the current density, and the generated heat value of the short-circuited coil in the unit time, calculates the operational time of the short-circuited coil according to the generated heat value and the electromagnetic structure parameter of the permanent magnet motor, and further optimizes and adjusts the permanent magnet motor to be adjusted according to the operational time. As such, the short-circuited coil of the permanent magnet motor may operate for a longer time, the permanent magnet motor is optimized, and the safety of the permanent magnet motor is improved. Based on the embodiments shown inFIGS.1to3, when the permanent magnet motor is used, the method for adjusting a permanent magnet motor may also be configured to adjust a rotational speed of the permanent magnet motor, to ensure that the train may smoothly arrive the destination. FIG.4is a flowchart of another method for adjusting a permanent magnet motor according to an embodiment of the disclosure. As shown inFIG.4, when a rail transit vehicle runs, the method of the embodiment may include the following operations. In operation S401, an electronic equipment acquires a CEMF parameter, information of an electromagnetic structure of a permanent magnet motor to be adjusted and a minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, here the CEMF parameter is a CEMF value of the permanent magnet motor at a maximum rotational speed. An implementation of the operation S401is similar to that of the operation S101in the embodiment shown inFIG.1, and will not be elaborated in the embodiment. In operation S402, the electronic equipment determines, according to the information of the electromagnetic structure, the minimum impedance value, and a target operational time of the permanent magnet motor, a target rotational speed of the permanent magnet motor to be adjusted. In the embodiment, after an inter-turn short circuit occurs to a coil of the permanent magnet motor to be adjusted, the rail transit vehicle needs to keep operating for the target time to arrive the destination, to reduce the influence on the operating order as much as possible. After acquiring information such as the information of the electromagnetic structure, the minimum impedance value, and the target operational time of the permanent magnet motor, etc. according to the operation S401, the electronic equipment may calculate the target rotational speed of the permanent magnet motor to ensure that the permanent magnet motor may operate for the target time at the rotational speed. In a specific embodiment, the rotational speed of the permanent magnet motor is calculated through the following operations. In a first operation, the electronic equipment acquires a maximum temperature reachable for the permanent magnet motor to be adjusted, and determines, according to the temperature and a current temperature of the permanent magnet motor, a temperature variation of the permanent magnet motor. In a second operation, the electronic equipment determines, according to the temperature variation and target operational time of the permanent magnet motor, a practical generated heat value of the short-circuited coil of the permanent magnet motor in a unit time. In a third operation, the electronic equipment determines, according to the practical generated heat value and a heat dissipation value of a cooling electromagnetic structure of the permanent magnet motor in the unit time in the information of the electromagnetic structure, a generated heat value of the short-circuited coil in the unit time. In a fourth operation, the electronic equipment determines, according to the generated heat value and the minimum impedance value, a CEMF value of the short-circuited coil. In a fifth operation, the electronic equipment determines, according to a coil parameter, the CEMF value, and the information of the electromagnetic structure of the permanent magnet motor, the target rotational speed of the permanent magnet motor. Here the coil parameter includes turns of the coil, resistivity of the wire, etc., and the information of the electromagnetic structure of the permanent magnet motor includes a permanent magnet parameter, etc. In operation S403, the electronic equipment determines, according to the target rotational speed, a parameter to be adjusted. In the embodiment, the electronic equipment may calculate, according to the target rotational speed calculated in the operation S402, a running speed of the rail transit vehicle at the target rotational speed, and determine, according to the running speed, a parameter required to be adjusted. In operation S404, the electronic equipment adjusts, according to the parameter to be adjusted, the operation of the permanent magnet motor to be adjusted. In the embodiment, the rail transit vehicle includes multiple permanent magnet motors. When the inter-turn short circuit occurs to the permanent magnet motor to be adjusted, power of the permanent magnet motor to be adjusted is cut off. In order to ensure the safety of the permanent magnet motor to be adjusted when the rail transit vehicle runs, the electronic equipment adjusts other permanent magnet motors of the rail transit vehicle according to the parameter to be adjusted determined in the operation S403, to reduce the running speed of the rail transit vehicle to a safe running speed, to further ensure that the rotational speed of the permanent magnet motor to be adjusted is less than or equal to the target rotational speed. According to the method for adjusting a permanent magnet motor provided in the embodiment, the electronic equipment determines, according to the information of the electromagnetic structure, the minimum impedance value, and a preset operational time of the permanent magnet motor, the target rotational speed of the permanent magnet motor to be adjusted, and further determines the parameter to be adjusted of the rail transit vehicle according to the target rotational speed, to make an adjusted rotational speed of the permanent magnet motor of the rail transit vehicle less than or equal to the target rotational speed. In the disclosure, for the rail transit vehicle in operation, when the inter-turn short circuit occurs to the permanent magnet motor of the rail transit vehicle, the electronic equipment adjusts the running speed of the rail transit vehicle to make the operational time of the permanent magnet motor where the inter-turn short circuit occurs equal to the target operational time, to ensure that the rail transit vehicle may keep running to the destination safely after the inter-turn short circuit failure occurs to the permanent magnet motor, to further ensure the safety of a person or goods in the rail transit vehicle, and improve the safety of the rail transit vehicle. FIG.5is a schematic diagram of electromagnetic structure of a device for adjusting a permanent magnet motor according to an embodiment of the disclosure. As shown inFIG.5, the device for adjusting a permanent magnet motor10of the embodiment is configured to implement the operations corresponding to the electronic equipment in any of the above-mentioned method embodiments. The device for adjusting a permanent magnet motor10of the embodiment may include a first acquisition unit11, a first processing unit12, a judgment unit13, and a first adjustment unit14. The first acquisition unit11is configured to acquire a CEMF parameter, information of an electromagnetic structure of a permanent magnet motor to be adjusted and a minimum impedance value of any short-circuited coil of the permanent magnet motor to be adjusted, here the CEMF parameter is a CEMF value of the permanent magnet motor at a maximum rotational speed. The first processing unit12is configured to determine, according to the CEMF parameter, the information of the electromagnetic structure and the minimum impedance value, an operational time of the short-circuited coil. The judgment unit13is configured to judge whether the operational time is consistent with a preset time. The first adjustment unit14is configured to transmit, to a production equipment, an adjustment instruction configured to instruct the production equipment to adjust the electromagnetic structure of the permanent magnet motor to be adjusted, when the operational time is inconsistent with the preset time, so that the production equipment adjusts the electromagnetic structure of the permanent magnet motor to be adjusted. The device for adjusting a permanent magnet motor10provided in the embodiment of the disclosure may execute the above-mentioned method embodiments, and a specific implementation principle and technical effect thereof may refer to the above-mentioned method embodiments, and will not be elaborated in the embodiment. FIG.6is a schematic diagram of electromagnetic structure of another device for adjusting a permanent magnet motor according to an embodiment of the disclosure. Based on the embodiment shown inFIG.5, as shown inFIG.6, the device for adjusting a permanent magnet motor10of the embodiment is configured to implement the operations corresponding to the electronic equipment in any of the above-mentioned method embodiments. In the device for adjusting a permanent magnet motor10of the embodiment, the first acquisition unit11, when configured to acquire the CEMF parameter, is specifically configured to: transmit an acquisition instruction to a permanent magnet motor, and receive the CEMF parameter returned by the permanent magnet motor according to the acquisition instruction; or, determine, through a simulation technology, the CEMF value of the permanent magnet motor at the maximum rotational speed. The first processing unit12includes a first determination module22and a second determination module23. The first determination module22is configured to determine, according to the CEMF parameter and the minimum impedance value, a generated heat value of the short-circuited coil. The second determination module23is configured to determine, according to the generated heat value of the short-circuited coil and the information of the electromagnetic structure, the operational time of the short-circuited coil. The device for adjusting a permanent magnet motor10provided in the embodiment of the disclosure may execute the above-mentioned method embodiments, and a specific implementation principle and technical effect thereof may refer to the above-mentioned method embodiments, and will not be elaborated in the embodiment. FIG.7is a schematic diagram of electromagnetic structure of another device for adjusting a permanent magnet motor according to an embodiment of the disclosure. Based on the embodiments shown inFIGS.5and6, as shown inFIG.7, the device for adjusting a permanent magnet motor10of the embodiment is configured to implement the operations corresponding to the electronic equipment in any of the above-mentioned method embodiments. The device for adjusting a permanent magnet motor10of the embodiment may further include a second acquisition unit15. The second acquisition unit15is configured to acquire a coil parameter of a coil of the permanent magnet motor to be adjusted. The first determination module22includes a first determination submodule33, a second determination submodule34, and a third determination submodule35. The first determination submodule33is configured to determine, according to the CEMF parameter and the minimum impedance value, a current value of the short-circuited coil. The second determination submodule34is configured to determine, according to the coil parameter and the current value, a current density of the short-circuited coil. The third determination submodule35is configured to determine, according to the current density and the minimum impedance value, the generated heat value of the short-circuited coil. In an example, the information of the electromagnetic structure of the permanent magnet motor to be adjusted includes one or more of: information of cooling electromagnetic structure of the permanent magnet motor, a melting point of an insulating material, a permanent magnet parameter, a slot shape parameter of the permanent magnet motor, an electromagnetic wire parameter, etc. In an example, the adjustment instruction is specifically configured to instruct the production equipment to perform one or more of: adjusting the permanent magnet parameter, replacing the insulating material, adjusting a cooling electromagnetic structure of the permanent magnet motor, adjusting the slot shape parameter of the permanent magnet motor, adjusting the electromagnetic wire parameter, etc., on the permanent magnet motor to be adjusted. The device for adjusting a permanent magnet motor10provided in the embodiment of the disclosure may execute the above-mentioned method embodiments, and a specific implementation principle and technical effect thereof may refer to the above-mentioned method embodiments, and will not be elaborated in the embodiment. FIG.8is a schematic diagram of electromagnetic structure of another device for adjusting a permanent magnet motor according to an embodiment of the disclosure. Based on the embodiment shown inFIG.5, as shown inFIG.8, when the rail transit vehicle runs, the device for adjusting a permanent magnet motor10of the embodiment is configured to implement the operations corresponding to the electronic equipment in any of the above-mentioned method embodiments. The device for adjusting a permanent magnet motor10of the embodiment may further include a second processing unit41, a determination unit42, and a second adjustment unit43. The second processing unit41is configured to determine, according to the information of the electromagnetic structure, the minimum impedance value, and a target operational time of the permanent magnet motor, a target rotational speed of the permanent magnet motor to be adjusted. The determination unit42is configured to determine, according to the target rotational speed, a parameter to be adjusted. The second adjustment unit43is configured to adjust, according to the parameter to be adjusted, the operation of the permanent magnet motor to be adjusted. The device for adjusting a permanent magnet motor10provided in the embodiment of the disclosure may execute the above-mentioned method embodiments, and a specific implementation principle and technical effect thereof may refer to the above-mentioned method embodiments, and will not be elaborated in the embodiment. In the disclosure, functional units of the device for adjusting a permanent magnet motor may be divided according to the above-mentioned method examples. For example, each functional unit may be divided by way of corresponding to each function, or two or more functions may be integrated into a processing unit. The integrated unit may be implemented in a hardware form, or may be implemented in form of a software functional unit. It is to be noted that the division of units in each embodiment of the disclosure is schematic and is only a logical function division, and other division manners may be adopted during practical implementation. FIG.9is a schematic diagram of hardware electromagnetic structure of an electronic equipment according to an embodiment of the disclosure. As shown inFIG.9, the electronic equipment50is configured to implement the operations corresponding to the electronic equipment in any of the above-mentioned method embodiments. The electronic equipment50of the embodiment may include a memory51, a processor52, and a communication interface54. The memory51is configured to store a computer program. The processor52is configured to execute the computer program stored in the memory to implement the method for adjusting a permanent magnet motor in the above-mentioned embodiments. References may specifically be made to the related descriptions in the above-mentioned method embodiments. In an embodiment, the memory51may be independent, or may be integrated with the processor52. When the memory51is a device independent of the processor52, the electronic equipment50may further include a bus53. The bus53is configured to connect the memory51with the processor52. In an embodiment, the embodiment further includes the communication interface54which may be connected with the processor51through the bus53. The processor52may control the communication interface54to realize receiving and transmitting functions of the electronic equipment50. The electronic equipment provided in the embodiment may be configured to execute the method for adjusting a permanent magnet motor, and an implementation and technical effect thereof are similar, and will not be elaborated in the embodiment. The disclosure also provides a computer-readable storage medium including a computer program which is configured to implement the method for adjusting a permanent magnet motor in the above-mentioned embodiments. In some embodiments provided by the disclosure, it is to be understood that the disclosed device and method may be implemented in other manners. For example, the device embodiment as described above is only schematic, and for example, the division of modules is only a logical function division, and other division manners may be adopted during practical implementation. For example, multiple modules may be combined or integrated into another system, or some characteristics may be neglected or may not be executed. In addition, coupling or direct coupling or communication connection between each displayed or discussed component may be indirect coupling or communication connection implemented through some interfaces, devices or modules, and may be electrical and mechanical or adopt other forms. The modules described as separate parts may be or may not be physically separated, and parts displayed as modules may be or may not be physical units, namely, they may be located in the same place, or may be distributed to multiple network units. Part or all of the modules may be selected to achieve the purposes of the solutions of the embodiments as required. In addition, each function module in each embodiment of the disclosure may be integrated into a processing module, or each module may physically exist independently, or two or more modules may be integrated into a module. The integrated module may be implemented in a hardware form, or may be implemented in form of hardware and software function unit. The integrated module implemented in form of a software function module may be stored in a computer-readable storage medium. The software function module is stored in a storage medium, including multiple instructions configured to enable a computer device (which may be a personal computer, a server, a network device, etc.) or a processor to execute part of the steps of the method in each embodiment of the disclosure. It is to be understood that the processor may be a Central Processing Unit (CPU), or another general-purpose processor, a Digital Signal Processor (DSP), an Application Specific Integrated Circuit (ASIC), etc. The general-purpose processor may be a microprocessor, or the processor may be any conventional processor, etc. The steps of the method disclosed in combination with the disclosure may be directly embodied to be executed and completed by a hardware processor, or executed and completed by a combination of hardware and software modules in the processor. The memory may include a high-speed Random Access Memory (RAM), or may include a Non-Volatile Memory (NVM), such as at least one disk memory, or may be a U disk, a mobile hard disk, a Read-Only Memory (ROM), a magnetic disk, an optical disk, etc. The bus may be an Industry Standard Architecture (ISA) bus, a Peripheral Component Interconnect (PCI) bus, an Extended Industry Standard Architecture (EISA) bus, etc. The bus may be divided into an address bus, a data bus, a control bus, etc. For ease of representation, the bus in the drawings of the disclosure is not limited to one bus or one type of bus. The computer-readable storage medium may be implemented by any type of volatile or nonvolatile storage devices or a combination thereof, such as a Static Random-Access Memory (SRAM), an Electrically-Erasable Programmable Read-Only Memory (EEPROM), an Erasable Programmable Read-Only Memory (EPROM), a Programmable Read-Only Memory (PROM), a ROM, a magnetic memory, a flash memory, a magnetic disk, or an optical disk. The storage medium may be any available medium accessible for a general-purpose or special-purpose computer. It may be understood by those of ordinary skill in the art that all or part of the steps of the above-mentioned method embodiments may be completed by instructing related hardware through a program. The program may be stored in a computer-readable storage medium. The program is executed to execute the steps in the above-mentioned method embodiments. The storage medium includes various media capable of storing program codes such as a ROM, a RAM, a magnetic disk, or an optical disk, etc. Finally, it is to be noted that the above-mentioned embodiments are not used to limit but used only to describe the technical solutions of the disclosure. Although the disclosure is described in detail with reference to the above-mentioned embodiments, it is to be understood by those of ordinary skill in the art that the technical solutions recorded in the above-mentioned embodiments may also be modified, or part or all of technical features therein may be equivalently replaced. The essences of corresponding technical solutions obtained by these modifications or replacements do not depart from the scopes of the technical solutions of the embodiments of the disclosure. | 52,281 |
11863110 | DETAILED DESCRIPTION OF THE DISCLOSURE Hereinafter, embodiments of the present disclosure will be described with reference to the accompanying drawings. FIG.1is a view illustrating a sensorless brushless direct current (BLDC) motor driver according to one embodiment of the present disclosure. Referring toFIG.1, a sensorless BLDC motor driver100according to one embodiment may include a BLDC motor200, an inverter300, a comparator400, and a BLDC motor controller500. The BLDC motor controller500may be abbreviated as a controller. The sensorless BLDC motor200includes a stator having 3-phase coils UC, VC, and WC with different phases and a rotor using permanent magnets, and inFIG.1, the rotor is omitted. The stator of the sensorless BLDC motor200may include a first coil UC with a U-phase (first phase), a second coil VC with a V-phase (second phase), and a third coil WC with a W-phase (third phase). The BLDC motor200is driven by a voltage supplied from the inverter300to each of the 3-phase coils UC, VC, and WC, and magnetic forces generated by the first to third coils UC, VC, and WC may rotate the rotor of the BLDC motor200. The inverter300may be operated according to control of the BLDC motor controller500to supply a first power voltage VDD or second power voltage VSS to each of the 3-phase coils UC, VC, and WC of the sensorless BLDC motor200through one of first to third nodes U, V, and W, or to float the corresponding coil without supplying the first and second power voltages VDD and VSS thereto. The inverter300receives the first power voltage VDD and the second power voltage VSS from a power supply part VDC. The inverter300may receive first-1 and first-2 coil control signals UP and UN, second-1 and second-2 coil control signals VP and VN, and third-1 and third-2 coil control signals WP and WN from the BLDC motor controller500. The coil control signals UP, UN, VP, VN, WP, and WN supplied from the BLDC motor controller500may be pulse width modulation (PWM) signals. The inverter300may include a first driving part which drives the first coil UC of the sensorless BLDC motor200, and the first driving part may include a first pull-up transistor Tup and a first pull-down transistor Tun which are disposed between and connected to a supply line of the first power voltage VDD and a supply line of the second power voltage VSS in series. A connection node between the first pull-up transistor Tup and the first pull-down transistor Tun may be connected to the first coil UC through a first node U. When the first-1 coil control signal UP supplied from the BLDC motor controller500has a gate-on voltage, the first pull-up transistor Tup may be turned on so that the first power voltage VDD may be applied to the first coil UC through the first node U. When the first-2 coil control signal UN supplied from the BLDC motor controller500is a gate-on voltage, the first pull-down transistor Tun may be turned on so that the second power voltage VSS may be applied to the first coil UC through the first node U. When both of the first-1 and first-2 coil control signals UP and UN supplied from the BLDC motor controller500have gate-off voltages, both of the first pull-up transistor Tup and the first pull-down transistor Tun may be turned off so that the first node U and the first coil UC may enter floating states. The inverter300may include a second driving part which drives the second coil VC of the sensorless BLDC motor200, and the second driving part may include a second pull-up transistor Tvp and a second pull-down transistor Tvn which are disposed between and connected to the supply line of the first power voltage VDD and the supply line of the second power voltage VSS in series. A connection node between the second pull-up transistor Tvp and the second pull-down transistor Tvn may be connected to the second coil VC through a second node V. When the second-1 coil control signal VP supplied from the BLDC motor controller500has a gate-on voltage, the second pull-up transistor Tvp is turned on so that the first power voltage VDD may be applied to the second coil VC through the second node V. When the second-2 coil control signal VN supplied from the BLDC motor controller500has a gate-on voltage, the second pull-down transistor Tvn may be turned on so that the second power voltage VSS may be applied to the second coil VC through the second node V. When both of the second-1 and second-2 coil control signals VP and VN supplied from the BLDC motor controller500have gate-off voltages, both of the second pull-up transistor Tvp and the second pull-down transistor Tvn are turned off so that both of the second node V and the second coil VC may enter floating states. The inverter300may include a third driving part which drives the third coil WC of the sensorless BLDC motor200, and the third driving part may include a third pull-up transistor Twp and a third pull-down transistor Twn which are disposed between and connected to the supply line of the first power voltage VDD and the supply line of the second power voltage VSS in series. A connection node between the third pull-up transistor Twp and the third pull-down transistor Twn may be connected to the third coil WC through a third node W. When the third-1 coil control signal WP supplied from the BLDC motor controller500has a gate-on voltage, the third pull-up transistor Twp may be turned on so that the first power voltage VDD may be applied to the third coil WC through the third node W. When the third-2 coil control signal WN supplied from the BLDC motor controller500has a gate-on voltage, the third pull-down transistor Twn may be turned on so that the second power voltage VSS may be supplied to the third coil WC through the third node W. When both of the third-1 and third-2 coil control signals WP and WN supplied from the BLDC motor controller500have gate-off voltages, both of the third pull-up transistor Twp and the third pull-down transistor Twn are turned off so that both of the third node W and the third coil WC may enter floating states. The comparator400may receive voltages Vcu, Vcv, and Vcw of the 3-phase coils UC, VC, and WC through the first to third nodes U, V, and W, respectively, and receive a neutral voltage Vneutral of a node, which is commonly connected to the 3-phase coils UC, VC, and WC, from the sensorless BLDC motor200. The comparator400may compare each of the voltages Vuc, Vvc, and Vwc of the 3-phase coils UC, VC, and WC with the neutral voltage Vneutral and output a comparison result to the BLDC motor controller500. Using the comparison result supplied from the comparator400, the BLDC motor controller500may detect a back-electro motive force (BEMF) of each of the 3-phase coils UC, VC, and WC and a zero crossing time point at which the BEMF of any one of the 3-phase coils UC, VC, and WC becomes zero. Whenever the zero crossing time point is detected at each of the 3-phase coils UC, VC, and WC, the BLDC motor controller500may control the inverter300by applying a transition time (first time) and a turn-on time (second time), which are predetermined based on the detected zero crossing time point, to generate the plurality of coil control signals UP, UN, VP, VN, WP, and WN to change phases. The 3-phase coils UC, VC, and WC of the sensorless BLDC motor200may alternately enter the floating states. When any one coil of the 3-phase coils UC, VC, and WC enters the floating state, a current flowing through the floated coil does not present, but among the voltages Vuc, Vvc, and Vwc of the 3-phase coils UC, VC, and WC, a voltage of the floated coil may be changed to be gradually increased or decreased. The BLDC motor controller500may detect a time point at which the voltage of the floated coil becomes the same as the neutral voltage Vneutral, that is, the zero crossing time point at which the BEMF of the floated coil becomes zero. Since the current flowing through the floated coil is not present, the BEMF of the floated coil may be detected using a difference between the voltage of the corresponding coil and the neutral voltage Vneutral. At a point at which the voltage of the floated coil becomes the same as the neutral voltage Vneutral, the BEMF of the floated coil may become zero. The BLDC motor controller500may control the inverter300using the zero crossing time point of the BEMF and the predetermined transition time and the predetermined turn-on time of each of the 3-phase coils UC, VC, and WC so that the first power voltage VDD or the second power voltage VSS is supplied to each of the 3-phase coils UC, VC, and WC of the sensorless BLDC motor200or the 3-phase coils UC, VC, and WC of the sensorless BLDC motor200are floated. The transition time is a time from a time point, at which any one coil of the 3-phase coils UC, VC, and WC is floated and zero crossing of the BEMF of the floated coil occurs, to a time point at which the first power voltage VDD or the second power voltage VSS starts to be supplied to the corresponding coil. The turn-on time is a time for which the first power voltage VDD or the second power voltage VSS is supplied to the corresponding coil, that is, from a time point, at which the first power voltage VDD or the second power voltage VSS starts to be supplied thereto, to a time point at which the floating state, in which both of the first and second power voltages VDD and VSS are not supplied thereto, starts. A floating time of the corresponding coil may be about two times the transition time. Specifically, among the voltages Vuc, Vvc, and Vwc of the 3-phase coils UC, VC, and WC, when the zero crossing time point, at which a voltage of a floated coil is increased gradually and becomes the same as the neutral voltage Vneutral, is detected, the BLDC motor controller500may control the first power voltage VDD to be supplied to the coil in the floating state for the turn-on time after the transition time from a zero crossing detection time point and may float the corresponding coil after the turn-on time. When the zero crossing time point, at which a voltage of a floated coil is decreased gradually and becomes the same as the neutral voltage Vneutral, is detected, the BLDC motor controller500may control the second power voltage VSS to be supplied to the coil in the floating state for the turn-on time after the transition time from a zero crossing detection time point and may float the corresponding coil after the turn-on time. For example, the BLDC motor controller500may control the first power voltage VDD to be applied to the first coil UC for the turn-on time through the first pull-up transistor Tup after the transition time from the zero crossing time point at which the voltage Vuc of the floated first coil UC is increased to be the same as the neutral voltage Vneutral and may float the first coil UC after the turn-on time. The BLDC motor controller500may control the second power voltage VSS to be applied to the first coil UC for the turn-on time through the first pull-down transistor Tun after the transition time from the zero crossing time point at which the voltage Vuc of the floated first coil UC is decreased to be the same as the neutral voltage Vneutral and may float the first coil UC after the turn-on time. The BLDC motor controller500may control the first power voltage VDD to be applied to the second coil VC for the turn-on time through the second pull-up transistor Tvp after the transition time from the zero crossing time point at which the voltage Vvc of the floated second coil VC is increased to be the same as the neutral voltage Vneutral and may float the second coil VC after the turn-on time. The BLDC motor controller500may control the second power voltage VSS to be applied to the second coil VC for the turn-on time through the second pull-down transistor Tvn after the transition time from the zero crossing time point at which the voltage Vvc of the floated second coil VC is decreased to be the same as the neutral voltage Vneutral and may float the second coil VC after the turn-on time. The BLDC motor controller500may control the first power voltage VDD to be applied to the third coil WC for the turn-on time through the third pull-up transistor Twp after the transition time from the zero crossing time point at which the voltage Vwc of the floated third coil WC is increased to be the same as the neutral voltage Vneutral and may float the third coil WC after the turn-on time. The BLDC motor controller500may control the second power voltage VSS to be applied to the third coil WC from the turn-on time through the third pull-down transistor Twn after the transition time from the zero crossing time point at which the voltage Vwc of the floated third coil WC is decreased to be the same as the neutral voltage Vneutral and may float the third coil WC after the turn-on time. Particularly, the BLDC motor controller500may operate in a test mode to detect an optimum driving mode, that is, the transition time and the turn-on time in which optimum efficiency is obtained, suitable for driving characteristics and a driving speed of the sensorless BLDC motor driver100. The BLDC motor controller500may set the transition time and the turn-on time detected in the test mode as the transition time and the turn-on time of the optimum driving mode of the sensorless BLDC motor driver100to maximize driving efficiency of the BLDC motor driver100. The BLDC motor controller500may start in a 120 degree driving mode, of which a basic rotation angle is 120 degrees, as a driving mode. The 120 degree driving mode may be set to have the turn-on time of 120 degrees corresponding to a 120 degree rotation section of the sensorless BLDC motor200and the transition time of 30 degrees corresponding to a 30 degree rotation section of the sensorless BLDC motor200. The BLDC motor controller500may repeatedly perform a test method of driving the sensorless BLDC motor200using the inverter300and detecting a driving error of the sensorless BLDC motor200while increasing the turn-on time from 120 degrees to 180 degrees step by step and decreasing the transition time from 30 degrees to 0 degrees step by step in the test mode. When the driving error of the sensorless BLDC motor200is detected, the BLDC motor controller500may set a turn-on time and a transition time, which are adjusted at a previous operation, as the turn-on time and the transition time of the optimum driving mode. In other words, the BLDC motor controller500may determine a driving mode having a maximum turn-on time and a minimum transition time among turn-on times and transition times, for which the driving error of the sensorless BLDC motor200is not detected, as the driving mode with optimum efficiency. FIG.2is a flowchart illustrating a method of testing the sensorless BLDC motor driver according to one embodiment of the present disclosure. Referring toFIG.2, the BLDC motor controller500may perform an initial driving operation (S202) of driving the sensorless BLDC motor driver100in the test mode and driving the BLDC motor driver100in the 120 degree driving mode which is a basic driving mode. The BLDC motor controller500adjusts a time by increasing the turn-on time of the test mode step by step and decreasing a transition time of a test mode step by step after rotating the sensorless BLDC motor200by one revolution in the 120 degree driving mode using the inverter300(S204). The BLDC motor controller500may drive the sensorless BLDC motor200and detect whether a driving error of the sensorless BLDC motor200occurs using an adjusted transition time and an adjusted turn-on time using the inverter300. The BLDC motor controller500may repeatedly perform a test operation of adjusting the time by increasing the turn-on time of the test mode step by step and decreasing the transition time of the test mode step by step, driving the sensorless BLDC motor200using an adjusted transition time and an adjusted turn-on time, and detecting the driving error (S206). The turn-on time of the test mode may be increased from 120 degrees to 180 degrees step by step, and the transition time of the test mode may be decreased from 30 degrees to 0 degrees step by step. When the zero crossing time point of the BEMF of each of the 3-phase coils UC, VC, and WC is not detected using the comparator400, since the operation of the BLDC motor controller500is not changed, it may be determined that a driving error, in which the sensorless BLDC motor200does not operate normally, occurs. While the BLDC motor controller500repeatedly performs the test operation of changing the transition time and the turn-on time of the test mode and detecting the driving error of the sensorless BLDC motor200, when the driving error is detected, the BLDC motor controller500may stop the test operation and set a transition time and a turn-on time in an operation, just before the driving error is detected, in the driving mode with optimum efficiency (S208). The driving mode with the optimum efficiency may be set to be suitable for driving characteristics and a driving speed of the BLDC motor driver100in the range of 120 degrees to less than 180 degrees. FIG.3is a flowchart illustrating a method of testing the sensorless BLDC motor driver according to one embodiment of the present disclosure. Referring toFIG.3, the BLDC motor controller500may perform an open loop driving operation (S302) of operating the sensorless BLDC motor driver100in a test mode, setting a transition time and a turn-on time corresponding to the 120 degree driving mode which is the basic driving mode, and driving the BLDC motor driver100in an open loop manner. The 120 degree driving mode may be set to have the turn-on time of 120 degrees and the transition time of 30 degrees. Driving in the open loop manner is to drive in a manner in which the BLDC motor controller500supplies the coil control signals UP, UN, VP, VN, WP, and WN, which each have a basic driving pattern, to drive the sensorless BLDC motor200using the inverter300without detecting the BEMF of each of the 3-phase coils UC, VC, and WC. The BLDC motor controller500may perform a closed loop driving operation (S304) of the 120 degree driving mode of driving the BLDC motor driver100in a closed loop manner after rotating the sensorless BLDC motor200at a speed sufficient to detect the BEMF of each of the 3-phase coils UC, VC, and WC. Driving in the closed loop manner is to drive in a manner in which the BLDC motor controller500detects the BEMF of each of the 3-phase coils UC, VC, and WC using the comparator400and drives the sensorless BLDC motor200using the zero crossing time point of the BEMF, the set transition time, and the set turn-on time using the inverter300. Hereinafter, the BLDC motor driver100is driven in a closed loop driving manner. The BLDC motor controller500may perform a test operation (S306) of the 120 degree driving mode of detecting the BEMF of each of the 3-phase coils UC, VC, and WC while rotating the sensorless BLDC motor200by a certain number of revolutions, for example, one mechanical revolution, in the 120 degree driving mode. The BLDC motor controller500may determine that a driving error of the sensorless BLDC motor200, in which the BEMF is not detected, occurs when one revolution of the sensorless BLDC motor200is not completed (NO) in the test operation S306of the 120 degree driving mode (S312) and may stop the test operation. Meanwhile, the BLDC motor controller500may repeat the open loop driving operation S302, the closed loop driving operation S304, and the 120 degree driving test operation S306, which are described above, after resetting an initial position of the rotor in the BLDC motor200or a driving speed of the BLDC motor driver100to be increased. In the test operation S306of the 120 degree driving mode, when at least one mechanical revolution of the sensorless BLDC motor200is completed (YES), that is, a BEMF detection error does not occur in the 120 degree driving mode, the BLDC motor controller500may proceed to a time adjustment operation (S308). In the time adjustment operation S308, the BLDC motor controller500performs time adjustment of decreasing the transition time step by step in the test mode and increasing the turn-on time step by step in the test mode. The BLDC motor controller500may perform a test operation (S310) of the adjusted time of detecting the BEMF of each of the 3-phase coils UC, VC, and WC while rotating the sensorless BLDC motor200by a certain number of revolutions, for example, one mechanical revolution, using an adjusted transition time and an adjusted turn-on time using the inverter300. In the test operation S310of the adjusted time, when one mechanical revolution of the sensorless BLDC motor200is completed (YES, see S310), the BLDC motor controller500may return to the time adjustment operation S308and may repeat the time adjustment operation S308and the test operation S310of the adjusted time. In the time adjustment operation S308, the BLDC motor controller500may increase the turn-on time in the test mode in increments of one degree from 120 degrees to less than 180 degrees and decrease the transition time in the test mode from 30 degrees to more than 0 degrees in increments of at least 0.5 degrees. In the test operation S310of the adjusted time, when one mechanical revolution of the sensorless BLDC motor200is not completed (NO), the BLDC motor controller500determines that a driving error of the sensorless BLDC motor200, in which the BEMF is not detected, occurs, and proceeds to an optimized time setting operation (S314). In the optimized time setting operation S314, the BLDC motor controller500may set an adjusted transition time and an adjusted turn-on time in an operation, just before the driving error is detected, as a transition time and a turn-on time optimized for driving characteristics and a driving speed of the BLDC motor driver100and stop the test operation. Through the above-described test method, the turn-on time is determined in the range of 120 degrees to less than 180 degrees and the transition time is determined in the range of 30 degrees to more than 0 degrees to be suitable for the driving characteristics and the driving speed of the BLDC motor driver100so that driving efficiency of the BLDC motor driver100may be maximized. For example, an optimum driving mode suitable for the BLDC motor driver100may be set to a 130 degree driving mode having a turn-on time of 130 degrees and a transition time of 25 degrees. The optimum driving mode suitable for the BLDC motor driver100may be set to a 140 degree driving mode having a turn-on time of 140 degrees and a transition time of 20 degrees. The optimum driving mode suitable for the BLDC motor driver100may be set to a 150 degree driving mode having a turn-on time of 150 degrees and a transition time of 15 degrees. The optimum driving mode suitable for the BLDC motor driver100may be set to a 160 degree driving mode having a turn-on time of 160 degrees and a transition time of 10 degrees. The optimum driving mode suitable for the BLDC motor driver100may be set to a 170 degree driving mode having a turn-on time of 170 degree and a transition time of 5 degrees. FIG.4is a signal waveform diagram showing a method of driving the sensorless BLDC motor according to one embodiment of the present disclosure. Referring toFIGS.1and4, a turn-on time T11of the 120 degree driving mode may be a 120 degree time corresponding to a 120 degree rotation section of the sensorless BLDC motor200, and a transition time T12may be a 30 degree time corresponding to a 30 degree rotation section of the sensorless BLDC motor200. The sensorless BLDC motor200may operate in the 120 degree driving mode including first to sixth operations (S401to S406). In the 120 degree driving mode, a 360 degree cycle of each of the coil control signals UP, UN, VP, VN, WP, and WN supplied to the inverter300from the BLDC motor controller500to drive one of the 3-phase coils UC, VC, and WC may include a 120 degree turn-on period, for which the gate-on voltage is supplied, and a 240 degree turn-off period for which the gate-off voltage is supplied. Referring toFIGS.1and4, in the first operation S401corresponding to a 0 degree to 60 degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The third coil UC enters the floating state due to the gate-off voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the first coil UC to the second coil VC. The third-2 coil control signal WN may be activated from the gate-off voltage to the gate-on voltage at a 60 degree time point, at which the transition time T12of 30 degrees elapsed from the zero crossing time point at which the voltage Vwc of the third coil WC in the floating state is decreased, so that the BEMF becomes zero. In the second operation S402corresponding to a 60 degree to 120 degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second coil VC enters the floating state due to the gate-off voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the first coil UC to the third coil WC. The second-1 coil control signal VP may be activated from the gate-off voltage to the gate-on voltage at a 120 degree time point at which the transition time T12of 30 degrees elapsed from the zero crossing time point at which the voltage Vvc of the second coil VC in the floating state is increased so that the BEMF becomes zero. In the third operation S403corresponding to a 120 degree to 180 degree rotation section of the sensorless BLDC motor200, the first coil UC enters the floating state due to the gate-off voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the second coil VC to the third coil WC. The first-2 coil control signal UN may be activated from the gate-off voltage to the gate-on voltage at a 180 degree time point at which the transition time T12of 30 degrees elapsed from the zero crossing time point at which the voltage Vuc of the first coil UC in the floating state is decreased so that the BEMF becomes zero. In the fourth operation S404corresponding to a 180 degree to 240 degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The third coil UC enters the floating state due to the gate-off voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the second coil VC to the first coil. The third-1 coil control signal WP may be activated from the gate-off voltage to the gate-on voltage at a 240 degree time point at which the transition time T12of 30 degree elapsed from the zero crossing time point at which the voltage Vwc of the third coil WC in the floating state is increased so that the BEMF becomes zero. In the fifth operation S405corresponding to a 240 degree to 300 degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The second coil VC enters the floating state due to the gate-off voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the third coil WC to the first coil UC. The second-2 coil control signal VN may be activated from the gate-off voltage to the gate-on voltage at a 300 degree time point at which the transition time T12of 30 degrees elapsed from the zero crossing time point at which the voltage Vvc of the second coil VC in the floating state is decreased so that the BEMF becomes zero. In the sixth operation S406corresponding to a 300 degree to 360 degree rotation section of the sensorless BLDC motor200, the first coil UC enters the floating state due to the gate-off voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the third coil WC to the second coil VC. The first-1 coil control signal UP may be activated from the gate-off voltage to the gate-on voltage at a 360 degree (0 degree) time point, at which the transition time T12of 30 degree elapsed from the zero crossing time point at which the voltage Vuc of the first coil UC in the floating state increases, so that the BEMF becomes zero. FIGS.5and6are signal waveform diagrams showing a method of driving the sensorless BLDC motor according to one embodiment of the present disclosure; and Referring toFIGS.1,5, and6, a turn-on time T21or T31of a “120+N” (N is a positive integer more than 0 degrees and less than 60 degrees) degree driving mode may be set to be a “120+N” degree time corresponding to a “120+N” degree rotation section of the sensorless BLDC motor200, and a transition time T22or T32may be set to be a 30−N/2 degree time corresponding to a “30−N/2” degree rotation section of the sensorless BLDC motor200. For example, inFIG.5, a 140 degree driving mode in which N=20 may be set to have the turn-on time T21of 140 degrees and the transition time 20 degrees. InFIG.6, a 160 degree driving mode in which N=40 may be set to have the turn-on time T31of 160 degrees, and the transition time of 10 degrees. In the “120+N” degree driving mode, a 360 degree rotation section of each of the coil control signals UP, UN, VP, VN, WP, and WN supplied to the inverter300from the BLDC motor controller500in order to drive each of the 3-phase coils UC, VC, and WC may include a “120+N” degree turn-on period for which the gate-on voltage is supplied and a “360−(120+N)” degree turn-off period for which the gate-off voltage is supplied. The sensorless BLDC motor200may operate in the “120+N” degree driving mode including first to twelfth operations S501to S512or S601to S612. Referring toFIGS.1,5, and6, in the first operation S501or S601corresponding to a 0 degree to N degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the first and third coil UC and WC to the second coil VC. In the second operation S502or S602corresponding to an N degree to 60 degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The third coil UC enters the floating state due to the gate-off voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the first coil UC to the second coil VC. The third-2 coil control signal WN may be activated from the gate-off voltage to the gate-on voltage at a 60 degree time point at which the transition time T22or T32of 30−N/2 degrees elapsed from the zero crossing time point at which the voltage Vwc of the third coil WC in the floating state is decreased so that the BEMF becomes zero. In the third operation S503or S603corresponding to a 60 degree to 60+N degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the first coil UC to the second and third coils VC and WC. In a fourth operation S504or S604corresponding to a 60+N degree to 120 degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second coil VC enters the floating state due to the gate-off voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the first coil UC to the third coil WC. The second-1 coil control signal VP may be activated from the gate-off voltage to the gate-on voltage at a 120 degree time point, at which the transition time T22or T32of 30−N/2 degrees elapsed from the zero crossing time point at which the voltage Vvc of the second coil VC in the floating state is decreased, so that the BEMF becomes zero. In the fifth operation S505or S605corresponding to a 120 degree to 120+N degree rotation section of the sensorless BLDC motor200, the first power voltage VDD is applied to the first coil UC due to the gate-on voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the first and second coils UC and VC to the third coil WC. In the sixth operation S506or S606corresponding to a 120+N degree to 180 degree rotation section of the sensorless BLDC motor200, the first coil UC enters the floating state due to the gate-off voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the second coil VC to the third coil WC. The first-2 coil control signal UN is activated from the gate-off voltage to the gate-on voltage at a 180 degree time point, at which the transition time T22or T32of 30−N/2 degrees elapsed from the zero crossing time point at which the voltage Vuc of the first coil UC in the floating state is decreased, so that the BEMF becomes zero. In the seventh operation S507or S607corresponding to a 180 degree to 180+N degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The second power voltage VSS is applied to the third coil UC due to the gate-off voltage of the third-1 coil control signal WP and the gate-on voltage of the third-2 coil control signal WN. A current flows from the second coil VC to the first and third coils UC and WC. In the eighth operation S508or S608corresponding to a 180+N degree to 240 degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The third coil UC enters the floating state due to the gate-off voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the second coil VC to the first coil UC. The third-1 coil control signal WP may be activated from the gate-off voltage to the gate-on voltage at a 240 degree time point, at which the transition time T22or T32of 30−N/2 degrees elapsed from the zero crossing time point at which the voltage Vwc of the third coil WC in the floating state is increased, so that the BEMF becomes zero. In the ninth operation S509or S609corresponding to a 240 degree to 240+N degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The first power voltage VDD is applied to the second coil VC due to the gate-on voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the second and third coils VC and WC to the first coil UC. In the tenth operation S510or S610corresponding to a 240+N degree to 300 degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The second coil VC enters the floating state due to the gate-off voltage of the second-1 coil control signal VP and the gate-off voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the third coil WC to the first coil UC. The second-2 coil control signal VN may be activated from the gate-off voltage to the gate-on voltage at a 300 degree time point, at which the transition time T22or T32of 30−N/2 degrees elapsed from the zero crossing time pointe at which the voltage Vvc of the second coil VC in the floating state is decreased, so that the BEMF becomes zero. In the eleventh operation S511or S611corresponding to a 300 degree to 300+N degree rotation section of the sensorless BLDC motor200, the second power voltage VSS is applied to the first coil UC due to the gate-off voltage of the first-1 coil control signal UP and the gate-on voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the third coil WC to the first and second coils UC and VC. In the twelfth operation S512or S612corresponding to a 300+N degree to 360 degree rotation section of the sensorless BLDC motor200, the first coil UC enters the floating state due to the gate-off voltage of the first-1 coil control signal UP and the gate-off voltage of the first-2 coil control signal UN. The second power voltage VSS is applied to the second coil VC due to the gate-off voltage of the second-1 coil control signal VP and the gate-on voltage of the second-2 coil control signal VN. The first power voltage VDD is applied to the third coil UC due to the gate-on voltage of the third-1 coil control signal WP and the gate-off voltage of the third-2 coil control signal WN. A current flows from the third coil WC to the second coil VC. The first-1 coil control signal UP may be activated from the gate-off voltage to the gate-on voltage at a 360 degree (0 degree) time point, at which the transition time T22or T32of 30−N/2 degrees elapsed from the zero crossing time point at which the voltage Vuc of the first coil UC in the floating state is increased, so that the BEMF becomes zero. As described above, in the BLDC motor driver and the method of driving the motor according to one embodiment of the present disclosure, a turn-on time and a transition time which are suitable for driving characteristics and a driving speed of the BLDC motor driver can be set by repeating a process of testing a driving error of the BLDC motor while increasing the turn-on time step by step and decreasing the transition time step by step, and driving efficiency of the BLDC motor driver can be maximized by applying the set turn-on time and the set transition time. It will be understood by those skilled in the art that the disclosure may be performed in other concrete forms without changing the technological scope and essential features. Therefore, the above-described embodiments should be considered in a descriptive sense only and not for purposes of limitation. The scope of the present disclosure is defined not by the detailed description but by the appended claims, and encompasses all modifications and alterations derived from meanings, the scope, and equivalents of the appended claims. | 44,488 |
11863111 | DESCRIPTION OF PREFERRED EMBODIMENTS First, embodiments of the present disclosure will be listed and described. Also, at least some of the embodiments described below may be combined with each other as appropriate. According to an aspect, a power supply control device for controlling power supply to a motor installed in a vehicle includes: a switching element configured to turn on and off the power supply to the motor; a current detection circuit configured to detect a current flowing to the motor; and a control unit configured to determine whether or not the motor is in a locked state based on the current detected by the current detection circuit, and control turning on and off the power supply to the motor at a duty cycle that corresponds to the current detected by the current detection circuit, if it is determined that the motor is in the locked state. With this aspect, when the motor is in the locked state, the control unit controls the switching element so as to be turned on and off at the duty cycle that corresponds to the current flowing to the motor. Accordingly, it is possible to change the drive time of the motor according to the magnitude of the current of the motor, and thus it is possible to efficiently resolve the locked state of the motor. As a preferable configuration, the duty cycle is set such that the smaller the current detected by the current detection circuit is, the longer on-time of the switching element is. With this aspect, the control unit controls the switching element so as to be turned on and off such that the smaller the current flowing to the motor is, the longer the on-time of the switching element is. Accordingly, the drive time of the motor can be made longer the smaller the current flowing to the motor is, and thus it is possible to efficiently resolve the locked state of the motor. Also, by setting shorter drive time of the motor the larger the current flowing to the motor is, it is possible to suppress heat generated by the motor. As a preferable configuration, the duty cycle is set such that heat generated by the motor is suppressed while the power supply to the motor is turned on and off. With this aspect, even if the motor is in the locked state, the control unit controls the switching element so as to be turned on and off at a duty cycle such that the heat generated by the motor is suppressed. By suppressing the heat generated by the motor, it is possible to suppress a decrease in the current flowing to the motor. Because the torque of the motor is proportional to the current flowing to the motor, suppressing the heat generated by the motor can prevent a reduction in the torque of the motor. Accordingly, it is possible to turn the switching element on and off while maintaining the torque of the motor, and efficiently resolve the locked state of the motor. As a preferable configuration, the duty cycle is set such that a torque of the motor is not reduced while the power supply to the motor is turned on and off. With this aspect, even if the motor is in the locked state, the control unit controls the switching element so as to be turned on and off at a duty cycle such that the torque of the motor is not reduced. Accordingly, it is possible to turn the switching element on and off while maintaining the torque of the motor, and efficiently resolve the locked state of the motor. As a preferable configuration, the power supply control device further includes a table in which a plurality of currents having different magnitudes are respectively associated with duty cycles, wherein the duty cycles in the table are set such that if, when each of the plurality of currents flows to the motor, the power supply is controlled with an on-period that is longer than an on-period indicated by the duty cycle associated with the corresponding current, a temperature of the motor increases. With this aspect, if the motor is in the locked state, the control unit references the table, and controls turning on and off the switching element at the duty cycle associated with the current flowing to the motor. Duty cycles in the table are set to values with which the temperature of the motor does not increase due to on/off control, that is, values with which the torque of the motor is not reduced. Accordingly, it is possible to turn the switching element on and off while maintaining a state in which the torque of the motor is at the maximum, and efficiently resolve the locked state of the motor. As a preferable configuration, the duty cycles in the table are set such that the torque of the motor is not reduced and the on-time of the switching element is the longest, at a predetermined ambient temperature in which the motor is used. With this aspect, it is possible to turn the switching element on and off while maintaining a state in which the drive time of the motor is long and the torque of the motor is at the maximum, thus making it possible to efficiently resolve the locked state of the motor. As a preferable configuration, the current detection circuit includes a shunt resistor connected in series to the switching element. With this aspect, since the power supply control device is configured to detect a current using the shunt resistor, it is possible to accurately detect the magnitude of the current. Accordingly, the control unit can appropriately control turning on and off of the switching element, and it is possible to efficiently resolve the locked state of the motor. As a preferable configuration, the control unit controls the switching element so as to be turned off if the current detected by the current detection circuit is greater than or equal to a first threshold or is less than or equal to a second threshold, and the control unit determines that the motor is in the locked state if the current is within a predetermined range between the first threshold and the second threshold. With this aspect, if the current of the motor is greater than or equal to the first threshold, that is, if the motor is in the short-circuit failure state, the control unit controls the switching element so as to be turned off, and interrupts the power supply path. Also, if the current of the motor is less than or equal to the second threshold, that is, if the motor is in the open-circuit failure state, the control unit controls the switching element so as to be turned off, and interrupts the power supply path. Furthermore, if the current of the motor is in the predetermined range between the first threshold and the second threshold, the control unit determines that the motor is in the locked state. Accordingly, the control unit can determine whether the motor is in a state in which it is possible to resolve the locked state, or in a dangerous failure state such as that of a short-circuit failure or an open-circuit failure, and the control unit can control the switching element so as to be turned on and off upon confirming that the motor is in a safe state. As a preferable configuration, the control unit controls the power supply to the motor that drives a windshield wiper provided in the vehicle. With this aspect, by performing on/off control on the switching element, it is possible to efficiently resolve the locked state of the windshield wiper. According to an aspect, a power supply control method for controlling power supply to a motor installed in a vehicle includes the steps of detecting a current flowing to the motor; determining whether or not the motor is in a locked state based on the detected current; and if it is determined that the motor is in the locked state, controlling turning on and off the power supply to the motor at a duty cycle that corresponds to the detected current. With this aspect, it is possible to change the drive time of the motor while suppressing the heat generated by the motor, and efficiently resolve the locked state of the motor. The following will describe specific examples of the power supply control device and the power supply control method according to embodiments of the present disclosure with reference to the drawings. Note that the present disclosure is not limited to the examples but is defined by the claims, and all modifications within the meaning and scope equivalent to the claims are intended to be included. The following will specifically describe the present disclosure with reference to the drawings showing an embodiment thereof. FIG.1is a circuit block diagram illustrating an example of a configuration of the power supply control device according to an embodiment. The power supply control device of the embodiment is a device that controls power supply from an unshown in-vehicle battery to a motor7for driving a windshield wiper7a. The power supply control device includes a control unit1, a drive circuit2, a switching element3, a shunt resistor4, a current detection circuit5, and an overcurrent detection circuit6. The switching element3is an N-channel power MOSFET, for example. The drain of the switching element3is connected to the plus terminal of the in-vehicle battery, and the source of the switching element3is connected to the plus terminal of the motor7. The minus terminal of the in-vehicle battery and the minus terminal of the motor7are grounded. The drive circuit2is connected to the gate of the switching element3. The drive circuit2drives and turns on the switching element3by stepping up the voltage of the gate of the switching element3relative to the ground potential. The current detection circuit5is a circuit that detects a current flowing through the shunt resistor4, that is, a current flowing to the motor7, by detecting voltages at both ends of the shunt resistor4, and outputs the voltage that corresponds to the detected current to the control unit1. An example of the current detection circuit5is described. The current detection circuit5includes a differential amplifier51, and constitutes an inverting amplifier circuit. The non-inverting input terminal of the differential amplifier51is connected to the minus terminal of the shunt resistor4. The inverting input terminal of the differential amplifier51is connected to the plus terminal of the shunt resistor4via an electric resistor52. The output terminal of the differential amplifier51is grounded via an electric resistor53. The output terminal of the differential amplifier51is also connected to the inverting input terminal, and realizes a negative feedback. The overcurrent detection circuit6is a circuit that performs control such that the switching element3is turned off if an overcurrent flows to the motor7. An example of the overcurrent detection circuit6is described. The overcurrent detection circuit6includes a differential amplifier61. The inverting input terminal of the differential amplifier61is connected to a reference voltage source62, which outputs a reference voltage for use in determining whether or not an overcurrent is flowing to the motor7. The output terminal of the current detection circuit5is connected to the non-inverting input terminal of the differential amplifier61. The output terminal of the differential amplifier61is connected to the gate of an interruption switching element63. The source of the interruption switching element63is grounded. The drain of the interruption switching element63is connected to one end portion of a resistor64, and the other end portion of the resistor64is connected to the cathode of a diode65. The anode of the diode65is connected to the gate of the switching element3. The differential amplifier61compares a voltage output from the current detection circuit5with the reference voltage. That is to say, the differential amplifier61compares a current flowing to the motor7with a predetermined current that corresponds to the reference voltage. If the voltage output from the current detection circuit5is greater than the reference voltage, the differential amplifier61turns the interruption switching element63on. That is to say, if an overcurrent is flowing to the motor7, the interruption switching element63is turned on. When the interruption switching element63is turned on, the switching element3is turned off, and the power supply path is interrupted. The control unit1stores a first threshold, a second threshold, and a third threshold for use when the state of the motor7is determined based on a current flowing to the motor7(seeFIG.2). Also, the control unit1includes a table11for use in executing switching control based on the current flowing to the motor7when the motor7is in a locked state. FIG.2is a graph illustrating relationships between the states and currents of the motor7. The horizontal axis indicates time, and the vertical axis indicates the current flowing to the motor7. The first threshold is a threshold for use in determining whether or not the motor7or a circuit involved in power supply has a short circuit failure. If, as indicated by a curve A1, the current flowing to the motor7is greater than or equal to the first threshold after a predetermined period of time has elapsed from when the switching element3was turned on, the control unit1determines that a short circuit failure has occurred. The second threshold is a threshold for use in determining whether or not the motor7or the circuit involved in power supply has an open circuit failure. If, as indicated by a curve A4, the current flowing to the motor7is less than or equal to the second threshold after a predetermined period of time has elapsed from when the switching element3was turned on, the control unit1determines that an open circuit failure has occurred. The second threshold is less than the first threshold. The third threshold is a threshold for use in determining whether or not the motor7is in the locked state. The third threshold is greater than the second threshold, and is less than the first threshold. If, as indicated by a curve A2, the current flowing to the motor7is greater than or equal to the third threshold and is less than the first threshold after a predetermined period of time has elapsed from when the switching element3was turned on, the control unit1determines that the motor7is in the locked state. If, as indicated by a curve A3, the current flowing to the motor7is greater than the second threshold and is less than the third threshold after a predetermined period of time has elapsed from when the switching element3was turned on, the control unit1determines that the motor7is in the normal state. FIG.3is a conceptual diagram illustrating an example of a configuration of the table11. The table11indicates the correspondence between currents flowing to the motor7that is in the locked state, and duty cycles of the switching element3. To efficiently resolve the locked state of the motor7, the switching element3needs to be turned on and off at an appropriate duty cycle that corresponds to the current. The duty cycle is set at least such that the smaller the current flowing to the motor7is, the longer the on-time of the switching element3is. It is preferable that the duty cycle be set in a range from 20% to 80% inclusive. For example, the duty cycle is set to 50[%] when the current is small, and is set to 20[%] when the current is large. Preferably, the duty cycle is set such that the torque is not reduced due to heat generated by the motor7while the power supply to the motor7is turned on and off. Preferable duty cycles will be described in detail later. The period in which the switching element3is turned on/off is from 0.5 to 3 seconds inclusive. In the present embodiment, a description is given assuming that the period in which the switching element3is turned on/off is 1 second. When the period is 1 second and the duty cycle is 50[%], the control unit1performs control such that the switching element3is on for 500 milliseconds, and is then off for 500 milliseconds. FIG.4is a flowchart illustrating a processing procedure of the power supply control, andFIG.5is a timing chart showing a power supply control method. InFIG.5, the horizontal axis indicates time, and the vertical axis indicates the current flowing to the motor7. In a state in which the windshield wiper7ais activated and a current flows to the motor7, the control unit1executes the following processing. The control unit1detects a current flowing to the motor7using the current detection circuit5(step S11). Then, the control unit1determines whether or not a short-circuit failure has occurred, based on the detected current (step S12). Specifically, the control unit1determines whether or not the current detected in step S11is greater than or equal to the first threshold. If it is determined that no short-circuit failure has occurred (No in step S12), the control unit1determines whether or not an open-circuit failure has occurred, based on the detected current (step S13). Specifically, the control unit1determines whether or not the current detected in step S11is less than or equal to the second threshold. If it is determined that a short-circuit failure has occurred (Yes in step S12), or it is determined that an open-circuit failure has occurred (Yes in step S13), the control unit1controls the switching element3so as to be turned off, and interrupts the power supply path (step S14), thereby ending the processing. If it is determined that no open-circuit failure has occurred (No in step S13), the control unit1stands by for a predetermined time period t0 (step S15). The predetermined time period t0 is, for example, about 10 milliseconds. The predetermined time period t0 is a time period that is longer than or equal to a transition time period until the value of the current returns to the value that corresponds to the state of the motor7after the switching element3has been turned on. When the predetermined time period t0 has elapsed, the control unit1determines whether or not the motor7is in the locked state, based on the current detected by the current detection circuit5(step S16). Specifically, the control unit1determines whether or not the current flowing to the motor7is greater than or equal to the third threshold, and is less than the first threshold. The current for use in determining whether or not the motor7is in the locked state needs only to be detected for a short time period ta. Specifically, it is sufficient that the current detection circuit5detects a current only once. The time period to is as long as one clock of the control unit1, e.g., 5 milliseconds. If it is determined that the motor7is not in the locked state (No in step S16), the control unit1controls the switching element3so as to continue to be on (step S17), and ends the processing. If it is determined that the motor7is in the locked state (Yes in step S16), the control unit1detects a current using the current detection circuit5a multiple number of times during a time period tb (step S18), and calculates the average of the currents flowing to the motor7(step S19). The time period tb is as long as four clocks of the control unit1, e.g., 20 milliseconds. Then, the control unit1references the table11, and determines the duty cycle that corresponds to the average current (step S20). Then, the control unit1controls the switching element3so as to be turned on and off at the determined duty cycle (step S21), and returns to the processing in step S11. Specifically, as shown inFIG.5, assuming that the duty cycle D=on-time/off-time [%], the control unit1controls the switching element3so as to be on for tc*D/100 seconds, then controls the switching element3so as to be off for tc*(100−D)/100, and controls the switching element3to be on again. FIGS.6A and6Bare diagrams illustrating an on/off control method based on an average current in the locked state.FIG.6Ashows the on/off state of the switching element3when the average current of the motor7is small, andFIG.6Bshows the on/off state of the switching element3when the average current of the motor7is large. As shown inFIGS.6A and6B, the control unit1controls the switching element3so as to be turned on and off at a larger duty cycle the smaller the average current of the motor7is. Accordingly, when the current flowing to the motor7is small, the drive time of the motor7can be prolonged, and it is thus possible to efficiently resolve the locked state of the motor7. Also, when the current flowing to the motor7is large, the drive time of the motor7can be shortened, thereby suppressing the heat generated by the motor7. FIGS.7A and7Bare graphs of simulation results illustrating the duty cycle at which it is possible to maintain the maximum torque. The horizontal axis indicates time, the vertical axis on the left side indicates the current, and the vertical axis on the right side indicates the temperature of the motor7.FIG.7Ashows a state in which the duty cycle is excessively large, andFIG.7Bshows a state in which the duty cycle is appropriate. The simulation results shown inFIGS.7A and7Bwere obtained by calculating the resistance of a motor coil, the length of the motor coil, the surface area of the coil, the mass of the coil, and the thermal capacity of the motor based on the average of currents flowing to the motor7, combining the heat generation amount and the heat discharge amount, and obtaining the increase in the temperature of the motor7and the decrease in the flowing current. The conditions of the simulation were as follows: the temperature of the motor7in the initial state was set to 25° C., and the average current was set to 25 [A]. A circumflex denotes exponentiation. T denotes a temperature. Applied voltage: Va(=12)[V] Copper resistance: ρ(=1.5475+0.0068725T)×10{circumflex over ( )}(·8)[Ω·m] Coil wire diameter: a(=1)[mm] Copper density: ρcu(=8.94)[g/cm{circumflex over ( )}3] Specific heat: A(=0.379)[J/g·K] Coefficient of heat transfer: λ(=10)[W/m{circumflex over ( )}2·K] Boltzmann constant: K(=5.67×10{circumflex over ( )}(˜8))[W/(m{circumflex over ( )}2K{circumflex over ( )}4)] Enamel emissivity: E(=0.37) As is clear fromFIGS.7A and7B, a long time period in which the motor7is on is not always preferable. In the graphs ofFIGS.7A and7B, thick lines indicate a change in current flowing to the motor7when the switching element3is kept on, and thick dotted lines indicate the temperature of the motor coil. When a current flows to the motor7, the temperature increases and the current decreases. Because the torque of the motor7is proportional to the current, the torque of the motor7is reduced. As shown inFIG.7A, if the on-time is long, the temperature of the motor7will increase. When the temperature of the motor7increases, the current flowing to the motor7decreases, resulting in a reduced torque.FIG.7Ashows this state. As shown inFIG.7B, if the duty cycle is appropriate, an increase in the temperature of the motor7can be suppressed even when the on/off control is continued, and it is thus possible to keep the torque of the motor7high. Preferably, the table11stores duty cycles at which an increase in the temperature of the motor7can be suppressed, or duty cycles at which the torque of the motor7is not reduced, as shown in the graph ofFIG.7B. More preferably, the duty cycles stored in the table11are set to values such that if power supply is controlled with an on-period that is longer than the on-periods indicated by the duty cycles corresponding to the currents stored in the table11, the temperature of the motor7increases as shown inFIG.7A. That is to say, the duty cycles are preferably set to values with which the on-time of the motor is the longest without reducing the torque of the motor7at a predetermined ambient temperature at which the motor7is used. In other words, it can be said that it is preferable to register duty cycles in association with a plurality of current values, the duty cycles being set such that if a predetermined ambient temperature is suitably set, and a time period longer than the on-time indicated by the duty cycle associated with each of the current values registered in the table11is set, the temperature of the motor7increases and the torque of the motor7decreases. Note that the longest on-time is not a theoretical value, and thus there is no exact boundary between the on-time in which the temperature of the motor7does not increase, and the temperature at which the temperature of the motor7starts to increase, and tolerance should be permitted to some extent due to the use environment of the motor7and other reasons. With the power supply control device according to the embodiment having the above-described configurations, it is possible to efficiently resolve the locked state of the motor7. Also, longer drive time of the motor7can be set the smaller the current flowing to the motor7is, and it is thus possible to efficiently resolve the locked state of the motor7. Also, by setting shorter drive time of the motor7the larger the current flowing to the motor7is, it is possible to suppress heat generated by the motor7. Furthermore, it is possible to turn the switching element3on and off while maintaining a state in which the torque of the motor7is at the maximum, and efficiently resolve the locked state of the motor7. Moreover, since the power supply control device is configured to detect a current using the shunt resistor4, it is possible to accurately detect a current compared to a configuration in which a current is detected based on the voltages at both ends of the switching element3. Accordingly, the control unit1can appropriately control turning on and off of the switching element3, and it is possible to efficiently resolve the locked state of the motor7. Moreover, in the present embodiment, especially, it is possible to resolve the locked state of the windshield wiper7aprovided in the vehicle. | 25,864 |
11863112 | DESCRIPTION OF EMBODIMENTS FIG.1shows a wind turbine100(WTG) comprising a tower101and a rotor102with at least one rotor blade103, such as three blades. The rotor is connected to a nacelle104which is mounted on top of the tower101and being adapted to drive a generator situated inside the nacelle via a drive train. The rotor102is rotatable by action of the wind. The wind induced rotational energy of the rotor blades103is transferred via a shaft to the generator. Thus, the wind turbine100is capable of converting kinetic energy of the wind into mechanical energy by means of the rotor blades and, subsequently, into electric power by means of the generator. The generator is connected with a power converter which comprises a generator side converter and a line side converter. The generator side converter converts the generator AC power into DC power and the line side converter converts the DC power into an AC power for injection into the utility grid. FIG.2Ashows an example of a power system200of a wind turbine100according to an embodiment. The power system comprises a generator or power source201and a power converter202. The power converter202comprises a machine side converter203, a line side converter204. The power converter202may further comprise a DC-link205and a resistor207connected with a controllable switch206. The resistor and switch forms a power dissipation device, also known as a chopper209, for dissipating active power. The DC-link205comprises one or more DC-link capacitors which are charged by the DC output current from the generator side converter203and which supplies DC power to the line side converter204. The output AC current from the line side converter204may be supplied via output inductors206and possibly via a wind turbine transformer208to the power line220. In this example, the output AC current is a 3-phase current output. Harmonic filter capacitors216may be arranged between the conductors of the output, which together with the inductors206, forms a harmonic filter which converts the square wave voltage signals from the line side converter204to voltage sinusoidal signals. Since the power system200also applies to other power generating units199configured with a full scale power converter202, the examples and embodiments of the present invention applies equally to other power generating units such as renewable power generating units, e.g. solar power units or photovoltaic power generating units. That is, the generator or power source201may be embodied by solar power sources such as photovoltaic power sources, wind turbine generators or other power sources or generators. It follows that the power generating unit199which comprises the power system200may be a wind turbine, a solar power plant or unit or other power units such as renewable power generating units. The power line220may be a medium voltage power bus which receives power from other wind turbines100. The power line220may be connected with a high voltage network, e.g. via further transformers. Thus, the power line220and one or more power systems200of corresponding wind turbines constitutes a wind power plant or park arranged to supply power to a utility grid for distribution of electrical power. The power converter202may be full-scale converter configured according to different principles including forced-commutated and line-commutated converters. The power system200is principally illustrated and therefore does not explicitly reveal that the system may be a three phase system. However, principles of the described embodiments apply both to single and multi-phase systems. The line side converter204uses some variant of pulse width modulation (PWM) for converting the DC power into AC power. The control system250is used for controlling the modulation of the line side converter204and for controlling the active power P and the reactive power Q generated by the line side converter204. FIG.2Ashows that the grid voltage Ugrid, here the voltage at the low voltage LV side of the transformer208, can be measured. The grid voltage Ugrid can be used for determining a virtual synchronous machine angle θVSM (as described elsewhere) and for controlling the power output of the converter, based on determining the active power Pgrid from grid voltage Ugrid and grid current Igrid. The reactive power Qgrid may similarly be determined from Ugrid and Igrid. Alternatively, the grid voltage Ugrid may be measured on the high voltage HV side of the transformer and corrected based on the turns ratio of the transformer, or the internal voltage magnitude reference Vqref is used instead of the measured voltage Ugrid. Thus, in an alternative, internal voltage magnitude reference such as Vqref, Vdqref or Vαβref may be used for determining Pgrid and consequently the synchronous machine angle θVSM. Thus, the grid current Igrid supplied to the grid can also be measured. FIG.2Bshows an example of control components260arranged for controlling the generation of active power Pgrid and reactive power Qgrid supplied to the grid at the power output270from the wind turbine100or power generating unit199. That is, the control components260may be arranged for controlling the output active power Pgrid and the output voltage magnitude at the low voltage side LV, alternatively for controlling the output active power Pgrid and the output reactive power Qgrid at the low voltage side LV. The control components260may form part of the control system250. Alternatively, the control components260receive control signals from the control system250. References for the active and reactive power may be received from a power plant controller, PPC, or a grid operator, TSO, or determined from active and reactive power references, e.g. from the grid operator. The active power, Pgrid, is controlled via the virtual synchronous machine angle θVSM. In short, the synchronous machine angle acceleration (the double-time derivative of θVSM) indicates a difference between a power reference Pref for a desired power output of the wind turbine and a grid power Pgrid supplied by the wind turbine to a power grid. Examples for determining the synchronous machine angle θVSM is given elsewhere. The synchronous machine angle θVSM may be used to transform the signals from the rotating DQ frame into a non-rotating frame such as the αβ or abc frame, or vice-versa. Based on the synchronous machine angle θVSM and voltage magnitude reference Vqref, control signals for the desired active power and reactive power are determined. Thus, the synchronous machine angle θVSM may be defined in a rotating DQ frame defined by the angular position θVSM. Based on the synchronous machine angle θVSM, control signals, i.e. the angle of the modulation voltage signals for the pulse-width-modulator PWM,265are determined and transformed into a non-rotating frame such as the αβ or abc frame. The modulation voltage reference signal controls the active and reactive power Pgrid and Qgrid. The frame conversion unit266transforms the control signal from the DQ frame into the op or abc frame and determines the sinusoidal voltage references for the PWM265. The frame converted output signals from the frame conversion unit266are converted by the pulse-width-modulator PWM,265into a modulation signal for the grid side converter204in order to generate the desired active power and reactive power and/or voltage magnitude. The reactive power Qgrid to be generated by the line side converter204can be controlled based on a voltage magnitude reference Vqref. The voltage magnitude reference Vqref may be defined in the DQ frame which rotates with the rotational speed ωVSM of the virtual synchronous machine, i.e. in steady state condition the fundamental frequency such as 50 Hz of the AC grid voltage. The voltage magnitude reference Vqref, or a modification thereof as described in the following, may be converted from the DQ frame to the op or abc frame and outputted from the frame conversion unit266as a control signal to the pulse-width-modulator PWM,265which determines the modulation signal for the grid side converter204. Due to the low thermal capacity of power converters in contrast to synchronous generators, overcurrent should be limited sufficiently fast to prevent damages of the switch semiconductors of the line side converter204. Embodiments of the invention, proposes a current limiting method implemented in a virtual synchronous generator which is activated during abnormal grid conditions and overcurrent situations. FIG.3A-3B,FIG.4andFIG.5show examples for implementing the virtual synchronous generator with the current limiting method. The idea of the current limiting method is to insert a virtual impedance Zvir between the output of the line side converter204and the grid or the grid connection such as the transformer208. The virtual impedance Zvir is principally illustrated inFIG.2A. The virtual impedance Zvir may be determined for one or more phases. Here the virtual impedance Zvir is determined for commonly for all phases, i.e. as the same virtual impedance. The virtual impedance could also be individually determined for each phase. By increasing the resistive and/or the reactive value of the virtual impedance Zvir, the output current drawn from the output of the line side converter can be reduced. As illustrated in connection withFIG.4andFIG.5, the power value Pvir of the virtual resistance part Rvir of the virtual impedance Zvir, i.e. the virtual power dissipation, is used in the swing equation of the virtual synchronous generator implementation to reduce acceleration of the virtual synchronous generator during overcurrent situations and thereby the output current of the line side converter204. An overcurrent may be defined as an output current of the line side converter which is greater than an overcurrent threshold value Imax. The overcurrent threshold value Imax may be a predetermined value, a data-sheet value of the power converter202, a value which is given as a factor greater than one multiplied with a nominal maximum output current of the power converter202. FIG.3AandFIG.3Bshow examples of control systems391for determining the synchronous machine angle θVSM of the virtual synchronous generator. The synchronous machine angle θVSM is determined based on a virtual synchronous machine control concept which aims at generating a power response which corresponds to the power response from a real synchronous generator, including the inertia of the synchronous generator. The power error ΔP is determined as the difference Pref-Pd-Pvsm, where Pref is a power reference for the desired active power output of the wind turbine, Pd is a damping power determined according to the virtual synchronous model301, and Pvsm is a virtual grid power. The virtual grid power Pvsm is given as the sum of the measured grid power Pgrid and the virtual power Pvir of the virtual resistor Rvir. Under steady state conditions, the value of the power error ΔP is zero. In response a change in the grid power Pgrid, e.g. due to an decrease in the grid voltage Ugrid and a corresponding increase in the grid current Igrid, the power error ΔP becomes non-zero, which causes the angle θVSM to increase or decrease to reduce the power error ΔP. For example, in an overcurrent situation where Pgrid decreases, e.g. due to a grid short-circuit, the power error value ΔP becomes positive and synchronous machine speed ωVSM will increase. Thus, in response to fluctuations in e.g. the grid power Pgrid, the synthetic inertial response value becomes non-zero, which causes the virtual machine to either accelerate or decelerate to reach a new equilibrium condition. The new equilibrium is reached when the virtual grid power Pvsm is again following Pref. The virtual synchronous machine control concept is utilized on the line side converter204using a swing equation to calculate θVSM. FIG.3Ashows an example of an implementation of the virtual synchronous model301. The virtual synchronous model301includes a closed loop where the virtual synchronous machine rotational speed ωVSM is determined based on a combination a feedback of a damping power Pd, a power reference Pref for the desired active power output of the wind turbine, the active grid power Pgrid supplied by the wind turbine to the grid via the power line220, the virtual resistor power Pvir and an inertial integration model311. The inertial integration model311is implemented as 1/(2Hs) where H is the inertia time constant and 1/s is the integration in s-domain. Accordingly, the combination of powers Pref−Pd−Pgrid−Pvir=ΔP is used as input for the inertial integration model311. Since the derivative of the synchronous machine rotational speed ωVSM is proportional to the deviation between the power reference Pref and the virtual grid power Pvsm, the integration of the difference ΔP gives the synchronous machine rotational speed ωVSM. The grid power Pgrid can be determined based on the measured grid voltage Ugrid—or internal voltage references such as the magnitude reference Vqref, or transformations thereof such as Vαβref or Vdqref—and the measured grid current Igrid. The damping power Pd is determined as the difference between the rotational speed of the grid ωL and the synchronous machine rotational speed ωVSM multiplied with the damping factor Dp. The rotational speed of the grid ωL, i.e. the grid frequency is determined from the measured grid voltage Ugrid. The synchronous machine angle θVSM is determined based on an integration of the synchronous machine rotational speed ωVSM according to ωr/s, where ωr is the rated synchronous generator speed. FIG.3Bshows an alternative virtual synchronous model301which is not based on a measured grid voltage Ugrid, but instead the rotational speed of the grid ωL is determined based on a high-pass filtering of the determined synchronous machine rotational speed ωVSM, i.e. by determining the rotational speed of the grid ωL as the output of the high-pass filter313which is arranged to filter the inputted synchronous machine rotational speed ωVSM. Thus, the alternative virtual synchronous model301is not based on a measured grid voltage Ugrid, but the damping part, e.g. the damping power Pd, is determined based on a high-pass filtering313of the synchronous machine rotational speed ωVSM. In general, the virtual synchronous model301determines the angle of the virtual machine θVSM based on the combination of powers Pref, Pd, Pgrid, Pvir, the inertial integration model311, e.g. implemented as 1/(2Hs) and a feedback of the damping power Pd determined based on ωVSM and an integration of ωVSM. In other words, the synchronous machine rotational speed ωVSM and the synchronous machine angle θVSM are determined so that they are indicative of an integrated deviation between a power reference Pref for a desired power output of the wind turbine and the virtual grid power Pvsm. The control systems391can be implemented based on power values Pref, Pd, Pgrid, Pvir but may equivalently be implemented based on corresponding torque values Tref, Td, Tgrid, Tvir based on the relationship where power equals torque times rotation frequency, e.g. the synchronous machine rotational speed ωVSM. FIGS.4and5show examples of determining the virtual impedance Zvir and feeding the virtual resistor power Pvir into the virtual synchronous model301. A current magnitude Im1or Im2is determined based on the grid current Igrid. The current magnitude Im1or Im2is determined so that the current magnitude represents the instantaneous current magnitude of the current. In this context, instantaneous may be the actual current amplitude per sample, or an average obtained over two or more samples, such as 10 samples. The current magnitude Im1, Im2may be determined based on the current components Ia, Ib, Ic in the reference frame, i.e. the current amplitudes of the available phases, based on the current components Id and Iq (FIG.4) and/or based on the current components Iα and Iβ (FIG.5). The current components Id, Iq are obtained by transforming the current signals Ia, Ib, Ic of the three-phase current signal Igrid from the measurement frame abc to the DQ frame which rotates with the virtual synchronous angular frequency. The abc/DQ transformation is performed based on the virtual synchronous machine angle θVSM. The abc/DQ transformation is the projection of the current quantities Ia, Ib, Ic—or other quantities such as voltages—onto the two-axis frame DQ which rotates according to the angle θVSM. The current components Iα, Iβ are obtained by transforming the current signals Ia, Ib, Ic of the three-phase current signal Igrid from the measurement frame abc to the op frame which is stationary. The abc/o transformation is the projection of the current quantities Ia, Ib, Ic—or other quantities such as voltages—onto the two-axis stationary frame op. The current magnitude Im1may be determined as the square root of the squares of the current components Iα, Iβ or the squares of the current components Id, Iq. Thus, the current components Iα, Iβ or Id, Iq may be determined for each sample, possibly as an average may be determined over a few samples. Alternatively, the current magnitude Im2may be determined based on the phase current magnitudes Ima, Imb, Imc, e.g. determined as the current amplitude or peak current value within a period, of the current signals. Further, the current magnitude Im2may be determined as the maximum of the determined current magnitudes Ima, Imb, Imc, or in other ways dependent on the current magnitudes. If a fast response is needed, i.e. if a fast compensation via the virtual impedance Zvir is needed, the first alternative of the current magnitude Im1may be preferred, due to the faster computation of Im1compared with Im2. If a better performance in unbalanced situations is needed, the second alternative of the current magnitude Im2may be preferred. In unbalanced cases, the magnitude of one phase, for example Ima, may be much larger than other phases (Imb and Imc), consequently Im2would be larger than Im1. So, If Im2is selected to limit the current, a suitable larger virtual impedance may be inserted. The virtual impedance value Zvir is determined by initially determining the virtual resistance part Rvir of the virtual impedance Zvir. The determination of the reactive part Xvir of the virtual impedance Zvir is determined based on the initially determined virtual resistance Rvir, but the determination of Xvir is optional since the method also works without determining the reactive part Xvir. Advantageously, Xvir may provide more independent active and reactive power control loops during overcurrent conditions. The virtual resistance Rvir is determined based on the current difference or current error Ierr, i.e. the difference between the current magnitude Im1or Im2and the overcurrent threshold value Imax. Consequently, the virtual reactance Xvir and the virtual impedance Zvir is also determined based on the current difference Ierr. When the current magnitude Im1, or Im2is lower than Imax, the current difference Ierr becomes negative and the virtual impedance Zvir is reduced (if not already equal to zero) until it is saturated to the zero value. Accordingly, in normal operation Zvir and Pvir equals zero values. In this example, the virtual resistance Rvir is determined using a controller. For example, the controller may be an integral controller (I-controller), proportional controller (P-controller) or a proportional-integral controller (PI-controller). Accordingly, the control algorithm of the controller may include a proportional part which determines a value which is proportional with the current difference Ierr and/or an integral part which determines a value which is proportional with a time integral of the current difference Ierr. Accordingly, the virtual resistance Rvir is obtained based the values determined by the P-controller and/or the I-controller, or the PI-controller. The integral part of the controller comprises an integral gain Ki which may be determined dependent on the current difference Ierr, such as the sign of the current difference Ierr. For example, in order to achieve a fast control action when the current difference Ierr is positive, i.e. when the current magnitude Im1, Im2is greater than Imax, the integral gain Ki may be set to a predetermined relative larger gain Ki1. For negative current differences Ierr, the integral gain Ki may be set to a predetermined relative smaller gain Ki2to achieve a more stable response during unbalanced situations, where Im1/Im2may oscillate. Furthermore, the integral gain Ki could also depend on the magnitude of the positive or negative difference Ierr, i.e. by having predetermined gain factors Ki for different signs and magnitudes of the current error Ierr. Therefore, the integral gain Ki may be determined from two or more predetermined gains. In the above example with two gains Ki1and Ki2, where Ki1is greater than Ki2. Thus in an example, when Ierr>0, Ki=Ki1is selected to limit the grid current fast. When Ierr<0, Ki=Ki2is selected to have a stable response for the unbalanced cases. The determined virtual resistance value Rvir may be limited by the limiter according to a lower minimum limit Rmin and/or an upper maximum limit Rmax to avoid determination of negative resistance values and/or very large values. The output from the limiter is referred to as Rvir_lim. As illustrated inFIG.4, the virtual reactance value Xvir may be determined based on the determined virtual resistance Rvir or the limited virtual resistance Rvir_lim. For example, the virtual reactance value Xvir may be determined dependent on a predetermined factor “Kx/r” which determines a ratio between the virtual reactance value Xvir and the virtual resistance value Rvir. The Kx/r factor may be an empirically determined factor, e.g. determined based required damping and grid conditions. For example, the Kx/r factor could be a value in the range from zero to ten. The voltage ΔVdq over the virtual impedance Zvir is determined as shown inFIG.4as ΔVd=Rvir_lim×Id−Xvir×Iq and ΔVq=Rvir_lim×Iq+Xvir×Id. For that purpose, the grid current Igrid is transformed into Idq in the DQ frame and the impedance voltage is determined in the DQ frame. If Xvir is not determined, Xvir is simply set to zero in the equations. In the example inFIG.5, the voltage ΔVαp over the virtual impedance Zvir is determined based on the grid current Igrid transformed into Iαβ in the αβ frame. The voltage may be determined as ΔVα=Rvir_lim×Iα+(Xvir/ωVSM)×d(Iα)/dt and ΔVβ=Rvir_lim×Iβ+(Xvir/ωVSM)×d(Iβ)/dt (not shown inFIG.5). Alternatively, as shown inFIG.5, in order to avoid the derivative part, only the virtual resistance Rvir or the limited resistance Rvir_lim is determined and the voltage over the virtual resistance is determined as Iαβ×Rvir_lim or Iαβ×Rvir. As another alternative the voltage ΔVabc over the virtual impedance Zvir may be determined by multiplying the current components Ia, Ib, Ic with the resistance and/or reactance components of the virtual impedance Zvir. In general as illustrated with examples inFIG.4andFIG.5, the virtual voltage ΔVαβ, ΔVdq over the virtual impedance Zvir is determined based on the virtual impedance value Zvir and the grid current Igrid, or a transformation thereof. InFIG.4, the virtual voltage ΔVdq is combined, here subtracted, with the voltage reference Vdgref to generate the modified voltage reference Vdq_m. The virtual power Pvir of the virtual resistor Rvir is determined as Pvir=1.5×Im1{circumflex over ( )}2×Rvir or 1.5×Im2{circumflex over ( )}2×Rvir. The virtual grid power Pvsm is determined as the sum of the virtual power Pvir and the grid power Pgrid=Ugrid×Igrid. Alternatively, the virtual grid power Pvsm may be determined as Pvsm=1.5×(Vαref×Iα+Vβref×Iβ) or Pvsm=1.5(Vdref×Id+Vqref×Iq), where Vαref and Vβref are the components of Vαβref, and where Vdref and Vqref are the components of Vdqref. Accordingly, the virtual grid power Pvsm may alternatively be determined based on the voltage magnitude reference Vqref and the grid current Igrid, or transformations thereof according to the equations. Thus, in general the virtual grid power Pvsm may be determined based on the virtual resistance Rvir and the grid current Igrid or transformations therefrom, or based on the voltage references (Vdqref, Vαref) and the grid current Igrid or transformations therefrom. As described in connection withFIGS.3A and3B, based on the deviation ΔP between the power reference Pref for a desired power output of the power generating unit and the virtual grid power Pvsm combined with a damping power (Pd), the virtual synchronous machine rotational speed ωVSM and/or the synchronous machine angle θVSM is determined by the virtual synchronous model301, where the derivative of the synchronous machine rotational speed (ωVSM) is indicative of the power deviation ΔP. The voltage reference Vabc for controlling the line side converter204to generate the desired reactive power Qgrid is determined by the frame conversion unit266based on the virtual synchronous machine rotational speed or angle ωVSM, θVSM, and a combination of the virtual voltage ΔVαβ, ΔVdq and the voltage magnitude reference Vdgref, here the difference Vdqref−ΔVdq. Here it is noted that Vdref may be zero, so that Vdqref equals Vαref. The voltage magnitude reference Vαref is provided to achieve a desired reactive power Qgrid to be generated by the line side converter204. Thus, the voltage reference Vdgref is the voltage reference in the DQ frame generated based on the voltage magnitude reference Vqref. The virtual synchronous machine angle θVSM is used by the frame conversion unit266to perform the transformation of the voltage signal Vdq_m from the DQ frame into the voltage signal Vabc in the abc frame. The voltage signal Vabc is provided as a control signal to the pulse-width-modulator PWM,265in order to generate the active power P based on θVSM as described in connection withFIG.2A. The virtual synchronous machine angle θVSM is determined as described in connection withFIG.3A-Band is used for determining the control signal Vabc via the frame conversion unit266. Accordingly, if a positive current error Ierr is generated, the virtual power Pvir determined from the virtual resistance Rvir is added to the grid power Pgrid and results in an increased virtual grid power Pvsm. The increased virtual grid power Pvsm results in reducing the error ΔP and consequently decreasing of θvsm. This causes that the angle difference between Vabc and Ugrid is decreased, which causes a decrease in the grid current Igrid supplied by the line side converter204to the grid. The filter compensated voltage reference Vdq_f is transformed from the DQ frame to the measurement frame abc similarly to the control signal derived from the synchronous machine angle θVSM FIG.5shows an alternative example for determining and implementing the virtual impedance Zvir and feeding the virtual resistor power Pvir into the virtual synchronous model301. Here the example, only shows determination of the virtual resistance Rvir. In this example, the grid current Igrid, i.e. the current components Ia, Ib, Ic, is transformed from the measurement frame abc to Iαβ in the stationary αβ frame. The abc/αβ transformation transforms the three vector values in the measurement frame abc to two vector values in the αβ frame. The voltage magnitude reference Vαβref is the op transformed version of Vdqref, where Vdqref is the voltage reference Vdgref in the DQ frame generated based on the voltage magnitude reference Vqref (not explicitly shown inFIG.5). The virtual synchronous machine angle θVSM is used by the frame conversion unit266for transforming the Vdgref from the DQ frame into Vαp in the αβ frame. The voltage control signal Vαβ is transformed by transform element621to the control signal Vabc in the measurement frame abc and is provided as a control signal to the pulse-width-modulator PWM. The determination of the virtual grid power Pvsm inFIG.5is equivalent to the determination of Pvsm inFIG.4. The virtual voltages ΔVαβ are combined, here subtracted, with the voltage reference Vαβref to generate the modified voltage reference Vαβ_m. The modified voltage reference Vαβ_m is transformed from the αβ frame to voltage control signal Vabc in the measurement frame abc by the αβ/abc transform element621. Accordingly, the response of the circuit inFIG.5due positive current error Ierr is equivalent to the control response of the circuit inFIG.4. FIG.6illustrates simulation results of an embodiment with the current limiting method. At t1the grid voltage magnitude Ugrid decreases due to a grid fault and consequently the grid power Pgrid decreases. The current magnitude Im, which could be Im1or Im2, increases due to the grid fault and causes an increasing virtual resistance Rvir and consequently an increasing virtual grid power Pvsm due to increasing virtual power Pvir. At t1the virtual synchronous speed ωVSM starts decreasing after an initial minor increase and, therefore, limits the increase of the virtual synchronous machine angle θVSM. As both the magnitude and angle difference between Vabc and Ugrid are decreased, the large values of the grid current Igrid or the grid current magnitude is prevented. Although the present invention has been described in connection with the specified embodiments, it should not be construed as being in any way limited to the presented examples. The scope of the present invention is to be interpreted in the light of the accompanying claim set. In the context of the claims, the terms “comprising” or “comprises” do not exclude other possible elements or steps. Also, the mentioning of references such as “a” or “an” etc. should not be construed as excluding a plurality. The use of reference signs in the claims with respect to elements indicated in the figures shall also not be construed as limiting the scope of the invention. Furthermore, individual features mentioned in different claims, may possibly be advantageously combined, and the mentioning of these features in different claims does not exclude that a combination of features is not possible and advantageous. | 30,374 |
11863113 | DETAILED DESCRIPTION FIGS.1and2show schematic views of an application employing the presently described invention. The application may comprise a number of electrical and mechanical components. The application comprises a variable frequency drive which may be a power converter, a motor, a gearbox and a rod pump as a reciprocating load driven by the motor. The motor, the gearbox and the rod pump are coupled mechanically such that power can be transmitted from the motor to the rod pump. The variable frequency drive provides an output signal to the motor for driving said motor. The application has the reciprocating load motor speed control integrated into the variable frequency drive. The reciprocating load motor speed control receives an input variable indicating the motor current or load, which is also indicative of the motor torque at any given time. A second input variables received by the drive is a user defined speed set-point which may be input by a user of the application. Based on the two input variables a reciprocating load motor speed control algorithm is provided at the drive for calculating the final speed reference, used by the variable frequency drive to output signals to the motor. As both, the user defined speed set-point and the actual motor current, or load, are considered as input variables by the drive, the drive may control the motor torque to be kept positive during a complete load cycle of the reciprocating load, thereby avoiding an input of regenerated energy from the load to the motor and the drive. The motor current indicative of the motor torque may be measured by at least one or exactly one current sensor, provided at or as part of the variable frequency drive. It is possible to carry out the present invention by only measuring current values. Accordingly, the variable frequency drive may only comprise current sensors or only one current sensor, and no other sensor equipment. FIG.3shows graphs indicating the performance of known state of the art applications, in which an electric motor is driving a reciprocating load. As indicated by the horizontal line in the central motor torque vs. time graph, negative torque may occur, wherein the reciprocating load effectively drives the motor, thereby creating regenerative power that may charge the DC-link of the variable frequency drive. In order to handle these reverse conditions, resistor braking is provided in applications known from the art. FIG.4shows graphs indicating the performance of an application employing the presently described method and/or variable frequency drive. The top speed reference vs. time graph shows that while the known solutions provide for a fixed speed reference, the present invention provides a load dependent and therefore varying speed reference. The load dependent speed reference varies as a function of the torque from the reciprocating load to the motor. The central motor torque vs. time graph shows that, according to the invention, no negative torque, i.e. torque from the reciprocating load to the motor is present anymore. Hence, the motor transmits net positive torque to the reciprocating load at all stages of its stroke, the corresponding curve is above the horizontal zero line. FIG.5shows graphs split into sections with focus on motor torque. The four sections Sector A, Sector B, Sector C, Sector D, are half stroke torque data of the reciprocating load to the motor. In Sector A and Sector C a decreasing curve of torque and energy to drive the motor is required. The motor is speeding up at the end of Sector A and Sector C while the motor torque is kept positive to avoid regenerative energy, which would normally require brake resistors, which again is wasted heat and energy. In Sector B and Sector D, an increasing curve of torque and energy to drive the motor is required. The motor is slowing down at the end of Sector B and Sector D to have a speed that when going into next sector, will not generate a negative torque which would normally require brake resistors and would be wasted heat and energy. | 4,082 |
11863114 | DETAILED DESCRIPTION The illustration in the drawings is in schematic form. It is noted that in different figures, elements similar or identical in structure and/or function are provided with the same reference signs or with reference signs, which differ only within the first digit. A description of an element not described in one embodiment may be taken from a description of this element with respect to another embodiment. FIG.1illustrates in a schematic form a wind turbine100according to an embodiment of the present invention as example of a power generation system, which provides electric energy to a utility grid101. The wind turbine comprises a hub103to which plural rotor blades105are connected. The hub is mechanically connected to a main shaft107whose rotation is transformed by a gear box108to a rotation of a secondary shaft109, wherein the gear box108may be optional. The main shaft107or the secondary shaft109drives a generator111which may be in particular a synchronous permanent magnet generator providing a power stream in the three phases or windings113,115and117to a converter119which comprises a generator side portion (AC-DC portion)121, a DC-link123and a grid side portion (DC-AC portion)125for transforming a variable AC power stream to a fixed frequency AC power stream which is provided in three phases or windings127,129,131to a wind turbine transformer133which transforms the output voltage to a higher voltage for transmission to the utility grid101. The converter119is controlled via a converter control signal135which is derived and supplied from an arrangement150for controlling a generator side converter portion according to an embodiment of the present invention, which receives at least one input signal137, such as one or more reference values or one or more quantities indicative of the operation of the generator111or any component of the wind turbine100. The generator inFIG.1comprises a single three-phase stator winding. Thereby, the winding113carries the stator current Ia, the winding115carries the stator current Iband the winding117carries the stator current Ic. The arrangement150is adapted to counteract torque and voltage harmonics (for example a harmonic corresponding to six times the electrical frequency of the generator111). The generator111, the converter119and the arrangement150together form a generator system according to an embodiment of the present invention. The arrangement150for controlling a generator side converter portion as included in the wind turbine power generation system100illustrated inFIG.1receives inputs signals137which may relate to stator voltage, stator current, may relate to a feedback signal, may relate to an operating point, as will be detailed with reference toFIGS.2to4. The control signal135in particular is a converter control signal for controlling the generator side converter portion121of the converter119. FIG.2schematically illustrates an arrangement250for controlling a generator side converter portion andFIGS.3and4illustrate respective embodiments350,450of the arrangement for controlling a generator side converter portion. The inputs to the arrangement250are collectively denoted with reference sign237. The arrangement250comprises a torque ripple controller (TRC)239which receives as inputs a feedback signal241and/or an optional signal243indicating the operating point. The torque ripple controller239derives therefrom a harmonic torque reference245(Th*). The arrangement250further comprises a harmonic voltage controller (HVC)247which receives as input a stator voltage indicating feedback signal249and derives therefrom a harmonic flux reference251(ψh*). The torque ripple controller239and the harmonic voltage controller247operate on a particular harmonic of a fundamental frequency. If more than one harmonic of the fundamental frequency are to be treated, for each harmonic, a respective torque ripple controller239and a respective harmonic voltage controller247may be provided. In case of several harmonics to be treated, all of the harmonic torque references245are added at an addition element254to a fundamental torque reference253(Te*) and an estimated generator torque255is subtracted to derive a torque error257. All of the harmonic flux references251are added at an addition element254to a fundamental flux reference259(ψs*) and an estimated generator flux261is subtracted to derive a flux error263. The desired flux change dψ and desired torque change dTe are obtained as output of the hysteresis controllers268,269. A converter control signal265is derived based on the desired torque change dTe and the desired flux change dψ. Thereby, the converter control signal265represents a switching state signal defining a switching state of plural controllable switches of the generator side converter portion121(illustrated inFIG.1). The stator voltage indicating feedback signal249is derived from a measured DC link voltage Vdc and the switching states265using a calculation module267. The calculated stator voltage may be bandpass-filtered to eliminate harmonics different from the harmonic the harmonic voltage controller247is working on. According to the embodiment250illustrated inFIG.2, the torque error257and the flux error263are supplied to respective hysteresis controllers268,269whose outputs, the desired flux change dψ and desired torque change dTe, are supplied to a switching table271that outputs the switching state265. Furthermore, the switching table271receives the stator flux position Θψs (273) and derives the switching states265also based on the stator flux position273. The estimated generator torque255and the estimated generator flux261are derived by a torque and flux estimator275based on the stator voltage vs, in particular reference stator voltage, and stator current is. The fundamental torque reference253and the fundamental flux reference259are derived by a fundamental torque/flux reference calculation module277based on one or several of the following quantities: stator voltage ‘vs’, stator current ‘is’ and DC link voltage Vdc. As can be taken fromFIG.2, the harmonic torque reference245(Th*) is derived by the torque ripple controller239further based on an operating point243of the generator111. The arrangements350,450illustrated inFIGS.3and4comprise modules similar or identical to modules of the arrangement250illustrated inFIG.2which modules are labelled with reference signs differing only in the first digit. However, the arrangement350for controlling a generator side converter portion according to an embodiment of the present invention does not comprise the hysteresis controllers268,269and the switching table271but instead comprises a predictive torque controller377which receives the torque error357and the flux error363and derives therefrom the converter control signal365. Instead of the predictive torque control377or the hysteresis controllers268,269and the switching table271, the arrangement450illustrated inFIG.4comprises a harmonic torque controller479and a fundamental torque controller481to which the torque error457is supplied and the outputs are added together using the addition element480to derive a first voltage reference483. Furthermore, the arrangement450comprises a fundamental flux controller485and a harmonic flux controller487to which the flux error463is supplied and whose outputs are added by an addition element480to arrive at a second voltage reference489. The first voltage reference483and the second voltage reference489are supplied to a dq-αβ-transformation module491that outputs a total voltage reference492uab*. The total voltage reference492is supplied to a space vector modulation module493which derives therefrom the switching state465. As can be taken fromFIG.4, the stator voltage indicating feedback signal449is derived by the module467based on the total voltage reference. The harmonic torque controller479inFIG.4and the torque ripple controller239,339,439inFIGS.2,3,4are distinct controllers, but they may have the same structure (such as any ofFIGS.5-7) in case suitable feedback signal are available. However, in case feedback signal is not available for torque ripple controller, it can be a look-up table having the operating point (OP) as input. The torque controller481and the flux controller485inFIG.4) may be PI controllers for fundamental torque and flux. The converter control signal inFIG.4is derived based on the inputs to the transformation module dq/αβ by Inverse Park transformation plus voltage modulator. High performance harmonic control in an electric drive is important as this is a requirement for permanent magnet generators for several reasons: (1) meet noise standards; (2) prevent excitation of structural modes and accelerated fatigue; (3) optimized DC link voltage usage and system efficiency. The torque ripple controller (TRC) illustrated inFIGS.2,3and4generates the harmonic torque reference (Th*) which is added to the fundamental torque reference (Te*), modifying the reference torque to contain the errors in the estimated torque at the harmonic frequencies of concern. The harmonic voltage controller (HVC) illustrated inFIGS.2,3and4generates a harmonic flux reference to minimize the respective harmonic voltage. Direct torque control merits are expected to be retained for the TRC and HVC, namely fast dynamics due to a decoupled control of torque and flux/voltage. Moreover, the simplicity of implementation of the LUT-based approach is obvious fromFIG.2, since the inner flux and torque controller (hysteresis controllers268,269) provide high bandwidth control without the need for additional controllers in parallel. The arrangement350illustrated inFIG.3is another attractive solution including the predictive torque control (PTC) method which is endowed with high bandwidth flux and torque control like the LUT-based DTC (inner loop composed of blocks for model-based prediction and cost function minimization). InFIG.4, the TRC and HVC are integrated into the direct torque control with space vector modulation. As opposed to the embodiments illustrated inFIGS.2and3, inFIG.4, voltage references are readily available at the modulator input, namely at the signal492, but the lower bandwidth of torque and flux controllers may require additional parallel controllers (HTC and HFC inFIG.4) for achieving high performance reference tracking of harmonic content and zero steady-state errors. Some of the approaches may be resonant controllers and PI controllers in multiple reference frames. The calculation of Vrms for the illustrated control method is as follows: Vrms=sqrt(Vd{circumflex over ( )}+Vq{circumflex over ( )}2) or Vrms=sqrt(Vα{circumflex over ( )}2+Vβ{circumflex over ( )}2). Vdq are readily available (sum of torque/flux controllers' outputs) in DTC-SVM, whereas for DTC-LUT voltages are reconstructed using switching states (Sabc) and dc-link voltage (Vdc): {ucα=23Vdc(Sa-Sb+Sc2)ucβ=13Vde(Sb-Sc) The harmonic of interest is extracted from Vrms by an adaptive BPF at the selected harmonic. It is emphasized that embodiments focus on the outer harmonic control loops shown inFIGS.2,3,4as TRC and HVC, which modify the reference torque and flux required by any DTC based control method. Accordingly, the different examples presented inFIGS.2,3,4(LUT, predictive, SVM) are given for the sake of completeness. However, fast inner control loops may be achieved in other manners. FIGS.5,6and7illustrate harmonic controllers as implementations of the control blocks HVC, TRC, HTC, HFC illustrated inFIGS.2,3and4. In particular, the torque ripple controllers239,339,439illustrated inFIGS.2,3,4may receive a harmonic torque demand signal240,340,440, respectively, representing the desired torque at the harmonic at consideration. This harmonic torque demand may for example be zero. Similarly, the harmonic voltage controller may receive as an input a harmonic voltage demand signal246,346,446inFIGS.2,3,4, respectively. This signal may represent a harmonic voltage demand, and this may be zero according to different applications. In all the implementations of the controller illustrated inFIGS.5,6and7, a harmonic error504is calculated from the demand harmonic value502and a feedback signal503. In the implementation illustrated inFIG.5, the error504is 90° shifted by a 90° phase-shifter506and the output signal is supplied to a frame transformation module508. The frame transformation module508transforms the error and the 90° shifted error according to a coordinate system rotating with the considered harmonic. The output is supplied to PI regulators510which derive an output signal such that the error reduces to zero. Downstream the regulators510, a further frame transformation module is arranged which back-transforms to obtain harmonic reference545. InFIG.6, the harmonic error604is supplied to trigonometric function612,614and the output is again supplied to PI regulators610whose output is multiplied with another trigonometric functions616,618and added together to obtain the harmonic reference645. In the implementation illustrated inFIG.7, the harmonic error704is supplied to a resonant regulator714, to obtain the harmonic reference745. Control blocks HVC, TRC, HTC, HFC inFIGS.2,3,4can all be implemented similarly and are next called harmonic controllers. Different options of harmonic controllers are shown inFIG.5,6,7. In all options, the harmonic error is calculated, and a close loop regulation is made. Thus, the regulators are used to control harmonic error to 0. The harmonic regulator based on vector control principle is shown inFIGS.5,6, enabling the implementation of simple PI controllers.FIG.7shows the harmonic regulator using resonant regulator with a typical transfer function as below, where f1 is the resonant frequency and ξ1 is the damping of the controller. The block diagrams shown inFIGS.5,6,7assume that adaptive band-pass filters (BPF) are applied to reference/feedback signals whenever needed and therefore only the harmonic order of interest is given to the controller input. Alternatively, BPFs may be implemented in the harmonic error (Vn_error) inFIGS.5,6and7. RR(s)=kp+ki4πf1ξ1Ss2+4πf1ξ1s+(2πf1)2 Application to PM machines may allow to reduced noise and vibration and also to optimize hardware utilization. Noise and vibration reduction by minimization of torque ripple may be straight forward to understand. On the other hand, increasing hardware usage by control of harmonic stator voltage/flux may not be so obvious. The latter is explained by the fact that in the presence of non-negligible voltage harmonics, a reduced (average) flux reference may need to be set to avoid converter over-modulation, resulting in entering the flux-weakening region at a lower speed level and operating with increased phase currents. The introduction of HVC enables the following possibilities: (1) increase of average stator flux reference; or (2) reduction of dc-link voltage. In other words, HVC allows to extend the operating range on MTPA (Maximum Torque Per Ampere), narrowing the flux weakening range, and therefore optimizing drive efficiency. One of the main advantages of the proposed control structure and respective feedback signals may be that the dependence on the accuracy of flux and torque estimators is eliminated with regards to harmonic control. It is to be noted that the input of TRC may not be limited to an accelerometer, and other signals such as microphone and different sensors may be options. Moreover, the closed-loop harmonic controller TRC may be replaced by a simple LUT with operating point (OP) information as input such as speed and torque, providing a cheap feed-forward solution. FIG.8illustrates simulation results of the modified space vector modulator based direct torque control as is illustrated inFIG.4. The control of the PM generator is accomplished by implementing a DC link voltage controller for calculation of the fundamental torque reference (Te*) and a flux weakening control for calculation of the fundamental stator flux reference (also called fundamental voltage controller, since it targets to keep the generator voltage below a given limit). Regarding harmonic control, measured or inferred torque ripple is used as feedback signal together with the controller inFIG.5for the torque ripple controller and the harmonic voltage control uses voltage ripple in Vrms as input and a controller structure as inFIG.5. The abscissas816denote the time and the ordinates818the strength of the signal. The curve820illustrates the feedback torque, the curve822illustrates the estimated torque, the curve824depicts the modulation index and the curve826indicates the estimated flux. The torque ripple controller is enabled at the time point t=3 s, i.e., at the time point828. Thus, it results in an effective reduction of the measured torque ripple (remaining oscillations are at a non-controlled lower harmonic frequency), whereas a sixth harmonic is imposed in the estimated/referenced torque. The harmonic voltage control is enabled at the time point830, i.e., at t=5 s, reducing the voltage ripple by imposing a sixth harmonic in the reference flux. Accordingly, the converter operates further from the flux weakening and/or overmodulation regions. According to an embodiment of the present invention, the fundamental references for the current/torque/flux are calculated by controllers or look-up tables, some examples are speed, power, torque, flux, voltage controllers and maximum torque per ampere methods. Such controllers provide Te* and ψs* which are usually DC signals during steady state operation. The torque ripple controller targets to control torque ripple by using a suitable sensor as feedback signal and generating a reference harmonic torque Th* which is a sinusoidal signal varying at a given frequency or a combination of sinusoidal signals with different frequencies. It may be composed of a variety of controllers (PI, search algorithms, etc.) and/or LUTs. The harmonic voltage controller targets to control voltage ripple by using the modulus of the reference voltage (Vrms) as feedback and generating a reference harmonic flux ψh* which is a sinusoidal signal varying at given frequencies or a combination of sinusoidal signals with different frequencies. It may be composed of a variety of controllers. The torque/flux controller may comprise PI controllers with given bandwidth, aiming to track DC content of torque and flux references. The harmonic torque/flux controller (HTC and HFC) may be controllers implemented by PI controllers in the harmonic reference frame, proportional-resonant controllers or any other suitable methods. Park transformation may transform between stationary frame (αβ) and synchronous rotating frame (dq) and vice versa. Similar techniques may be applied for transformations between synchronous rotating frame and harmonic reference frame. Voltage modulator uses the reference voltages in the stationary frame for generating PWM (pulse width modulation) pattern. The PWM signals are used to control power electronic switches such as IGBT in the generator side converter portion. Hysteresis (bang-bang controller) may be considered as controllers used to define LUT entries in DTC-LUT, 2 and/or 3 level controllers are typically used. Controllers output determine if torque/flux is to increase, decrease or remain unchanged. Switching table determines the optimized voltage according to the desired action stated at the LUT entries by the hysteresis controllers. Stator flux angle is also an entry of the LUT, defining a given number of sectors. Torque and flux estimators may employ machine models together with current measurements to estimate electromagnetic torque and stator flux. Reference voltages are typically used instead of measured voltages. A variety of models and observer structures may be used for estimation purposes. Embodiments of the present invention may provide a control method for incorporating harmonic control capability in direct torque control drives. The harmonic control may include torque ripple control and voltage ripple control. A control method for improving harmonic control performance in electrical drives may be provided. The harmonic control may have little dependency on the accuracy of torque ripple and flux linkage ripple estimation and thus may have a good robustness. A control method may be well suited for the control of the permanent magnet generators for wind turbines which may provide an alternative to the more commonly employed vector control methods. Embodiments of the present invention may reduce noise and vibration and increasing voltage control range and drive efficiency. A control method suitable for implementation in the controller of a frequency converter may be provided. Although the present invention has been disclosed in the form of preferred embodiments and variations thereon, it will be understood that numerous additional modifications and variations could be made thereto without departing from the scope of the invention. For the sake of clarity, it is to be understood that the use of “a” or “an” throughout this application does not exclude a plurality, and “comprising” does not exclude other steps or elements. | 21,343 |
11863115 | All drawings are schematic and not necessarily to scale. Parts numbered in one figure but appearing un-numbered in another figure are the same parts unless explicitly noted otherwise herein. DETAILED DESCRIPTION The features and benefits of the invention are illustrated and described herein by reference to example (“exemplary”) embodiments. This description of example embodiments is intended to be read in connection with the accompanying drawings, which are to be considered part of the entire written description. Accordingly, the disclosure expressly should not be limited to such embodiments illustrating some possible non-limiting combination of features that may exist alone or in other combinations of features. In the description of embodiments disclosed herein, any reference to direction or orientation is merely intended for convenience of description and is not intended in any way to limit the scope of the present invention. Relative terms such as “lower,” “upper,” “horizontal,” “vertical,”, “above,” “below,” “up,” “down,” “top” and “bottom” as well as derivative thereof (e.g., “horizontally,” “downwardly,” “upwardly,” etc.) should be construed to refer to the orientation as then described or as shown in the drawing under discussion. These relative terms are for convenience of description only and do not require that the apparatus be constructed or operated in a particular orientation. Terms such as “attached,” “affixed,” “connected,” “coupled,” and “interconnected,” refer to a relationship wherein structures are secured or attached to one another either directly or indirectly through intervening structures, as well as both movable or rigid attachments or relationships, unless expressly described otherwise. FIGS.1-13depict a solar panel mounting system and its components according to the present disclosure.FIG.1is a perspective view of a solar panel mounting system used for mounting a plurality of panels to support structure. The mounting system includes a plurality of horizontally oriented and axially elongated longitudinal main support runners or rails20. Multiple parallel rows of spaced apart rails may be provided for mounting an array of solar panels30thereto in side-by-side series fashion. Multiple parallel horizontal rows of solar panels may be installed in some applications using the main support rails in typical manner for form the array. The main support rails20each define a respective longitudinal axis LA and corresponding axial directions for convenience of reference. Rails20are attached to a support structure on which the solar panels30are to be mounted, such as a building roof21. The roof may be a pitched roof such as an A-frame inclined to horizontal as illustrated, or a flat roof. In other possible embodiments, the main support rails20may be attached to a frame mounted at ground level from grade, or other types of grade-level or elevated support structures used in a solar panel array. Accordingly, the present invention is not limited in its applicability to roof mounted installations alone and may be used with equal benefit in various different solar panel mounting situations. FIG.2is an perspective end view of one non-limiting embodiment of a main support rail20usable for mounting solar panels. Any suitable configuration of a support rail may be used for this purpose. The support rail20may be coupled to the roof21via suitable brackets, such as L-shaped brackets22and fasteners28(e.g. threaded bolt-nut sets) which are commonly used for this purpose in the art. The support rails20each include a lower base portion20-1for engaging the roof or spacer structure between the rail and roof to raise the rails above the roof tiles or shingles for proper drainage, and an upper panel mounting portion20-2configured for coupling the solar panels30thereto. The upper panel mounting portions of main support rails20each comprise a longitudinally-extending internal fastening channel23formed immediately adjacent to the top of the rail. Channels23are each defined by a horizontal closed or solid bottom wall24, partially open top wall25, and pair of solid lateral sidewalls27extending between the top and bottom walls. The top wall25defines an upwardly open and longitudinally-extending mounting slot26configured for inserting the head of a panel mounting fastener therethrough slot into the channel23. The opposing ends29of the channels23may be open in one embodiment or closed such as via a permanent or removable end cap. The channel23may have a rectangular configuration in transverse cross section as shown, or other suitable cross-sectional shape. In one embodiment, the main support rails20may be formed of an extruded metallic material, such as aluminum or other metallic materials. Other configurations of main support rails may be used and does not limit the invention. FIGS.3and4show assembled and exploded views respectively of a rectilinear solar panel30having a unique frame construction according to the present disclosure.FIGS.5-8show additional details of the peripheral frame, mounting hardware, and main support rails. Referring toFIGS.1-8, solar panel30may have a generally planar/flat square or more commonly a rectangular configuration. Solar panel30includes an interior photovoltaic cell array31comprising a plurality of photovoltaic solar cells31-1. This forms the active solar energy absorbing portion or region of the panel which converts solar energy into electricity. The photovoltaic cell array31may have any type of composite construction which may include encapsulation of the solar cells between protective layers such as transparent glass or plastic on the top and bottom of the cells, a sheet of top cover glass for protection from the elements or debris, a backing sheet on the bottom of the panel for additional support, and other elements. Numerous variations of the cell array construction are possible and well known in the art without further undue elaboration. The photovoltaic cell array construction is not limiting of the present invention. The solar panels30each include a peripheral frame32which defines a rectilinear perimeter of the assembly panel. Peripheral frame32is formed by a plurality of orthogonally arranged and perimetrically extending frame members. The frame members include a first pair of opposing right and left frame members40,41and a second pair of opposing top and bottom frame members42,43arranged perpendicularly to the first pair. Each of the frame members are linearly elongated in shape. Adjacent frame members are oriented perpendicularly to each other at the corners of the solar panel. The three frame members41-43may be considered first basic or primary frame elements that may be formed by structural C-channels in transverse cross sectional shape in one embodiment. Each C-channel defines an inwardly and laterally open cavity44for receiving and retaining the peripheral sides of the photovoltaic cell array therein. Cavity44is formed by orthogonally intersecting horizontal top flange or wall44-1, horizontal opposing bottom flange or wall44-2, and vertical sidewall44-3of each C-channel extending therebetween (orientations assuming the panel is in a horizontal or angled position for convenience of reference). The outward facing surface or side47of the sidewalls44-3of frame members41-43opposite cavity44may be generally plain and flat without any cavity (see, e.g.FIG.7), and oriented vertically in one embodiment (assuming the panel is in a horizontal or angled position for convenience of reference). Bottom wall44-2may be laterally elongated and wider than top wall44-1to provide extra support for the solar array. Top wall44-1is preferably laterally narrower by contrast to maximize the surface area available for absorbing solar radiation. The remaining fourth frame member40(right side) is configured differently for coupling an adjacent solar panel30thereto. In one embodiment, frame member40may be a deformable locking frame member formed of a structural H-channel in transverse cross sectional shape defining both an inwardly and laterally open inner cavity45and an opposite outwardly and laterally open outer cavity46(see, e.g.FIG.7). Cavity45receives the remaining peripheral side edge of the photovoltaic cell array therein. The outwardly open cavity46is configured for receiving a peripheral side edge of one of the C-shaped side frame members of an adjacent solar panel30at least partially therein (e.g. left frame member41inFIG.3in this example). Cavities45and46are formed by orthogonally intersecting horizontal top flange or wall45-1, horizontal opposing bottom flange or wall45-2, and vertical wall or web45-3of each H-channel extending therebetween (orientations assuming the panel is in a horizontal or angled position for convenience of reference). Bottom wall45-2may be laterally elongated and wider than top wall45-1to provide extra support for the solar array. Top wall45-1is laterally narrower by contrast to maximize the surface area available for absorbing solar radiation. The portion of locking frame member40which defines the outwardly open cavity46forms the deformable and compressible “clamping portion” of the member which produces a clamping action onto an adjacent second solar panel when mounted to the first solar panel. Tightening the captive T-bolt sets50further described herein deforms and compresses the top and bottom walls45-1,45-2together against the second solar panel. This firmly clamps and locks the peripheral side edge of the second solar panel in the outwardly open cavity46which may be considered to define a “locking cavity.” It bears noting that the secure engagement between the clamping portion of locking frame member40and the second solar panel is important to provide a stable solar panel mounting system which can resist wind loadings imposed on the solar array. The wind loading and forces may be significant particularly if the solar array is to be mounted on the roof of a building and at an angle. In some preferred but non-limiting embodiments, the H-shaped deformable locking frame member40may have a stepped shape in transverse cross section. The opposing top and bottom walls45-1,45-2of locking frame member40may each comprise a step-shaped transition45-4between inner portions and outer portions of the walls. The inner and outer portions are defined with respect to the vertical web45-3of the frame member40(i.e. the outer portion is outboard of the sidewall and the inner portion is inboard of the sidewall). The vertical distance D1between the top and bottom walls45-1,45-2of the outer portion is greater than the vertical distance D2between the top and bottom walls45-1,45-2of the inner portion. The outwardly open outer cavity46of locking frame member40in this embodiment thus has a greater height (corresponding to D2) than the height of inwardly open inner cavity45(corresponding to D1). This serves two important functions. First, the vertically shorter inner cavity45needs to be narrower and preferably has the same relative height as the inward facing cavities44of the C-channel shaped frame members41-43for receiving the peripheral edges of photovoltaic cell array31therein. Conversely, the vertically taller out cavity46must be larger to receive the left frame member41as noted above which has a greater vertical height than the height/thickness of the photovoltaic cell array31. The step-shaped transitions45-4of locking frame member40therefore makes possible the different height requirements of the inner and outer cavities45,46. It bears noting that the photovoltaic cell array31may have a height/thickness substantially less than the inner cavity45to permit routing of the control/electrical cabling thereunder, as further described herein. In one embodiment, the frame members40-43may be formed of extruded aluminum or another suitable preferably metallic material. Non-metallic materials such as fiber reinforced plastic (FRP) may alternatively be used. Accordingly, the frame material does not limit the invention. All frame members may be mounted and assembled to the photovoltaic cell array31in the shop prior to shipment to the jobsite, thereby providing a complete solar panel for shipment and handling. Any suitable method may be used for securing the frame members together. In some non-limiting examples, the frame may be welded together or assembled via a suitable adhesive such as adhesive silicone (RTV) as some non-limiting examples. It bears noting that although the locking frame member40is described above for convenience of description only as being the right frame member of the solar panel frame, the locking frame member may become the left frame member if desired by simply rotating the panel 180 degrees. Whether the locking frame member40is located on the right or left side of the panel depends on the direction from which the solar panels will be coupled together (i.e. starting from left to right or right to left as shown inFIG.1) since the next adjacent panel in the array to be placed on the support rails20is slideably inserted partially into the locking frame member40of previously placed panel as described above. Preferably, the H-shaped locking frame member40is located on the side of the solar panel facing laterally outwards to receive and lock the C-shaped frame member of the next panel thereto. Accordingly, the use of the foregoing “left” and “right” terminology is therefore for convenience of description only and not limiting of the invention in any way. According to another advantageous aspect of the invention, the present peripheral frame includes a captive T-bolt assembly50for securing the solar panel30to the main support rails20and forming an interlock with adjacent panels during installation of the solar array.FIG.6shows an example of captive T-bolt assembly prior to mounting to the solar panel frame.FIGS.5,7, and13show the T-bolt assembly after pre-mounting to the peripheral frame preferably in the shop before shipment to the jobsite for ease of panel installation and to prevent having to manually handle the hardware. The pre-mounted captive bolt assembly advantageously provides accurate and pre-set spacing between the modules, thereby eliminating the need for panel-to-panel spacing adjustment and alignment in the field per previous installation systems. Alternatively, however, it is still possible to couple the T-bolt assembly50to the locking frame member40in the field if necessary or desired for some reason as an alternative. With continuing reference toFIG.6, T-bolt assembly50includes a T-bolt51comprising a T-shaped locking head52on one end of the shank56and an opposite threaded end53of the shank, and a threaded nut54rotatably coupled to the threaded end. The threads may be formed on only the end portion of the shank proximate to end53, or the entire shank may but need not be threaded in other embodiments as shown. Any suitable threaded conventional or lock nut may be used. The head52may be rectangular and elongated in shape including two opposite short ends or sides52-2and two opposite long sides52-1as illustrated. A pair of locking corners52-3of the head are formed at two diagonally opposite corners of the head52. The locking corners52-3may be angle chamfered in some embodiments as shown to facilitate both rotating the T-head once inside the fastening channel23of the main support rails20, and importantly to enhance and optimize both locking frictional engagement with the opposing longitudinally-extending lateral sidewalls27of the channel and electrical grounding of the locking frame member40. The angle chamfered locking corners52-3of T-bolt head52each define a substantially linear or straight surface or face lying in a locking plane Lp which is obliquely angled (e.g. about 30-60 degrees) to the adjacent short sides52-2and long sides52-1that meet at the corner. The T-bolt head51is dimensioned and angle of chamfered locking corners52-3are selected such that a substantially flat-to-flat interface is created between the locking corners and lateral sidewalls27of channel23for optimum frictional engagement. In some embodiments, the corners52-3may have a textured surface such as serrated (shown), knurled, or other to increase frictional engagement between the locking corners and sides of the frame channel23. In one embodiment, a serrated surface is preferred for positive electrical grounding between the solar panel30and the longitudinal main support rails20via the T-bolt assemblies50. Advantageously, this can eliminate or minimize any additional provisions necessary for grounding the panels to the rails. It bears noting that in contrast to typical square corner which optionally may be provided and used on T-bolt head52, the diagonally opposed pair of angle chamfered locking corners52-3allows the T-bolt to be rotated farther without interference from square corners which advantageously increases securement of the T-bolt in the fastening channel23of the main support rails20. The T-bolt assemblies50(i.e. T-bolt51) are each rotatable between an unlocked position shown inFIG.6B, and a locked position shown inFIG.6C. To accomplish this, the T-shaped head52of T-bolt51and each fastening channel23of main support rails20are cooperatively configured and dimensioned to allow the head of the T-bolt to be partially but preferably not fully rotated inside the channel to engage its lateral sidewalls27for locking the T-bolt in place. Similarly, the longitudinal mounting slot26in the main support rail20which opens downwards into channel23is configured and dimensioned in lateral width (i.e. measured transversely and perpendicularly to longitudinal axis LA) to allow insertion of the T-shaped head52of the T-bolt through the slot into the channel when the long sides of the T-shaped head are oriented parallel to the longitudinal axis LA (see, e.g.FIG.6B). When the T-shaped head of the T-bolt is rotated to the locked position (FIG.6C) so that the long sides of the head are oriented obliquely or perpendicularly to the longitudinal axis LA, the head which is longer than wide is preferably prevented from being vertically withdrawn through the slot26from the fastening channel23of the main support rail20by the top wall25segments on each side of slot26. This arrangement allows the T-bolt assembly50to be tightened for securing and locking the solar panels30to the main support rails20, as further described herein. In one embodiment, the fastening channel23of main support rail20may have a lateral width (i.e. measured transversely to longitudinal axis LA) which is less than the length of the T-bolt head52measured parallel along the long sides between the ends defining the short sides52-2of the head (i.e. length from short side to short side). When the T-bolt head is positioned in the channel23with the long sides oriented parallel to the longitudinal axis LA of the main support rail20, rotating the bolt51to the locked position will engage the sidewalls27of the channel23with the chamfered locking corners52-3and prevent further rotation when the head52is oriented obliquely to the longitudinal axis (e.g. between about 30 and 60 degrees in some cases, or about 45 degrees in one embodiment). This is the locked positon of the T-bolt, versus the unlocked position when the T-bolt head long sides are parallel to longitudinal axis LA and insertable through mounting slot26of the rail20. Engagement between the locking corners52-3of T-bolt head52and sidewalls27of channel23is substantially one of a flat-to-flat interface as shown inFIG.6C. In the unlocked position of the T-bolt assemblies50(i.e. T-bolt51), the locking planes Lp defined by the angle chamfered locking corners52-3of each T-bolt head52are oriented at an oblique angle A1to the longitudinal axis La of the main support rails20(see, e.g.FIG.6B). In the locked position of the T-bolts assemblies, the locking plane Lp by contrast are oriented parallel to the longitudinal axis LA of the rails20(see, e.g.FIG.6C). Advantageously, the “captive” T-bolt assembly50is pre-installed on the H-shaped locking frame member40of the peripheral frame32in one preferred embodiment. As previously noted, this prevents having to handle loose solar panel/module mounting hardware (e.g. bolts, clips, washers, and nuts used in prior panel systems) in the field, thereby expediting the solar panel mounting process and preventing the hardware from being dropped and falling beneath other panels or off the roof. To provide “captive” pre-mounted T-bolt assemblies50preferably mounted to the locking frame members40of the solar panel peripheral frame in the fabrication shop before shipping to the job site, mounting holes45-5(see, e.g.FIGS.5and8) are provided in the parallel top and bottom flanges or walls45-1,45-2of frame member40which receive the at least partially threaded shank of the T-bolt51therethrough. Preferably, the T-bolt is mounted through holes45-5which are located in the outer clamping portion of the flanges or walls45-1,45-2bounding the outward facing cavity46of each locking frame member40. To provide maximum insertion depth of the adjacent solar panel into the cavity46, the T-bolt assembly50may be mounted more proximally towards the central vertical web45-3of frame member40than the outer peripheral free edges of the flanges. In some installation approaches if not shop-mounted, the locking T-bolt assemblies may be coupled to the clamping portion of the locking frame member40on the ground before lifting each solar panel onto the roof. It bears noting that one skilled in the art will clearly understand that the term “captive” is used herein to connote that the locking T-bolt assemblies are captured by and extend directly through a portion of the locking frame members40via mounting holes45-5cited above. In prior solar panel mounting approaches previously described in the “Background,” bolt assemblies were placed between adjacent panels but did not penetrate the peripheral frame members of the panels, thereby resulting in the myriad of problems noted before. In the preferred installed position shown inFIGS.7and8, the T-shaped head52is located below the bottom flange or wall45-2of locking frame member40, the shank56extends upwards therefrom through the outward facing cavity46, and the threaded end53of the shank protrudes upwards beyond the top flange or wall45-1. The nut54is loosely threaded onto this upper exposed threaded end of the shank56until tightened later during the solar panel installation process. In a preferred embodiment, the solar panel30may included two spaced apart captive T-bolt assemblies50pre-installed on the peripheral frame of the panel since each panel is normally supported by two vertically spaced apart main support rails20. More or less numbers of T-bolt assemblies may be provided to meet the needs of a particular installation. In some embodiments, a nylon torque patch55may be permanently bonded onto a select upper portion of the threaded end53of the shank of the T-bolt51. The torque patch creates frictional resistance between the nut54and shank threads53, such that rotating the nut will cause the entire T-bolt to rotate concurrently therewith as long as the T-bolt head is free to rotate. This is somewhat analogous to resistance created by a nylon locknut, which in some alternative embodiments may be used in lieu of the torque patch. The nylon torque patch makes it convenient for rotating the head of the T-bolt from the unlocked to locked positon with one hand after head52is located within the rail fastening channel23described herein during the solar panel mounting process. Once the T-bolt head is positively engaged with the sidewalls of the rail fastening channel23in the locked position and cannot rotate freely any further, continued rotation of the bolt will now tighten the entire T-bolt assembly50causing the nut to travel downward on the threaded end53of the T-bolt. The torque patch also ensures the T-bolt head is properly longitudinally aligned with the mounting slot26of the main support rail20when being initially inserted into the fastening channel23. In addition, the torque patch55ensures that the nut54does not come loose from the T-bolt51during shipping and handling prior to installation of the solar panels. A non-limiting example of a suitable commercially available torque patch which may be used is Nylok® Blue Nylon Torq-Patch® Tuflok®. A method for installing a solar panel system with solar panels30to form a solar array will now be briefly described using the solar panel frame construction and mounting hardware according to the present disclosure.FIGS.8-11show sequential views in the process of coupling an adjacent second solar panel30to a first solar panel30, which are referenced below. A first solar panel30with pre-installed “captive” T-bolt assemblies50in the peripheral frame32is provided (FIG.8). The method includes mounting at least one main support rail20on a support structure such as a roof21or other support structure transversely and perpendicularly to the longitudinal axis A of the rail (see, e.g.FIGS.1and5). Preferably, at least two main support rails20are mounted and run horizontally/laterally in parallel spaced apart relationship. A starter channel, which may be a loose one of the locking frame members40with H-channel configuration, may first be placed transversely/perpendicularly across the pair of rails20and mounted thereto as shown inFIG.1. The starter locking frame member40may be removably coupled to the pair of main support rails20using the T-bolt assemblies50in the same manner to be described below for the locking frame members40which are part of the solar panel peripheral frame32. This process will therefore not be described here at present for the starter rail, recognizing that the mounting procedure for the starter locking frame member40is the same. The starter locking frame member40may have a length which exceeds the length of those on the peripheral frame, and in some embodiments may span transversely across more than a single pair of main support rails20as seen inFIG.1. After mounting the starter locking frame member40to the main support rails, the method continues and further includes placing the first solar panel30against and onto the pair of main support rails20. Preferably, the first solar panel is oriented so that the outward facing outer cavity46of H-shaped frame member40is facing horizontally/laterally. The method continues by inserting the elongated locking head52of each T-bolt51(which is exposed on the bottom of the peripheral frame32) vertically through the top mounting slot26in the main support rails20and into fastening channels23of the rails. This may occur concurrently during and as part of the prior placing step above. The elongated locking head52of each bolt is aligned with its long sides52-1oriented and aligned parallel to rail fastening channel23along the longitudinal axis LA of the rail20and lengthwise with the mounting slot26during the insertion step. Accordingly, the T-bolt assemblies50are in the “unlocked” position during the insertion step in which the locking planes Lp defined by the angle chamfered locking corners52-3of each T-bolt head52are oriented at an oblique angle A1to the longitudinal axis LA of the main support rails20(see, e.g.FIG.6B). The first solar panel30may be adjusted in lateral/horizontal position on the rails as needed to insert the lateral C-shaped frame member41opposite the locking frame member40on the panel into the awaiting outward facing outer cavity46of H-shaped frame member40of the starter locking frame member40which was previously mounted to the pair of main support rails20. The method further continues and includes rotating each T-bolt51until the long sides of the T-bolt head52are no longer parallel to the rail longitudinal axis or mounting slot26(e.g. obliquely or perpendicularly angled thereto) and in the T-bolt assemblies50are in “locked” position. The locking planes Lp by contrast are now oriented parallel to the longitudinal axis LA of the main support rails20(see, e.g.FIG.6C). This prevents the T-bolt head from being withdrawn upwards through the slot26associated with the fastener channel23of the rails20. In some embodiments, this step may be performed by simply rotating the nut54, which will concomitantly rotate the T-bolt when a friction patch55is added to the threads of the T-bolt as previously described herein (or alternatively a nylon lock nut). It bears noting that the nut need not be rotated to an extent which fully tightens the T-bolt assembly50. Accordingly, the T-bolt assembly50may still be in a somewhat slightly loosened condition at this point in the installation process and is slideable laterally/horizontally in the fastening channels23along the rails20with the solar panel30even while the T-bolt assembly is in the loosely locked position. This permits the lateral/horizontal adjustment of the first solar panel30on the rails20. It further bears particular noting that once the T-bolt head52has been inserted into the fastening channel23of the support rail20, the solar panel30advantageously will be automatically held temporarily in positon and prevented from sliding vertically down the roof relative to the support rails201even though the T-bolt assembly has not been fully tightened to any significant degree with the assemblies in the locked rotational position. This is particularly significant for an angled/sloped roof installation. With the T-bolt assemblies50in the loosely locked position, wind gusts which may work their way beneath the solar panel30will be prevented from lifting the panels off of the main support rails20adding to the safety of the installer. It bears noting that insertion of the T-bolt heads52into the rails even while the T-bolt assemblies50may be in the unlocked position of the T-bolt assemblies described above still prevents the solar panels30from sliding downwards across the rails20and roof. Either the locked or unlocked scenarios of the T-bolt assemblies advantageously therefore provides significant installation flexibility and advantages by eliminating the past problem of juggling with the mounting hardware until the bolt assembly and clamp/washer have been securely tightened against a pair of adjacent solar panels as previously described herein. A single installer may handle, place, and mount the solar panel in lieu of at least two installations as used in past panel mounting approaches described heretofore. In addition, the solar panel30may be slid horizontally along the rails to adjust the desired position on the rail or to make electrical connections. Once the solar panel is in the desired position on the pair of main support rails20, the T-bolts may then be rotated via turning the nuts54on exposed top of the first solar panel30which are tightened to a first degree sufficient to positively lock the T-bolt head in the fastening channel23of the rail20without preferably substantially deforming the clamping portion of the locking frame member40on the peripheral frame32. With the first solar panel30now in place on the main support rails, the method continues by aligning and inserting the peripheral side edge frame of a second solar panel30into the outward facing cavity46of H-shaped locking frame member40of the first solar panel (see, e.g.FIGS.9and10). The second solar panel30may be inserted until its peripheral side edge (e.g. defined by basic C-shaped frame member41opposite the H-shaped frame member40of the second panel) abuttingly engages the shank of the captive T-bolt assembly50in the first solar panel30. Once the second solar panel is fully in proper inserted position, the T-bolt assemblies50on the first solar panel30may be fully tightened to a second degree greater than the first degree by rotating the nut to the desired degree of tightness or torque value (e.g. inch-lbs.) beyond the point necessary to previously lock the T-bolts51to the main support rails20by using an electric driver drill or manual socket wrench as non-limiting examples. This tightening step deforms the clamping portion of locking frame member40on the first solar panel and displaces the top flange of frame member40above the outer cavity46downwards, thereby imparting a compressive clamping force F1acting on the frame member41of peripheral frame32of the second solar panel30(see, e.g.FIG.10). The second panel is now fully engaged and clamped between the top and bottom flanges45-1,45-2of the locking frame member41of the first solar panel30(see, e.g.FIG.11). It bears noting that the opposing top and bottom flanges of first solar panel locking frame member40act as compressible jaws which apply the compressive force to retain the second member, not a clamp/washer of the mounting hardware as in the past practices previously described herein. The presently disclosed T-bolt assemblies50therefore advantageously provide a more positive clamping action and more uniformly distributed clamping force F1to the second solar panel30. The process of fully tightening the T-bolt assembly50to lock the second solar panel30in position also concurrently and positively locks the peripheral frame of the first solar panel in horizontal/longitudinal position along the main support rails20. However, there is no compressive clamping force applied to the peripheral frame of the first solar panel by its H-shaped frame member40because the T-bolt assembly50only acts within the outward facing cavity46in the clamping portion of the locking frame member40outboard of the vertical web45-3, not the inward facing cavity45coupled to the photovoltaic cell array. During placement of the second solar panel30, the second solar panel may be temporarily secured to the main support rails20via its captive T-bolt assemblies50using the same methodology described above for the first solar panel. This allows adjustment of the second solar panel in position along the rails20as needed to insert the peripheral side edge of the second solar panel into the clamping portion of the first solar panel. A third solar panel30may then be coupled to the second solar panel in the same manner used to couple the second solar panel to the first solar panel, and so on for each success solar panel mounted in the array. It bears noting that the present T-bolt assembly50eliminates the need for the clamp or washer used in previous solar panel mounting system hardware s in which the mounting hardware is shipped loose and not captured or pre-installed on the panel frame as in the present invention. Advantageously, the present disclosure captive T-bolt assemblies50not only eliminate the loss of the panel mounting hardware at the installation site, but increases the speed with which the installation process can be completed in an efficient and convenient manner in the field. According to other aspects of the invention, a cable management system is provided which is configured to both minimize time required for making electrical/control cable connections and eliminate or minimize post-installation damage to control/electrical cables and electrical equipment caused by rodents. Referring toFIGS.5,12, and13-16each solar panel30may include a covered rear cable compartment60with detachable removable cover64for enclosing electrical equipment such as electrical junction boxes67and routing power/electrical cables61from and through one lateral peripheral frame member40to the opposite lateral peripheral frame member41. Because a series of solar panels are typically mounted in horizontal rows in series fashion, and the H-shaped locking frame member40is intended to be on one vertical side of the solar panel, this arrangement allows adjacent panels in the row to be electrically connected together. The side peripheral lateral frame members (including the H-channel locking frame member40and opposing C-channel frame member41) may each include a laterally open cable aperture or hole62communicating with the enclosed cable compartment60and fitted with a rubber/elastomeric sealing grommet62for extending the cables61outward beyond the frame to make the inter-panel electrical cable connections. The grommets62are preferably configured to provide a tight leak-resistant and snug fit to the cable to prevent ingress of water and rodents. The two opposing ends of the cable61may be terminated with suitable male/female electrical couplers such as end plugs63for forming the electrical connection to and between the adjacent solar panels. A simplistic standard pronged electric male end plug63is shown for convenience as an example, recognizing that any type of mating male/female electrical end plugs or couplings including locking/latching type plug sets may be used to electrically couple the cables of adjacent solar panels together. Advantageously, routing the cables61internally within the panel30through the covered cable compartment60and grommeted openings eliminates the past need for field routing cables externally to and underneath the solar panels and using a multitude of cable clips which is a time-consuming process. The internal cable routing further denies access to rodents eager to chew on exposed cables. In one embodiment, as shown inFIGS.12-16, cable compartment cover64may have a longitudinally elongated body oriented parallel to longitudinal axis LA of solar panel30. The cover body may have an obtusely angled configuration in one embodiment as shown and including opposing longitudinal mounting edges65,66. One edge65is configured for detachable mounting to the bottom flange or wall of C-channel frame member43while the opposite edge65is configuration for mounting to the underside of the photovoltaic cell array31. Any suitable type of edge constructions, configurations, and methods may be used for removably attaching cover64to the underside/rear of the solar panel30to preferably fully enclose cable compartment60, such as for example without limitation snap fitted interfaces, threaded fasteners, or combinations thereof. In one, mounting edge65may be configured as a grounding type WEEB (“washer, electrical equipment bonding”) type interface similar to those commercially available from Wiley Electronics which includes provisions such as a threaded post for grounding the solar frame. As shown inFIG.14, the lateral frame members such as the H-channel locking frame member40and opposing C-channel frame member41may include an intermediate support flange70oriented perpendicularly to the lateral sides of the frame members to support the photovoltaic cell array31where the array has a thickness less than the height or depth between the top and bottom flanges or walls of the frame members. This may be used with any of the frame constructions disclosed herein regardless of whether a covered cable compartment is provided or not. While the foregoing description and drawings represent exemplary (“example”) embodiments of the present invention, it will be understood that various additions, modifications and substitutions may be made therein without departing from the spirit and scope and range of equivalents of the accompanying claims. In particular, it will be clear to those skilled in the art that the present invention may be embodied in other forms, structures, arrangements, proportions, sizes, and with other elements, materials, and components, without departing from the spirit or essential characteristics thereof. In addition, numerous variations in the methods/processes as applicable described herein may be made without departing from the spirit of the invention. One skilled in the art will further appreciate that the invention may be used with many modifications of structure, arrangement, proportions, sizes, materials, and components and otherwise, used in the practice of the invention, which are particularly adapted to specific environments and operative requirements without departing from the principles of the present invention. The presently disclosed embodiments are therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being defined by the appended claims and equivalents thereof, and not limited to the foregoing description or embodiments. Rather, the appended claims should be construed broadly, to include other variants and embodiments of the invention, which may be made by those skilled in the art without departing from the scope and range of equivalents of the invention. | 40,576 |
11863116 | DETAILED DESCRIPTION OF THE DRAWINGS FIGS.1to4Cshow various embodiments of the present floating solar photovoltaic array10, and its system of powering various accessory devices. In its various aspects, as seen inFIG.1, the system comprises: a plurality of PV modules40; a plurality of floating pontoons30for supporting PV modules40above the water; an inverter50or262for receiving DC power from PV modules40and converting the DC power to AC power. As will be explained, inverter262has an AC power limit such that any power received above the AC Power limit would be clipped by the inverter. Also included are at least one powered accessory device80; a power260line running from the floating solar array10to an on-shore grid; and an energy management power control system400configured to send power that has been clipped by the inverter to the at least one powered accessory device80. In preferred aspects, energy management power control system400is further configured to send power that has not been clipped by the inverter to the at least one powered accessory device. In still further aspects, energy management power control system400is further configured to receive power through the power line260running from the floating solar array to the on-shore grid to send power to the at least one powered accessory device80. As such, energy management power control system400can be configured to send power to the at least one powered accessory device80by adjustably changing the amounts of power received from each of the following power sources over a period of time: (i) power received from the PV modules that has been clipped by the inverter, (ii) power received from the PV modules that has not been clipped by the inverter, and (iii) power received from the on-shore grid. In various preferred aspects, powered accessory device80may be a water quality device including any one or more of the surface aerator200, the dredger201, the air compressor202, the ozone treatment device203or the water sensor204illustrated inFIG.1; or the aerator200, diffuser210, sub-surface agitator220, or sub-surface water circulator230illustrated inFIG.4A; the mooring/positional system250illustrated inFIG.4Bor the panel washer270or bird removal system280illustrated inFIG.4C. As will be further explained, when the powered accessory80is air compressor202, the air compressor can be used for inflating the plurality of pontoons. In optional embodiments, the powered accessory could also include debris collectors, UV treatment equipment, desalination equipment, or electrolyzers. Turning next toFIGS.2A to3, various exemplary embodiments of the present array10are seen. It is to be understood that for clarity of understanding these figures are only simplified illustrations, and that not all structural components are illustrated. FIG.2Ashows module40facing in a southward direction. In commercial embodiments, a plurality of the systems illustrated inFIG.2Aare positioned side by side (for example as seen inFIG.3).FIG.2Bshows alternating rows of modules40facing in either an east or west direction. As seen inFIGS.2A and3, array10comprises: a plurality of inflatable upper support pontoons20with upper mounting hardware/mounts22thereon; a plurality of lower support pontoons30with lower mounting hardware/mounts32thereon; and a plurality of solar photovoltaic modules40mounted therebetween. As seen inFIG.2B, two PV modules40may share the same upper support pontoon20. Each solar photovoltaic module40has an upper end41that is connected to the mounting hardware/mounts22on one of the inflatable upper support pontoons20and a lower end43that is connected to the mounting hardware/mounts32on one of the lower support pontoons30. As can be seen, the mounting hardware/mounts22on inflatable upper support pontoon20is higher (i.e.: farther above the water) than the mounting hardware/mounts32on lower support pontoon30. This preferred design holds each of the solar photovoltaic modules40at an inclined angle, as shown. In other embodiments, the mounting hardware22on each of the inflatable upper support pontoons20includes a U-ring connector thermally welded or adhesively connected to the inflatable upper support pontoon. In preferred aspects, upper support pontoon20may be an inflatable cylindrical tube made of materials including, but not limited to, High Density Polyethylene (HDPE), Thermoplastic Olefin (TPO), Polyvinycl Chloride (PVC), Ethylene tetrafluoroethylene (ETFE), or a PVC-coated fabric. Preferably, upper support pontoons20have a thickness of between 50 um to 25 mm, or more preferably between 0.5 and 2.5 mm. Lower support pontoons30may be made of similar materials and may also be inflatable. Also in preferred aspects, the lower support pontoons30have a flattened top surface31that can function as a walkway for operators to gain access to the PV modules. In optional aspects, a wire management chamber can be positioned on or in the lower support pontoons30. As explained above, the present array10also includes an air manifold system100(shown schematically inFIG.3). System100preferably comprises an air compressor202(SeeFIG.1) (or any other air source including an air tank), and pneumatic tubing140(see alsoFIGS.10A to10E) connecting air compressor202to each of the plurality of inflatable upper support pontoons20. Pressure sensors150can be included for determining air pressures in each of the inflatable upper support pontoons20. Lastly, an air manifold control system160can be used for measuring the output of pressure sensors150and controlling the air pressures in each of the inflatable upper support pontoons20. Most preferably, air manifold system100is completely (or at least partially) powered by the photovoltaic modules40in the solar photovoltaic array. In preferred aspects, the inclined angle of each of the solar photovoltaic modules40can be adjusted simply by adjusting an inflation level in one of the inflatable upper support pontoons20. Specifically, as an upper support pontoon20is inflated, the top end41of a solar PV module40will be raised, thereby placing PV module40into a somewhat more vertical orientation. Conversely, deflating upper support pontoon20will place the PCV module40into a somewhat more horizontal orientation. Therefore, by changing the inflation pressures within upper support pontoons20over the course of a day, the angle of tile of the PV modules can be made to better track the motion of the sun. As can be appreciated, the present floating mounting system uses substantially fewer components than traditional floating solar PV arrays. Instead, with the present system, so few components are required that the center portion of each solar photovoltaic module40can be positioned directly above water with no mechanical structure positioned directly thereunder (as seen inFIGS.2A and2B). As such, the only mechanical connection between any of the inflatable upper support pontoons and any of the lower support pontoons is through one of the solar photovoltaic modules. Next,FIG.4Aillustrates the present solar PV array10showing a variety of optional powered accessories (e.g.: devices80inFIG.1) that may be included therewith. Most preferably, these various powered accessories are powered by the PV modules40in the solar photovoltaic array. It is to be understood, however, that these powered accessories can be powered from a battery on the array (which may be recharged by the PV modules). As such, the powered accessory can be powered directly from the PV modules during the day and through the battery during the night (after the battery has been re-charged by the PV modules during the day). In various aspects, the powered accessories can optionally include an aerator200, a diffuser210, sub-surface agitator220, a sub-surface water circulator230, and a water quality sensor (204inFIG.1). It is to be understood that the present system can include any number or combination of these accessories. Placing large, floating solar arrays onto bodies of water has the advantage of not requiring large amounts of terrestrial real estate for array deployment. Unfortunately, covering a comparatively large body of water with a floating solar array can have undesirable effects. For example, stratification of the water can be a problem. Floating solar arrays also interfere with natural wave motion and partially block the sun from reaching the water, thereby darkening the water below the array. Accessories200,210,220and230(and203inFIG.1) can be used to remediate or improve water quality, and water quality sensor (204inFIG.1) can be used for measuring water quality. For example, aerator200can be a floating surface fountain as illustrated that sprays water upwards. Diffuser210can be a bottom resting device that releases bubbles of air (i.e.: air is pumped air down in a tube from above the array and released underwater it so that it bubbles upwards). Both aerator200and diffuser210assist in aerating the water. Sub-surface agitator230can be a propeller/turbine device mounted to the underside of the array that stirs the water under array10. Sub-surface water circulator230can be a bottom mounted propeller/turbine device that stirs the water under array10. These powered accessories help repair stratified water bodies, prevent algae blooms, and support desired flora and fauna. Ideally, accessories200,210,220,230and203can be powered by PV modules40, thereby permitting their operation during the daytime (when power is being generated by the array). Since accessories200,210,220,230and203typically do not need to be operating 24 hours/day to provide benefits, it is possible to operate these accessories solely relying upon power generated from the PV modules40. This provides a fully self-contained water quality remediation system. When water quality remediation devices such as these are integrated into the present solar array, installation costs are minimal. In addition, another advantage of using these powered accessories/water quality remediation devices is that it reduces the future costs of maintenance programs to reduce pond scum and toxic gasses. However, although these various devices may be powered solely by array10, it is to be understood that the present system also encompasses variations with accessories200,210,220and230powered by PV modules40, an on-board battery, a power line260running to shore or any combination thereof. FIG.4Bis a side elevation view of the present solar PV array showing an optional sub-surface mooring/positioning system250. Sub-surface mooring system250comprises a plurality of different propeller/turbines that move array10to a desired location (or keep array10at a desired location on the body of water). Although sub-surface positioning system250may be powered by PV modules40, it is to be understood that its power may also be supplemented by an on-board battery or by a power line260running to shore (to power sub-surface positioning system250during the night).FIGS.11A and11Bshow another preferred embodiment of mooring system250, as follows. InFIG.11A, a plurality of separate turbine/propeller systems251and252. Propellers251point outwardly from the bottom of array10. Conversely, propellers252point inwardly under the bottom of array10. By selectively turning on and off any of these propellers/turbines251and252, it is possible to move the array10in any desired direction. This includes both moving the array to a desired location and keeping it there. For example, on a calm day, propellers/turbines251and252may be turned off. However, on a windy day, those propellers/turbines251and252that are pointing in a direction opposite to the wind may be turned on to keep the array in a desired position. Propellers/turbines251and252can also be selectively turned on and off to rotate array10on the body of water such that PV modules40can track the movement of the sun. The present system encompasses embodiments in which propellers/turbines251and252are individually steerable and embodiments where propellers/turbines251and252are operated at different intensities (for example, with a strong horizontal “pushback” on a windy day to keep the array at a desired location on the body of water, together with a smaller rotational “push” causing the array to rotate to track the sun over the course of the day). As seen inFIG.11B, propeller/turbines251A in system250can be angled slightly downwards to further assist in keeping array10buoyant (as compared to more horizontal directed propeller/turbine251B). In various aspects, the present system also includes a plurality of mooring cables connected to at least one of the plurality of inflatable upper support pontoons20or lower support pontoons20for mooring the array at a desired location on a body of water. FIG.4Cis a perspective view of the present solar PV array10showing an optional panel washer270and an optional bird removal system280. Panel washer system270may simply comprise a sprayer275than can be directed to suck up water from below the array (with submersible pump276) and spray the water onto the surfaces of PV modules40to periodically clean the modules. Sprayer275can be automatically controlled to point in various directions to cover the surfaces of the different PV modules. The various cleaning routines can be programmed into the control system such that sprayer275sprays the surfaces of PV modules40one after another. Optional bird removal system280can function similar to panel washer270. Specifically, bird removal system280suck up water from below the array and spray the water onto the surfaces of PV modules40. However, the modules40are only sprayed when camera/motion sensor285detects a bird sitting on one of the PV modules40. When a bird is viewed sitting on one of the PV modules, the sprayer275is aimed at the bird. FIG.5is an exemplary graph of array power generated over time showing the portion of inverter-clipped power directed to the powered accessory. Specifically, over a 24 hour period, power output from PV modules40peaks mid-day, and is zero overnight. However, in this example, the maximum power the inverter is able to send to the grid (via power line260to shore inFIG.4A or4B) is 2500 KW. Accordingly, the power in region500can be sent to the on-shore grid. However, the power in region520will be “clipped” by the inverter and cannot be sent to shore. Accordingly, in accordance with the present energy management power control system400, the power in region520is instead sent directly to power an accessory80such as aerator200. Accordingly, the aerator is operated between about 8 am and 3 pm. Should it be desirable to operate an accessory80at extended periods of time, energy management power control system400can use different power balancing approaches as explained inFIGS.6A to8Bas follows. FIG.6Ais an exemplary graph showing inverter-clipped power520sent to a powered accessory80over a period of time. In this illustration, accessory80will only be operated during daylight hours when inverter-clipped power520is available. FIG.6Bis an exemplary graph showing the portions of both inverter-clipped power520and power500that has not been clipped by the inverter being sent to the powered accessory (or accessories)80over a period of time. In this illustration, non-clipped power500is used at the end of the day to power accessory80when inverter-clipped power has tapered off. Finally,FIG.6Cis an exemplary graph showing the portions of inverter-clipped power520, non-inverter-clipped power500and shore-received power540all being sent to the powered accessory (or accessories)80over a period of time. This specific illustration is taken over a period of a full year and shows the situation where some power from the grid (i.e.: power540) is used to power accessory80throughout the course of the year. FIG.7Ais an exemplary graph showing various sources of power being generated by the solar PV array over a continuous 24 hour period.FIG.7Bshows the power being sent to the powered accessory (or accessories) corresponding toFIG.7Aduring the continuous 24 hour period. Specifically, inverter-clipped power520is only sent to accessory80when such power is available (between about 6 am and 12 pm and 1 pm to 6 pm). Accordingly, power500(which has not been clipped by the inverter) will also be sent to accessory80from about 6 am to 6 pm such that the accessory has sufficient power for its operation (i.e.: such that the combined power regions500and520total the necessary power to run the device—identified as “WT Load” inFIG.7B). Before 6 am and after 6 pm, the PV modules40won't be generating any power. Thus, power540will be drawing directly from the grid to keep the accessories running. As can be seen, the relative contributions of power regions/sources500/520/540will change over time. Early in the morning as the day starts, grid power540is phased out as non-clipped power500comes online. By mid-day, clipped power520starts to come online (as the PV modules40exceed the “PV AC Limit” seen inFIG.7A), and the amount of non-clipped power500can be reduced. Later in the day, clipped power520starts to decrease until all power is supplied by non-clipped power source500(between about 5 pm and 6 pm). Finally, as non-clipped power source500starts to fall off, then grid power540will begin to take up the slack and will be the final sole power source overnight. Region600represents the power that array10supplies to the on-shore grid over the course of the day. FIGS.8A and8Bare similar toFIGS.7A and7B, however,FIGS.8A and8Bdeal with the situation where the powered accessory80need only be operated between about 10 am and 9 pm. Specifically, at around 10 am, power is supplied to the powered accessory from regions/sources500and520. As can be seen, the relative proportions of these two amounts will vary somewhat over the course of the day. After about 6 pm, the powered accessory will rely solely upon grid-supplied power540. At about 9 pm, the device80will be turned off and not turned on again until about 10 am the next morning. Power region521is “lost power” that has been clipped by the inverter but is not required to power accessory80at that particular time. FIGS.9A to9Cshow different schematics of powering the powered accessories. Specifically,FIG.9Ashows powering using a DC bus;FIG.9Bshows powering using an AC bus; andFIG.9Cshows powering using an electrically isolated system. InFIG.9A, an on-shore inverter (262inFIG.1) converts DC output from the solar photovoltaic modules40into AC power, and a DC bus300sends the DC power to the on-shore inverter. Advantages of using DC Bus300include lower DC ohmic losses, and the ability to use clipped power520with a constrained AC connection more easily. In addition, voltage droop control can be used. InFIG.9B, a dedicated on-board inverter55converts DC output from each of the solar photovoltaic modules into AC power, and an AC bus320is connected to the inverter for sending AC power to shore. Advantages of using AC bus300include the fact that any type of solar inverter55can be used, and off-the-shelf VFDs and AC motor drives can be used to power the aerators. Lastly, inFIG.9C, an advantage of an electrically isolated system is that, again, any type of solar inverter can be used. Next,FIGS.10A to10Eshow various layouts of the PV modules40using the present floating mounting system. Specifically,FIG.10A(which corresponds toFIGS.2A and3) shows PV modules40in a portrait, south facing orientation. As can be seen, each PV module40has its own dedicated upper support pontoon20.FIG.10B(which also corresponds toFIGS.2A and3) also shows the PV modules in a portrait, south facing orientation, but two PV modules40are sharing each upper support pontoon20.FIG.10C(which also corresponds toFIGS.2A and3) shows the PV modules40laid out in a landscape, south facing orientation.FIG.10D(which corresponds toFIG.2B) shows the PV modules40mounted in portrait, but laid out in in an east-west facing orientation. Specifically, a two rows of upper support pontoons20are next to one another. For a large array, two rows of lower support pontoons30would be positioned next to one another. As seen inFIG.10D, each of the PV modules40have their own dedicated upper support pontoon20. Lastly,FIG.10E(which corresponds toFIG.2B) shows PV modules40laid out in a portrait, east-west facing orientation, with the individual PV modules40sharing upper support pontoons20. As can be appreciated, a wide variety of different array configurations are possible with the present system (depending upon where the successive rows of pontoons20and30are positioned, and whether the PV modules40are positioned in portrait or landscape). Finally,FIGS.12A and12Bshow various systems for attaching two lower support pontoons30together. Specifically,FIG.12Ashows top and side views of a system for attaching pontoons30together using elastic connectors35.FIG.12Bshows top and side views of a system for attaching pontoons30together using mechanical plates37. | 21,033 |
11863117 | DETAILED DESCRIPTION Before any independent embodiments of the invention are explained in detail, it is to be understood that the invention is not limited in its application to the details of construction and the arrangement of components set forth in the following description or illustrated in the following drawings. The invention is capable of other embodiments and of being practiced or of being carried out in various ways. Also, it is to be understood that the phraseology and terminology used herein is for the purpose of description and should not be regarded as limiting. The use of “including,” “comprising,” or “having” and variations thereof herein is meant to encompass the items listed thereafter and equivalents thereof as well as additional items. Unless specified or limited otherwise, the terms “mounted,” “connected,” “supported,” and “coupled” and variations thereof areused broadly and encompass both direct and indirect mountings, connections, supports, and couplings. Further, “connected” and “coupled” are not restricted to physical or mechanical connections or couplings. Also, it is to be understood that phraseology and terminology used herein with reference to device or element orientation (such as, for example, terms like “central,” “upper,” “lower,” “front,” “rear,” etc.) are only used to simplify description of embodiments of the present invention and do not alone indicate or imply that the device or element referred to must have a particular orientation. In addition, terms such as “first” and “second” are used herein for purposes of description and are not intended to indicate or imply relative importance or significance. FIGS.1and2illustrate a roof mount system10including a roof12, a plurality of solar panels14, a plurality of mounting brackets16, flashing18and a skirt21. The solar panels14are mounted to the roof12via the plurality of mounting brackets16. In the illustrated embodiment, the solar panels14are orientated in a grid array. The solar panels14are coupled to the mounting brackets16located in a corresponding grid array on the roof12. In particular, the mounting brackets16are coupled to the solar panels14on the periphery of the major (e.g., horizontal) length of the solar panels14. Furthermore, the mounting brackets16are secured to the roof12by a roof fastener20(shown inFIGS.9and10). In other embodiments, the roof mount system10may secure other components to the roof12in addition to or in lieu of the solar panels14. The illustrated mounting brackets16are constructed to connect to a single solar panel14to the roof12or to connect to a plurality of solar panels14to the roof12. The illustrated mounting brackets16can also couple a plurality of solar panels14together. In addition, the mounting brackets16can support the skirt21on the roof12. The illustrated skirt21may extend along the entire length of the solar panels14at the bottom of the array of solar panels14. For example, the skirt21is located near a gutter system (not shown) or outer perimeter of the roof12. The skirt21provides a barrier between the ambient environment surrounding the solar panels14and the area located between the roof12and the solar panels14. For example, the skirt21is utilized to inhibit a substantial pressure differential between the ambient environment and an area under the solar panels14. The flashing18is positioned between the mounting brackets16and the roof12to inhibit leakage of fluids (e.g., rain water, snow, etc.) through the roof12while providing a structure to which the mounting brackets16are securely mounted. The flashing18is described in detail in U.S. Pat. No. 8,209,914, issued Jul. 3, 2012, the contents of which are herein incorporated by reference. With reference toFIGS.3-10, the mounting brackets16include a standoff30, a first clamp portion44, a second clamp portion46and an adjustment assembly48. The standoff30is coupled to the flashing18whereas the first clamp portion44, the second clamp portion46and the adjustment assembly48are coupled to the standoff30. The illustrated standoff30generally defines a 90 degree cross sectional construction including a first portion52that is substantially parallel with the roof12, and a second portion56that is substantially perpendicular with the roof12. The illustrated first portion52includes a mounting aperture60located closer to an edge of the standoff30rather than being centered on the standoff30. The mounting aperture60is sized to receive the roof fastener20and a portion of the flashing18. In the illustrated embodiment, the mounting aperture60is frustoconically shaped. In other words, a first opening of the mounting aperture60that is adjacent the flashing18includes a greater diameter than a second opening of the mounting aperture60positioned away from the flashing18(FIGS.3and8). The second portion56includes a channel64and a void68that both extend the entire length of the standoff30. The channel64generally defines a “T” shaped opening. Ridges66extend along both sides of the channel64and face downwardly towards the first portion52. The void68is configured to reduce the weight without compromising the structural integrity of the standoff30. The illustrated standoff30is manufactured from a nonferrous material (e.g., aluminum). The first clamp portion44is positioned above the standoff30. The first clamp portion44includes a first support surface70aand a second support surface70bseparated by walls72. The first and second support surfaces70a,70bare operable to support a portion of a solar panel14. The first support surface70aincludes a recess74that defines as a curvilinear depression on the first support surface70a. The second support surface70bincludes a channel78that protrudes below the second support surface70b. In other words, the channel78faces the standoff30. Similar to the channel64, the channel78includes ridges80. The illustrated channel78defines a “T” shaped opening that extends entirely through the length of the second support surface70b. The walls72are substantially perpendicular to the first and second support surfaces70a,70b. Located between the walls72is an aperture82(FIG.5). In addition, the first clamp portion44is moveable relative to the standoff30along a first axis86(FIG.6). The first axis86is generally perpendicular to the plane defined by the roof12. In the illustrated embodiment, the first clamp portion44is manufactured from a nonferrous material (e.g., aluminum). The second clamp portion46is sized to be received between the walls72and includes a first aperture90(FIG.5), a second aperture94and clamping protrusions98. Also, the second clamp portion46is moveable along the first axis86relative to the first clamp portion44. The first aperture90has a diameter that is less than the diameter of the second aperture94. The illustrated clamping protrusions98extend substantially parallel to and in the same direction as the first and second support surfaces70a,70b. In the illustrated embodiment, the first and second support surfaces70a,70bextend beyond the clamping protrusions98. In some embodiments, the second clamp portion46is manufactured from a nonferrous material (e.g., aluminum). The adjustment assembly48includes a stud102, a jam nut104, an adjustment nut108and a securing nut112. The stud102includes external threads and has a head116(FIG.5) which is sized to be received within the channel64. The jam nut104threadably engages the stud102. Likewise, the adjustment nut108and the securing nut112threadably engage the stud102, but the adjustment nut108includes a protrusion120and the securing nut112includes a protrusion122. The illustrated nuts108,112are substantially constructed as hexagonal cylinders with the respective protrusions120,122radially extending therefrom. The illustrated protrusions120,122are offset to one end of the nuts108,112. In other words, the illustrated protrusions120,122are not centered along the length of the nuts108,112. The protrusions120,122are offset to different locations in the configuration illustrated inFIGS.3-8than in the configuration illustrated inFIGS.9-10. The jam nut104, the adjustment nut108and the securing nut112are sized to receive a standard wrench or socket wrench. For example, the jam nut104, the adjustment nut108and the securing nut112may each be a ½″ nut able to receive a ½″ wrench. Additionally, the adjustment assembly48is manufactured from a ferrous material (e.g., steel, stainless steel, etc.). The skirt21shown inFIGS.1and2may be coupled to the first clamp portion44via the channel78. Specifically, a head portion of a fastener (e.g., a bolt) is received within the channel78and engages the ridges80(similar to the head116engaging the ridges66). The fastener is attached to a portion of the skirt21by either a threaded nut or the fastener engages a threaded aperture of the skirt. As the skirt21is coupled to the first clamp portion44, the head of the fastener is forced into the ridges80to enable a locking mechanism to inhibit the fastener from moving along or rotating within the channel78. With reference toFIGS.9and10, the standoff30is positioned on the flashing18and secured to the roof12by the roof fastener20. Before the roof fastener20is fully secured, the standoff30is able to rotate relative to the flashing18. The stud102is secured to the standoff30by sliding the head116to a desired location along the channel64. The jam nut104secures the stud102to the standoff30. Consequently, the stud102is rotationally and laterally fixed relative to the standoff30. Because the head116is manufactured from a ferrous material (e.g., stainless steel), the head116includes a material hardness greater than that of the nonferrous material (e.g., aluminum) of the standoff30. As the jam nut104is tightened, the head116is forced into and deforms the ridges66to provide an additional mechanism to inhibit rotational or sliding movement of the stud102relative to the standoff30. The adjustment nut108is positioned on the stud102and extends through the aperture82to support the first clamp portion44on the stud102. Consequently, the protrusion120of the adjustment nut108abuts the first clamp portion44, and a hexagonal portion of the adjustment nut108extends through the aperture82and is located between walls72(FIG.6). The second clamp portion46is also positioned on the stud102. The securing nut112is positioned on the stud102and extends through the first aperture90. In particular, the protrusion122of the securing nut112abuts a portion of the second clamp portion46adjacent the first aperture90to support the second clamp portion46on the stud102. Also, a hexagonal portion of the securing nut112extends between the first and second apertures90,94. In operation of securing the solar panels14to the mounting brackets16, the first clamp portion44is adjusted to a desired height above the roof12by rotating the adjustment nut108. By rotating the adjustment nut108, the first clamp portion44translates along the first axis86. In addition, the securing nut112is positioned at the top end of the stud102(i.e., away from the standoff30) to allow for the maximum clearance between the first and second support surfaces70a,70band the clamping protrusions98. The solar panels14are then easily received on either of the first and second support surfaces70a,70b. Once the height of the first clamp portion44is properly adjusted and a solar panel14is seated on either of the first and second support surface70a,70b, the securing nut112is rotated about the stud102such that the clamping protrusion98of the second clamp portion46engages a top surface of the solar panel14. In particular, the protrusions122of the securing nut112contact a portion of the second clamp portion46adjacent the first aperture90to translate the second clamp portion46downwardly along the first axis86. The securing nut112rotates about the stud102to move the second clamp portion46along the stud102in the direction of the first axis86. In addition, the recess74provides enough clearance between the first clamp portion44and the second clamp portion46to insert, pivot and secure a solar panel14therebetween. For example, if a first solar panel14is seated on the second support surface70band the second clamp portion46is tightened against the first solar panel14, a second solar panel14is able to be rotated via the recess74onto the first support surface70a. FIGS.11-18illustrate a roof mount system210according to another embodiment. The roof mount system210is similar to the roof mount system10; therefore, like components have been given like reference numbers incremented by 200 and only the differences between the roof mount systems will be discussed in detail. In addition, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated roof mount system210includes a mounting bracket216that couples at least one solar panel214to a roof212, and flashing218(shown inFIGS.17and18) is coupled between the mounting bracket216and the roof212. The illustrated mounting bracket216includes a standoff230, a first clamp portion244, a second clamp portion246, and an adjustment assembly248. The illustrated standoff230includes a first portion252and a second portion256. The first portion252includes a mounting aperture260and the second portion256includes a channel264and a void268. Ridges266extend along both sides of the channel264and face downwardly towards the first portion252. The first clamp portion244includes a support surface270and a hub232. The illustrated support surface270is defined as a circular surface surrounding the hub232. The hub232includes an internally threaded hollow core with an upper projection234extending above the support surface270and a lower projection238extending below the support surface270. A hexagonal protrusion242extends from the upper projection234away from the support surface270. The hexagonal protrusion242is sized to fit a standard sized wrench or socket wrench. In other embodiments, one or both of the projections234,238may be omitted, and the hexagonal protrusion242may directly abut the support surface270. The first clamp portion244is moveable along a first axis286relative to the standoff230. The second clamp portion246includes an aperture250and clamping protrusions298. Also, the second clamp portion246is moveable along the first axis286relative to the first clamp portion244. The illustrated aperture250is constructed as a countersunk aperture. The illustrated clamping protrusions298extend away from the aperture250. The adjustment assembly248includes a stud302, a jam nut304and a securing fastener254. The stud302includes a head316which is sized to be received within the channel264. Likewise, the securing fastener254also includes a head258that is sized to receive a standard sized wrench or socket wrench. The stud302and the securing fastener254are sized to threadably engage the hub232. With reference toFIGS.17and18, the standoff230is positioned on the flashing218and secured to the roof212by a roof fastener220. The stud302is secured to the standoff230by sliding the head316along the channel264to a desired location. The jam nut304secures the stud302to the standoff230. As the jam nut304is tightened, the head316deforms the ridges266to provide an additional mechanism to inhibit rotational and lateral movement of the stud302relative to the standoff230. As best shown inFIGS.13and16, the first clamp portion244is threaded onto the stud302. In particular, the lower projection238receives a portion of the stud302. The second clamp portion246is secured to the first clamp portion244when the securing fastener254is inserted into the aperture250and threadably engaged to the upper projection234. In operation of securing the solar panels214to the mounting brackets216, the first clamp portion244is adjusted to a desired height above the roof212by rotating the first clamp portion244via the hexagonal protrusion242. By rotating the hexagonal protrusion242with a wrench or socket wrench, the first clamp portion244translates along the first axis286. Because the hexagonal protrusion242receives a standard sized wrench socket, additional or specialty tools are not required to adjust the height of the first clamp portion244. The solar panels214are then seated on the support surface270such that an edge of the solar panel214is adjacent an outer circumference of the upper projection234. Once the height of the first clamp portion244is properly adjusted and the solar panel214is seated on the support surface270, the second clamp portion246is installed to clamp onto the solar panel214. Particularly, the securing fastener254is tightened to the hub232such that the clamping protrusions298clamp the solar panel214against the support surface270. FIGS.19-22illustrate a roof mount system according to another embodiment. The roof mount system is similar to the roof mount system10; therefore, like components have been given like reference numbers incremented by 400 and the only differences between the roof mount systems will be discussed in detail. In addition, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated roof mount system includes a mounting bracket416, rails401and flashing218. The illustrated rails401(only one is illustrated inFIG.21) are a different method of securing solar panels, such as the solar panels14, to the mounting bracket416. The illustrated mounting bracket416includes a standoff430, a first clamp portion444, a second clamp portion446, and an adjustment assembly448. The illustrated standoff430includes a first portion452and a second portion456. The first portion452includes a mounting aperture460and the second portion456includes a channel464and a void468. Ridges466extend along both sides of the channel464and face downwardly towards the first portion452. The first clamp portion444includes a base403that includes a channel sized to receive a protrusion of the rail401. The rails401are configured to support the solar panels14. The base403includes apertures482with one of the apertures being a threaded aperture. The illustrated first clamp portion444is moveable along a first axis486relative to the standoff430. The second clamp portion446includes an aperture450and clamping protrusions498. Also, the second clamp portion446is moveable along the first axis486relative to the first clamp portion444. The illustrated clamping protrusions498extend away from the aperture450. The adjustment assembly448includes a stud502, a jam nut504, an adjustment nut508, a locking nut513and a securing fastener454. The stud502includes a head516which is sized to be received within the channel464. Likewise, the securing fastener454also includes a head458that is sized to receive a standard sized wrench or socket wrench. The adjustment nut508includes a protrusion520that is offset to one end of the adjustment nut508. The securing fastener454engages the threaded aperture482whereas the adjustment nut508is received through the other aperture482. The locking nut513includes a threaded aperture and is slidably received within the first clamp portion444so that the threaded aperture aligns with the aperture482. The engagement between the first clamp portion444and the locking nut513inhibits the locking nut513from rotating relative to the first clamp portion444. With reference toFIG.22, the standoff430is positioned on the flashing418and secured to the roof12by a roof fastener420. The stud502is secured to the standoff430by sliding the head516along the channel464to a desired location. The jam nut504secures the stud502to the standoff430. As the jam nut504is tightened, the head516deforms the ridges466to provide an additional mechanism to inhibit rotational and lateral movement of the stud502relative to the standoff430. With continued reference toFIG.22, the first clamp portion444, the locking nut513and the adjustment nut508are received on the stud502. In particular, the protrusion520of the adjustment nut508is located between the locking nut513and the first clamp portion444. As a result, the first clamp portion444, the locking nut513and the adjustment nut508move along the first axis486together. In addition, the second clamp portion446and the securing fastener454are coupled to the threaded aperture482. In operation of securing the rail401directly to the mounting brackets416(e.g., a rail based system), the rails401are connected to the base403such that each rail401engages one channel of the base403. The first clamp portion444is adjusted to a desired height above the roof12by rotating the adjustment nut508. By rotating the adjustment nut508(e.g., with a wrench or socket wrench), the first clamp portion444translates along the first axis286. The rails401and the solar panels14are then seated on the first clamp portion444. Once the height of the first clamp portion444is properly adjusted and the solar panel14is seated on the rail401, the second clamp portion446is installed to clamp onto the solar panel14. Particularly, the securing fastener454is tightened such that the clamping protrusions498clamp the solar panel14against the rail401. FIGS.23-27illustrate a mounting bracket616according to another embodiment. The mounting bracket616is similar to the mounting bracket16; therefore, like components have been given like reference numbers incremented by 600 and only the differences between the mounting brackets will be discussed in detail. In addition, the mounting bracket616includes similar components to the mounting brackets; therefore, like components have been given like reference numbers. Furthermore, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated mounting bracket616includes a slide617, a standoff630, a first clamp portion644, a second clamp portion646, and an adjustment assembly648. The illustrated slide617abuts a flashing618and both are secured to the roof12by a roof fastener620. The illustrated standoff630is coupled to the slide617and moves along a second axis631(FIG.24) relative to the slide617. The second axis631extends substantially parallel to the roof12. The slide617includes a mounting aperture sized to receive the roof fastener620and upper and lower protrusions. The standoff630includes upper grooves and lower grooves that are sized to receive the respective upper and lower protrusions of the slide617(seeFIGS.25-26). The standoff630is selectively fixed relative to the slide617by a locking bolt607. The illustrated locking bolt607may also function as a grounding bolt to electrically connect the slide617to the standoff630to enable electrical current to flow therebetween. The illustrated first clamp portion644includes a first support surface670aand a second support surface670bseparated by walls672. The first support surface670aincludes a recess674that defines as a curvilinear depression on the first support surface670a. The second support surface670bincludes a channel678having ridges680that protrude into the channel below the second support surface670b. Located between the walls672are apertures682(FIG.26) with one aperture being a threaded aperture. In addition, the first clamp portion644is moveable relative to the standoff630along a first axis686(FIG.24). The first axis686is generally perpendicular to the plane defined by the roof12. The second clamp portion646includes a plurality of first apertures690, a plurality of second apertures694and clamping protrusions698. The illustrated first apertures690are constructed with different diameters, and the first apertures690substantially align with a respective second aperture694(FIG.26). Also, the second clamp portion646is moveable along the first axis686relative to the first clamp portion644. With reference toFIG.27, the illustrated adjustment assembly648includes a stud702, an adjustment nut708having a protrusion720, a securing fastener754, and a locking nut713. The stud702is threadably coupled to and extends from the standoff630. The first clamp portion644and the second clamp portion646are received on the stud702. The adjustment nut708is received in one of the first apertures690, for example, the first aperture690including the larger diameter. The locking nut713is slidably coupled to the second clamp portion646below the adjustment nut708. In other words, the locking nut713engages side walls of the second clamp portion646and is slid in place to align with the adjustment nut708with the protrusion720of the adjustment nut708located between the locking nut713and the aperture690. The illustrated locking nut713holds the adjustment nut708in place within the first aperture690. The adjustment nut708and the locking nut713are both received on the stud702. The securing fastener754is received through the other first aperture690and threadably engages the first clamp portion644. In operation of securing solar panels14to the roof12with the mounting bracket616, the securing fastener754clamps the first and the second clamp portions644,646onto the solar panels14. The securing fastener754is tightened by a socket wrench that is received through the corresponding second aperture694. The first and the second clamp portions644,646, and ultimately the solar panels14, are adjusted to a desired height above the roof12. In particular, the socket wrench that was utilized to tighten the securing fastener754can now be used to engage the adjustment nut708. By rotating the adjustment nut708, the adjustment nut708moves relative to the stud702along the first axis686. Because the locking nut713wedges the adjustment nut708against the second clamp portion646, the second clamp portion646moves with the adjustment nut708. In addition, because the securing fastener754couples the first and the second clamp portions644,646together, the first clamp portion644also moves with the adjustment nut708relative to the stud702. If desired, the height of the first clamp portion644can be adjusted before clamping the solar panel14between the first and second clamp portions644,646. FIGS.28-39illustrate a mounting bracket816according to another embodiment. The mounting bracket816is similar to the mounting brackets16and616; therefore, like components have been given like reference numbers and only the differences between the mounting brackets will be discussed in detail. Furthermore, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated mounting bracket816includes a slide817, a standoff830, a first clamp portion844, a second clamp portion846, and an adjustment assembly848. The illustrated slide817abuts a flashing818and both are secured to the roof12by a roof fastener820. The illustrated standoff830is coupled to the slide817and moves along a second axis831(FIG.30) relative to the slide817. The standoff830is selectively fixed relative to the slide817by a locking bolt807. The illustrated locking bolt807may also function as a grounding bolt to directly connect the slide817to the standoff830to enable electrical current to flow therebetween. The illustrated first clamp portion844includes a first support surface870aand a second support surface870bseparated by walls872. The first support surface870ais constructed as a hook that defines a channel809. Stated another way, the first support surface870ais discontinuous due to a gap created by the channel809. The illustrated gap functions similar to the recesses74,674to provide sufficient clearance between the first clamp portion844and the second clamp portion846to allow a solar panel814to rotate into position. A portion of the first support surface870ais also flexible to bend or deform generally along a first axis886. The illustrated channel809is configured and sized to receive one or more wires or cords. The second support surface870bincludes a channel878that extends below the second support surface670b. In contrast to the channels78,678, the channel878openings away from the standoff830. Located between the walls872are apertures882. In other embodiments, one of the apertures882may be a threaded aperture. In addition, the first clamp portion844is moveable relative to the standoff830along the first axis886(FIG.30). In the illustrated embodiment, the second clamp portion846includes a first aperture890, a second aperture894, a third aperture811and pins815. The illustrated pins815extend towards the first clamp portion844. Each pin815is located adjacent a respective clamping protrusion898. With reference toFIG.32, the illustrated adjustment assembly848includes a stud902, an adjustment nut908having a protrusion920, a securing fastener954, a biasing member903and a locking nut913. The stud902is threadably coupled to and extends from the standoff830. The first clamp portion844and the second clamp portion846are received on the stud902. The adjustment nut908is received in one of the first apertures890, for example, the first aperture890including the larger diameter. The locking nut913is slidably coupled to the first clamp portion844below the adjustment nut908. The locking nut913includes two apertures that respectively align with the apertures882of the first clamp portion844. The illustrated locking nut913holds the adjustment nut908in place within the first aperture890. The adjustment nut908and the locking nut913are both received on the stud902. The securing fastener954is received through the other first aperture890and threadably engages the first clamp portion844. The biasing member903is concentric with the securing fastener954and is located between the first and the second clamp portions844,846to bias the second clamp portion846away from the first clamp portion844. Operation of securing the solar panels814to the mounting bracket816is a similar process to the mounting bracket616, as described above. However, the support surface870aacts as a spring to bias the solar panel814against the second clamp portion846. Once the second clamp portion846is clamped onto the solar panel814, the pins815engage a top surface of the solar panel814to provide an electrical bond therebetween. With reference toFIGS.35and36, a skirt821may be coupled to the mounting bracket816by a bonding screw919. The bonding screw919is threaded into the third aperture811of the second clamp portion846. In the illustrated embodiment, the skirt821is orientated at an angle 8 relative to the flashing816(e.g., the surface of the roof12). The illustrated angle 8 is about 60 degrees, but in other embodiments the angle 8 can range from about 20 degrees to about 80 degrees. The illustrated skirt821includes a snow fence protrusion923, a top protrusion925and a bottom protrusion927. The illustrated snow fence protrusion923extends above the second clamp portion846at a distance X. The illustrated distance X is about one inch, but in other embodiments, the distance X may be greater than an inch. The bottom protrusion927slides along the channel878of the first clamp portion844. The bottom protrusion927can be vertically moved into and out of the channel878upon height adjustment of the second clamp portion846with respect to the first clamp portion844. The illustrated top protrusion925engages a lip on the top of the second clamp portion846. The bonding screw919abuts the top protrusion925and upon tightening of the bonding screw919presses the top protrusion925against the top surface of the second clamp portion846. In addition, the skirt821includes a channel880having ridges, which function similar to the channel78and the ridges88of the mounting bracket16, to further secure the skirt821to the mounting bracket816by a bolt (not shown). In addition, the pins815are sized to engage the skirt821to mechanically and electrically connect the mounting bracket816to the skirt821. Upon rotation of the bonding screw919, the skirt821is pressed against the pins815and deformed to create a mechanical and electrical bond between the skirt821and the mounting bracket816. In some embodiments, the bonding screws919and the pins815cut into the skirt821to form an electrical bond. In embodiments that include two solar panels814, the bonding screws919cut into one or more of the solar panel frames814to form an electrical bond. As shown inFIGS.37-39, the first and second clamp portions844,846can include a plurality of first cushions929that contact the solar panel814. As shown inFIG.39, a second cushion933is positioned between the solar panel814and the first and second clamp portions844,846. In some embodiments, the cushions929,933inhibit scratching of the solar panels814by the first and second clamp portions844,846. In some embodiments, the cushions929,933can inhibit the solar panels814from rattling when installed in a windy climate. The cushions929,933can be included on one or more of the clamps described and illustrated herein. FIGS.40and41illustrate a mounting bracket1016according to another embodiment. The mounting bracket1016is similar to the mounting brackets16,616and816; therefore, like components have been given like reference numbers and only the differences between the mounting brackets will be discussed in detail. Furthermore, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated mounting bracket1016includes a slide1017, a standoff1030, a first clamp portion1044, a second clamp portion1046, and an adjustment assembly1048. The illustrated slide1017abuts a flashing1018and both are secured to the roof12by a roof fastener1020. The illustrated standoff1030is coupled to the slide1017and moves parallel to a second axis1031relative to the slide1017. The standoff1030is selectively fixed relative to the slide1017by a locking bolt (similar to the locking bolt807). The illustrated first clamp portion1044includes a first support surface1070aand a second support surface1070bseparated by walls1072. The first support surface1070aincludes a recess1074that defines as a curvilinear depression on the first support surface1070a. The second support surface1070bincludes a channel1078having ridges1080that protrudes below the second support surface1070b. Located between the walls1072are apertures1082(only one is illustrated inFIGS.40and41) with one aperture being a threaded aperture (e.g., the aperture not shown inFIGS.40and41). In addition, the first clamp portion1044is moveable relative to the standoff1030along a first axis1086. The second clamp portion1046includes a first aperture, a second aperture1094and clamping protrusions1098. The illustrated first aperture is constructed with a smaller diameter than the second aperture1094, and the first aperture substantially aligns with the second aperture1094. Also, the second clamp portion1046is moveable along the first axis1086relative to the first clamp portion1044. With reference toFIG.40, the illustrated adjustment assembly1048includes a stud1102, an adjustment nut1108ahaving snap rings1135, and a securing fastener1154. The adjustment nut1108athreadably engages the stud1102and is received through one of the apertures1082. As such, the snap rings1135are coupled to grooves defined on the adjustment nut1108asuch that one snap ring1135is located above the first clamp portion1044, and the other snap ring1135is located below the first clamp portion1044. In other embodiments, the adjustment assembly1048may include a biasing member (e.g., a spring) to bias the second clamp portion1046away from the first clamp portion1044. FIG.41illustrated an alternative embodiment of an adjustment nut that can be utilized in place of the adjustment nut1108ashown inFIG.40. An adjustment nut1108b, as illustrated inFIG.41, threadably engages the stud1102and is received through one of the apertures1082. Collars1137are fixed to the adjustment nut1108bsuch that one collar1137is located above the first clamp portion1044and one collar1137is located below the first clamp portion1044. In other embodiments, the adjustment assembly1048may include a biasing member (e.g., a spring) to bias the second clamp portion1046away from the first clamp portion1044. Operation of securing the solar panels14to the mounting bracket1016is similar to the mounting bracket816. In particular, the adjustment nuts1108a,1108bare rotatable about the stud1102but are inhibited from axial movement with respect to the first clamp portion1044via the snap rings1135and the collars1137to adjust the first clamp portion1044to a desired height above the roof12. The securing fastener1154threadably engages the first clamp portion1044to clamp the solar panels14between the clamp portions1044,1046. FIGS.42-46illustrate a mounting bracket1216according to another embodiment. Components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated mounting bracket1216includes a skirt bracket1273coupled to a rail1201. The illustrated rail1201is similar to the rail401. In other embodiments, the mounting bracket1216may couple the skirt bracket1273to other embodiments of the mounting bracket16,216,416,616,816. For example, the skirt bracket1273may directly connect to the first clamp portion44,244,444,644,844. The illustrated rail1201includes side fingers1291and top fingers1293. The side fingers1291define an opening to channels located on a side of the rail1201, and the top fingers1293define an opening to a channel located on a top of the rail1201. The rail1201is configured to be utilized in a rail-based system to couple the solar panels14to the roof12. The illustrated skirt bracket1273includes a first end1275and a second end1277spaced from the first end1275. The first end1275includes a first, a second and a third flange1279,1281,1283. The illustrated first flange1279includes a threaded aperture (FIG.45). A hook portion1285is located near the second end1277. In addition, the skirt bracket1273includes a top plate1287and a securing fastener1254. The skirt1221is similar to the skirt21includes a top edge, a bottom edge, a first side edge, a second side edge and a plurality of perforations1289. In some embodiments, the perforations1289are spaced across substantially the entire skirt1221, whereas in other embodiments the perforations1289are positioned only on portions of the skirt1221. In the illustrated embodiment, the perforations1289are ¼ inch holes. In some embodiments, the skirt1221is constructed from aluminum, such as extruded aluminum. In other embodiments, the skirt1221may be a solid member. To assemble the skirt bracket1273onto the rail1201, the skirt bracket1273is pivotally coupled to the rail1201. In particular, the third flange1283engages one side finger1291. Then, the second flange1281is able to rotate towards one of the top fingers1293to engage with the top finger1293. As such, the skirt bracket1273snaps onto the rail1201by engagement between the top finger1293and the second flange1281. In the illustrated embodiment, the skirt brackets1273are positioned periodically along a base of the roof12. To assembly the skirt1221to the skirt bracket1273, the skirt1221is received into the hook portions1285of adjacent skirt brackets1273extending along the roof12. As a result, the skirt1221is seated flush with the skirt bracket1273. The top plate1287is fixedly coupled to the first flange1279by the securing fastener1254such that the skirt1221is secured between the top plate1287and the skirt bracket1273. In addition, the first flange1279is positioned above the rail1201, which functions as a snow fence protrusion, to inhibit snow and ice from sliding off the solar panels14and the roof12. The skirt1221is oriented at a non-parallel, non-perpendicular angle with respect to the rail1201and ultimately the roof12. The shape of the skirt bracket1273and the angle of the solar panel14and the roof12that the skirt bracket1273is attached to affect the angle between the skirt1221and the roof12. The angle is selected to inhibit a viewer on the ground surface from viewing the area between the solar panels14and the roof12through the perforations1289in the skirt1221. However, air is permitted to flow through the area between the solar panels14and the roof12because of the perforations1289. This air flow allows cooling of the solar panels14but inhibits small animals and birds from entering the area between the solar panel14and the roof12. The skirt1221also inhibits debris from gathering between the solar panels14and the roof12. FIGS.47-51illustrate a mounting bracket1416according to another embodiment. The mounting bracket1416is similar to the mounting brackets16and816; therefore, like components have been given like reference numbers and only the differences between the mounting brackets will be discussed in detail. Furthermore, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. The illustrated mounting bracket1416includes a first clamp portion1444, a second clamp portion1446and an adjustment assembly1448. The illustrated first clamp portion1444includes a first support surface1470adefining a channel1409and a second support surface1470bhaving a channel1478. Walls1472separate the support surfaces1470a,1470bwith first apertures1482located between the walls1472. In other embodiments, the first apertures1482may be threaded apertures. In the illustrated embodiment, the second clamp portion1446includes first apertures1490, second apertures1494, third apertures1411, pins1415and clamping protrusions1498. The pins1415are located on one of the clamping protrusions1498. The second clamp portion1446is moveable relative to the first clamp portion1444along a first axis1486. The illustrated adjustment assembly1448includes securing fasteners1554, a locking nut1513and biasing members1503. The locking nut1513engages with the first clamp portion1444and includes threaded apertures that align with respective first apertures1482. The securing fasteners1554are concentric with the biasing members1503to bias the second clamp portion1446away from the first clamp portion1444. The securing fasteners1554threadably engage the locking nut1513. In operation, the mounting bracket1416can couple two solar panels1414together, four solar panels1414together, or two solar panels1414in combination with two skirts1421together (FIGS.50and51). The mounting bracket1416provides support to adjacent solar panels1414and/or adjacent skirts1421without coupling the mounting bracket1416directly to the roof12. When clamping the mounting bracket1416onto solar panels1414, the process is similar to the mounting bracket816as described above. The illustrated skirt1421includes a snow fence protrusion1523, a top protrusion1525, a bottom protrusion1527and a channel1478having ridges1480. As discussed above in reference to the skirt821, the bottom protrusion1527slides along the channel1478. The bottom protrusion1527can be vertically moved into and out of the channel1478upon height adjustment of the second clamp portion1446with respect to the first clamp portion1444. The top protrusion1525engages a lip on the top of the second clamp portion1446. Bonding screws1519abut the top protrusion1525and upon tightening of the bonding screws1519, the top protrusion1525is pressed against the top surface of the second clamp portion1446and the pins1415. FIGS.52-59illustrate a mounting bracket1616according to another embodiment. The mounting bracket1616is similar to the mounting bracket16; therefore, like components have been given like reference numbers incremented by 1600 and only the differences between the mounting brackets will be discussed in detail. In addition, components or features described with respect to only one or some of the embodiments described herein are equally applicable to any other embodiments described herein. In some embodiments, the flashing18may be illustrated as flashing1618(FIGS.52and59). The illustrated flashing1618is substantially defined as a rectangular member of sheet metal and is coupled between the roof12and the mounting bracket1616. In particular, the flashing1618is positioned (e.g., sandwiched) between adjacent roof shingles (not shown) of the roof12. The illustrated flashing1618includes a first portion1624, a second portion1628, an intermediate portion1639and a coupling mechanism1641. The first portion1624and the second portion1628are separated by the intermediate portion1639, and the illustrated first portion1624includes a larger surface area than the second portion1628. The illustrated intermediate portion1639is a curvilinear protrusion that extends towards the mounting bracket1616(e.g., away from the roof12). In other embodiments, the intermediate portion1639may be differently constructed to distinguish the first portion1624from the second portion1628. The illustrated coupling mechanism1641is coupled to a bottom surface of the second portion1628and located between the roof12and the flashing1618. In the illustrated embodiment, the coupling mechanism1641is doubled sided tape to secure the flashing1618to the roof12without penetrating the roof (e.g., with a fastener); however, in other embodiments, the coupling mechanism1641may be any suitable adhesive. In further embodiments, the coupling mechanism1641may be a clip that couples the second portion1628to a single roof shingle. The illustrated mounting bracket1616includes a first clamp portion1644, second clamp portions1646, an adjustment assembly1648and a support structure1643. The mounting bracket1616is coupled to the roof12by the support structure1643. The first clamp portion1644is moveable relative to the flashing1618in an upward direction and a downward direction parallel to a first axis1686(FIG.59). The illustrated first clamp portion1644includes a first support surface1670aand a second support surface1670b. In the illustrated embodiment, the first support surface1670aincludes a concave recess1674. The illustrated first clamp portion1644includes a tool aperture1645, a central aperture1626and first and second threaded apertures1682. The illustrated tool aperture1645is located between a threaded aperture1682and the central aperture1626whereas the central aperture1626is located between the first and the second threaded apertures1682. In other embodiments, the first clamp portion1644may include two tool apertures1645located on respective sides of the central aperture1626. The illustrated first clamp portion1644also includes walls1672that extend substantially normal to the first and second support surfaces1670a,1670b. The walls1672abut the second clamp portions1646so that the second clamp portions1646are inhibited from rotating relative to the first clamp portion1644. In addition, the second support surface1670bincludes a channel1678having ridges1680. In some embodiments, the skirt21is coupled to the mounting bracket1616by engagement with the channel1678and the ridges1680. The second clamp portions1646are illustrated as two distinct members that respectively align with the first and the second threaded apertures1682. The second clamp portions1646are moveable relative to the first clamp portion1644in the upward direction and the downward direction parallel to the first axis1686. Each of the illustrated second clamp portions1646include a first aperture and a second aperture1694positioned above the first aperture. In addition, each of the illustrated second clamp portions1646include clamping protrusions1698. In other embodiments, the second clamp portions1646may be constructed as one integral member with an aperture aligning with the central aperture1626and an aperture aligning with the tool aperture1645. The adjustment assembly1648includes securing fasteners1654. The securing fasteners1654connect the second clamp portions1646to the first clamp portion1644. The securing fasteners1654substantially align with a respective one of the first and second threaded apertures1682. The illustrated support structure1643supports the first clamp portion1644at a desired height above the flashing1618and ultimately the roof12. The illustrated support structure1643includes a base1649, a stud1651and an adjustment nut1609. The illustrated base1649is generally defined as a cylindrical disk that is configured to abut the flashing1618and includes a bottom portion1653and a plurality of grooves1655. Particularly, the bottom portion1653is in direct contact with the flashing1618and is constructed from material that increases the coefficient of friction between the base1649and the flashing1618. For example, the bottom portion1653may be manufactured as an adhesive, sand paper type material, or the like to grip a surface of the flashing1618. The illustrated grooves1655are radially spaced on a circumference of the base1649. In the illustrated embodiment, the base1649includes six grooves equally spaced apart; however, in other embodiments, the base1649may include more or less than six grooves. Additionally, the base1649is rotatable such that one of the grooves1655is moveable into alignment with the tool aperture1645. The illustrated stud1651includes external threads and couples the base1649to the first clamp portion1644. The illustrated stud1651is substantially centrally attached and rotatably fixed to the base1649. The central aperture1626is sized to receive the stud1651. The illustrated adjustment nut1609extends through the central aperture1626and is rotatable relative to the first clamp portion1644. However, the adjustment nut1609is inhibited from movement relative to the first clamp portion1644along the first axis1686by projections1620. The illustrated projections1620are located on opposite surfaces (i.e., top and bottom) of the first clamp portion1644adjacent the central aperture1626. The projections1620radially extend from the adjustment nut1609. In addition, the adjustment nut1609is sized to receive a standard sized wrench or socket wrench. The illustrated mounting bracket1616also includes a tool1659. The tool aperture1645is sized to receive the tool1659. In addition, the grooves1655are sized to receive a portion of the tool1659. In the illustrated embodiment, the tool1659is a round rod with a bent end. In some embodiments, the tool1659is an Allen wrench, a threaded rod or a fastener. In operation, the support surfaces1670a,1670bsupport the corners of the solar panels14at a desired height above the flashing1618and ultimately the roof12. The first clamp portion1644is moveable relative to the flashing1618and the roof12along the first axis1686upon rotation of the adjustment nut1609. In particular, the tool1659is received within the tool aperture1645so a portion of the tool1659engages one of the grooves1655. Consequently, the tool1659inhibits relative rotation between the base1649and the first clamp portion1644. As the adjustment nut1609is rotated (by a standard socket wrench or the like), the projections1620move the first clamp portion1644along the first axis166. To install the solar panels14onto the mounting bracket1616, one of the solar panels14is positioned on the second support surface1670bin abutting relationship with the wall1672. Each of the second clamp portions1646engages a portion of the respective solar panel14via the clamping protrusions1698. The securing fasteners1654are tightened to clamp the solar panels14between the clamp portions1644,1646to secure the solar panels14to the mounting bracket1616. Another solar panel14is positioned on the first support surface1670ain abutting relationship with the other wall1672by utilizing the concave recess1674. In particular, an edge of the solar panel1614is positioned within the concave recess1674such that the solar panel14is orientated at an acute angle relative to the first support surface1670a. The solar panel14is then rotated towards the flashing1618to abut the wall1672and the first support surface1670a. The flashing1618is retained on the roof12because the intermediate portion1639is positioned upslope of the base1649. The bottom portion1653of the base1649and the coupling mechanism1641also retain the flashing1618on the roof12. Therefore, the flashing1618is retained on the roof12without using a fastener that penetrates the flashing1618and roof12. With reference toFIGS.60and61, a roof mount system1810can be electrically grounded (e.g., earth grounding) via a grounding lug1861and grounding wire1863. The grounding lug1861is directly coupled to one of the solar panels1814, and the grounding wire1863is electrically coupled to the grounding lug1861and is coupled to the earth (e.g., ground). The illustrated solar panels1814are coupled to the roof12by mounting brackets1816a. The illustrated mounting brackets1816amay be anyone or a combination of the mounting brackets16,216,416,616,816,1016,1216. In addition, the illustrated solar panels1814are coupled to each other by mounting brackets1816b. The illustrated mounting brackets1816bmay be anyone or a combination of the mounting brackets1416,1616. Also, the roof mount system1810includes a skirt1821, which is similar to the skirt21,821,1221,1421. The mounting brackets1816a,1816band the skirt1821provide a grounding path1865between adjacent solar panels1814to electrically couple every solar panel1814to the grounding lug1861. With reference toFIG.61, two adjacent roof mount systems1810are electrically coupled and grounded. Each roof mount system1810includes a grounding lug1861coupled to a single solar panel1814. The illustrated grounding path1865electrically couples every solar panel1814together within a respective roof mount system1810. Both grounding lugs1861are electrically coupled together by the grounding wire1863, and the grounding wire1863also couples one grounding lug1861to the earth to completely ground both roof mount systems1810. In some embodiments, electrical bonding is achieved by cutting into one or more of the solar panel frames with one or more pins or screws coupled to the mounting bracket16,216,416,616,816,1016,1216,1416,1616,1816a,1816b. In some embodiments, electrical bonding is achieved by cutting into one or more of the solar panel frames by press-fitting the solar panel frame into the mounting bracket16,216,416,616,816,1016,1216,1416,1616,1816a,1816band/or deforming the solar panel frame with the mounting bracket16,216,416,616,816,1016,1216,1416,1616,1816a,1816b. In some embodiments a stainless steel element pin, screw or protrusion is utilized to form an electrical bond between the respective solar panel and mounting bracket16,216,416,816,1016,1216,1416,1616,1816a,1816b. In some embodiments, the skirt21,821,1221,1421,1821is a snow guard that inhibits snow from sliding down the solar panels. In some embodiments, the skirt21,821,1221,1421,1821is a snow guard and an electrical ground for at least one of the solar panels. | 55,262 |
11863118 | DETAILED DESCRIPTION The descriptions of the various embodiments have been presented for purposes of illustration, but are not intended to be exhaustive or limited to the embodiments disclosed. Many modifications and variations will be apparent to those of ordinary skill in the art without departing from the scope and spirit of the described embodiments. The terminology used herein was chosen to best explain the principles of the embodiments, the practical application or technical improvement over technologies found in the marketplace, or to enable others of ordinary skill in the art to understand the embodiments disclosed herein. In one or more embodiments, a specialized bearing half-race that attaches to a support post or pier in tandem with a second identical half-bearing race and forms a full bearing race for a simple bearing for torque tubes on a single-axis tracker. In some embodiments, metal straps are formed to create an attachable half of a bearing race. In one or more embodiments, two identical halves of a lower bearing component are coupled or interleaved with each other to create a full bearing race that holds a rotating torque tube as a journal. One embodiment provides an apparatus for a single axis tracker that includes a first strap component formed into a first portion of a bearing race. A second strap component is formed into a second portion of the bearing race. The first portion of the bearing race and the second portion of the bearing race removably interleave to hold a torque tube. The first basic concept of a bearing for a large single-axis tracker row is to maintain alignment of the torque tube axle while allowing the torque tube to rotate, journaling the solar panels between an east west tilt orientations. One significant advantage of one or more embodiments is that the bearing race is made from a single flat piece of metal, such as steel, with no welding required, but only cutting and cold forming. Two identical items of the bearing half-race unique interleave such that the two identical halves form a full bearing race for containing the journal section of the torque tube. Another advantage of the one or more embodiments is that the bearing halves pivot, or scissor, to allow the torque tube system to drop into the bearing race from above, facilitating field assembly of the torque tube system. There are several innovations for one or more embodiments, which include various methods to attach bearing halves to support posts or piers. The lower portion of a bearing may have vertical slots for attaching to a flat surface web of the support post (or pier), such as an I-Beam, using fasteners (e.g., bolts, washers and nut hardware, etc.) and for providing vertical adjustment of the bearing relative to the post or pier. The posts or piers may have slots (or openings) such that the formed bearing may be adjusted (e.g., laterally, vertically, etc.). A lower cost embodiment forgoes mounting slots in both the posts or piers and the bearing race-halves, and relies on field welding of the bearing race-halves to the piers or posts. FIG.1shows a perspective view of a first lower portion1of a bearing race component for a solar tracker system, according to one embodiment. In some embodiments, the lower part of the first lower portion1includes multiple (e.g., two, three, four, etc.) slots (e.g., vertical openings, through-holes, etc.) for attaching to a flat side of a pier3(FIG.9), such as a web of an I-beam. The middle portion of the first lower portion1is twisted about ninety (90) degrees and forms a transition neck. In one embodiment, the upper portion is rolled into a partial (e.g., a quarter, etc.) circular race ending with an upper tab folded about ninety (90) degrees outward and including a through-hole or opening (having a circular, polygonal, etc., shape) for attaching a top bearing race or bearing top cap2(FIGS.8A-B). In one embodiment, the first lower portion1may be formed from a metal strap having a significant thickness and width to perform as half of the bottom component of an attachable simple bearing race for a journaling single axis solar tracker, which is interleaved or congruent with a second identical formed metal strap to create the bottom half of a bearing race. The form of the first lower portion1includes a lower section for attachment to the flat surface of a support pier3(FIG.9), either by field welding, with fastening hardware (e.g., bolt and nut, screws, etc.), etc., in one or more openings (e.g., holes, slots, etc.). Positioned above the openings is an off-center twist (e.g., about ninety (90) degrees) followed by about a ninety (90) degree bend outwards to form a semi-circular half-race with an inside radius just slightly larger than the outside radius of the torque tube journal (e.g., round torque tube, round journal, round journal-coupler, etc.) that rotates within to form half of the bottom half of a bearing race, or a bottom quarter of the full bearing race, followed by a ninety (90) degree bend outward to form an upper tab for securing the bearing top cap2(FIGS.8A-B) half-race. FIG.2shows a perspective view of another first portion10of another bearing race component for a solar tracker system, according to one embodiment. The first portion10may be formed from a similar metal strap as the first portion1(FIG.1). The bottom and middle portions of the first portion10are the same as the first portion1(FIG.1). The top portion is rolled into a semi-circular race ending with an upper tab folded about ninety (90) degrees upward and including a through-hole or opening (having a circular, polygonal, etc., shape) for attaching to a second portion10(seeFIG.16). FIG.3shows multiple (manufacturing) forming zones (10A-D) on the first portion10of the bearing race component ofFIG.2, according to one embodiment. In one embodiment, the forming zone10A pertains to the attachment slots. The forming zone10B pertains to the off-center twist of about ninety (90) degrees. The forming zone10C pertains to the semi-circular race. The forming zone10D pertains to the adjoining upper tab. In one embodiment, the forming of the zones10A-D may be made manually, by metal bending machinery, robotics, etc. FIG.4shows the multiple (manufacturing) forming zones (41A-D) on a flat metal strap40before forming into the first lower portions1or10of the different bearing race components shown inFIGS.1and2, according to some embodiments. In some embodiments, the forming zone41A pertains to the attachment slots for either the first lower portions1or10. The forming zone41B pertains to the off-center twist of about ninety (90) degrees for either the first lower portions1or10. The forming zone41C pertains to the race portion for either the first lower portions1or10. The forming zone41D pertains to the adjoining upper tab for either the first lower portions1or10. In one embodiment, the forming of the zones41A-D may be made manually, by metal bending machinery, robotics, etc. In one embodiment, the flat metal strap40may be made of a metal, metal alloy, etc., that may be rust inhibitive, coated or bonded with rust preventive coating or material. FIG.5shows multiple metal straps40patterned on a single sheet of metal before cutting out the patterns into multiple single straps40as shown inFIG.4, according to some embodiment. In one embodiment, the single sheet may be made of metal, metal alloy, etc. In some embodiments, the single sheet may be cut using precision laser or cutting machinery, tools, etc. FIG.6shows a side view of the multiple (manufacturing) forming zones10A-D on the first portion10of the bearing race component ofFIG.2, according to one embodiment. As shown, the forming zone10A includes the forming of the attachment slots for either first or second lower portion10. The forming zones10B-D includes the bending or twisting of a single strap40(FIG.5) for either first or second lower portion10. FIG.7shows a side view of the multiple (manufacturing) forming zones1A-D on the first lower portion of the bearing race component ofFIG.1, according to one embodiment. As shown, the forming zone1A includes the forming of the attachment slots for either first or second lower portion1. The forming zones1B-D includes the bending or twisting of a single strap40(FIG.5) for either first or second lower portion1. FIG.8Ashows an exploded view of a bearing race system comprised of the first and second lower portions1of the bearing race component ofFIG.1and a bearing race cap2, according to one embodiment. In one embodiment, the bearing race cap2may be made of a metal, metal alloy, etc., that may be rust inhibitive, coated or bonded with rust preventive coating or material. FIG.8Bshows a view of a formed bearing race system comprised of the first and second lower portions1of the bearing race component ofFIG.1and the bearing race cap2, according to one embodiment. As shown, the bearing race cap2attaches to the first and second lower portions1that are connected or interleaved together before fastened as described below. FIG.9shows two interleaved first and second lower portions1of the bearing race component ofFIG.1attached to a post (or pier)3via hardware4at a crossing of vertical and horizontal slots, according to one or more embodiments. As shown, the combined first and second lower portions1result in alignment of the slots (e.g., vertical slots, openings, etc.) of each of the first and second lower portions1. Connecting the post (or pier)3with the combined first and second lower portions1results in alignment of the post (or pier)3horizontal openings (through-holes, etc.) with the aligned slots of the first and second lower portions1, such that the hardware4fasteners (e.g., bolts and nuts, etc.) may be inserted into the combined union of the fastening horizontal and vertical slots for coupling. In one embodiment, the post (or pier)3may be an I-beam post (or pier), or similar structure for support of torque tubes for a solar tracking system that supports multiple solar panels. FIG.10shows the assembly ofFIG.9with a torque tube5(journal) component being placed into the formed lower bearing race component (from first and second lower portions1) in the direction of the arrow, according to one embodiment. As shown, the combined first and second lower portions1form a concave-up cradle of the bottom bearing race system. FIG.11shows the torque tube5(journal) component placed into the formed lower bearing race component (from first and second lower portions1), according to one or more embodiments. The torque tube5(journal) when placed onto the formed lower race component rests in the formed cradle without additional components, elements or materials between the torque tube5(journal) and the formed lower race component. FIG.12shows the bearing race cap2being lowered onto the formed lower bearing race (fromFIG.11) in the direction of the arrow in order to complete a full race with the bearing race cap2for retaining a torque tube5system (e.g., multiple torque tubes serially coupled to one another) in alignment, according to one or more embodiments. In one embodiment, the bearing race cap2is lowered over the torque tube5(journal) such that the upper tabs of the lower bearing race align with attaching tabs of the bearing race cap2(see, e.g.,FIG.13). FIG.13shows the formed complete bearing race continuing fromFIG.12after fastening, according to one or more embodiments. As shown, once the bearing race cap2is lowered such that the upper tabs of the lower bearing race align with attaching tabs of the bearing race cap2for retaining the torque tube system in alignment, which is especially needed during wind uplift of solar panels connected to the torque tubes5, according to one or more embodiments. In one embodiment, the bearing race cap2is fastened to the formed lower bearing race using hardware4fasteners. FIG.14shows a first portion10and a second portion10of a bearing race component for a solar tracker system facing each other, according to one or more embodiments. In one embodiment, the first and second lower portions10are placed together or intertwined to form one full bearing race. FIG.15shows the first portion10and the second portion10of a bearing race component moving in the direction of the arrow to form a lower portion of the bearing race component, according to one or more embodiments. FIG.16shows the formed lower portion continuing fromFIG.15for the bearing race component, according to one or more embodiments. As shown, the first portion10and the second portion10are placed together or intertwined about the forming zone10B (FIG.3) that provides for the first portion10and the second portion10to be congruently placed together. FIG.17shows the first lower portion1of a bearing race ofFIG.1coupled to a post3(or pier), according to one or more embodiments. As shown, the first lower portion1is superimposed with an I-Beam3to highlight the vertical and horizontal adjustability due to the crisscrossing of the vertical and horizontal slots. As shown, the first lower portion1is aligned such that the slots (e.g., vertical slots, openings, etc.) of the first lower portion1aligns with the post (or pier)3horizontal openings (through-holes, etc.). FIG.18shows the formed complete bearing race formed with the lower portion ofFIG.16that is fastened and coupled with a post3(or pier), according to one or more embodiments. As shown, the complete bearing race is assembled using the first and second portions10with a torque tube5(journal) placed within the completed bearing race and the lower portion having the openings aligned with those of the post3(or pier) and fastened at the lower aligned openings and at the upper tabs.FIG.18also shows the critical twisting formation at the forming zone10B to attain alignment of the two openings in the two connection tabs at forming zone10D. The hardware fasteners (e.g., bolts and nuts, etc.) may be inserted into the combined union of the fastening horizontal and vertical slots for coupling. FIG.19shows details of theFIG.13without the torque tube5(journal), according to one or more embodiments. As shown, the first and second lower portions1form the bottom half of the bearing race and highlights the ability of the bearing race cap2that may compensate for misalignment of the tabs at the forming zone1D due to an imperfect or off-center twist in the forming zone1B. FIG.20shows a plan view of a first and second lower portions1of a bearing race, according to one or more embodiments. As shown, the post3(or pier) and the lower openings of the first and second lower portions1are positioned on either side of the post3(or pier). The center location7of the first and second lower portions1are shown as being offset from the center of the first and second lower portions1. FIGS.21A-Cshow the first and second lower portions1of a bearing race ofFIG.20as moved (or rotated) together to interleave congruently to form a bearing race as shown inFIGS.18and19, according to one or more embodiments.FIG.21Ashows the view ofFIG.20. For ease of showing the transitions.FIG.21Bshows a higher level of the view inFIG.21Awhere the ninety (90) degree twist (see, e.g., forming zone1B ofFIG.7) of the first and second lower portions1are commencing to be interleaved or connected congruently.FIG.21Cshows the next transition where the interleaving or congruent connection is completed. In one embodiment, the twists or forming zones1B of the first and second lower portions1pair are each slightly offset from each other, and are symmetrically interleaved without interference to each other (as a result of the offset twisting, such as a rotation about the center point7). FIG.22shows the bearing race ofFIG.18with a rotation scissor feature for accommodating/removal of a torque tube5(journal), according to one or more embodiments. As shown, in one embodiment an upper hardware4fasteners set is removed and the lower hardware4set remains attached but loose enough to provide for the first and second portions10to pivot or rotate open and then coming to rest against the shoulders3A of flanges of the post3(or pier), which provides for ease of closing the first and second portions10to close against one another to form the bearing race. This aspect provides for ease of field assembly and disassembly with the torque tubes5of a solar tracking system. FIG.23shows the bearing race ofFIG.22with a torque tube5(journal) segment entering the scissored-open bearing in the direction of the arrow, according to one or more embodiments. As shown, once the first and second portions10are rotated apart, the torque tube5(journal) may be lowered into the formed opening of the bearing race. FIG.24shows the continuation fromFIG.23with the torque tube5(journal) segment suspended in the approximate center of an opened bearing race, according to one or more embodiments. In one embodiment, the suspension (e.g., manually, via a crane or other machinery, robotically, etc.) of the torque tube5(journal) provides for ease of installing multiple torque tubes5(journals) in the field for assembling and disassembling solar tracking systems. FIG.25shows the continuation fromFIG.24with two bearing half-races (first and second portions10) pivoting-close in the direction of the arrows to capture the torque tube5(journal) segment, according to one or more embodiments. Once the bearing race is formed by rotating the first and second portions10towards each other, the tabs may be fastened together (e.g., via hardware4, welding, etc.) and the post3(or pier) hardware4may be fastened such that the bearing race is completed with the torque tube5(journal). References in the claims to an element in the singular is not intended to mean “one and only” unless explicitly so stated, but rather “one or more.” All structural and functional equivalents to the elements of the above-described exemplary embodiment that are currently known or later come to be known to those of ordinary skill in the art are intended to be encompassed by the present claims. No claim element herein is to be construed under the provisions of 35 U.S.C. section 112, sixth paragraph, unless the element is expressly recited using the phrase “means for” or “step for.” The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the embodiments. As used herein, the singular forms “a”, “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprises” and/or “comprising,” when used in this specification, specify the presence of stated features, integers, steps, operations, elements, and/or components, but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or groups thereof. The corresponding structures, materials, acts, and equivalents of all means or step plus function elements in the claims below are intended to include any structure, material, or act for performing the function in combination with other claimed elements as specifically claimed. The description of the present embodiments has been presented for purposes of illustration and description, but is not intended to be exhaustive or limited to the embodiments in the form disclosed. Many modifications and variations will be apparent to those of ordinary skill in the art without departing from the scope and spirit of the embodiments. The embodiment was chosen and described in order to best explain the principles of the embodiments and the practical application, and to enable others of ordinary skill in the art to understand the various embodiments with various modifications as are suited to the particular use contemplated. | 19,871 |
11863119 | DESCRIPTION OF THE EMBODIMENTS FIG.1is a schematic diagram showing the disassembly of the three-dimensional part in an embodiment of the present invention. Please refer toFIG.1, disclosed in the present embodiment is a solar panel bracket with a water conducting function for carrying a plurality of solar photovoltaic panels (90), comprising: a plurality of first brackets (10), each of the first brackets (10) is arranged in parallel with each other, each of the first brackets (10) has a first water conducting groove (11); a plurality of second brackets (20), each of the second brackets (20) is arranged in parallel with each other, each of the second brackets (20) has a second water conducting groove (21), the second brackets (20) and the first brackets (10) are arranged perpendicular to each other, and the second brackets (20) and the first brackets (10) surround to form a plurality of square spaces (80), the solar photovoltaic panels (90) are arranged on the square spaces (80); and a plurality of third water conducting groove groups (30), each of the third water conducting groove groups (30) is disposed on the side of each of the first brackets (10), and each of the third water conducting groove group (30) has a third water conducting groove (31), and the second water conducting grooves (21) communicate with the third water conducting grooves (31). Taking the direction ofFIG.1as an example, when rainwater falls into a longitudinal gap between the solar photovoltaic panels (90), the rainwater would fall into the first water conducting groove (11). When the rainwater falls into a horizontal gap between the solar photovoltaic panels (90), the rainwater would fall into the second water conducting groove (21), and since the second water conducting groove (21) communicates with the third water conducting groove (31), rainwater would then fall into the third water conducting groove (31). Thereby, the embodiment of the present invention has the functions of water conduction and waterproof regardless of being in the longitudinal and horizontal directions. FIG.2is a three-dimensional exploded schematic diagram in the embodiment ofFIG.1. Please refer toFIG.2, taking the direction ofFIG.2as an example, preferably, the third water conducting groove group (30) may include a left third water guiding groove (311) and a right third water conducting groove (312), the left third water conducting groove (311) is disposed on the left side of the first bracket (10), and the right third water conducting groove (312) is disposed on the right side of the first bracket (10). Thereby, the third water conducting groove group (30) can receive the rainwater from the left and right second water conducting grooves (21), so that all the rainwater in the longitudinal and horizontal directions is in the direction of the first water conducting groove (11), so as to facilitate subsequent centralized flow conduction and management. Preferably, the first bracket (10) has two first bracket female parts (12), the third water conducting groove group (30) has two third water conducting groove male parts (32), and the third water conducting groove male parts (32) are detachably clipped into the first bracket female parts (12). Thereby, the first bracket (10) and the third water conducting groove group (30) can be temporarily and stably combined before being locked with cooperative screws, thereby improving the speed, convenience and safety of assembly. Preferably, the second bracket (20) is disposed above the third water conducting groove group (30), and the second water conducting groove (21) is in communication with the third water conducting groove (31) in a high and low manner. Thereby, the second water conducting grooves (21) communicate with the third water conducting grooves (31). FIG.3is a schematic diagram of a position to be enlarged in the embodiment ofFIG.1.FIG.4is an enlarged three-dimensional schematic diagram of position A in the embodiment ofFIG.3.FIG.5is an enlarged cross-sectional schematic diagram of position A in the embodiment ofFIG.3. Please refer toFIGS.3to5simultaneously, preferably, the present embodiment may further include a plurality of fixing devices (40), and the fixing devices (40) cooperate with the first brackets (10) to fix the solar photovoltaic panels (90). Thereby, the solar photovoltaic panels (90) can be temporarily and stably fixed before being locked with the cooperative screws, thereby improving the speed, convenience and safety of the assembly. Preferably, the first bracket (10) has two sliding grooves (13), the fixing device (40) has two sliding sheets (41) and two fixing sheets (42), and the two sliding sheets (41) are slidably disposed in the two sliding grooves (13). Thereby, the fixing position of the fixing device (40) can be easily adjusted before being locked with the cooperative screw to provide better stress. FIG.6is a three-dimensional exploded schematic diagram in another embodiment of the present invention.FIG.7is an enlarged cross-sectional schematic diagram in the embodiment ofFIG.6. Please refer toFIG.6andFIG.7again, preferably, the fixing device (40) may further have a hook plate (43), and the hook plate (43) is U-shaped with two wings (431) extending inwards. Preferably, the solar photovoltaic panel (90) has two legs (91), and the two wings (431) are movably hooked on the legs (91). Thereby, with the two wings (431) and the legs (91), the solar photovoltaic panels (90) can be temporarily and stably fixed before being locked with the cooperative screws, thereby improving the speed, convenience and safety of the assembly. In addition, the fixing device (40) can be integrally formed, or as shown in the figure, it can be composed of components in different parts. Please refer toFIG.1again, preferably, the present embodiment may further include a basic bracket group (70) with a plurality of uprights, wherein the heights of the uprights connected to the same second bracket (20) are all the same, and the heights of the uprights connected to the different second brackets (20) are different. Describing from a different perspective, preferably, the basic bracket group (70) includes a first upright group (71) and a second upright group (72), the first upright group (71) and the second upright group (72) are connected to different second brackets (20), and the height of the first upright group (71) is greater than the second upright group (72). Thereby, the embodiment of the present invention can provide the solar panel bracket with a slope and a water conducting function, which has achieved a better water conducting effect. By analogy, preferably, the basic bracket group (70) may include a third upright group (73), or more. Thereby, the embodiment of the present invention can stably provide the slope and support. FIG.8is a three-dimensional exploded schematic diagram in another embodiment of the present invention.FIG.9is an enlarged cross-sectional schematic diagram in the embodiment ofFIG.8. Please refer toFIGS.8and9, in order to strengthen the waterproof effect of the second bracket (20), preferably, the outer side of the bottom of the second bracket (20) may have a second bracket side wing (201), the second bracket side wing (201) can be screwed through by a screw. Thereby, the overall waterproof effect can be strengthened by avoiding directly passing through the second water conducting groove (21). Similarly, the first bracket (10) and the third water conducting groove group (30) can also have similar flanks, which will not be repeated here. In addition, the outer side of the top of the second bracket (20) may have a second bracket blocking wing (202), which can movably block the two legs (91), so that the solar photovoltaic panels (90) can be temporarily and stably fixed, thereby improving the speed, convenience and safety of the assembly. To sum up, through the cooperation of the first bracket, the second bracket and the third water conducting groove group, the embodiment of the present invention can provide the solar panel bracket with the water conducting function, which can conduct water longitudinally and horizontally at the same time, or collect longitudinal and horizontal falling water together into the same direction. In addition, the solar panel bracket with the water conducting function provided in the embodiment of the present invention is capable of quick or safe assembly. DESCRIPTION OF SYMBOLS a first bracket (10)a first water conducting groove (11)a first bracket female part (12)groove (13)a second bracket (20)a second bracket side wing (201)a second bracket blocking wing (202)a second water conducting groove (21)a third water conducting groove group (30)a third water conducting groove (31)a left third water conducting groove (311)a right third water conducting groove (312)a third water conducting groove male part (32)a fixing device (40)a sliding sheet (41)a fixing sheet (42)a hook plate (43)a swing (431)a basic bracket group (70)a first upright group (71)a second upright group (72)a third upright group (73)a square space (80)a solar photovoltaic panel (90)a leg (91) | 9,186 |
11863120 | DETAILED DESCRIPTION The illustrations presented herein are not meant to be actual views of any particular solar array or component thereof, but are merely idealized representations employed to describe illustrative embodiments. The drawings are not necessarily to scale. As used herein, the term “substantially” in reference to a given parameter means and includes to a degree that one skilled in the art would understand that the given parameter, property, or condition is met with a small degree of variance, such as within acceptable manufacturing tolerances. For example, a parameter that is substantially met may be at least about 90% met, at least about 95% met, at least about 99% met, or even at least about 100% met. In another example, a substantially straight line may be a line wherein an angle between individual segments of the line vary from 180° by less than about 10°, such as less than about 5° or even less than about 1°. As used herein, relational terms, such as “first,” “second,” “top,” “bottom,” etc., are generally used for clarity and convenience in understanding the disclosure and accompanying drawings and do not connote or depend on any specific preference, orientation, or order, except where the context clearly indicates otherwise. As used herein, the term “and/or” means and includes any and all combinations of one or more of the associated listed items. As used herein, the terms “vertical” and “lateral” refer to the orientations as depicted in the figures. Solar arrays may be formed from multiple composite solar panels. The size of the solar array may be limited by the volume (e.g., thickness or height) required for the multiple composite solar panels, when in the stowed position. For example, a typical composite solar panel may have a thickness of between about 1 centimeters (cm) and about 2 cm. Moreover, when in a stowed or folded position adjacent composite solar panels may require an additional 0.5 cm to about 1 cm of space between the adjacent composite solar panels. The solar array may be deployed on equipment for use in space. The equipment may be transported to space on a vehicle configured to travel from the surface of the earth to space, such as a rocket or other spacecraft. The vehicle may have significant limitations on available payload volume and weight. The volume and weight restrictions may limit the size of the equipment or accessories that may be included with the equipment. Thus, reducing the volume, and specifically the height required by the solar arrays in a stowed position may allow additional accessories to be included on the equipment. The size of the equipment may also be able to increase if the volume required by the solar arrays in the stowed position is decreased. In other cases, reducing the volume and/or weight required for the solar arrays in the stowed position may allow additional equipment to be transported in the same vehicle. FIG.1illustrates a solar array100extending from a body102of a piece of equipment, such as a space craft, a satellite, etc. The solar array100may include a base composite panel104, a top composite panel106, and multiple panel modules108extending between the base composite panel104and the top composite panel106. The panel modules108may be substantially thinner and lighter than the base composite panel104and the top composite panel106, as described in more detail below in the descriptionFIG.3. The panel modules108may each have a surface area that is substantially the same as at least one of the base composite panel104or the top composite panel106. Thus, the solar array100may have an active surface area substantially the same as if each of the panel modules108were also composite panels; however, the assembly may weigh much less than a similar sized solar array entirely formed from composite solar panels. The base composite panel104and the top composite panel106may be coupled to opposite ends of a mast110(e.g., boom, arm, etc.). For example, the base composite panel104may be coupled to the mast110with a bracket112. In some embodiments, the bracket112may be part of the mast110. For example, the bracket112may be one or more flanges extending laterally from the sides of the mast110. In some embodiments, the bracket112may be an additional element configured to interface with the mast110. For example, the bracket112may extend over the mast110with connection points extending on either end to interface with the base composite panel104. In other embodiments, the bracket112and the mast110may include complementary geometry configured to interlock coupling the mast110to the bracket112. The bracket112may be coupled to the base composite panel104through a hardware connection (e.g., screws, bolts, rivets, studs, nuts, etc.) or an adhesive connection (e.g., epoxy, glue, tape, etc.). The top composite panel106may be coupled to a distal end of the mast110with a movable connection, such as a top hinge114. The top hinge114may couple the distal end of the mast110to a top edge of the top composite panel106. As described in further detail below with reference toFIGS.16A-16D, the top hinge114may allow the top composite panel106to rotate relative to the distal end of the mast110as the solar array100extends from the body102. The mast110may be coupled to the body102through another moveable connection between a proximal end of the mast110and the body102. The movable connection may be a root hinge116. As described in further detail below with reference toFIGS.17A and17B, the root hinge116may allow the mast110and the solar array100, through its connection to the mast110, to rotate relative to the body102when the solar array100is being deployed. The root hinge116may also be configured to latch the mast110and the solar array100in position once the solar array100is deployed. FIG.2illustrates a composite solar panel200, such as the base composite panel104or the top composite panel106. The composite solar panel200may include multiple solar cells202arranged on a surface of the composite solar panel200. The solar cells202may be coupled to the composite solar panel200with an adhesive layer204. The adhesive layer204may couple the solar cells202to a substrate206. The substrate206may include a top film208, commonly referred to as a facesheet, and a bottom film210sandwiching a core212. The top film208may include an electrically insulating coating or film, such as KAPTON® configured to substantially prevent electrical connections between the solar cells202through the top film208. The sandwich construction may improve strength and stiffness characteristics of the substrate206. For example, the core212may be formed from a flexible honeycomb structured material, such as aluminum honeycomb and the top film208and the bottom film210may be formed from a thin material having a relatively high tensile strength and stiffness, such as a graphite. Spacing the top film208and the bottom film210with the core212may increase a bending resistance of the substrate206, increasing the rigidity of the substrate206. The increased rigidity may allow the substrate206to substantially protect the solar cells202from damage due to bending. In some embodiments, the substrate206may include multiple sandwich layers. For example, the substrate206may include two layers of core212having the top film208above the first layer of core212and the bottom film210below the second layer of core212and an additional intermediate film layer between the first layer of core212and the second layer of core212. The substrate206may allow additional structural elements, such as hinges, latches, tensioners, etc., to be mounted to the composite solar panel200. For example, some areas of the composite solar panel200may be free from solar cells202. These areas may include holes passing through the substrate206sized and configured to receive hardware connections, electrical devices, electrical connectors or wiring, etc. In some embodiments, the additional structural elements may be coupled to the substrate206with an adhesive (e.g., epoxy, glue, etc.). As described above, the composite solar panel200may have a thickness of between about 1 cm and about 2 cm. The majority of the thickness of the composite solar panel200may result from the thickness of the substrate206. FIG.3illustrates an exploded view of a panel module108. The panel module108may be much thinner than the composite solar panel200. For example, the panel module108may not include a substrate206or similar structure. Thus the panel module108may be relatively flexible (e.g., not rigid). The panel module108may include multiple solar cells302secured to a flexible cell support304. The cell support304may be a thin sheet of material, such as an open weave scrim material. The cell support304may be formed from synthetic fiber materials, such as VECTRAN®, or KEVLAR® or fiber glass. The material of the cell support304may be stretched flat by a front frame306and a rear frame308. The front frame306and the rear frame308may substantially surround the cell support304and sandwich the cell support304between the front frame306and the rear frame308, such that the cell support304may remain substantially planar as defined by the front frame306and/or the rear frame308. The front frame306and the rear frame308may be formed from thin strips of material, such as a graphite or an epoxy configured to provide structural support to the solar cells302and the cell support304. The thin strips of material may have a width that is greater than a thickness of the thin strips of material. For example, the thin strips may have a width of between about 0.10 inches (in) (2.54 millimeters (mm)) and about 0.20 in (5.08 mm), such as between about 0.15 in (3.81 mm) and about 0.19 in (4.83 mm) or about 0.18 in (4.57 mm). The thin strips may have a thickness of between about 0.020 in (0.508 mm) and about 0.030 in (0.762 mm), such as between about 0.022 in (0.559 mm) and about 0.026 in (0.660 mm) or about 0.024 in (0.610 mm). The greater width may allow the thin strips to provide greater strength in an outward direction to maintain the tension in the cell support304, while maintaining a relatively thin profile for the panel module108. The thin strips of material may be coupled to one another to form the front frame306and rear frame308through multiple corner supports310. The corner supports310may be configured to form a substantially rectangular frame by joining the thin strips of material at substantially 90° angles. In some embodiments, the corner supports310may further allow the panel modules108to be connected to one another and may place the majority of the loads (e.g., stress, tension, impact, etc.) in the thin strips of material forming the front frame306and the rear frame308rather than in the cell support304or the solar cells302. For example, the corner supports310may have a substantially complementary geometry with one another, such that each corner support310may be joined with an adjacent corner support310to form a hinged connection. In some embodiments, one or more sides of the front frame306and/or the rear frame308may be formed from a cable (e.g., lanyard, string, fiber, etc.). For example, lateral sides316of at least one of the front frame306or the rear frame308may be formed from a cable, such as a glass fiber lanyard, carbon fiber lanyard, graphite lanyard, etc. The lateral sides316may be the sides of the panel module108substantially aligned with the extension direction of the solar array100. For example, the lateral side316may be substantially perpendicular to the joint side318, which may be the side of the panel module108configured to abut against a joint side318of a proximate panel module108or to the base composite panel104or top composite panel106. As described in further detail below with respect toFIG.4, the cable may form a substantially continuous cable connection between the base composite panel104and the top composite panel106passing along the lateral sides316of each of the intervening panel modules108and coupled to each of the corner supports310. The front frame306and/or rear frame308may further include a wire support312on at least two sides of the panel module108. The wire support312may be configured to position wires (e.g., electrical wiring, control wiring, transmission wiring, etc.) along the lateral sides316of the front frame306and the rear frame308. The wire support312may be positioned in a central portion of the thin strips of material forming the respective sides of the panel module108. In some embodiments, the wire support312may be positioned on the lateral side316of the panel module108. The portions of the front frame306and the rear frame308in the lateral sides316may be configured to bear the tension loads exerted on the panel module108. The tension loads may be configured to maintain the base composite panel104, top composite panel106, and the panel modules108of the solar array100in substantially the same plane. The panel module108may include multiple cushion strips314arranged on a rear side of the panel module108(e.g., a side of the panel module108opposite the solar cells302). The cushion strips314may be configured to allow the panel module108, while in a stowed position, to rest on an adjacent panel module108without damaging the adjacent panel module108while reducing a distance between the panel modules108to substantially a thickness of the cushion strips314. The cushion strips314may have a thickness of between about 0.020 in (0.508 mm) and about 0.030 in (0.762 mm), such as between about 0.022 in (0.559 mm) and about 0.026 in (0.660 mm) or about 0.024 in (0.610 mm). In some embodiments, the cushion strips314may be formed from a resilient material, such as a foam, a rubber or other elastomer material, a polymer etc. The cushion strips314may be coupled to the cell support304with an adhesive, such as a tape (e.g., polymide adhesive tape, KAPTON®, etc.), an epoxy, glue, etc. The cushion strips314may be arranged on the panel module108such that the cushion strips314are parallel to and offset from the cushion strips314on the adjacent panel module108. Arranging the cushion strips314to be parallel and offset from the adjoining cushion strips314may allow the panel modules108to rest against one another with a cushioned contact area substantially equal to twice that of the cushion strips314of each individual panel module108because the cushion strips314of each of the individual panel modules108may contact surfaces of an adjacent panel module108. Furthermore, as the cushion strips314of the adjacent panel modules108may not contact cushion strips314on the opposing panel modules108the standoff between the adjacent panel modules108when stowed is minimized. FIG.4illustrates two panel modules108joined together at a joint402. The joint402may be formed along the joint sides318of the two adjacent panel modules108. The panel modules108may be joined together at the joint402with hinges404formed from an interface between the respective corner supports310. For example, as described above, the corner supports310may include complementary geometry along the joint sides318of the respective panel modules108configured to form a hinged connection between the adjoining corner supports310and therefore the adjoining joint sides318of the respective panel modules108. The hinges404may allow the panel modules108to rotate relative to one another about an axis defined along the joint402. The hinges404may also restrict translational movement between the adjacent panel modules108in both a direction along the axis defined by the joint402and a direction transverse to the axis defined by the joint402. A cable406may be coupled between the corner supports310of each panel module108and the wire support312on each lateral side316of each panel module108. In some embodiments, the cable406may form a side of one or more of the front frame306or the rear frame308as described above. In some embodiments, the cable406may be distinct from the front frame306and the rear frame308. For example, the cable406may be configured to provide additional support along the lateral sides316of the panel modules108. In some embodiments, the cable406may have similar geometry to the thin strips of material used to form the front frame306and the rear frame308. Tension in the cable406may be transmitted between the adjacent panel modules108through the hinges404such that the connections between the cable406, and the corner supports310may form a substantially constant tension member along the length of the joined panel modules108. In some embodiments, the cable406may pass through the hinges404such that the cable406is substantially constant between the base composite panel104and the top composite panel106forming the tension member without the hinges404. Electrical wiring408for carrying solar cell power, signals, and/or control data, such as telemetry, temperature, etc., to the spacecraft may run along the lateral sides316of the panel modules108. The wiring408may be secured to the wire support312and to the corner supports310. For example, the corner supports310and the wire support312may be configured to protect the wiring408at interface points between the panel modules108and other components of the solar array100. In some embodiments, the corner supports310and/or the wire support312may be positioned at wiring interfaces of the respective panel module108, such that the wiring408may form electrical connections with the panel module108at points adjacent to the corner supports310and/or the wire support312. In some embodiments, the corner supports310and/or the wire support312may be configured to substantially protect the connection point from damage, such as fatigue damage, damage from movement, damage from impacts, etc. FIG.5illustrates the connection point between the base composite panel104and the first panel module108. The base composite panel104may connect to the first panel module108through a base connection502. The base connection502may be coupled to the corner support310adjacent to the base composite panel104. The base connection502may be configured to form a hinged connection between the base composite panel104and the first panel module108. For example, the base connection502may allow the panel module108to rotate relative to the base composite panel104about an axis defined substantially along the joint side318of the panel module108adjacent to the base composite panel104. The base connection502may include a preload assembly504configured to maintain tension through the tension member formed by the cable406or the cable406and the corner supports310. For example, when deployed in space the solar array100may experience significant changes in temperature. Such significant changes in temperature may change the tension in the tension member as the components of the tension member expand and/or contract in response to the temperature changes. As the components expand or contract the tension may require adjustment to maintain the base composite panel104, top composite panel106, and the intervening panel modules108in substantially the same plane. FIGS.6A and6Billustrate enlarged views of the preload assembly504in two different positions.FIG.6Aillustrates the preload assembly504in an unloaded (e.g., no pre-load, resting, etc.) position. The preload assembly504may include a stationary flange602and a moving flange604coupled together by multiple tensioner elements606. The cable406may be coupled to the moving flange604through a cable connection608. The moving flange604may be configured to translate relative to the stationary flange602. The translation of the moving flange604may be limited by a top detent610and a bottom detent612. For example, the base composite panel104may include a stop614configured to interface with the top detent610and/or the bottom detent612substantially preventing the moving flange604from translating a distance greater than the distance between the top detent610and the bottom detent612. In some embodiments, the stop614may include a substantially round surface configured to allow the moving flange604to freely move against the stop614until the stop614encounters the top detent610or the bottom detent612. For example, the stop614may include a roller, such as a wheel or a bearing. The stationary flange602may remain substantially stationary relative to the base composite panel104. The stationary flange602may be secured to the base composite panel104through one or more anchors616. The anchors616may be a hardware connection, such as a pin, screw, clamp, stud, etc. The anchors616may be configured to maintain the stationary flange602in substantially a same position, such that movement of the stationary flange602in any direction is substantially prevented. The tensioner elements606may be configured to restrict movement of the moving flange604relative to the stationary flange602. For example, the tensioner elements606may act as a biasing element biasing the moving flange604towards a resting position as illustrated inFIG.6A. FIG.6Billustrates the preload assembly504in a loaded position. In the loaded position the moving flange604may be translated relative to the stationary flange602such that the tensioner elements606are applying a load (e.g., preload, biasing load) on the moving flange604, effectively biasing the moving flange604in a direction away from the panel modules108. The tensioner elements606may be configured to increase the load as the moving flange604moves away from the resting position and decrease the load as the moving flange604moves toward the resting position. Changing the load may effectively maintain the tension in the cable406as the components of the solar array100expand and contract due to changing thermal conditions in space. The stationary flange602may include multiple adjustment apertures618configured to adjust the magnitude of the preload on the cable406. For example, the adjustment apertures618may allow the position of the stationary flange602to be changed by moving the location where the anchors616are coupled to the stationary flange602. Moving the position of the stationary flange602may move the resting position of the preload assembly504. As described above, the load exerted by the tensioner elements606may be proportional to the distance of the moving flange604from the resting position. Thus, moving the resting position away from the panel module108may increase the preload exerted on the cable406by the preload assembly504and moving the resting position toward the panel module108may decrease the preload exerted on the cable406by the preload assembly504. FIGS.7-10illustrate the solar array100in a stowed configuration. In the stowed configuration each of the base composite panel104, top composite panel106, and the panel modules108may be positioned such that the base composite panel104, top composite panel106, and panel modules108are substantially mutually parallel and substantially parallel to the plane of the body102. For example, the mast110may be positioned in a direction substantially parallel to the plane of the body102. As described in further detail below, the mast110may be configured to extend and retract, such that in the stowed configuration the mast110may be substantially the same length as the base composite panel104and/or the top composite panel106. The mast110may be positioned between the base composite panel104and the body102. The mast110may be coupled to the body102by the root hinge116. The connection between the root hinge116and the mast110may effectively couple the solar array100to the body102. In the stowed configuration, the root hinge116may be configured to secure at least one end of the solar array100through the connection to the mast110. The body102may further include side latches704and a top latch702. The side latches704may be configured to secure the sides of the solar array100to the body102in the stowed configuration and the top latch702may be configured to secure an end of the solar array100opposite the root hinge116to the body102. The combination of the root hinge116, the side latches704, and the top latch702may position the base composite panel104, top composite panel106, and the panel modules108in a substantially parallel orientation relative to the body102. The base composite panel104may be the lowermost panel (e.g., closest panel to the body102). A void may be defined between the body102and the base composite panel104in the stowed position. For example, the void may be at least sufficient to allow the mast110to be positioned between the base composite panel104and the body102. The connections between the side latches704, top latch702, and the root hinge116may be configured to secure the solar array100during high stress and/or high load conditions, such as launch, loading, etc. The side latches704, top latch702, and root hinge116combined with the substantially rigid base composite panel104and top composite panel106may secure and protect the panel modules108during the high stress and/or high load conditions. The rigidity of the base composite panel104and the top composite panel106may substantially prevent damage to the solar cells302of the panel modules108when in the stowed position. In the stowed position the top composite panel106may be positioned such that the solar cells202are directed (e.g., facing) outward away from the body102. This may allow the solar array100to generate power even when in the stowed position. Power generated when in the stowed position may charge batteries, power controllers, and/or power components of the equipment such as to extend the solar array100, etc. The top composite panel106may include latch receivers802configured to receive the side latches704. The side latches704may be configured to pass through the base composite panel104and secure the top composite panel106through the latch receivers802, such that the panel modules108are sandwiched between the base composite panel104and the top composite panel106. In some embodiments, the base composite panel104may include a latch receiver configured to receive a portion of an associated side latch704. For example, the side latch704may be configured to position the base composite panel104and the top composite panel106relative to the body102and relative to one another. In some embodiments, the side latch704may be configured to independently release the top composite panel106and the base composite panel104, such that one of the top composite panel106and the base composite panel104may be released before the other base composite panel104or top composite panel106is released. In the stowed configuration the solar array100may include connection assemblies804on opposing corners of the solar array100. The connection assemblies804may be positioned in substantially the same position as the hinges between the base composite panel104, the panel modules108, and the top composite panel106. Thus, the connection assemblies804may be positioned on the ends of the solar array100that are parallel with the joint sides318of the panel modules108. The joint sides318of the panel modules108may be on the same sides of the solar array100as the top hinge114and/or the root hinge116. FIG.10illustrates an enlarged view of a connection assembly804. The connection assembly804may include a stop1002and a stop plate1004. The stop1002and the stop plate1004may be configured to maintain a separation between the base composite panel104and the top composite panel106. For example, the separation may be configured to allow the panel modules108to be folded between the base composite panel104and the top composite panel106. The stop1002may include fingers1006configured to extend from the stop1002into the space defined between the base composite panel104and the top composite panel106. For example, the fingers1006may be configured to extend between each of the adjacent panel modules108, securing each of the panel modules108relative to the base composite panel104and the top composite panel106. The fingers1006may substantially prevent movement of the panel modules108relative to one another in the stowed position. Preventing movement of the panel modules108may substantially prevent damage due to impacts, and/or vibrations during high stress and/or high load conditions. As illustrated inFIG.10, the cable406may follow the panel modules108through the hinged points between the panel modules108. For example, the cable406may bend around the hinges404between adjacent panel modules108. The panel modules108may be folded in a “Z” pattern between the base composite panel104and the top composite panel106such that the hinges between adjacent panel modules108are on opposing ends of the solar array100. The “Z” pattern may cause the panel modules108to be arranged in a front-to-front or back-to-back arrangement with adjacent panel modules108. The cable406may enter the top composite panel106through the stop plate1004. The cable406may enter the base composite panel104on an opposite end of the solar array100. The top composite panel106may include a panel module clip1008extending from the top composite panel106and configured to secure the panel modules108to the top composite panel106in the stowed position. The panel module clip1008may include a latch bar1010. The latch bar1010may be configured to interface with a bottom strike1012extending from a hinge404between two panel modules108. For example, the bottom strike1012may extend from the lowermost hinge404in the stack of panel modules108of the solar array100in the stowed position. The bottom strike1012may be configured to release from the latch bar1010when the lowermost panel module108begins to form an angle with the top composite panel106moving a position or orientation of the bottom strike1012. The stack of panel modules108may further include one or more intermediate strikes1014configured to interface with the latch bar1010after the bottom strike1012is released from the latch bar1010. The intermediate strikes1014may be configured to stage the panel modules108when the solar array100is moved from the stowed position to the extended position. FIGS.11A through11Cillustrate views of the top latch702. The top latch702may be configured to interface with the top hinge114. For example, the top latch702may be configured to latch to the top composite panel106through the connection between the top composite panel106and the top hinge114. The top hinge114may include a latching interface1102configured to receive a latching component associated with the top latch702and to rest against the top latch702. The latching component may be a fracture bolt1112. The fracture bolt1112may be threaded into the latching interface1102of the top hinge114. The top latch702may include a release mechanism1104configured to fracture the fracture bolt1112releasing the top hinge114from the top latch702. The release mechanism1104may be a solenoid or other electronic component, such as a FRANGIBOLT® actuator, configured to increase a tensile force on the fracture bolt1112to beyond the fracture limit of the fracture bolt1112on an electrical signal. The top latch702may also include a latching mechanism1106configured to latch the fracture bolt1112within the top latch702. For example, the fracture bolt1112may include a head1114configured to interface with the latching mechanism1106. In some embodiments, the head1114may be a fitting threaded onto the fracture bolt1112, such as a nut, wingnut, or other threaded component. In other embodiments, the head1114may be formed into the fracture bolt1112, such as the head of a bolt. The latching mechanism1106may secure the top hinge114to the top latch702through the fracture bolt1112. The fracture bolt1112may be secured to the top hinge114through the threaded connection between the fracture bolt1112and the latching interface1102of the top hinge114. The fracture bolt1112may also be secured to the latching mechanism1106through the interface between the head1114of the fracture bolt1112and the latching mechanism1106. Thus, until the fracture bolt1112is broken by the release mechanism1104, the fracture bolt1112may secure the top hinge114to the latching mechanism1106of the top latch702. The top latch702and the top hinge114may include complementary geometry configured to locate the top hinge114relative to the top latch702. For example, the top latch702may include a mating surface1110. The mating surface1110may include a recess, such as a frustoconical recess, configured to locate the top hinge114relative to the top latch702. In some embodiments, the mating surface1110may be configured to align the fracture bolt1112in the release mechanism1104, such that the tensile forces for fracturing the fracture bolt1112may be substantially aligned with a longitudinal axis of the fracture bolt1112enabling a substantially clean break of the fracture bolt1112. The top latch702may include a retainer1116configured to retain the broken elements of the fracture bolt1112after the fracture bolt1112is broken by the release mechanism1104. The retainer1116may be a space between the latching mechanism1106and the body102. The retainer1116may be configured to retain the head1114of the fracture bolt1112within a void defined by the retainer1116. Retaining the broken elements of the fracture bolt1112may substantially prevent damage to other components of the equipment and/or damage to the solar array100that may be caused by allowing the broken components of the fracture bolt1112to freely float, fall, etc., after being broken to release the top hinge114from the top latch702. The top latch702may include a wiring connection1108configured to connect the components of the top latch702to a central controller. For example, the central controller may send a release signal. The release signal may be provided to the release mechanism1104through the wiring connection1108. The release mechanism1104may then fracture the fracture bolt1112upon receipt of the release signal through the wiring connection1108. In some embodiments, the top latch702may send signals back to the controller through the wiring connection1108, such as status signals. FIGS.12A through12Cillustrate views of the side latches704. The side latch704may be configured to interface with the latch receivers802in the base composite panel104and the top composite panel106. The latch receivers802may include a base plate1214and a top plate1216. The base plate1214and the top plate1216may be configured to sandwich the respective base composite panel104or top composite panel106. The latch receiver802may also include an interface1226configured to interface with a latching component. In some embodiments, the interface1226may include a threaded interface. In other embodiments, the interface1226may include a tapered surface, such as a frustoconical surface. The latching component may be a fracture bolt1210. The fracture bolt1210may be threaded into the interface1226the latch receiver802of at least one of the base composite panel104or the top composite panel106. In some embodiments, the interface1226of the latch receiver802of the top composite panel106may be a threaded interface and the interface1226of the latch receiver802of the base composite panel104may be a tapered surface. The fracture bolt1210may be threaded into the interface1226of the latch receiver802of the top composite panel106. The fracture bolt1210may have a complementary taper configured to interface with the tapered surface of the interface1226of the latch receiver802of the base composite panel104. The space between the threads and the complementary taper of the fracture bolt1210may define a space between the base composite panel104and the top composite panel106wherein the folded panel modules108may reside in the stowed position. Once released the fracture bolt1210may remain threaded into the interface1226of the latch receiver802of the top composite panel106and may be slidably removed from the tapered interface1226of the latch receiver802of the base composite panel104, such that the fracture bolt1210no longer forms a latching connection between the base composite panel104and the top composite panel106after the side latch704is released. The side latch704may include a release mechanism1208configured to fracture the fracture bolt1210releasing the base composite panel104and the top composite panel106from the side latch704. The release mechanism1208may be a solenoid or other electronic component, such as a FRANGIBOLT® actuator, configured to increase a tensile force on the fracture bolt1210to beyond the fracture limit of the fracture bolt1210responsive to an electrical signal. In some embodiments, the fracture bolt1210may include a fracture point1212(e.g., a weak point) defined in an area of the fracture bolt1210. The fracture point1212may be configured to create stress concentrations in a specific part of the fracture bolt1210, such that the fracture bolt1210may break at the fracture point1212when the release mechanism1208breaks the fracture bolt1210. The fracture point1212may be defined in the fracture bolt1210at a point below or within the latch receivers802in the base composite panel104. The side latch704may also include a latching mechanism1206configured to latch the fracture bolt1210within the side latch704. For example, the fracture bolt1210may include a head1224configured to interface with the latching mechanism1206. In some embodiments, the head1224may be a fitting threaded onto the fracture bolt1210, such as a nut, wingnut, or other threaded component. In other embodiments, the head1224may be formed into the fracture bolt1210such as the head of a bolt. The latching mechanism1206may secure the base composite panel104and the top composite panel106to the side latch704through the fracture bolt1210. The fracture bolt1210may be secured to the latch receivers802in at least one of the base composite panel104and the top composite panel106through the threaded connection between the fracture bolt1210and the latch receivers802of the base composite panel104and/or the top composite panel106. The fracture bolt1210may also be secured to the latching mechanism1206through the interface between the head1224of the fracture bolt1210and the latching mechanism1206. Thus, until the fracture bolt1210is broken by the release mechanism1208, the fracture bolt1210may secure the latch receivers802of the base composite panel104and/or the top composite panel106to the latching mechanism1206of the side latch704. The side latch704and the latch receiver802of the base composite panel104may include complementary geometry configured to locate the latch receiver802of the base composite panel104relative to the side latch704. For example, the side latch704may include a mating surface1220. The mating surface1220may include a recess, such as a frustoconical recess, configured to locate the latch receiver802of the base composite panel104relative to the side latch704. In some embodiments, the mating surface1220may be configured to align the fracture bolt1210in the release mechanism1208, such that the tensile forces for fracturing the fracture bolt1210may be substantially aligned with a longitudinal axis of the fracture bolt1210enabling a substantially clean break of the fracture bolt1210. The latch receiver802of the base composite panel104and the latch receiver802of the top composite panel106may include an intermediate interface1218similar to the mating surface1220. The intermediate interface1218may be configured to locate the top composite panel106relative to the base composite panel104. For example, the intermediate interface1218may include a recess, such as a frustoconical recess configured to locate the latch receiver802of the top composite panel106relative to the latch receiver802of the base composite panel104. The side latch704may include a retainer1222configured to retain the broken elements of the fracture bolt1210after the fracture bolt1210is broken by the release mechanism1208. The retainer1222may be a void defined within the side latch704in an area between the latching mechanism1206and the body102. The retainer1222may be configured to retain the head1224of the fracture bolt1210within the void defined by the retainer1222. Retaining the broken elements of the fracture bolt1210may substantially prevent damage to other components of the equipment and/or damage to the solar array100that may be caused by allowing the broken components of the fracture bolt1210to freely float, fall, etc., after being broken to release the base composite panel104and the top composite panel106from the side latch704. The side latch704may include a wiring connection1204configured to connect the components of the side latch704to a central controller. For example, the central controller may send a release signal. The release signal may be provided to the release mechanism1208through the wiring connection1204. The release mechanism1208may then fracture the fracture bolt1210upon receipt of the release signal through the wiring connection1204. In some embodiments, the side latch704may send signals back to the controller through the wiring connection1204, such as status signals. The side latch704may include a base1202configured to mount the side latch704to the body102. The base1202may be configured to space the side latch704from the body102to position the base composite panel104and the top composite panel106relative to the body102. For example, the base1202may be configured to position the base composite panel104and the top composite panel106such that the base composite panel104and the top composite panel106are at substantially the same level as defined by the top latch702and the root hinge116connections between the body102and the solar array100. FIGS.13,14, and15illustrate acts of extending the solar array100from the stowed position described above to an extended position.FIG.13illustrates the solar array100after a first extension movement. The first extension movement may occur after the side latches704and the top latch702release the solar array100from the stowed position in the manner described above. The base composite panel104may rotate about the root hinge116relative to the body102until the base composite panel104is substantially perpendicular to the body102(e.g., extending away from the body102at about a 90° angle). The mast110may be coupled to both the root hinge116and the base composite panel104. For example, the base composite panel104may be coupled to the root hinge116through the mast110. Thus, the mast110may also rotate about the root hinge116relative to the body102until the mast110extends away from the body102at about a 90° angle. In some embodiments, the root hinge116may include a biasing element, such as a spring, configured to bias the mast110toward the perpendicular extended position. The top hinge114coupled between the mast110and the top composite panel106may cause the top composite panel106to extend at about a 90° angle to the base composite panel104, such that the top composite panel106is substantially parallel to the body102and spaced a distance from the body102defined by the base composite panel104. The panel modules108may be retained on the bottom surface of the top composite panel106by the connection assemblies804. As described above, the connection assemblies804may include a panel module clip1008configured to secure the panel modules108to the top composite panel106with the latch bar1010and the bottom strike1012and/or the intermediate strikes1014. Therefore, the panel modules108may be substantially parallel to the body102and spaced a distance from the body102defined by the base composite panel104. FIG.14illustrates the solar array100during a second extension movement. The second extension movement may include extending the mast110. As described below with respect toFIG.18, the mast110may be a telescoping shaft including multiple nested segments. As the mast110extends away from the body102, the top hinge114connecting the top composite panel106to the mast110may cause the top composite panel106to move away from the body102and the base composite panel104. As the top composite panel106moves away from the base composite panel104, the panel modules108may unfold from beneath the top composite panel106. For example, the panel module clip1008may release the bottom strike1012and the intermediate strikes1014as the panel modules108align below the top composite panel106. The panel modules108may align with one another (e.g., be arranged in a substantially straight line) by rotating about the hinges404between the respective panel modules108. The panel modules108may all align with one another to form a flexible array of panel modules108, such as a solar blanket. The panel modules108may be tied together by the hinges404. In some embodiments, the cables406may pass through the hinges404tying the panel modules108together as well. The cables406may be coupled to the base composite panel104and the top composite panel106, such that the base composite panel104may be coupled to the top composite panel106through the cables406and/or the array of panel modules108. As illustrated inFIG.14, the top composite panel106may be maintained in a position substantially perpendicular to the base composite panel104by the top hinge114. Thus as the panel modules108unfold from beneath the top composite panel106, the panel modules108may align at an angle relative to the base composite panel104. FIG.15illustrates the solar array100fully extended after the second extension movement. The mast110may continue to extend until all of the panel modules108are unfolded and aligned. Once the panel module108are all unfolded and aligned the top composite panel106may pivot about the top hinge114to align with the base composite panel104. The mast110may then continue to extend until the entire solar array100is aligned with the base composite panel104. The top composite panel106and the base composite panel104may be coupled to the mast110as described above. The solar array100may be coupled to the body102through the mast110, such as through the root hinge116connection between the body102and the mast110. Tension in the cables406may pull the panel modules108into alignment with the base composite panel104and the top composite panel106. The tension in the cables406may also absorb loads, such as shocks, impacts, etc., protecting the solar cells302of the panel modules108from such loads. The length of the extended solar array100may be substantially determined by the number of panel modules108included between the base composite panel104and the top composite panel106. In some example embodiments, the solar array100may include between about 1 panel module108and about 13 panel modules108between the base composite panel104and the top composite panel106, such as between about 3 panel modules108and about 13 panel modules108, or between about 5 panel modules108and about 9 panel modules108. FIGS.16A through16Dillustrate enlarged views of the top hinge114through the different positions of the solar array100described above.FIG.16Aillustrates the top hinge114when the solar array100is in the stowed position. The top hinge114may include a mounting flange1606configured to secure the top hinge114to the top composite panel106. The top hinge114may be configured to couple the mast110to the top composite panel106through the mounting flange1606. The top hinge114may include one or more biasing elements1602, such as springs, torsion springs, spring clips, etc. The biasing elements1602may be operatively coupled between the mast110and the top composite panel106. For example, the biasing elements1602may be coupled to the mast110and the mounting flange1606. The biasing elements1602may be configured to bias the top composite panel106away from the mast110. For example, in the stowed position, the biasing elements1602may be exerting a force on the top composite panel106away from the mast110and the base composite panel104. As described above the top latch702may be configured to interface with the top hinge114to secure the solar array100in the stowed position in conjunction with the side latches704. The side latches704and the top latch702may be configured to oppose the force exerted on the top composite panel106by the biasing element1602when in the stowed position. FIG.16Billustrates the top hinge114after the top latch702and the side latches704are released. The biasing elements1602may cause the top composite panel106to move to a position substantially perpendicular to the base composite panel104. For example, the top hinge114may include a stop or detent configured to prevent the top composite panel106from rotating more than about 90° relative to the base composite panel104. The top hinge114may include a damper1604configured to control the motion of the top composite panel106under the force of the biasing elements1602. For example, the damper1604may be configured to provide resistance to the biasing elements1602, such that the rotation of the top composite panel106about the top hinge114may be slowed under the force of the biasing elements1602. Slowing the motion of the top composite panel106may reduce forces due to impact and shock in the top composite panel106when the top hinge114contacts the stop or detent. FIG.16Cillustrates the top hinge114in the position illustrated inFIG.14during the second movement. The biasing elements1602may maintain the position of the top composite panel106relative to the end of the mast110. Thus, due to the biasing elements1602acting on the top composite panel106, the top composite panel106may be maintained at about a 90° angle relative to the mast110and the base composite panel104as the mast110extends. The biasing elements1602may continue to maintain the relative position between the mast110and the top composite panel106until the panel modules108are all unfolded and aligned such that the cables406begin to have tension therein. As the tension builds in the cables406the tension may begin to overpower the biasing elements1602reducing the angle between the top composite panel106and the mast110until the top composite panel106is substantially aligned with the mast110as illustrated inFIG.16D. FIG.17Aillustrates the root hinge116in the stowed position. The root hinge116may be coupled between the body102and the mast110. A base plate1712may be coupled to the body102and a bottom plate1714may be coupled to the mast110. The root hinge116may include biasing elements1706configured to bias the root hinge116to the extended position. In some embodiments, the biasing elements1706may be a spring, such as a constant force spring (e.g., clock spring). A constant force spring may be a roll of pre-stressed strip that exerts a nearly constant force to resist uncoiling. In the stowed position a substantial length of the constant force spring may be unwound from the biasing elements1706, such that the constant force spring may be exerting a constant force to re-coil the unwound portion of the constant force spring. The root hinge116may also include a damper1702configured to resist motion of the root hinge116, such that movement of the root hinge116may be slowed to substantially prevent impact and shock stresses from damaging the solar array100. The damper1702may be operatively coupled to the root hinge116through offset gears1704. The offset gears1704may reduce the profile of the root hinge116. For example, the offset gears1704may allow the damper1702to be placed in a more advantageous position. In some embodiments, the offset gears1704may be configured as a force multiplier, such that a smaller damper1702may be used to generate a similar resistance. The base plate1712may include an electrical connection1716configured to provide an electrical connection between the solar array100and the body102of the spacecraft. The electrical connection1716may be operatively coupled to the wiring408through the base plate1712. For example, the wiring408may enter the base plate1712and be coupled to an electrical connection within the base plate1712. The electrical connection1716within the base plate1712may be configured to be received by a complementary electrical connection in the body102of the space craft, such that the wiring408may be electrically coupled to the spacecraft through the connection between the base plate1712and the body102. The electrical connection1716may be a connector, such as a multi-pin connector, configured to secure multiple electrical connections between the root hinge116and the body102. For example, the electrical connection may enable power generated by the solar array100to pass from the solar array100to the spacecraft. The electrical connection may further enable signals and data, such as control signals, sensor data, control data, etc., to pass between the spacecraft and the solar array100. In some embodiments, the electrical connection1716may be a latching connector, such as a female connector and/or male connector including latching features, such as clips, detents, screws, threaded apertures, etc., configured to secure the electrical connection1716substantially preventing the electrical connection from separating under vibration and movement of the root hinge116and the body102. The base plate1712may include one or more latches1708. The bottom plate1714may include strikes1710configured to interface with the one or more latches1708. The latches1708and the strikes1710may include complementary features configured to latch the bottom plate1714to the base plate1712once the biasing elements1706move the root hinge116into the extending position. FIG.17Billustrates the root hinge116in the extending position. The root hinge116may be in the extending position after the first movement and throughout the second movement as illustrated inFIGS.13through15. In the extending position the strikes1710may be secured in the latches1708securing the bottom plate1714in a position adjacent and parallel to the base plate1712. FIGS.18and19illustrate the mast110assembly and related drive systems. The mast110may be a telescoping mast110configured to extend. The mast110may include multiple nested segments1808configured to extend individually telescoping as the mast110extends. The mast110may include a screw drive1804driven by a drive assembly1802. The screw drive1804may be substantially the same length as each of the individual nested segments1808. The screw drive1804may drive a drive plate1806the length of the screw drive1804. The drive plate1806may impinge upon each of the nested segments1808individually starting with the smallest nested segment1808. Once the associated nested segment1808is extended the drive plate1806may be returned to the proximal end of the mast110and engage the next nested segment1808extending each nested segment1808in turn until the mast110is fully extended or until the solar array100is fully extended. The mast110may be constricted from composite materials configured to reduce the weight of the mast110while maintaining similar strength to heavier materials. For example, the segments of the mast110may be formed from composite materials, such as carbon fiber or fiberglass. FIG.19illustrates the drive assembly1802. The drive assembly1802may be included in the root hinge116. For example, the drive assembly1802may be housed within the bottom plate1714of the root hinge116adjacent to the mast110. The drive assembly1802may be configured to move with the bottom plate1714. The drive assembly1802may receive power and/or control signals through the electrical connection1716in the base plate1712. In some embodiments, the drive assembly1802may be configured to send signals, such as status and/or sensor signals to the body102through the electrical connection1716. The drive assembly1802may include a motor1902coupled to a gear box1904. The gear box1904may be operatively coupled to a drive gear1906. The drive gear1906may be operatively coupled to the screw drive1804. The motor1902may be configured to rotate gears1908within the gear box1904. The gears1908may be operatively engaged with the drive gear1906, such that rotation of the gears1908may cause the drive gear1906to rotate. Rotating the drive gear1906may cause the screw drive1804to rotate extending the nested segments1808of the mast110. The embodiments of the present disclosure may allow the volume and weight requirements for transporting a solar array to be reduced. Equipment used in space and transmitted to space may have strict payload requirements due to volume and weight restrictions and the expense of transporting equipment to space. Reducing the volume and weight requirements of a solar array may allow larger solar arrays to be installed on the equipment, which may increase the power available to the equipment. In some embodiments, reducing the volume and weight requirements of the solar array may allow additional components or equipment to be transported. In other embodiments, reducing the volume and weight requirements of the solar array may allow the transportation vehicles carrying the equipment and solar array or arrays to be reduced in size, which may reduce the amount of fuel required and the expense of transporting the equipment. The embodiments of the disclosure described above and illustrated in the accompanying drawing figures do not limit the scope of the invention, since these embodiments are merely examples of embodiments of the invention, which is defined by the appended claims and their legal equivalents. Any equivalent embodiments are intended to be within the scope of this disclosure. Indeed, various modifications of the present disclosure, in addition to those shown and described herein, such as alternative useful combinations of the elements described, may become apparent to those skilled in the art from the description. Such modifications and embodiments are also intended to fall within the scope of the appended claims and their legal equivalents. | 58,072 |
11863121 | DETAILED DESCRIPTION OF THE INVENTION The present disclosure may be embodied in many different forms without departing from the spirit and significant characteristics of the present disclosure. Therefore, the embodiment of the present disclosure is disclosed only for illustrative purposes and should not be construed as limiting the present disclosure. It will be understood that, although the terms first, second, etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element, from another element. For instance, a first element discussed below could be termed a second element without departing from the teachings of the present disclosure. Similarly, the second element could also be termed the first element The term “and/or” includes any and all combinations of one or more of the associated listed items. It will be understood that when an element is referred to as being “coupled” or “connected” to another element, it can be directly coupled or connected to the other element or intervening elements may be present therebetween. In contrast, it should be understood that when an element is referred to as being “directly coupled” or “directly connected” to another element, there are no intervening elements present. The terminology used herein is for the purpose of describing a particular embodiment only and is not intended to be limiting. As used herein, the singular forms “a”, “an”, and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprise”, “include”, “have”, etc. when used in this specification, specify the presence of stated features, integers, steps, operations, elements, components, and/or combinations of them but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or combinations thereof. Unless otherwise defined, all terms including technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which the present disclosure belongs. It will be further understood that terms, such as those defined in commonly used dictionaries, should be interpreted as having a meaning that is consistent with their meaning in the context of the relevant art and the present disclosure, and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein. Hereinbelow, preferred embodiments of the present disclosure will be described in detail with reference to the accompanying drawings, the same reference numerals will be used throughout the drawings and the description to refer to the same or like elements or parts, and a detailed description of those elements will be omitted. In the following description, when the functions of conventional elements and the detailed description of elements related with the present disclosure may make the gist of the present disclosure unclear, a detailed description of those elements will be omitted. An entire system including a complex energy generation device using sunlight and solar heat (hereinbelow, the device will refer to ‘complex energy generation device’) according to an embodiment of the present disclosure includes a heat storage tank1, a heating tube2, a transfer tube3, and a complex energy generation device100,200,300, as shown inFIG.1. The heat storage tank1is a water reservoir having a predetermined inside space, and an upper portion thereof is connected to a supply port to be supplied with domestic water and a lower portion thereof is connected to a discharge port to discharge hot water. The heating tube2is provided to heat the domestic water in the heat storage tank1to generate hot water. The heating tube2may be arranged in a zigzag array in the heat storage tank1. The transfer tube3is a tube connecting a first side portion of the complex energy generation device to the heating tube2. The transfer tube3allows heat medium oil that has passed through the complex energy generation device and the heating tube2to flow into the complex energy generation device again, so that the heat medium oil is circulated. Meanwhile, the complex energy generation device is a device configured to perform both power generation using sunlight and water heating using solar heat. Hereinbelow, the complex energy generation device will be described in detail with reference to accompanying drawings. As shown inFIG.1, the complex energy generation device includes a heat storage tube100, a solar panel200, and a heat radiation panel300. The heat storage tube100has an inlet portion101at a first side portion thereof to receive the heat medium oil and an outlet portion103at a second side portion thereof to discharge the heat medium oil. As shown inFIG.1, the first side portion of the heat storage tube100is connected to the transfer tube3and the second side portion of the heat storage tube100is connected to the heating tube2. The heat medium oil flowing into the heat storage tube100through the transfer tube3performs heat-exchange and then is discharged into the heating tube2. A lower surface of the heat storage tube100has a slit100S formed in a longitudinal direction thereof. An upper portion of the heat radiation panel300is inserted into the slit100S. In the structure of the heat storage tube100as described above, the heat medium oil may perform heat-exchange while circulated into the heat storage tube100and being brought into contact with the heat radiation panel300. The solar panel200has a plurality of solar cells201performing power generation using sunlight. Specifically, the solar cells201are provided on a front surface of the solar panel200, as shown inFIG.3. A transparent protective glass210may be laminated on the front surface of the solar panel200. The solar cells201provided on the solar panel200are an element generating energy using sunlight, and are configured to supply generated electric energy to the outside through a separate electric wiring (not shown) or to store the generated electric energy in a battery. The heat radiation panel300has the upper portion that is inserted into the heat storage tube100through the slit100S of the heat storage tube100while sealing the slit100S and a lower portion that is laminated on a rear surface of the solar panel200. When the solar panel200receives sunlight, the temperature of the solar panel200gradually is increased. When the temperature of the solar panel200is increased above a predetermined temperature, efficiency of energy generation may be reduced. The heat radiation panel300serves to prevent the temperature of the solar panel200from being increased above the predetermined temperature. The form of the slit100S and the sectional form of the heat radiation panel300are formed identically. Accordingly, the slit100S may be sealed when the upper portion of the heat radiation panel300is inserted into the slit100S. In order to increase a sealing force between surfaces of the slit100S and the heat radiation panel300, an O-ring or a separate sealing means may be provided. As shown inFIG.5, an upper end surface300aof the heat radiation panel300inserted into the heat storage tube100through the slit100S is extended to reach an inner surface of the heat storage tube100. The upper end surface300aof the heat radiation panel300is formed in a curved surface so as to be in close contact with the inner surface of the heat storage tube100. As shown inFIG.6, it is preferable that a first end portion300band a second end portion300cof the heat radiation panel300inserted into the heat storage tube100through the slit100S have streamlined cross-sections, so that the heat medium oil may be efficiently circulated. The heat radiation panel300is laminated to be in close contact with the rear surface of the solar panel200and serves to transmit heat of the solar panel200to the heat medium oil. When the heat-exchange is performed between the solar panel200and the heat medium oil circulated in the heat storage tube100, the temperature of the solar panel200is prevented from being increased above the predetermined temperature, so that the efficiency of energy generation may be maintained. Thermal grease may be disposed between the heat radiation panel300and the solar panel200, and the thermal grease may increase the efficiency of heat transfer between the heat radiation panel300and the solar panel200. The heat radiation panel300described above may be formed of a material with the same coefficient of thermal expansion as the heat storage tube100in order to prevent a gap formed between the slit100S and the heat radiation panel300due to a difference in the coefficient of thermal expansion. According to the structure of the heat radiation panel300as described above, the heat radiation panel300may be configured to prevent the temperature of the solar panel200from being increased above the predetermined temperature and to transmit solar heat to the heat medium oil. As shown inFIG.5, an exposed surface of the heat radiation panel300to the outside of the heat storage tube100may be covered by an insulation material310. As shown inFIGS.7A to7C, the thickness of the heat radiation panel300may vary. For example, as shown inFIG.7A, the heat radiation panel300may have a uniform thickness. Furthermore, for example, as shown inFIG.7B, a thickness of a first side portion of the heat radiation panel300corresponding to the first side portion of the heat storage tube100is formed thicker than a thickness of a second side portion of the heat radiation panel300corresponding to the second side portion of the heat storage tube100. In other words, the thickness of the heat radiation panel300is formed in a multi-stepped shape, so that the thickness of the heat radiation panel300is thinner stepwisely as the heat radiation panel300goes from the first side portion to the second side portion thereof. The slit100S is formed in a shape corresponding to the thicknesses of the heat radiation panel300. A contact surface of the heat radiation panel300in contact with the rear surface of the solar panel200is formed flat and an opposite surface to the contact surface is formed in the multi-stepped shape, so variation of the thickness of the heat radiation panel300is realized. For example, as shown inFIG.7C, a thickness of the first side portion of the heat radiation panel300corresponding to the first side portion of the heat storage tube100may be formed thicker than a thickness of the second side portion of the heat radiation panel300corresponding to the second side portion of the heat storage tube100. In other words, the thickness of the heat radiation panel300may be formed gradually thinner as the heat radiation panel300goes from the first side portion to the second side portion thereof. InFIG.7C, the slit100S is formed in a shape corresponding to the thickness of the heat radiation panel300. The contact surface of the heat radiation panel300in contact with the rear surface of the solar panel200is formed flat, and the opposite surface to the contact surface is formed to be inclined, so the thickness of the heat radiation panel300is changed. As described above, when the thickness of the first side portion of the heat radiation panel300corresponding to the first side portion of the heat storage tube100is formed thicker than the thickness of the second side portion of the heat radiation panel300corresponding to the second side portion of the heat storage tube100, the heat-exchange efficiency between the heat radiation panel300and the heat medium oil may be increased. The thickness shape of the heat radiation panel300is intended to increase a contact area with the heat medium oil flowing in the heat storage tube100. In addition, the temperature of the heat medium oil flowing into the inlet portion101of the heat storage tube100is increased while the heat medium oil passes through the heat storage tube100, so the heat radiation panel300is formed to be thicker to increase an amount of heat-exchange, in order to promote heat-exchange in the inlet portion101of the heat storage tube100. The complex energy generation device according to the described above embodiment of the present disclosure may perform both power generation using sunlight and heating water using solar heat. As the thickness variation of the heat radiation panel300increases a contact area between the heat radiation panel300and the heat medium oil, and increases the amount of heat-exchange at the inlet portion side100aof the heat storage tube100, the efficiency of heat-exchange is increased. Although a preferred embodiment of the present disclosure has been described for illustrative purposes with respect to the accompanying drawings, those skilled in the art will appreciate that various modifications, additions and substitutions are possible, without departing from the scope and spirit of the present disclosure as disclosed in the accompanying claims. Therefore, the scope and spirit of the present disclosure should be interpreted by the accompanying claims disclosed with the various modifications. | 13,296 |
11863122 | DESCRIPTION OF EXAMPLE EMBODIMENTS The degradation of a photosensitive device may result in an unexpected failure of a power system and may be expensive to address if not known before installation. Thus, it is important to know the degradation rate of a photosensitive device. Testing may be useful and reduce overall expenses for a given design or configuration. The degradation rate for a given photosensitive device is inversely related to, for example, the power produced by the photosensitive device. That is, the higher the degradation rate, the less power produced over time. Also, the degradation rate is directly proportional to the failure rate. That is, the higher the degradation rate, the more likely it is that a given photosensitive device will fail. A photosensitive device may be considered to have failed when the photosensitive device has degraded by 20% of the photosensitive device's original performance metric. The failure threshold may be adjusted up or down according to the given criteria for a particular photosensitive device configuration or installation. While testing is important, it is also important to reduce the testing time to ensure prompt implementation of a new design or configuration or installation of a photosensitive device. As photosensitive devices may be designed to last for several years or even decades, accelerated degradation is needed to reduce overall expenses and improve performance. The present disclosure provides a system and method for providing accelerated degradation and performance measurement for a given photosensitive device. The example embodiments herein may utilize a single information handling system local to a user. In certain embodiments more than one information handling system may be utilized. In other embodiments, one or more information handling systems may be remote, such as a server. In one or more embodiments, the methods and systems disclosed may be performed in conjunction with other photosensitive device degradation testing techniques. The teachings of the present disclosure are intended to encompass any combination of embodiments. While specific advantages are discussed, various embodiments may include all, some, or none of the enumerated advantages. Embodiments of the present disclosure and its advantages are best understood by referring toFIGS.1through4, wherein like numerals refer to like and corresponding parts of the various drawings. FIG.1illustrates an example information handling system100for implementing one or more embodiments disclosed herein. The information handling system100may include one or more elements, components, instrumentalities, etc. or any combination thereof operable to perform any functionality for implementing any embodiment disclosed herein. An information handling system100may be an embedded information handling system, a system-on-chip (SOC), a single-board information handling system, a mainframe, an interactive device such as a kiosk, a client device, a server (for example, blade server or rack server), personal computer (for example, desktop or laptop), tablet computer, mobile device (for example, personal digital assistant (PDA) or smart phone), a consumer electronic device, a network storage device, printer, switch, router, data collection device, virtual machine, or any other suitable computing device known to one of ordinary skill in the art. In one or more embodiments, information handling system100may be a single information handling system100or may be multiple information handling systems100, may be self-contained or distributed (for example, may span multiple data centers), may be hosted in a cloud, may be part of one or more other computing devices or may be any other suitable configuration known to one of ordinary skill in the art. Information handling system100may perform one or more operations in real-time, at timed intervals, in batch mode, at a single information handling system100or at multiple information handling systems100, at a single location or multiple locations, or in any other sequence or way known to one of ordinary skill in the art. The information handling system100may be any number of suitable components and is not limited to the number or the arrangement of components shown inFIG.1. Information handling system100may include a processor102, a memory104, a storage106, an input output (I/O) interface108, a display110, a bus112, and a network connectivity device114. Bus112may couple processor102, memory104, storage106, I/O interface108, and network connectivity device114to each other. Bus112may also couple any one or more of any other appropriate components of information handling system100to any other one or more components of information handling system100. Bus112may include hardware, software or any combination thereof for coupling any one or more components of information handling system100. Bus112may be any type of bus or combination of buses known to one of ordinary skill in the art. Information handling system100may include a processor102that is in communication with memory devices memory104and storage106. Processor102may be a general processing unit (GPU), a microprocessor, a central processing unit (CPU), multiple CPUs, single-core, dual-core, multi-core, or any other suitable processor known to one of ordinary skill in the art. Processor102may include one or more of internal read-only memory (ROM) (and any variation thereof), random access memory (RAM) (and any variation thereof), cache, internal registers, buffer, any other type of suitable storage component known to one of ordinary skill in the art, an arithmetic logic unit (ALU), and any other appropriate components known to one of ordinary skill in the art. Processor102includes hardware for executing one or more instructions or modules, for example, a software program or computer program. It is understood that by programming and/or loading executable instructions onto the information handling system100, at least one of the processor102, memory104, and storage106are changed, transforming the information handling system100in part into a particular machine or apparatus having the novel functionality taught by the present disclosure. It is fundamental to the electrical engineering and software engineering arts that functionality that can be implemented by loading executable software into an information handling system100can be converted to a hardware implementation by well known design rules. Decisions between implementing a concept in software versus hardware typically hinge on considerations of stability of the design and numbers of units to be produced rather than any issues involved in translating from the software domain to the hardware domain. Generally, a design that is still subject to frequent change may be preferred to be implemented in software, because re-spinning a hardware implementation is more expensive than re-spinning a software design. Generally, a design that is stable that will be produced in large volume may be preferred to be implemented in hardware, for example in an application specific integrated circuit (ASIC), because for large production runs the hardware implementation may be less expensive than the software implementation. Often a design may be developed and tested in a software form and later transformed, by well known design rules, to an equivalent hardware implementation in an application specific integrated circuit that hardwires the instructions of the software. In the same manner as a machine controlled by a new ASIC is a particular machine or apparatus, likewise a computer that has been programmed and/or loaded with executable instructions may be viewed as a particular machine or apparatus. Memory104may be internal or external to processor102. Memory104may be RAM, dynamic RAM (DRAM), static RAM (SRAM) or any other suitable type of memory known to one of ordinary skill in the art. While only one memory104is shown, the present disclosure contemplates any number of memory104. Memory104may include main memory for storing one or more instructions executed by processor102. Information handling system may load one or more instructions from storage106or any other information handling system100to memory104. Processor102may load one or more instructions from memory104to an internal memory of processor102for execution, for example, to an internal register or internal cache. Storage106may include mass storage for data, one or more instructions, one or more modules, or any other type of suitable information known to one of ordinary skill in the art. Storage106may be a hard disk drive (HDD), floppy disk drive, flash memory, optical disc drive, magneto-optical disc drive, magnetic tape, universal serial bus (USB) drive, non-volatile solid-state memory, read-only memory (ROM), mask-programmed ROM, programmable ROM (PROM), erasable PROM (EPROM), electrically erasable PROM (EEPROM), electrically alterable ROM (EAROM), any other type of ROM known to one of ordinary skill in the art, flash memory, any other storage known to one of ordinary skill in the art, or any combination of two or more of these. Storage106may include one or more storage106. Storage106is typically used for non-volatile storage and as over-flow storage for memory104. Storage106may store executable programs, such as software programs or computer programs which may be loaded into memory104when such programs are selected for execution. Memory104and storage106may be referred to in some contexts as computer readable storage media and/or non-transitory computer readable storage media. Network connectivity device114may be any or more network connectivity devices114and may take the form of modems, modem banks, Ethernet cards, USB interface cards, serial interfaces, token ring cards, fiber distributed data interface (FDDI) cards, wireless local area network (WLAN) cards, radio transceiver cards such as code division multiple access (CDMA), global system for mobile communications (GSM), long-term evolution (LTE), worldwide interoperability for microwave access (WiMAX), and/or other air interface protocol radio transceiver cards, and other well-known network devices. These network connectivity devices114may enable the processor102to communicate with the Internet or one or more intranets. With such a network connection, it is contemplated that the processor102might receive information from the network (for example, network210ofFIG.2), or might output information to the network in the course of performing the above-described method steps. Such information, which is often represented as a sequence of instructions to be executed using processor102, may be received from and outputted to the network, for example, in the form of a computer data signal embodied in a carrier wave. Such information, which may include data, instructions, or modules to be executed using processor102, for example, may be received from and outputted to the network, for example, in the form of a computer data baseband signal or signal embodied in a carrier wave. The baseband signal or signal embodied in the carrier wave generated by the network connectivity device114may propagate in or on the surface of electrical conductors, in coaxial cables, in waveguides, in an optical conduit, for example an optical fiber, or in the air or free space. The information contained in the baseband signal or signal embedded in the carrier wave may be ordered according to different sequences, as may be desirable for either processing or generating the information or transmitting or receiving the information. The baseband signal or signal embedded in the carrier wave, or other types of signals currently used or hereafter developed, may be generated according to several methods well known to one skilled in the art. The baseband signal and/or signal embedded in the carrier wave may be referred to in some contexts as a transitory signal. The processor102executes instructions, codes, computer programs, scripts which it accesses from memory104, storage106or the network connectivity device114. While only one processor102is shown, multiple processors may be present. Thus, while instructions may be discussed as executed by a processor, the instructions may be executed simultaneously, serially, or otherwise executed by one or multiple processors. Instructions, codes, computer programs, scripts, and/or data that may be accessed from the storage106, for example, hard drives, floppy disks, optical disks, and/or other device, ROM, and/or the RAM may be referred to in some contexts as non-transitory instructions and/or non-transitory information. I/O interface108may be hardware, software, or any combination thereof. I/O interface108provides one or more interfaces for communication between information handling system100and one or more I/O devices. In one embodiment, I/O interface108couples to display110and may communicate information to and from display110. While only a display110is shown, the present invention contemplates any number of internal or external I/O devices coupled to the I/O interface108such as one or more of video monitors, liquid crystal display (LCDs), touch screen displays, printers, keyboards, keypads, switches, dials, mice, track balls, voice recognizers, card readers, paper tape readers, thumb drives, hard disk drives, optical disk drives, microphones, video cameras, stylus, tablets, still cameras, speakers, sensors, or any other devices known to one of ordinary skill in the art. Information handling system100may also include one or more communication ports (not shown) for communicating with external devices. I/O interface108may also include one or more device drivers for any one or more I/O devices coupled to the information handling system100. In an embodiment, the information handling100may comprise two or more information handling systems100in communication with each other that collaborate to perform a task. For example, but not by way of limitation, an application may be partitioned in such a way as to permit concurrent and/or parallel processing of the instructions of the application. Alternatively, the data processed by the application may be partitioned in such a way as to permit concurrent and/or parallel processing of different portions of a data set by the two or more computers. In an embodiment, virtualization software may be employed by the information handling100to provide the functionality of a number of servers that is not directly bound to the number of information handling systems100in given configuration. For example, virtualization software may provide twenty virtual servers on four physical computers. In an embodiment, the functionality disclosed above may be provided by executing the application and/or applications in a cloud computing environment. Cloud computing may comprise providing computing services via a network connection using dynamically scalable computing resources. Cloud computing may be supported, at least in part, by virtualization software. A cloud computing environment may be established by an enterprise and/or may be hired on an as-needed basis from a third party provider. Some cloud computing environments may comprise cloud computing resources owned and operated by the enterprise as well as cloud computing resources hired and/or leased from a third party provider. In an embodiment, some or all of the functionality disclosed above may be provided as a computer program or software product. The computer program product may comprise one or more computer readable storage medium having computer usable program code embodied therein to implement the functionality disclosed above. The computer program product may comprise data structures, executable instructions, and other computer usable program code. The computer program product may be embodied in removable computer storage media and/or non-removable computer storage media. The removable computer readable storage medium may comprise, without limitation, a paper tape, a magnetic tape, magnetic disk, an optical disk, a solid state memory chip, for example analog magnetic tape, compact disk read only memory (CD-ROM) disks, floppy disks, jump drives, digital cards, multimedia cards, and others. The computer program product may be suitable for loading, by the information handling system100, at least portions of the contents of the computer program product to the storage106, to the memory104, and/or to other non-volatile memory and volatile memory of the information handling system100. The processor102may process the executable instructions and/or data structures in part by directly accessing the computer program product, for example by reading from a CD-ROM disk inserted into a disk drive peripheral of the information handling system100. Alternatively, the processor102may process the executable instructions and/or data structures by remotely accessing the computer program product, for example by downloading the executable instructions and/or data structures from a remote server through the network connectivity device114. The computer program product may comprise instructions that promote the loading and/or copying of data, data structures, files, and/or executable instructions to the storage106, to the memory104, and/or to other non-volatile memory and volatile memory of the information handling system100. In some contexts, a baseband signal and/or a signal embodied in a carrier wave may be referred to as a transitory signal. In some contexts, the storage106and the memory104may be referred to as a non-transitory computer readable medium or a computer readable storage media. A dynamic RAM embodiment of the memory104, likewise, may be referred to as a non-transitory computer readable medium in that while the dynamic RAM receives electrical power and is operated in accordance with its design, for example during a period of time during which the information handling system100is turned on and operational, the dynamic RAM stores information that is written to it. Similarly, the processor102may comprise an internal RAM, an internal ROM, a cache memory, and/or other internal non-transitory storage blocks, sections, or components that may be referred to in some contexts as non-transitory computer readable media or computer readable storage media. FIG.2is a block diagram illustrating an example networked configuration for one or more information handling systems100. In one embodiment, one or more clients220are coupled to one or more servers240via network210. Network210may be a public network, private network, wireless network, local area network (LAN), wide-area network (WAN), the Internet, extranet, intranet, or any other network known to one of ordinary skill in the art. In one embodiment, network210may include one or more routers for routing information between one or more clients220and one or more servers240. Client220may be any type of information handling system100. In one embodiment, client220may be a thin-client having limited processing and storage capabilities. Server240may be any type of information handling system100. In one embodiment server240may be a virtual machine or a desktop session. One or more servers240may provide access to software and/or hardware to one or more clients220. For example, a server240may provide access to a client220to a virtual device and/or a virtual application. Any one or more clients240may communicate with one or more servers240via any of one or more protocols known to one of ordinary skill in the art. One or more clients220may be coupled to one or more degradation testing systems230. While only one degradation testing system230is shown coupled to a given client220, the present disclosure contemplates any one or more degradation systems230coupled to a single client220or to multiple clients220. In one embodiment one or more degradation testing systems230may be coupled to the same one or more clients230. It is contemplated by the present disclosure that any combination of degradation testing systems230may be coupled in any number of configurations to any one or more clients220. In one or more embodiments, client220may communicate information received from any one or more degradation testing systems230via network210to any one or more servers240. FIG.3is a block diagram illustrating an exemplary degradation testing system230according to one or more embodiments of the present disclosure. While only certain components are depicted, the present disclosure contemplates that a degradation testing system230may comprise any number of components. While one or more components are depicted within degradation testing system230, the present disclosure contemplates that any one or more of the components may be contained within a single structure or unit or within multiple structures or units. A degradation testing system230provides an efficient way to test the degradation of photosensitive devices. Degradation testing system230may comprise a light power source302, a multiplexor (mux)304, an electrical source measure device (or measuring device)306, and a photosensitive device test system308. In one or more embodiments, light power source302, mux304, measuring device306, and photosensitive device test system308may be separate devices or within a single device, housed within one or more racks or within a single rack, or any combination thereof. Light power source302may be a programmable power supply which allows for controlling one or more of current, voltage, time stamps, or any other parameters associated with supplying power to one or more light sources. In one embodiment, light power source302may be a Keithley 2231A-30-3 Triple Channel DC Power Supply, any other light power source302known to one of ordinary skill in the art, or any combination of light power sources302. Light power source302controls the light intensity emitted by the light source plate312. Light power source302may have one or more local controls to allow a user to adjust (manually, automatically, or programmatically) any one or more parameters of the light power source302. Light power source302may be coupled to client220to allow for bi-directional communication between light power source302and client220. Any of the one or more parameters associated with the light power source302may be controllable by client220. Light power source302may transmit values for any of the one or more parameters to the client220. Based, at least in part, on the one or more parameters associated with the light power source302, client220may alter any of the one or more parameters associated with the light power source302. For example, any one or more of the one or more parameters may be compared to a threshold value and based, at least in part, on that comparison, the client220may communicate to the light power source302a command to alter or change one or more of these parameters. For example, client220may receive a parameter indicative of the voltage level being output by the light power source302and that parameter may be compared with a predefined threshold or limit whereupon client220may send a command to the light source302to adjust the voltage so as to attain the threshold (such as sending a command to the light power source302to either increase, decrease, or maintain the current voltage level). Degradation testing system230may also include a mux304. The mux304is a multiplexor for multiplexing the pixels of photosensitive device318to a coupled measuring device306. In one embodiment, the mux304may be an Agilent34792or any other suitable switch unit known to one of ordinary skill in the art. In one embodiment the measuring device306may be a Keithley 2450 source meter unit or any other measuring device known to one of ordinary skill in the art. The measuring device306may only measure one pixel of a photosensitive device318at a time. The measuring device306may send a signal or command to the mux304requesting information or a measurement for a selected pixel. In response, the mux304sends the measurement associated with a selected pixel to the measuring device306. In such a manner, each pixel of each photosensitive device318may be tested. While only one mux304is shown, any number of muxes304may be utilized according to the number of inputs allowed by the mux304and the number of pixels of photosensitive devices318required to be measured. In one embodiment, a first set of muxes304(where a set may be one or more) may be coupled to a first measuring device306while a second set of muxes (where a set may be one or more) may be coupled to a second measuring device306. Any combination of muxes304and measuring devices306may be utilized according to the specific requirements of a given testing configuration. The mux304and the measuring device306are also coupled to the client220. The client220communicates to the mux304the particular pixel of a photosensitive device318selected for testing (the pixel of photosensitive device318for measuring). For example, the client220may communicate to the mux304to close or open one or more relays associated with the mux304so as to complete, open or other otherwise connect the necessary circuitry associated with the selected pixel. The client220may then request a measurement for the selected pixel from the measuring device306. The degradation testing system230may also include a photosensitive device test system308. Photosensitive device test system308includes the components necessary to source, house, cool, maintain, access, communicate with, or perform any other operations for the photosensitive device318designated or selected for testing. For example, photosensitive device test system308may include a light source plate temperature control device310, light source plate312, cell interface plate314, container316, and cell interface temperature control device326. While light source plate temperature control device310, light source plate312, cell interface plate314, container316, and cell interface temperature control device326are shown within photosensitive device test system308, any one or more may be external to photosensitive device test system308. Light source plate temperature control device310heats, cools, or both heats and cools the light source plate312and subsequently any light sources mounted thereon. In one embodiment, the thermoconductive compound320is a dielectric material. In one embodiment the thermoconductive compound320is one of thermally-conductive grease or epoxy, carbon nano tubes, graphite, carbon black, CHO-THERM pads, any other suitable thermoconductive material known to one of ordinary skill in the art, or any combination thereof. The light source plate temperature control device310may be a thermoelectric cooler, a water circulating bath, dry ice, flame, any source that provides heating or cooling as known to one of ordinary skill in the art, or any combination thereof. In one embodiment, the light source plate temperature control device310is external to the photosensitive device test system308. In one embodiment, the light source plate temperature control device310couples to an external source that controls the temperature of the light source plate312. The light source plate temperature control device310is generally in close enough proximity to light source plate312to provide the required heating/cooling. Light source plate312provides a mounting surface for the light source, such as for one or more bulbs. Light source plate312is coupled to light power source302. Light source plate312may include one or more light sources. The one or more light sources may be any device that produces photons. For example, the light source may be fluorescent, incandescent, laser, thermo ionic emitter, light emitting diode (LED), or any other type of light source known to one of ordinary skill in the art. In one embodiment, one or more LED bulbs are utilized as the light source as the intensity may be modulated by only changing the power wattage input. The light source plate312intensity is typically measured in a unit of measurement known as a sun equivalent (for example, 1,000 W/m2) but any other applicable unit of measurement known to one of ordinary skill in the art may also be used. Light power source302may send a signal or command to light source plate312to increase or decrease the intensity of light source plate312. For example, the intensity may be altered in increments of 1 sun or a partial sun. In one embodiment, the photosensitive device318is exposed to an emission of 10 sun equivalents from light source plate312. Cell interface plate314may include a container316. Container316may be a chuck, holder, or any other container for housing or supporting a photosensitive device318such that photosensitive device318is exposed to emissions from the light source plate312. The photosensitive device318may be any one or more of photovoltaics (PVs), solar cells, photodiodes, photoresistors, photocapacitors, phototransducers, phototransistors, any other photosensitive device known to one of ordinary skill in the art, or any combination thereof. Photosensitive device318may include any number of individual photosensitive devices (also herein referred to as ‘pixels’) according to a given configuration. The container316may be constructed of a thermoconductive material, for example, aluminum. The container316includes pins that mate to form an electrical connection with the pads of the photosensitive devices318. A lid may be placed on top of the container316to provide stability and to apply a pressure to the photosensitive device318to ensure that the pads of the photosensitive device318electrically connect to the pins of the container316. While only certain components are shown, the present disclosure contemplates that container316may include any number of components known to one of ordinary skill in the art. The photosensitive device318sits on or above a thermoconductive compound320to provide heat transfer. While thermoconductive compound320is depicted below photosensitive devices318, the present disclosure contemplates that the thermoconductive compound320may be above or below, completely surround, or any combination thereof the photosensitive devices318. For example, in one embodiment, a thermoconductive compound320may be above and below photosensitive device318. Photosensitive device318may include one or more substrates where each substrate includes one or more individual photosensitive devices. In one embodiment, the photosensitive device318includes four substrates with six individual photosensitive devices per substrate. In one embodiment, photosensitive device test system308includes multiple containers316and each container316may include multiple substrates within each photosensitive device318. In one embodiment, photosensitive device test system308includes four containers316, each having a photosensitive device318where photosensitive device318includes four substrates with six individual photosensitive devices per substrate for a total of ninety-six individual photosensitive devices. Light metering device322measures the intensity of the emission from light source plate312. The light metering device322may be a photo diode, thermistor, any light measuring device322known to one of ordinary skill in the art, or any combination thereof. Light metering device322measures any fluctuations of the performance of the light intensity from the light source plate312. The fluctuations of the performance of the configuration of photosensitive devices318may be due to fluctuations of the performance of the photosensitive devices318themselves or to fluctuations of the light source plate312. While light metering device322is depicted within the container316, the present disclosure contemplates light metering device322being external to the container316. The light metering device322may communicate one or more light intensity measurements based, at least in part, on one or more light intensity measurement criteria for the testing configuration. For example, the light metering device322may communicate one or more light intensity measurements to the mux304based, at least in part, on a request for a light intensity measurement from the mux304, a timed interval, an interrupt, a manual command or input by a user, a determination that a threshold or a range has been exceeded (above or below), any other criteria known to one of ordinary skill in the art, or any combination thereof. While light metering device322is depicted within container322, the present disclosure contemplates light metering device322being external to the container316but proximate to the light source plate312such that light metering device322can accurately measure the light intensity exposed to the photosensitive devices318. Light metering device322may be any distance from the light source plate312but for accurate measurement must be within the tolerance for measuring emissions from the light source plate312exposed to the photosensitive device318. In one embodiment, light metering device322is coupled to a photosensitive device318on either side of thermoconductive compound320. In one embodiment, light metering device322is in between photosensitive devices318and light source plate312but does not obstruct any light or degrade the light intensity of light source312to photosensitive devices318. Temperature metering device324monitors the temperature of the photosensitive devices318. While temperature metering device324is shown within the container316, the present disclosure contemplates that temperature metering device324may be external to the container316, within the photosensitive device test system308or external to the photosensitive device test system308. The temperature metering device324is in close proximity to the photosensitive devices318so as to provide an accurate measurement of the photosensitive devices318where the proximity may be determined based, at least in part, on the sensitivity of the temperature metering device324, the accuracy required of the testing configuration, the type of photosensitive devices318, or any other criteria known to one of ordinary skill in the art. The temperature metering device324communicates via an interface of the cell interface plate314to the mux304. The temperature metering device324may communicate one or more temperature measurements based, at least in part, on one or more temperature measurement criteria for the testing configuration. For example, the temperature metering device324may communicate one or more temperature measurements to the mux304based, at least in part, on a request for a temperature measurement from the mux304, a timed interval, an interrupt, a manual command or input by a user, a determination that a threshold or a range has been exceeded (above or below), any other criteria known to one of ordinary skill in the art, or any combination thereof. The photosensitive device test system308may also include a cell interface temperature control device326. The cell interface temperature control device326controls the temperature of the cell interface plate314and the container316including the photosensitive device318. The cell interface temperature control device326may be a thermoelectric cooler, a water circulating bath, dry ice, flame, any source that provides heating or cooling as known to one of ordinary skill in the art, or any combination thereof. In one embodiment, the cell interface temperature control device326is external to the photosensitive device test system308. In one embodiment, the cell interface temperature control device326couples to an external source (for example, programmable logic controller and power supply) that controls the temperature of the cell interface plate314. Cell interface temperature control device326is generally in close proximity to cell interface plate314so as to provide the specified or required heating and/or cooling. FIG.4is a flowchart illustrating an example method400for a degradation testing system230. At step402, the degradation testing system230is initialized and configured. One or more degradation testing parameters or configurations may be initialized or set at client220. The degradation testing parameters or configurations may be indicative of the configuration and type of testing for the degradation testing system230. One or more of the degradation testing parameters or configurations may be initialized via a graphical user interface (GUI), a command-line interface (CLI), automatically via an expert system that polls one or more components of the degradation testing system230, for example photosensitive devices318, or any combination thereof, or any other way known to one of ordinary skill in the art. The one or more degradation testing parameters or configurations may be initialized or set by a user or automatically by one or more other clients220or severs240. In one embodiment, a user remotely logs in to the client220(shown inFIG.3) and sets or initializes the one or more degradation testing parameters. In another embodiment, a user locally sets or initializes the one or more degradation testing parameters at the client220(shown inFIG.3). In one or more embodiments, client220(shown inFIG.3) is local to the degradation testing system230. In one or more embodiments, client220(shown inFIG.3) is remote to the degradation testing system230. In one embodiment the degradation testing parameters may include a photosensitive device pin lookup table. The photosensitive device pin lookup table may include unique entries or an address map for each photosensitive device318. Each pin of each individual photosensitive device of photosensitive devices318may have a unique address that is stored in the photosensitive device pin lookup table. The photosensitive device pin lookup table may be a flat file, a database, a linked list, an addressed value stored in a memory location (such as memory104or storage106), any other suitable form known to one of ordinary skill in the art, or any combination thereof. The photosensitive device pin lookup table may be initialized by a user via a graphical user interface (GUI), a command-line interface (CLI), automatically via an expert system that polls each individual photosensitive device of photosensitive devices318, the degradation testing system230, or any combination thereof, or any other way known to one of ordinary skill in the art for obtaining the identification or addresses for each individual pin of an individual photosensitive device of photosensitive devices318. The photosensitive device pin lookup table may correlate to the wiring from the mux304to each pin of each photosensitive device of the photosensitive devices318. Also at step402, one or more degradation testing thresholds may be set. The one or more degradation testing thresholds may include one or more of photosensitive device failure threshold, a pixel performance rating, pixel failure threshold, light intensity threshold, light intensity time interval, a temperature threshold, a humidity threshold, a voltage threshold, a current threshold, an atmospheric threshold (for example, set levels for oxygen, nitrogen, argon, or any other atmospheric criteria known to one of ordinary skill in the art), a testing duration threshold (for example, 1 day, 10 days, or any other suitable unit of measurement known to one of ordinary skill in the art) or any other thresholds or combinations thereof known to one of ordinary skill in the art. For example, the degradation testing system230may be configured to test photosensitive devices318at a predefined baseline light intensity threshold of 1 sun so as to establish a baseline. In another example, after a baseline is established, the degradation testing system230may be configured to test photosensitive devices318at a light intensity threshold of 10 suns. Also at step402, the degradation testing system230may be configured to obtain one or more types of measurements over a range of data points and at a specified interval within that range. In one embodiment, the range is set to −0.2 Volts to +1.3 Volts by the measure device306with performance measurements of photosensitive devices318taken at each 0.1 V interval. An interval duration may also be associated with each interval. In one embodiment the interval duration may be based on a frequency such that measurements are taken at a time period measured in Hertz. In another embodiment, the duration of an interval may also be measured in days or any other suitable unit of measurement known to one of ordinary skill in the art. The scan direction may also be specified such that the measurements are taken beginning at a negative voltage to positive voltage or a positive voltage to a negative voltage. At step402, one or more other configurations or parameters that may be initialized or set may include the number of degradation testing systems230, the number of containers316within each degradation testing system230, the number of photosensitive devices318within each container316, the number of individual photosensitive devices within each photosensitive devices318, the process used to create each individual photosensitive device of photosensitive devices318, a file name or other unique identifier for each individual substrate, identification of which pins of each individual photosensitive device of each photosensitive devices318will be measured (or tested), testing temperature, testing atmosphere (for example, water vapor, air, pure nitrogen, pure oxygen, pure argon, etc., or any combination thereof) and any other parameters known to one of ordinary skill in the art. At step404, the light intensity is set based, at least in part, on the light intensity threshold (or if a baseline, the baseline light intensity threshold). In one embodiment, client220sends a command to the light power source302(for example, a programmable power source) to output a particular voltage or current to the light source plate312. The command may be based on any one or more of the degradation testing parameters. For example, in one embodiment a light intensity threshold is set to 10 suns and the duration for testing at 10 suns is set to every 10 days with an interval set to adjust the light intensity to 1 sun and to maintain the 1 sun light intensity during the photosensitive device318testing cycle, and returned to 10 suns upon testing cycle completion. In this embodiment, the client220sends a corresponding voltage or current command to the light power source302so as to set the light intensity of the light source plate312to the required level. At step406it is determined if a measurement should be requested. For example, one or more of the degradation testing parameters may indicate when a measurement is requested, a user may request a measurement or client220may request a measurement based on any number of criteria, degradation testing parameters, or any combination thereof. In one embodiment, it is determined if a specific interval has passed or a duration has been reached. For example, the degradation testing system230may be configured to take a performance measurement of any one or more pixels of the photosensitive devices318at the expiration of a certain time interval or duration. For example, performance measurements (or any other requested measurements) may be taken daily, twice a day, after the expiration of a timer (for example, at the expiration of a set time period), as a result of an interrupt, or based on any other interval of time. The interval of time may be stored as a duration threshold or an interval threshold such that when the threshold is exceeded, an interrupt is triggered, or client220may continuously poll to determine if the threshold has been exceeded, or by any other way known to one of ordinary skill in the art. If one or more degradation system parameters or conditions are not met such that a measurement is not requested, the system may continuously loop at406. The process may spawn a separate thread to continuously poll for an interrupt or any other indication that on one or more of the degradation system parameters or conditions (for example, a duration threshold or an interval threshold) have been met. Such polling need not be performed in a separate thread but rather may be performed in a single thread or in any manner known to one of ordinary skill in the art. In one embodiment, a measurement may be requested of the performance of one or more pixels (corresponding to an individual pin) of one or more individual photosensitive devices of photosensitive devices318for any of the one or more containers316as described above with respect toFIG.3. A measurement may be requested for any measurable degradation testing system condition including any condition associated with any one or more degradation testing parameters. For example, in addition to obtaining a measurement of a pixel, the humidity, temperature, atmosphere, or any other suitable condition may be measured. The one or more conditions may be measured separately from the performance of a given pixel. For example, client220may request measurements or automatically receive measurements for one or more conditions utilizing one or more measuring devices including, but not limited to, measuring device306, temperature metering device324, and light metering device322. One or more conditions may be associated with each type of requested measurement. For example, a performance measurement for a particular pixel may have an associated duration threshold, an interval threshold, a range threshold, or any other suitable condition known to one of ordinary skill in the art. Step406determines if any such associated conditions have been met before requesting that the specified measurement be requested. If a measurement is requested, then at step408, client220sends a request for the particular measurement to the appropriate device. For example, client220sends a request for a performance measurement for a particular pixel. The request (or command) is sent to mux304. The request may be based, at least in part, on an address of the pixel (that corresponds to a particular pin of an individual photosensitive device of photosensitive device318) to be measured where the address may be obtained from the photosensitive device pin lookup table, identification of the container316, the identification of the substrate containing the particular pixel of interest, the identification of the individual photosensitive device within the photosensitive devices318, an identification of the particular degradation testing system230, or any other criteria or identifier known to one of ordinary skill in the art. The mux304makes the appropriate electrical connections so as to receive the performance measurement associated with the identified pixel. At step410, the mux304based, at least in part, on the address received from the client220obtains a performance measurement for the identified pixel. For example, typically a voltage across a range is applied to the photosensitive device318(or to an individual photosensitive device of photosensitive device318) by the measurement device306via mux304and the current generated at each interval is measured by the measurement device306via mux304. These measurements may then be used to generate a current/voltage (or I-V) curve from which all information may be derived. For example, resistance, maximum power, capacitance, open-circuit voltage, short-circuit current, or any other related information known to one of ordinary skill in the art may be derived. In one embodiment, the measuring device306may convert the performance measurement to a form suitable for consumption by client220and communicates the result to the client220. In one embodiment, the measuring device306communicates the performance measurement via one or more suitable interfaces, components or devices to the client220. In one embodiment, the client220stores the measurement in the substrate file associated with the measured pixel. The measurement may be stored as an entry in a flat file, a database, a linked list, an addressed value stored in a memory location (such as memory104or storage106), any other suitable manner known to one of ordinary skill in the art, or any combination thereof. At step412, the client220determines based, at least in part, on the result received from step410for the performance measurement if a failure of an individual photosensitive device of photosensitive device318has occurred. If no photosensitive device failure has occurred, the process continues at step416. A photosensitive device failure may be determined based, at least in part, on the performance measurement of any one or more pixels of the particular photosensitive device. For example, if the performance measurement of any one or more pixels falls below a certain pixel performance rating (for example, below a certain percentage) then the particular photosensitive device may be determined to have failed. In one example, the pixel failure threshold is set to one such that if one pixel does not meet the specified pixel performance rating, the entire individual photosensitive device is determined to have failed. In another embodiment, the pixel failure threshold is a specified number or percentage of pixels and once that threshold is met a particular photosensitive device is determined to have failed. If it is determined at step412that a particular photosensitive device or pixel has failed, the photosensitive device or the pixel may be marked with a testing indicator at step414such that no further testing is performed on that particular photosensitive device or pixel within photosensitive devices318. The testing indicator may be a single bit where one setting is indicative of a failure and another setting is indicative of a pass, a non-failure, or that testing should continue for the particular pixel or photosensitive device. In another embodiment, a user is notified that a particular photosensitive device has failed and needs to be replaced. A user may be notified via an electronic mail, a GUI, a CLI, a warning message, an alarm, an light indicator, or any other way known to one of ordinary skill in the art. In one embodiment, the failure is recorded in the substrate file associated with the particular photosensitive device. At step416it is determined if further testing of any of the one or more degradation testing systems should continue. For example, the determination of step416may be made based, at least in part, on the number of failed pixels, the number of particular photosensitive devices marked as failures, or any other degradation testing thresholds or any combination thereof. In one or more embodiments, the process may end if the number of individual photosensitive devices of photosensitive device318exceeds the photosensitive device failure threshold. For example, in one embodiment the photosensitive device failure threshold may be set to one such that even if more than one photosensitive device is included within photosensitive devices318if a single photosensitive device fails the test ends. In one or more embodiments, two or more degradation testing systems230exist such that even if testing for one degradation testing system230ends the others may continue. Whether to continue testing may be based, at least in part, on one or more of a duration threshold (for example, testing may end at the expiration of a predetermined time limit), suitability of the testing environment (for example, testing may end if the humidity, temperature, atmosphere, etc. are not at acceptable levels), pixel failure rate, photosensitive device failure rate, number of photosensitive devices marked as failures, number of pixels marked as failures, user input (for example, the user via a GUI, CLI, or other input indicates whether the testing should continue), one or more evaluations of one or more measured parameters, or any other criteria known to one of ordinary skill in the art. If at step416, further testing is determined to be needed, then at step418it is determined whether the light intensity should be altered. For example, when obtaining a baseline, the light intensity may initially be set and maintained or held at the initial level for the duration of the baseline test. If the light intensity does not need to be altered the process continues at step406. If the light intensity does need to be altered the process continues at step404. The alteration of the light intensity may be determined based, at least in part on any one or more of a light intensity time interval, a light intensity threshold, at certain measurement intervals (for example, after each measurement, after each second measurement, etc.), duration intervals, or any other suitable parameter known to one of ordinary skill in the art. In one embodiment, at step418any other configurations associated with the degradation testing system230may also be altered. For example, it may be determined that the temperature, humidity, atmosphere, or any other condition of the degradation testing system230environment should be altered. In one embodiment, the process shown at400is exercised to obtain a baseline measurement. The baseline measurement may be established using any one or more degradation testing threshold parameters and one or more values for the degradation testing threshold parameters. For example, a baseline may be run for a duration of 1 day with a light intensity threshold of 1 sun. Subsequent to establishing a baseline measurement, the process shown at400may be ran in normal operation for any given period of time and for any light intensity threshold (for example, 10 days at a light intensity of 10 suns). In one or more embodiments, client220may shut down the testing of degradation testing system230based on any one or more alarms. The one or more alarms may be based, at least in part, on any one or more of a smoke detector, a carbon monoxide detector, a temperature measurement, a humidity measurement, an atmospheric measurement, a voltage measure, a current measurement, a power measurement, a vibration detector (for example, a device that detects vibration or movement in the structure housing the degradation testing system230, for example, vibrations due to an earthquake), a short circuit, an open circuit, or any other alarm known to one of ordinary skill in the art. Herein, “or” is inclusive and not exclusive, unless expressly indicated otherwise or indicated otherwise by context. Therefore, herein, “A or B” means “A, B, or both,” unless expressly indicated otherwise or indicated otherwise by context. Moreover, “and” is both joint and several, unless expressly indicated otherwise or indicated otherwise by context. Therefore, herein, “A and B” means “A and B, jointly or severally,” unless expressly indicated otherwise or indicated otherwise by context. This disclosure encompasses all changes, substitutions, variations, alterations, and modifications to the example embodiments herein that a person having ordinary skill in the art would comprehend. Similarly, where appropriate, the appended claims encompass all changes, substitutions, variations, alterations, and modifications to the example embodiments herein that a person having ordinary skill in the art would comprehend. Moreover, reference in the appended claims to an apparatus or system or a component of an apparatus or system being adapted to, arranged to, capable of, configured to, enabled to, operable to, or operative to perform a particular function encompasses that apparatus, system, component, whether or not it or that particular function is activated, turned on, or unlocked, as long as that apparatus, system, or component is so adapted, arranged, capable, configured, enabled, operable, or operative. Any of the steps, operations, or processes described herein may be performed or implemented entirely with hardware or entirely with software (including firmware, modules, instructions, micro-code, etc.) or with any combination of hardware and software. In one embodiment, a software module is implemented with a computer program product comprising a computer-readable medium containing computer program code, which can be executed by a computer processor for performing any or all of the steps, operations, or processes described. Embodiments of the invention may also relate to an apparatus for performing the operations herein. This apparatus may be specially constructed for the required purposes, and/or it may comprise a general-purpose computing device, such as an information handling system, selectively activated or reconfigured by a computer program stored in the information handling system. Such a computer program may be stored in a tangible computer readable storage medium or any type of media suitable for storing electronic instructions, and coupled to an information handling system bus. Furthermore, any computing systems referred to in the specification may include a single processor or may be architectures employing multiple processor designs for increased computing capability. Although the present invention has been described with several embodiments, a myriad of changes, variations, alterations, transformations, and modifications may be suggested to one skilled in the art, and it is intended that the present invention encompass such changes, variations, alterations, transformations, and modifications as fall within the scope of the appended claims. Moreover, while the present disclosure has been described with respect to various embodiments, it is fully expected that the teachings of the present disclosure may be combined in a single embodiment as appropriate. | 57,905 |
11863123 | DESCRIPTION OF EXEMPLARY EMBODIMENTS Thereafter, a preferred embodiment of the present disclosure will be described in detail. The present embodiment to be described below does not unduly limit contents described in the claims, and all configurations described in the present embodiment are not necessarily essential constituent elements. 1. Circuit Device and Oscillator FIG.1is a configuration example of an oscillator200and a circuit device100according to the present embodiment. The oscillator200includes a resonator10and the circuit device100. The resonator10is an element that generates mechanical oscillation according to an electric signal. The resonator10can be implemented by a resonator element such as a quartz crystal resonator element. For example, the resonator10is a tuning fork type quartz crystal resonator element. Alternatively, the resonator10can be implemented by a quartz crystal resonator element or the like that has a cut angle of AT cut, SC cut, or the like and that performs thickness-shear vibration. The resonator10according to the present embodiment can be implemented by various resonator elements such as a resonator element other than a tuning fork type or thickness-shear vibration type resonator element, or a piezoelectric resonator element made of a material other than quartz crystal. For example, the resonator10may be a SAW resonator, or a MEMS resonator serving as a silicon resonator formed using a silicon substrate. SAW is an abbreviation for surface acoustic wave, and MEMS is an abbreviation for micro electro mechanical systems. The circuit device100is electrically coupled to the resonator10and oscillates the resonator10by driving the resonator10. Coupling in the present embodiment is electrical coupling. The electrical coupling refers to coupling in which an electric signal can be transmitted, and refers to coupling in which information can be transmitted by the electric signal. The electrical coupling may be coupling performed via a passive element, an active element, or the like. In addition, the circuit device100performs a temperature compensation process of causing an oscillation frequency of the oscillator200to be constant regardless of a temperature. The circuit device100is an integrated circuit device referred to as IC. The circuit device100is an IC manufactured by a semiconductor process, and is a semiconductor chip in which a circuit element is formed at a semiconductor substrate. The circuit device100includes a temperature detection circuit105, a temperature compensation circuit135, an adjustment circuit154, an oscillation circuit160, and a temperature detection rate control circuit190. The oscillator and the circuit device are not limited to the configuration inFIG.1, and various modifications such as omitting a part of constituent elements of the oscillator and the circuit device or adding other constituent elements are possible. The temperature detection circuit105measures an environmental temperature of the resonator10and outputs a result of the measurement as temperature detection data ETD. The temperature detection data ETD is data that monotonically increases or monotonically decreases with respect to the temperature in an operation temperature range of the circuit device100. The temperature detection circuit105includes a counter and a ring oscillator whose oscillation frequency has temperature dependency. The counter counts an oscillation signal of the ring oscillator in an enable period defined according to a clock signal CLK output from the oscillation circuit160, and outputs a count value as the temperature detection data ETD. Alternatively, the temperature detection circuit105may include an analog temperature sensor that outputs a temperature detection voltage using temperature dependence of a forward voltage of a PN junction, and an A/D converter that performs A/D conversion on the temperature detection voltage and that outputs the temperature detection data ETD. As described later with reference toFIG.6, the temperature detection circuit105may further include a calculation circuit that outputs the temperature detection data ETD by adjusting temperature sensitivity for the output data of the counter or the output data of the A/D converter. The temperature compensation circuit135outputs, based on the temperature detection data ETD, adjustment data QCL for temperature compensation on an oscillation frequency of the oscillation circuit160. The adjustment data QCL is data for canceling or reducing temperature properties of the oscillation frequency. The temperature compensation circuit135obtains the adjustment data QCL based on the temperature detection data ETD using a look-up table. Alternatively, the temperature compensation circuit135may obtain the adjustment data QCL based on the temperature detection data ETD by executing calculation using a polynomial that approximates the temperature properties of the oscillation frequency. The adjustment circuit154is coupled to the oscillation circuit160, and adjusts the oscillation frequency of the oscillation circuit160to an oscillation frequency corresponding to the adjustment data QCL. The oscillation frequency is controlled to be constant regardless of a temperature by adjusting the oscillation frequency using the adjustment data QCL for canceling or reducing the temperature properties of the oscillation frequency. The adjustment circuit154is a capacitance array circuit coupled to one end or the other end of the resonator10. Alternatively, the adjustment circuit154may include a D/A converter that performs D/A conversion on the adjustment data QCL, and a variable capacitor coupled to one end or the other end of the resonator10. A capacitance value of the variable capacitor is controlled according to an output voltage of the D/A converter. The oscillation circuit160generates an oscillation signal using the resonator10. Specifically, the oscillation circuit160oscillates the resonator10by driving the resonator10, and generates an oscillation signal by the oscillation. An example of the oscillation circuit160is a colpitts type oscillation circuit described later, and the oscillation circuit160is not limited thereto. Alternatively, various types of oscillation circuits may be used as long as the oscillation frequency can be adjusted by the adjustment circuit154. The clock signal CLK is output based on the oscillation signal. For example, the oscillation circuit160may output the oscillation signal as the clock signal CLK. Alternatively, the circuit device100may include an output circuit, and the output circuit may output the clock signal CLK by buffering or dividing the oscillation signal. The temperature detection rate control circuit190calculates a variation in the temperature detection data ETD, and adaptively controls the temperature detection rate according to the variation. The temperature detection rate is a rate at which the temperature detection circuit105detects a temperature, that is, a rate at which the temperature detection circuit105updates the temperature detection data ETD. The temperature detection rate control circuit190enables an enable signal ENR at the controlled temperature detection rate, and the temperature detection circuit105detects the temperature when the enable signal ENR is enabled. The temperature detection rate can be set to n stages of a first rate to a n-th rate when n is an integer of two or more. The k-th rate is a rate higher than the (k−1)-th rate when k is an integer of two or more and n or less. The temperature detection rate control circuit190sets the temperature detection rate as the n-th rate when the variation in the temperature detection data ETD is a first threshold value or more, and decreases the temperature detection rate in a stepwise manner when the variation in the temperature detection data ETD is a second threshold value or less. The second threshold value is smaller than the first threshold value. The temperature detection rate and the enable period of the enable signal ENR are defined based on a frequency or a cycle of the clock signal CLK. In the present embodiment described above, the circuit device100includes the oscillation circuit160that generates an oscillation signal using the resonator10, the temperature detection circuit105that outputs the temperature detection data ETD, the temperature compensation circuit135, and the temperature detection rate control circuit190. The temperature compensation circuit135performs temperature compensation on an oscillation frequency of the oscillation signal based on the temperature detection data ETD. The temperature detection rate control circuit190controls a temperature detection rate at which the temperature detection circuit105executes temperature detection. At this time, the temperature detection rate control circuit190controls the temperature detection rate based on a variation in the temperature detection data ETD. According to the present embodiment, when an environmental temperature changes, the temperature detection rate is controlled based on the variation in the temperature detection data ETD. Accordingly, it is possible to cause the temperature compensation to follow the temperature variation caused by various factors and at various timings, and a highly accurate oscillation frequency is maintained. In addition, since the temperature detection rate can be decreased when there is no or small temperature variation, both high accuracy of the oscillation frequency and low power consumption of the circuit can be achieved. 2. Temperature Detection Rate Control Circuit FIG.2is a detailed configuration example of the temperature detection rate control circuit190.FIG.3is a timing chart showing an operation of the temperature detection rate control circuit190. As shown inFIG.2, the temperature detection rate control circuit190includes a temperature adjustment rate generation counter191, a selector192, a rate control circuit193, a rate setting circuit194, and a temperature variation calculation circuit195. The temperature adjustment rate generation counter191divides the clock signal CLK to output a rate control clock signal CNTQ having the same frequency as a maximum rate of a temperature adjustment rate. As shown inFIG.3, the rate control clock signal CNTQ has the same pulse width as a pulse width of the enable signal ENR. Here, it is assumed that a high level of the enable signal ENR indicates enable, and the pulse width of the enable signal ENR is a high width.FIG.3shows an example in which the pulse width is set to seven cycles of the clock signal CLK. The temperature variation calculation circuit195calculates a variation HNQ in the temperature detection data ETD based on the temperature detection data ETD. As shown inFIG.3, the temperature detection circuit105detects a temperature when a pulse of the enable signal ENR is input, and outputs the temperature detection data ETD which is a result of the detection. The temperature variation calculation circuit195calculates, as the variation HNQ in the temperature detection data ETD, a difference between the temperature detection data ETD and the previously output temperature detection data ETD. For example, in a waveform diagram in which RTQ=1, when the temperature detection circuit105outputs temperature detection data ETD1bfollowing temperature detection data ETD1a, the temperature variation calculation circuit195outputs the variation HNQ=ETD1b−ETD1ain the temperature detection data ETD. The rate setting circuit194adaptively sets a rate setting value RTQ according to the variation HNQ in the temperature detection data ETD. Thereafter, it is assumed that the rate setting circuit194selects a rate setting value from RTQ=1, 2, 5, and 10. Options of the rate setting value are not limited to the above, and the number of the options is not limited to four and may be two or more. The rate control circuit193outputs a rate control signal SEL based on the rate control clock signal CNTQ and the rate setting value RTQ. The selector192outputs the rate control clock signal CNTQ as the enable signal ENR when the rate control signal SEL is at a high level, and outputs the enable signal ENR at a low level when the rate control signal SEL is at a low level. When a pulse rate of the rate control clock signal CNTQ is Rmax, the rate control circuit193outputs the rate control signal SEL such that a pulse rate of the enable signal ENR becomes Rmax/RTQ. FIG.3is a waveform example in which RTQ=1, and a waveform example in which RTQ=2. When RTQ=1, the rate control circuit193fixes the rate control signal SEL to a high level, and the selector192outputs the rate control clock signal CNTQ as the enable signal ENR. The temperature detection circuit105outputs the temperature detection data ETD=ETD1a, ETD1b, ETD1c, ETD1d, . . . at a temperature detection rate Rmax. When RTQ=2, the rate control circuit193outputs the rate control signal SEL having a frequency Rmax/RTQ=Rmax/2, and the selector192outputs a pulse of the rate control clock signal CNTQ once as the pulse of the enable signal ENR when RTQ=2. The temperature detection circuit105outputs the temperature detection data ETD=ETD2a, ETD2b, . . . at a temperature detection rate Rmax/2. The same applies to the cases of RTQ=5 and 10, and the temperature detection rates are Rmax/5 and Rmax/10. FIG.4is a flowchart of a process executed by the temperature detection rate control circuit190. In step S1, the rate control circuit193sets the rate setting value to RTQ=1. That is, an initial value of the temperature detection rate is a maximum rate Rmax. As shown inFIG.2, the rate control circuit193includes a rate control counter196that counts the number of pulses of the rate control clock signal CNTQ. In step S2, the rate control counter196resets a count value. Here, it is assumed that the rate control counter196counts up to 1, 2, 3, . . . with1as an initial value. In step S3, the rate control circuit193determines whether the count value reaches the rate setting value RTQ. When the count value is smaller than the rate setting value RTQ, in step S4, the rate control counter196increments the count value when the pulse of the rate control clock signal CNTQ is input. Thereafter, step S3is executed. When the count value is the rate setting value RTQ or more, in step S5, the rate control counter196resets the count value. In step S6, the temperature compensation is performed. That is, the temperature detection rate control circuit190outputs the pulse of the enable signal ENR, the temperature detection circuit105detects a temperature, and the temperature compensation circuit135outputs the adjustment data QCL based on the temperature detection data ETD. By executing steps S3to S6described above, a temperature is detected at a temperature detection rate Rmax/RTQ, and the temperature compensation is performed. In step S7, the temperature variation calculation circuit195calculates a variation in the temperature detection data ETD. That is, the temperature variation calculation circuit195calculates, as the variation in the temperature detection data ETD, a difference between the temperature detection data ETD detected in step S6and the temperature detection data ETD detected when step S6is previously performed. In step S8, the rate setting circuit194determines whether the variation in the temperature detection data ETD is the first threshold value or more. For example, the first threshold value is 6 LSB. When the variation is 6 LSB or more, in step S9, the rate setting circuit194sets the rate setting value to RTQ=1. That is, the temperature detection rate is set to the maximum rate Rmax. Thereafter, step S3is executed. In step S8, when the variation is less than 6 LSB, in step S10, the rate setting circuit194determines whether the variation in the temperature detection data ETD is the second threshold value or less. For example, the second threshold value is 2 LSB. When the variation is 2 LSB or less, in step S11, the rate setting circuit194lowers the rate setting by one stage. That is, when the rate setting values are RTQ=1, 2, and 5, the rate setting circuit194changes the rate setting values to RTQ=2, 5, and 10, respectively. Accordingly, when the temperature detection rates are Rmax, Rmax/2, and Rmax/5, the temperature detection rates decrease to Rmax/2, Rmax/5, and Rmax/10, respectively. When the rate setting value is RTQ=10, RTQ=10 is maintained. When the variation is larger than 2 LSB in step S10, in step S12, the rate setting circuit194maintains the rate setting value RTQ. Thereafter, step S3is executed. By executing steps S7to S12described above, the temperature adjustment rate is adaptively controlled according to the variation in the temperature detection data ETD. FIG.5is a waveform example showing a change in a temperature detection rate when an environmental temperature of the circuit device100changes. An upper part ofFIG.5is a waveform showing a temporal variation in the temperature, and a lower part is a waveform of a temperature detection rate RT and a waveform of a movement average RTave of the temperature detection rate RT. Here, a waveform when the temperature rises from 25° C. to 125° C., stays at 125° C. for a while, and then falls to −50° C. is shown. When the temperature starts to rise from 25° C. and the variation in the temperature detection data ETD exceeds the first threshold value, the temperature detection rate control circuit190sets the temperature detection rate RT to the maximum rate Rmax. As the temperature detection rate becomes high, an update interval of the temperature detection data ETD becomes short. Therefore, the variation of the temperature detection data ETD, which is the difference from the previous data, becomes small. When the variation in the temperature detection data ETD is the second threshold value or less, the temperature detection rate control circuit190decreases the temperature detection rate RT to Rmax/2 by one stage. Since the variation in the temperature detection data ETD becomes large when the temperature detection rate decreases, the temperature detection rate RT becomes the maximum rate Rmax again, and then Rmax and Rmax/2 are repeated. The movement average RTave of the temperature detection rate RT is Rmax and Rmax/2. Since the variation in the temperature detection data ETD is the second threshold value or less when the temperature reaches 125° C. and a temperature change stops, the temperature detection rate control circuit190decreases the temperature detection rate RT step by step to a minimum rate Rmax/10. Thereafter, similarly, the temperature detection rate control circuit190adaptively controls the temperature detection rate according to the temperature variation. In the present embodiment described above, the temperature detection circuit105executes an intermittent operation of executing temperature detection in an intermittent temperature detection period. The temperature detection rate control circuit190controls the temperature detection rate by controlling a rate of the intermittent operation. In the example inFIG.3, the enable period of the enable signal ENR corresponds to the intermittent temperature detection period, and the temperature detection in the enable period corresponds to the intermittent operation. Further, the temperature detection rate control circuit190controls a rate in the enable period, which corresponds to controlling the rate of the intermittent operation. According to the present embodiment, the temperature compensation is intermittently performed by detecting a temperature by the intermittent operation. Accordingly, power consumption of the temperature compensation is reduced. Further, both high accuracy of an oscillation frequency and low power consumption of a circuit can be achieved by adaptively controlling the rate of the intermittent operation according to the variation in the temperature detection data ETD. In the present embodiment, the temperature detection rate control circuit190sets the temperature detection rate to a n-th rate that is the highest in a first rate to the n-th rate when the variation in the temperature detection data ETD is the first threshold value or more. n is an integer of two or more. In the examples inFIGS.4and5, n=4, and the temperature detection rates Rmax/10, Rmax/5, Rmax/2, and Rmax when the rate setting values RTQ=10, 5, 2, and 1 correspond to a first rate, a second rate, a third rate, and a fourth rate. At this time, Rmax that is the fourth rate is the highest temperature detection rate. According to the present embodiment, when there is a temperature variation in which the variation in the temperature detection data ETD is the first threshold value or more, that is, when it is determined that a rapid temperature variation occurs, the temperature detection rate is set to the maximum n-th rate. Accordingly, the temperature compensation follows a variation in an oscillation frequency due to the rapid temperature variation, and a highly accurate oscillation signal with a reduced frequency deviation is obtained. In the present embodiment, the temperature detection rate control circuit190decreases the temperature detection rate from the k-th rate to the (k−1)-th rate among the first rate to the n-th rate when the variation in the temperature detection data ETD is the second threshold value or less. The second threshold value is smaller than the first threshold value. k is an integer of two or more and n or less. According to the present embodiment, when it is determined that a temperature variation occurring between a previous temperature compensation and a current temperature compensation at a current temperature adjustment rate is small, that is, the temperature adjustment rate is too high with respect to the temperature variation, the temperature detection rate decreases by one stage. Accordingly, the temperature adjustment rate can be set to be as low as possible in a range in which the temperature compensation can follow the variation in the oscillation frequency due to the temperature variation. In the present embodiment, the initial value of the temperature detection rate is the n-th rate. Since the variation in the temperature detection data ETD is unknown in an initial state, a followability of the temperature compensation can be ensured from a start of temperature adjustment by setting the n-th rate as the initial value. In addition, when the variation in the temperature detection data ETD is small, the temperature adjustment rate adaptively decreases as described above. In the present embodiment, the temperature detection rate control circuit190controls the temperature detection rate based on the difference between the previous temperature detection data ETD and the current temperature detection data ETD. The difference between the previous temperature detection data ETD and the current temperature detection data ETD indicates the temperature variation occurring between the previous temperature compensation and the current temperature compensation, and relates to the frequency deviation of the oscillation frequency occurring between the previous temperature compensation and the current temperature compensation. By controlling the temperature adjustment rate according to the difference between the previous temperature detection data ETD and the current temperature detection data ETD, the temperature adjustment rate is controlled such that the frequency deviation of the oscillation frequency occurring between the previous temperature compensation and the current temperature compensation is in an appropriate range. In the present embodiment, the temperature compensation circuit135performs the temperature compensation at the temperature detection rate. For example, inFIG.4, in step S6, the temperature compensation is performed together with the temperature detection. In addition,FIG.7described later shows that the adjustment data QCL is updated when the temperature detection data ETD is updated. According to the present embodiment, the temperature adjustment rate that is a rate of the temperature compensation can be controlled by the temperature detection rate control circuit190controlling the temperature detection rate. Accordingly, since an overall rate of the temperature compensation from a time point when the temperature detection circuit105detects a temperature to a time point when the adjustment circuit154adjusts the oscillation frequency is controlled, power consumption of an entire circuit relating to the temperature compensation decreases when the temperature detection rate is reduced. 3. Temperature Detection Circuit FIG.6is a detailed configuration example of the temperature detection circuit105.FIG.7is a timing chart showing operations of the temperature detection circuit105and the temperature compensation circuit135. As shown inFIG.6, the temperature detection circuit105includes a temperature sensor circuit110, a calculation circuit120, and a register170. The temperature sensor circuit110measures an environmental temperature of the resonator10and outputs a result of the measurement as output data TD. The output data TD is data that monotonically increases or monotonically decreases with respect to the temperature in an operation temperature range of the circuit device100. The temperature sensor circuit110includes a ring oscillator112and a counter113. As shown inFIG.7, the ring oscillator112oscillates during the enable period of the enable signal ENR, and outputs an oscillation signal RNGQ. The ring oscillator112includes, for example, a NAND circuit and an even number of inverters coupled in series between an output and a first input of the NAND circuit. The enable signal ENR is input to a second input of the NAND circuit. In this case, a high level corresponds to active, the ring oscillator112oscillates when the enable signal ENR is at the high level, and the oscillation of the ring oscillator112stops when the enable signal ENR is at a low level. The above configuration is an example, and the configuration of the ring oscillator112is not limited to the above. The counter113executes a count operation based on the oscillation signal RNGQ of the ring oscillator112, and outputs the output data TD based on a count value. Specifically, the counter113counts the number of pulses of the oscillation signal RNGQ output in the above enable period, and outputs the count value as the output data TD. InFIG.7, the counter113outputs the output data TD=TDa, TDb, . . . in time series for each pulse of the enable signal ENR. The counter113may execute the count operation based on the oscillation signal RNGQ, and may count, for example, the number of pulses of a signal obtained by dividing the oscillation signal RNGQ. In addition, the counter113may output the output data TD based on the count value, and may output, for example, the output data TD by smoothing the count value. The calculation circuit120is a logic circuit that converts, into the temperature detection data ETD, the output data TD output by the temperature sensor circuit110. The temperature detection data ETD is data that monotonically increases or monotonically decreases with respect to the temperature in the same manner as the output data TD, and a slope of the temperature detection data ETD is converted from a slope of the output data TD according to the temperature range. As shown inFIG.7, the calculation circuit120converts the output data TDa into temperature detection data ETDa when the temperature sensor circuit110outputs the TDa and converts the output data TDb into temperature detection data ETDb when the temperature sensor circuit110outputs the TDb. Details of the conversion will be described below. The register170stores a parameter for the conversion executed by the calculation circuit120. The parameter stored in the register170is input to the calculation circuit120, and the calculation circuit120converts the output data TD into the temperature detection data ETD based on the parameter. The temperature compensation circuit135outputs adjustment data QCLa based on the temperature detection data ETDa when the calculation circuit120outputs the ETDa, and outputs adjustment data QCLb based on the temperature detection data ETDb when the calculation circuit120outputs the ETDb. In this way, since the temperature compensation is performed at the temperature detection rate, the temperature adjustment rate is the same as the temperature detection rate. FIG.8is an example of the conversion executed by the calculation circuit120. The calculation circuit120obtains the temperature detection data ETD by multiplying the output data TD by a coefficient and adding an offset. InFIG.8, the calculation circuit120changes the coefficient to be multiplied to the output data TD with temperatures Ta, Tb, and Tc as boundaries. The coefficient is smaller than 1 in a temperature range TRA lower than Ta, is 1 in a temperature range TRB of Ta or higher and lower than Tb, is larger than 1 in a temperature range TRC of Tb or higher and lower than Tc, and is further larger in a temperature range TRD of Tc or higher. The temperature Ta is set near a room temperature of 25° C., and the slope of the temperature detection data ETD increases in a temperature range higher than the room temperature of 25° C. FIG.9is an example of the frequency sensitivity. The frequency sensitivity is sensitivity of an oscillation frequency for a temperature change, and inFIG.9, is indicated by a frequency deviation when the output data TD or the temperature detection data ETD changes by 1 LSB. FS indicates a property of the frequency sensitivity when temperature data is not corrected by the calculation circuit120, and FScol indicates a property of the frequency sensitivity when the temperature data is corrected by the calculation circuit120. When the temperature data is not corrected by the calculation circuit120, the frequency sensitivity is relatively low near the room temperature or in a temperature range lower than the room temperature. In such a temperature range, since the oscillation frequency is less likely to change even if the temperature changes, a lower temperature adjustment rate is desirable in terms of lower power consumption. On the other hand, in the temperature range higher than the room temperature, as the temperature becomes higher, the frequency sensitivity becomes higher. In such a temperature range, since the oscillation frequency is likely to greatly change when the temperature changes, a higher temperature adjustment rate is desirable in terms of the followability of the temperature compensation. However, in the output data TD of the temperature sensor circuit110shown inFIG.8, a slope in the temperature range lower than the room temperature is relatively larger than a slope in the temperature range higher than the room temperature. In this way, the output data TD is likely to variate with respect to a temperature change at a low temperature, and the temperature adjustment rate is relatively high at the low temperature. Therefore, as shown inFIG.8, the calculation circuit120adjusts the temperature sensitivity of the temperature detection data ETD, so that a slope of the temperature detection data ETD in the temperature range higher than the room temperature is relatively larger than the slope thereof in the temperature range lower than the room temperature. Accordingly, the temperature detection data ETD is likely to variate with respect to a temperature change at a high temperature, and the temperature adjustment rate is likely to relatively increase at the high temperature. Accordingly, the temperature adjustment rate is low in the temperature range that is near the room temperature or that is lower than the room temperature, the temperature adjustment rate is high in the temperature range higher than the room temperature, and the followability of the temperature compensation for the frequency variation due to the temperature change is uniform. In other words, the calculation circuit120adjusts the temperature sensitivity of the temperature detection data ETD, so that the frequency sensitivity is uniform. As shown inFIG.9, the frequency sensitivity FScol when the temperature data is corrected by the calculation circuit120is more uniform than the frequency sensitivity FS when the temperature data is not corrected by the calculation circuit120. When the temperature data is not corrected by the calculation circuit120, it is required to increase the temperature adjustment rate to ensure the followability of the temperature compensation in a high temperature region where the frequency sensitivity FS is high. In the present embodiment, since the frequency sensitivity FScol in the high temperature region is reduced by the calculation circuit120correcting the temperature data, the temperature adjustment rate can be reduced as compared with the case in which the temperature data is not corrected by the calculation circuit120. Accordingly, the power consumption due to the temperature compensation can be reduced. In the present embodiment described above, the temperature detection circuit105includes the temperature sensor circuit110that executes the temperature detection, and the calculation circuit120that outputs, based on the output data TD of the temperature sensor circuit110, the temperature detection data ETD in which the temperature sensitivity is adjusted. The temperature detection rate control circuit190controls the temperature detection rate based on the temperature detection data ETD in which the temperature sensitivity is adjusted by the calculation circuit120. The temperature detection rate control circuit190controls the temperature detection rate based on a variation in the temperature detection data ETD. According to the present embodiment, the calculation circuit120adjusts the temperature sensitivity of the temperature detection data ETD, so that a degree of the temperature variation at which the variation of the temperature detection data ETD is determined is adjusted. By adjusting the temperature sensitivity of the temperature detection data ETD according to the frequency sensitivity as described above, the temperature adjustment rate can be increased in a temperature region where the frequency sensitivity is high, and the temperature adjustment rate can be decreased in a temperature region where the frequency sensitivity is low. Accordingly, the accuracy of the oscillation frequency can be improved, and power consumption of the temperature compensation can be reduced. In the present embodiment, the temperature properties of the oscillation frequency include a first sensitivity in a first temperature range, and a second sensitivity higher than the first sensitivity in a second temperature range lower or higher than the first temperature range. The calculation circuit120sets the temperature sensitivity of the temperature detection data ETD as a first temperature sensitivity in the first temperature range and as a second temperature sensitivity higher than the first temperature sensitivity in the second temperature range. In the example inFIG.8, when the first temperature range is any one of the temperature ranges TRA, TRB, and TRC, the second temperature range is the temperature range TRD. In addition, the frequency sensitivity FS corresponds to the temperature properties of the oscillation frequency in the example inFIG.9. As shown inFIGS.8and9, the frequency sensitivity FS in any one of the temperature ranges TRA, TRB, and TRC corresponds to the first sensitivity, and the frequency sensitivity FS in a temperature range TDR corresponds to the second sensitivity higher than the first sensitivity. At this time, as shown inFIG.8, the temperature sensitivity of the temperature detection data ETD in any one of the temperature ranges TRA, TRB, and TRC corresponds to the first temperature sensitivity, the temperature sensitivity of the temperature detection data ETD in the temperature range TRD corresponds to the second temperature sensitivity, and the second temperature sensitivity is set to be higher than the first temperature sensitivity.FIGS.8and9show examples in which the second temperature range is higher than the first temperature range, and the second temperature range may be lower than the first temperature range according to the temperature properties of the oscillation frequencies of the resonator and the oscillation circuit. According to the present embodiment, the temperature adjustment rate can be increased in a second temperature region where the frequency sensitivity is high, and the temperature adjustment rate can be decreased in a first temperature region where the frequency sensitivity is low. Accordingly, the followability of the temperature compensation can be increased and the accuracy of the oscillation frequency can be improved in the second temperature region where the frequency sensitivity is high, and the temperature adjustment rate can be decreased and the power consumption of the temperature compensation can be reduced in the first temperature region where the frequency sensitivity is low. 4. Calculation Circuit FIG.10is a detailed configuration example of the calculation circuit120. The calculation circuit120includes start point setting circuits KSA, KSB, and KSC, multiplication circuits MLA, MLB, and MLC, and an addition circuit126. The start point setting circuit KSA sets a start temperature Ta of the temperature range TRA inFIG.8. The start temperature Ta is a boundary between the adjacent temperature ranges TRA and TRB, and is an upper limit of the temperature range TRA. Specifically, the start temperature Ta is set according to the output data TD=TDa of the temperature sensor circuit110corresponding to the start temperature Ta. The start point setting circuit KSA outputs differential temperature data KSAQ=TD−TDa when TD≥TDa, and outputs differential temperature data KSAQ=0 when TD<TDa. The multiplication circuit MLA outputs output data MLAQ=−KSAQ×GA=−(TD−TDa)×GA. When TD<TDa, MLAQ=0. The start point setting circuit KSB sets a start temperature Tb of the temperature range TRC inFIG.8. The start temperature Tb is a boundary between the adjacent temperature ranges TRB and TRC, and is a lower limit of the temperature range TRC. Specifically, the start temperature Tb is set according to the output data TD=TDb of the temperature sensor circuit110corresponding to the start temperature Tb. The start point setting circuit KSB outputs differential temperature data KSBQ=−(TD−TDb) when TD≤TDb, and outputs differential temperature data KSBQ=0 when TD>TDb. The multiplication circuit MLB outputs output data MLBQ=−KSBQ×GB=(TD−TDb)×GB. When TD>TDb, MLBQ=0. The start point setting circuit KSC sets a start temperature Tc of the temperature range TRD inFIG.8. The start temperature Tc is a boundary between the adjacent temperature ranges TRC and TRD, and is a lower limit of the temperature range TRD. Specifically, the start temperature Tc is set according to the output data TD=TDc of the temperature sensor circuit110corresponding to the start temperature Tc. The start point setting circuit KSC outputs differential temperature data KSCQ=−(TD−TDc) when TD≤TDc, and outputs differential temperature data KSCQ=0 when TD>TDc. The multiplication circuit MLC outputs output data MLCQ=−KSCQ×GC=(TD−TDc)×GC. A gain GC satisfies GC>0. when TD>TDc, MLCQ=0. The addition circuit126adds the output data TD, the output data MLAQ, the output data MLBQ, the output data MLCQ, and an offset value EQOF, and outputs a result of the addition as the temperature detection data ETD. In the temperature range TRA, ETD=TD−(TD−TDa)×GA+EQOF=(1−GA)×TD+(TDa×GA+EQOF), and the temperature sensitivity of the temperature detection data ETD is lower than the temperature sensitivity of the output data TD of the temperature sensor circuit110. In the temperature range TRB, ETD=TD+EQOF, and the temperature sensitivity of the temperature detection data ETD is the same as the temperature sensitivity of the output data TD of the temperature sensor circuit110. In the temperature range TRC, ETD=TD+(TD−TDb)×GB+EQOF=(1+GB)×TD+(−TDb×GB+EQOF), and the temperature sensitivity of the temperature detection data ETD is higher than the temperature sensitivity of the output data TD of the temperature sensor circuit110. In the temperature range TRD, ETD=TD+(TD−TDb)×GB+(TD−TDc)×GC+EQOF=(1+GB+GC)×TD+(−TDb×GB−TDc×GC+EQOF), and the temperature sensitivity of the temperature detection data ETD is further higher than the temperature sensitivity in the temperature range TRC. The offset value EQOF is set such that a lower limit of the temperature detection data ETD in an operating temperature range does not become negative. That is, the offset value EQOF is set such that the lower limit of the temperature detection data ETD is 0 or larger than 0. FIG.11is a detailed configuration example of the start point setting circuit KSA and the multiplication circuit MLA. Configurations of the start point setting circuits KSB and KSC and the multiplication circuits MLB and MLC are also the same. The start point setting circuit KSA includes an addition circuit ADa, a sign inversion circuit SRa1, a selector SLa1, and a ReLU circuit RLa. ReLU is an abbreviation for rectified linear unit. The addition circuit ADa adds an offset OFFa to the output data TD of the temperature sensor circuit110. The sign inversion circuit SRa1inverts a sign of output data of the addition circuit ADa. The selector SLa1selects the output data of the addition circuit ADa when a sign selection signal ISGa is 0, and selects output data of the sign inversion circuit SRa1when the sign selection signal ISGa is 1. The ReLU circuit RLa outputs0when the output data of the selector SLa1is smaller than 0, and outputs the data as it is when the output data of the selector SLa1is 0 or larger. The offset OFFa and the sign selection signal ISGa are stored in the register170. In the example inFIG.8, it is set that the offset OFFa=−TDa and the sign selection signal ISGa=0. At this time, the output data of the addition circuit ADa is TD−TDa, the selector SLa1selects TD−TDa, and the ReLU circuit RLa outputs KSAQ=TD−TDa when TD−TDa≥0, and outputs KSAQ=0 when TD−TDa<0. The start temperature Ta of the temperature range RTA is specified according to the offset OFFa=−TDa, and an absolute value TDa of the offset OFFa is the output data of the temperature sensor circuit110corresponding to the start temperature Ta. As described above, an operation of the start point setting circuit KSA described with reference toFIG.10is achieved. The multiplication circuit MLA includes a bit shift circuit BSa, a sign inversion circuit SRa2, and a selector SLa2. The bit shift circuit BSa bit-shifts the differential temperature data KSAQ from the start point setting circuit KSA to multiply the differential temperature data KSAQ by the gain GA. A shift direction and a shift amount of the bit shift are specified according to a bit shift value GAa. The shift direction is a LSB direction or a MSB direction, and the shift amount is the number of bits to be shifted. The gain GA of the bit shift is 2, 4, 8, . . . in the MSB direction, and is 0.5, 0.25, 0.125, . . . in the LSB direction. In addition, when the shift amount is 0, the gain of the bit shift is GA=1. The sign inversion circuit SRa2inverts a sign of output data of the bit shift circuit BSa. The selector SLa2selects the output data of the bit shift circuit BSa when a sign selection signal QSGa is 0, and selects output data of the sign inversion circuit SRa2when the sign selection signal QSGa is 1. The bit shift value GAa and the sign selection signal QSGa are stored in the register170. In the example inFIG.8, a shift direction of the bit shift value GAa is set in the LSB direction. That is, the gain of the bit shift is GA<1. In addition, it is set that the sign selection signal QSGa=1. At this time, the output data of the bit shift circuit BSa is KSAQ×GA, and the selector SLa2selects −KSAQ×GA. Accordingly, MLAQ=−(TD−TDa)×GA is output when TD≥TDa, and MLAQ=0 is output when TD<TDa. As described above, an operation of the multiplication circuit MLA described with reference toFIG.11is achieved. 5. Temperature Compensation Circuit, Adjustment Circuit, and Oscillation Circuit FIG.12shows a detailed configuration example of the temperature compensation circuit135and the adjustment circuit154, and a coupling configuration example of the resonator10, the oscillation circuit160, and the adjustment circuit154. Hereinafter, it is assumed that n is an integer of 1 or more and the temperature detection data ETD is n+1 bit data ETD [n:0]. First, the detailed configuration example of the temperature compensation circuit135will be described. The temperature compensation circuit135includes a storage circuit130and an interpolation circuit152. The storage circuit130stores a look-up table131indicating correspondence between the temperature detection data ETD [n:0] and frequency adjustment data. Specifically, an upper bit ETD [n:i+1] of the temperature detection data ETD [n:0] is input to the storage circuit130as an address of the look-up table131. i is an integer of 1 or more and n or less. The look-up table131stores the frequency adjustment data at each address, and the storage circuit130outputs frequency adjustment data CLa at the address specified according to the upper bit ETD [n:i+1] and frequency adjustment data CLb at an adjacent address. The storage circuit130is, for example, a semiconductor memory such as a nonvolatile memory or a RAM, or a register including a latch circuit or the like. The nonvolatile memory is, for example, an OTP memory such as a FAMOS memory, and is not limited thereto. Alternatively, the nonvolatile memory may be an EEPROM such as a MONOS memory, or a fuse ROM, or the like. FAMOS is an abbreviation for floating gate avalanche injection metal oxide semiconductor. MONOS is an abbreviation for metal-oxide-nitride-oxide-silicon. The interpolation circuit152outputs the adjustment data QCL by interpolating between the frequency adjustment data CLa and the frequency adjustment data CLb based on a lower bit ETD [i:0] of the temperature detection data ETD [n:0]. The frequency adjustment data stored in the look-up table131is data for reducing a temperature dependence of the oscillation frequencies of the oscillator circuit160and the resonator10, and the oscillation frequencies become constant regardless of the temperature by adjusting the oscillation frequencies using the frequency adjustment data. Next, the coupling configuration example of the resonator10, the oscillation circuit160, and the adjustment circuit154will be described. One end of the resonator10is coupled to a terminal TX1, and the other end of the resonator10is coupled to a terminal TX2. The terminals TX1and TX2are terminals of the circuit device100, and are, for example, pads that are provided at a semiconductor substrate. One end of a capacitor CX1is coupled to the terminal TX1, and the other end of the capacitor CX1is coupled to a ground node. One end of a capacitor CX2is coupled to the terminal TX2, and the other end of the capacitor CX2is coupled to the ground node. The capacitors CX1and CX2are provided, for example, as external components of the circuit device100. The resonator10, the oscillation circuit160, and the capacitors CX1and CX2constitute a so-called colpitts type oscillation circuit. The oscillation circuit160generates a drive signal SDR by inverting and amplifying a signal SIN received from the other end of the resonator10via the terminal TX2, and outputs the drive signal SDR to the one end of the resonator10via the terminal TX1. The oscillation circuit160is, for example, an inverter, and an input node of the inverter is coupled to the terminal TX2, and an output node of the inverter is coupled to the terminal TX1. However, the oscillation circuit160is not limited thereto, and may be various amplifier circuits such as an amplifier circuit using a bipolar transistor. The oscillation signal is, for example, the drive signal SDR. The drive signal SDR may be output as the clock signal CLK inFIG.1, and the circuit device100may include an output circuit that outputs the clock signal CLK by buffering or dividing the drive signal SDR. Next, the detailed configuration example of the adjustment circuit154will be described. The adjustment circuit154includes capacitance array circuits CAC1and CAC2. The capacitance array circuit CAC1is coupled to the terminal TX1, and the capacitance array circuit CAC2is coupled to the terminal TX2. Hereafter, a configuration is described using the capacitance array circuit CAC1as an example, and the capacitance array circuit CAC2also has the same configuration. The capacitance array circuit CAC1includes first to m-th capacitors and first to m-th switches. m is an integer of two or more. The j-th capacitor and the j-th switch are coupled in series between the terminal TX1and the ground node. j=1, 2, . . . , and m. The first to m-th switches are controlled to be turned on or off according to the adjustment data QCL from the interpolation circuit152. Accordingly, a capacitance value of the capacitance array circuit CAC1is controlled according to the adjustment data QCL, and the oscillation frequency is adjusted. A circuit device according to the present embodiment described above includes an oscillation circuit that generates an oscillation signal using a resonator, a temperature detection circuit that outputs temperature detection data, a temperature compensation circuit, and a temperature detection rate control circuit. The temperature compensation circuit performs temperature compensation on an oscillation frequency of the oscillation signal based on the temperature detection data. The temperature detection rate control circuit controls a temperature detection rate at which the temperature detection circuit executes temperature detection. The temperature detection rate control circuit controls the temperature detection rate based on a variation in the temperature detection data. According to the present embodiment, when an environmental temperature changes, the temperature detection rate is controlled based on the variation in the temperature detection data. Accordingly, it is possible to cause the temperature compensation to follow the temperature variation caused by various factors and at various timings, and a highly accurate oscillation frequency is maintained. In addition, since the temperature detection rate can be decreased when there is no or small temperature variation, both high accuracy of the oscillation frequency and low power consumption of the circuit can be achieved. In the present embodiment, the temperature detection circuit may include a temperature sensor circuit that executes the temperature detection, and a calculation circuit that outputs, based on output data of the temperature sensor circuit, temperature detection data in which the temperature sensitivity is adjusted. The temperature detection rate control circuit may control the temperature detection rate based on the temperature detection data in which the temperature sensitivity is adjusted by the calculation circuit. The temperature detection rate control circuit controls the temperature detection rate based on a variation in the temperature detection data. According to the present embodiment, the calculation circuit adjusts the temperature sensitivity of the temperature detection data, so that a degree of the temperature variation at which the variation of the temperature detection data is determined is adjusted. By adjusting the temperature sensitivity of the temperature detection data according to the frequency sensitivity, the temperature adjustment rate can be increased in a temperature region where the frequency sensitivity is high, and the temperature adjustment rate can be decreased in a temperature region where the frequency sensitivity is low. Accordingly, the accuracy of the oscillation frequency can be improved, and power consumption of the temperature compensation can be reduced. In the present embodiment, temperature properties of the oscillation frequency may include a first sensitivity in a first temperature range, and a second sensitivity higher than the first sensitivity in a second temperature range lower or higher than the first temperature range. The calculation circuit may set the temperature sensitivity of the temperature detection data as a first temperature sensitivity in the first temperature range and as a second temperature sensitivity higher than the first temperature sensitivity in the second temperature range. According to the present embodiment, the temperature adjustment rate can be increased in a second temperature region where the frequency sensitivity is high, and the temperature adjustment rate can be decreased in a first temperature region where the frequency sensitivity is low. Accordingly, the followability of the temperature compensation can be increased and the accuracy of the oscillation frequency can be improved in the second temperature region where the frequency sensitivity is high, and the temperature adjustment rate can be decreased and the power consumption of the temperature compensation can be reduced in the first temperature region where the frequency sensitivity is low. In the present embodiment, the temperature detection circuit may execute an intermittent operation of executing temperature detection in an intermittent temperature detection period. The temperature detection rate control circuit may control the temperature detection rate by controlling a rate of the intermittent operation. According to the present embodiment, the temperature compensation is intermittently performed by detecting a temperature by the intermittent operation. Accordingly, the power consumption of the temperature compensation is reduced. Further, both high accuracy of an oscillation frequency and low power consumption of a circuit can be achieved by adaptively controlling the rate of the intermittent operation according to the variation in the temperature detection data. In the present embodiment, the temperature detection rate control circuit may set the temperature detection rate to a n-th rate that is the highest in a first rate to the n-th rate when the variation in the temperature detection data is a first threshold value or more. n is an integer of two or more. According to the present embodiment, when there is a temperature variation in which the variation in the temperature detection data is the first threshold value or more, that is, when it is determined that a rapid temperature variation occurs, the temperature detection rate is set to the maximum n-th rate. Accordingly, the temperature compensation follows a variation in an oscillation frequency due to the rapid temperature variation, and a highly accurate oscillation signal with a reduced frequency deviation is obtained. In the present embodiment, the temperature detection rate control circuit may decrease the temperature detection rate from the k-th rate to the (k−1)-th rate in the first rate to the n-th rate when the variation in the temperature detection data is the second threshold value or less. The second threshold value is smaller than the first threshold value. k is an integer of two or more and n or less. According to the present embodiment, when it is determined that a temperature variation occurring between a previous temperature compensation and a current temperature compensation at a current temperature adjustment rate is small, that is, the temperature adjustment rate is too high with respect to the temperature variation, the temperature detection rate decreases by one stage. Accordingly, the temperature adjustment rate can be set to be as low as possible in a range in which the temperature compensation can follow the variation in the oscillation frequency due to the temperature variation. In the present embodiment, an initial value of the temperature detection rate may be the n-th rate. Since the variation in the temperature detection data is unknown in an initial state, a followability of the temperature compensation can be ensured from a start of temperature adjustment by setting the n-th rate as the initial value. In addition, when the variation in the temperature detection data is small, the temperature adjustment rate adaptively decreases as described above. In the present embodiment, the temperature detection rate control circuit may control the temperature detection rate based on a difference between the previous temperature detection data and the current temperature detection data. The difference between the previous temperature detection data and the current temperature detection data indicates the temperature variation occurring between the previous temperature compensation and the current temperature compensation, and relates to the frequency deviation of the oscillation frequency occurring between the previous temperature compensation and the current temperature compensation. By controlling the temperature adjustment rate according to the difference between the previous temperature detection data and the current temperature detection data, the temperature adjustment rate is controlled such that the frequency deviation of the oscillation frequency occurring between the previous temperature compensation and the current temperature compensation is in an appropriate range. In the present embodiment, the temperature compensation circuit may perform the temperature compensation at the temperature detection rate. According to the present embodiment, the temperature adjustment rate that is a rate of the temperature compensation can be controlled by the temperature detection rate control circuit controlling the temperature detection rate. Accordingly, since an overall rate of the temperature compensation from a time point when the temperature detection circuit detects a temperature to a time point when the adjustment circuit adjusts the oscillation frequency is controlled, power consumption of an entire circuit relating to the temperature compensation decreases when the temperature detection rate is reduced. An oscillator according to the present embodiment includes a resonator and a circuit device. The circuit device includes an oscillation circuit that generates an oscillation signal using a resonator, a temperature detection circuit that outputs temperature detection data, a temperature compensation circuit, and a temperature detection rate control circuit. The temperature compensation circuit performs temperature compensation of an oscillation frequency of the oscillation signal based on the temperature detection data. The temperature detection rate control circuit controls a temperature detection rate at which the temperature detection circuit executes temperature detection. The temperature detection rate control circuit controls the temperature detection rate based on a variation in the temperature detection data. In the present embodiment, the temperature detection circuit may include a temperature sensor circuit that executes the temperature detection, and a calculation circuit that outputs, based on output data of the temperature sensor circuit, temperature detection data in which the temperature sensitivity is adjusted. The temperature detection rate control circuit may control the temperature detection rate based on the temperature detection data in which the temperature sensitivity is adjusted by the calculation circuit. In the present embodiment, temperature properties of the oscillation frequency may include a first sensitivity in a first temperature range, and a second sensitivity higher than the first sensitivity in a second temperature range lower or higher than the first temperature range. The calculation circuit may set the temperature sensitivity of the temperature detection data as a first temperature sensitivity in the first temperature range and as a second temperature sensitivity higher than the first temperature sensitivity in the second temperature range. Although the present embodiment has been described in detail as described above, it will be readily apparent to those skilled in the art that multiple modifications may be made without departing substantially from novel matters and effects according to the present disclosure. Therefore, all such modifications are intended to be included within the scope of the present disclosure. For example, a term cited with a different term having a broader meaning or the same meaning at least once in the specification or in the drawings can be replaced with the different term in any place in the specification or in the drawings. In addition, all combinations of the present embodiment and the modifications are also included in the scope of the present disclosure. Further, the configurations and operations of the circuit device, the resonator, and the oscillator are not limited to those described in the present embodiment, and various modifications can be made. | 61,535 |
11863124 | DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, the present embodiment will be described. The present embodiment to be described below does not unduly limit contents described in the claims. All configurations described in the present embodiment are not necessarily essential constituent elements. 1. Circuit Device FIG.1is a diagram illustrating a configuration example of a circuit device20according to the present embodiment. The circuit device20according to the present embodiment includes an oscillation circuit30, a temperature compensation circuit40, and a frequency control circuit50. An oscillator4according to the present embodiment includes a resonator10and the circuit device20. The resonator10is electrically coupled to the circuit device20. The resonator10is an element that generates mechanical oscillation according to an electrical signal. The resonator10can be implemented by, for example, a resonator element such as a quartz crystal resonator element. For example, the resonator10can be implemented by a quartz crystal resonator element that has a cut angle of AT cut, SC cut, or the like and that performs thickness-shear oscillation, a tuning fork type quartz crystal resonator element, or a double-tuning fork type quartz crystal resonator element. For example, the resonator10may be a resonator built in a temperature compensated crystal oscillator (TCXO) not provided with an oven, or may be a resonator built in an oven controlled crystal oscillator (OCXO) provided with an oven. The vibrator10according to the present embodiment can be implemented by various resonator elements such as a resonator element other than a thickness-shear oscillating type, a tuning fork type, or a double-tuning fork type, and a piezoelectric resonator element formed of a material other than quartz crystal. For example, a surface acoustic wave (SAW) resonator, or a micro electro mechanical systems (MEMS) resonator as a silicon resonator formed using a silicon substrate may be employed as the resonator10. The circuit device20is an integrated circuit device called an integrated circuit (IC). For example, the circuit device20is an IC manufactured by a semiconductor process, and is a semiconductor chip in which a circuit element is formed on a semiconductor substrate. InFIG.1, the circuit device20includes the oscillation circuit30, the temperature compensation circuit40, and the frequency control circuit50, and may also include a temperature sensor48. The oscillation circuit30includes a first variable capacitance circuit31and a second variable capacitance circuit32. The oscillation circuit30is a circuit that oscillates the resonator10. For example, the oscillation circuit30oscillates the resonator10to generate an oscillation signal. The oscillation signal is an oscillation clock signal. For example, the oscillation circuit30can be implemented by an oscillation drive circuit electrically coupled to one end and the other end of the resonator10, and a passive element such as a capacitor and a resistor. The drive circuit can be implemented by, for example, a CMOS inverter circuit or a bipolar transistor. The drive circuit is a core circuit of the oscillation circuit30, and the drive circuit oscillates the resonator10by driving the resonator10with a voltage or a current. As the oscillation circuit30, various types of oscillation circuits such as an inverter type, a Pierce type, a Colpitts type, and a Hartley type can be used. Note that coupling in the present embodiment is electrical coupling. The electrical coupling is coupling in which electrical signals can be transmitted, and is coupling in which information can be transmitted by the electrical signals. The electrical coupling may be coupling established via a passive element or the like. The oscillation circuit30includes the first variable capacitance circuit31and the second variable capacitance circuit32. The first variable capacitance circuit31and the second variable capacitance circuit32are, for example, circuits that change a capacitance formed at least at one of one end and the other end of the resonator10, and an oscillation frequency of the oscillation circuit30can be adjusted by adjusting capacitances of the first variable capacitance circuit31and the second variable capacitance circuit32. The first variable capacitance circuit31and the second variable capacitance circuit32can be implemented by a variable capacitance element such as a varactor. For example, each of the first variable capacitance circuit31and the second variable capacitance circuit32includes at least one variable capacitance element. The temperature compensation circuit40is a circuit that performs temperature compensation for the oscillation frequency of the oscillation circuit30. For example, the temperature compensation circuit40outputs a temperature compensation voltage VCP for temperature compensating the oscillation frequency of the oscillation circuit30, based on a temperature detection result of the temperature sensor48. The temperature detection result is a temperature detection signal, for example, a temperature detection voltage. The temperature compensation is, for example, processing of performing compensation by reducing a fluctuation in oscillation frequency caused by a temperature fluctuation. That is, the temperature compensation circuit40performs temperature compensation for the oscillation frequency of the oscillation circuit30such that the oscillation frequency is constant even when a temperature fluctuation occurs. The temperature sensor48is a sensor that detects a temperature. Specifically, the temperature sensor48outputs, as a temperature detection voltage, a temperature dependent voltage that changes in accordance with an environmental temperature. For example, the temperature sensor48generates a temperature detection voltage, which is a temperature detection signal, by using a circuit element having temperature dependency. Specifically, the temperature sensor48outputs the temperature detection voltage, which changes depending on the temperature by using, for example, temperature dependence of a forward voltage of a PN junction. Although the temperature sensor48is provided in the circuit device20inFIG.1, a modification may be made in which the temperature sensor48is provided outside the circuit device20and the temperature compensation circuit40performs temperature compensation based on a temperature detection signal such as a temperature detection voltage input from the outside. A modification in which a digital temperature sensor circuit is used as the temperature sensor48is also possible. In this case, the temperature detection voltage may be generated by performing D/A conversion on temperature detection data. The frequency control circuit50is a circuit that controls the oscillation frequency of the oscillation circuit30. Specifically, the frequency control circuit50outputs a frequency control voltage VFC for the oscillation frequency. For example, the frequency control circuit50generates the frequency control voltage VFC based on a control voltage input from the outside, and outputs the generated frequency control voltage VFC. Alternatively, the frequency control circuit50may generate the frequency control voltage VFC based on a control voltage obtained by performing D/A conversion on control data input from the outside. By providing such a frequency control circuit50, it is possible to implement control of setting the oscillation frequency of the oscillation circuit30to a desired frequency. For example, by providing the temperature compensation circuit40and the frequency control circuit50, it is possible to set the oscillation frequency to a desired frequency according to the control voltage and the control data input from the outside while performing temperature compensation for the oscillation frequency. The oscillation circuit30includes the first variable capacitance circuit31whose capacitance change characteristic with respect to the capacitance control voltage is a positive characteristic, and the second variable capacitance circuit32whose capacitance change characteristic with respect to the capacitance control voltage is a negative characteristic. For example, the first variable capacitance circuit31and the second variable capacitance circuit32have different polarities of capacitance change characteristic with respect to the capacitance control voltage, that is, one has a positive capacitance change characteristic and the other has a negative capacitance change characteristic. The positive capacitance change characteristic refers to, for example, a change characteristic that the capacitance increases as the capacitance control voltage increases, as will be described later with reference toFIG.3. The negative capacitance change characteristic refers to, for example, a change characteristic that the capacitance decreases as the capacitance control voltage increases, as will be described later with reference toFIG.4. The capacitance can also be referred to as a capacitance value. In the present embodiment, as illustrated inFIG.1, the temperature compensation circuit40supplies the temperature compensation voltage VCP as a capacitance control voltage to the first variable capacitance circuit31. The frequency control circuit50supplies the frequency control voltage VFC as a capacitance control voltage to the second variable capacitance circuit32. Since the first variable capacitance circuit31is a variable capacitance circuit having a positive characteristic, the capacitance of the first variable capacitance circuit31increases when the temperature compensation voltage VCP from the temperature compensation circuit40increases, and the capacitance of the first variable capacitance circuit31decreases when the temperature compensation voltage VCP decreases. Since the second variable capacitance circuit32is a variable capacitance circuit having a negative characteristic, the capacitance of the second variable capacitance circuit32decreases when the frequency control voltage VFC from the frequency control circuit50increases, and the capacitance of the second variable capacitance circuit32increases when the frequency control voltage VFC decreases. In this way, by providing the oscillation circuit30with the first variable capacitance circuit31having a positive characteristic to which the temperature compensation voltage VCP from the temperature compensation circuit40is supplied as the capacitance control voltage, for example, when the temperature rises, the capacitance of the first variable capacitance circuit31increases, and the oscillation frequency of the oscillation circuit30decreases. Accordingly, for example, in a high temperature range, when the oscillation frequency of the resonator10increases, since the capacitance of the first variable capacitance circuit31increases, it is possible to implement temperature compensation that cancels the increase in the oscillation frequency. By providing the first variable capacitance circuit31having a positive characteristic in the oscillation circuit30, for example, an amplifier circuit for a class A operation can be used as an output amplifier of the temperature compensation voltage VCP of the temperature compensation circuit40. Therefore, as compared with a case where an amplifier circuit for a class AB operation is provided as an output amplifier, it is possible to perform appropriate temperature compensation for the oscillation frequency in a wide temperature range while achieving a reduction in circuit scale. In addition, by providing the oscillation circuit30with the second variable capacitance circuit32having a negative characteristic to which the frequency control voltage VFC from the frequency control circuit50is supplied as the capacitance control voltage, when the frequency control voltage VFC increases, the capacitance of the second variable capacitance circuit32decreases, and the oscillation frequency of the oscillation circuit30increases. Therefore, the oscillation frequency can be variably controlled by using the frequency control voltage VFC. As a result, it is possible to implement control of the oscillation frequency based on the frequency control voltage VFC while implementing appropriate temperature compensation for the oscillation frequency in a wide temperature range by the temperature compensation circuit40. For example, since the frequency control circuit50does not necessarily need to be provided with an amplifier circuit, a reduction in circuit scale can be achieved. FIG.2is a diagram illustrating a detailed configuration example of the circuit device20and the oscillator4according to the present embodiment. InFIG.2, the circuit device20includes the oscillation circuit30, the temperature compensation circuit40, the temperature sensor48, the frequency control circuit50, a logic circuit60, a nonvolatile memory70, an output circuit80, and a power supply circuit90. The oscillator4includes the resonator10and the circuit device20. The resonator10is electrically coupled to the circuit device20. For example, the resonator10is electrically coupled to the circuit device20by using an internal wiring of a package that accommodates the resonator10and the circuit device20, a bonding wire, or a metal bump. The circuit device20and the oscillator4are not limited to the configuration inFIG.2, and various modifications such as omitting some of the components, adding other components, and replacing some of the components with other components can be made. The circuit device20includes pads PVDD, PGND, PX1, PX2, PVC, and PCK. The pad is a terminal of the circuit device20that is a semiconductor chip. For example, in a pad region, a metal layer is exposed from a passivation film that is an insulating layer, and the exposed metal layer forms the pad that is a terminal of the circuit device20. The pads PVDD and PGND are a power supply pad and a ground pad, respectively. A power supply voltage VDD from an external power supply device is supplied to the pad PVDD. The pad PGND is a pad to which GND, which is a ground voltage, is supplied. GND may be referred to as VSS, and the ground voltage is, for example, a ground potential. In the present embodiment, the ground voltage is appropriately described as GND. For example, VDD corresponds to a high potential side power supply, and GND corresponds to a low potential side power supply. The pads PX1and PX2are pads for coupling to the resonator10. The pad PVC is a pad for inputting a control voltage VC, and the pad PCK is a pad for outputting a clock signal CK. The pads PVDD, PGND, PVC, and PCK are electrically coupled to terminals TVDD, TGND, TVC, and TCK, respectively, which are external terminals for external coupling to the oscillator4. For example, each pad is electrically coupled to a corresponding terminal using an internal wiring of a package, a bonding wire, or a metal bump. The oscillation circuit30is electrically coupled to the resonator10via the pads PX1and PX2. The pads PX1and PX2are pads for coupling to the resonator. The oscillation drive circuit of the oscillation circuit30is provided between the pad PX1and the pad PX2. The oscillation circuit30includes the first variable capacitance circuit31and the second variable capacitance circuit32. Since the first variable capacitance circuit31and the second variable capacitance circuit32are electrically coupled to at least one of the pads PX1and PX2, a load capacitance of the oscillation circuit30can be variably adjusted. The temperature compensation circuit40performs analog temperature compensation according to polynomial approximation, for example. For example, when the temperature compensation voltage VCP for compensating a frequency-temperature characteristic of the resonator10is approximated by using a polynomial, the temperature compensation circuit40performs the analog temperature compensation based on coefficient information of the polynomial. The analog temperature compensation is, for example, temperature compensation implemented by addition processing of a current signal or a voltage signal that is an analog signal. For example, when the temperature compensation voltage VCP is approximated by using a high-order polynomial, a zero-order coefficient, a linear coefficient, and a high-order coefficient of the polynomial are stored in a storage unit implemented by, for example, the nonvolatile memory70as zero-order correction data, linear correction data, and high-order correction data, respectively. The high-order coefficient is, for example, a coefficient of an order higher than the first order, and the high-order correction data is correction data corresponding to the high-order coefficient. For example, when the temperature compensation voltage VCP is approximated by using a cubic polynomial, a zero-order coefficient, a linear coefficient, a quadratic coefficient, and a cubic coefficient of the polynomial are stored in the storage unit as zero-order correction data, linear correction data, quadratic correction data, and cubic correction data. Then, the temperature compensation circuit40performs temperature compensation based on the zero-order correction data to the cubic correction data. In this case, the quadratic correction data and temperature compensation based on the quadratic correction data may be omitted. For example, when the temperature compensation voltage VCP is approximated by using a quintic polynomial, a zero-order coefficient, a linear coefficient, a quadratic coefficient, a cubic coefficient, a quartic coefficient, and a quintic coefficient of the polynomial are stored in the storage unit as zero-order correction data, linear correction data, quadratic correction data, cubic correction data, quartic correction data, and quintic correction data. Then, the temperature compensation circuit40performs temperature compensation based on the zero-order correction data to the quintic correction data. In this case, the quadratic correction data or the quartic correction data, and the temperature compensation based on the quadratic correction data or the quartic correction data may be omitted. The order of polynomial approximation is any, and for example, polynomial approximation of an order higher than the fifth order may be performed. The zero-order correction may be performed by the temperature sensor48. A control voltage VC from the outside is input to the frequency control circuit50. For example, the control voltage VC from an external system implemented by a microcomputer or various ICs is input to the frequency control circuit50via the terminal TVC and the pad PVC. As an example, the oscillation circuit30of the circuit device20functions as a voltage controlled oscillator, and a feedback loop of PLL is formed by an external system. The frequency control circuit50outputs the frequency control voltage VFC corresponding to the control voltage VC from the outside. For example, the frequency control circuit50outputs the frequency control voltage VFC obtained by gain-adjusting the control voltage VC. Digital control data may be input to the circuit device20via an interface circuit (not illustrated), and the control voltage VC obtained by D/A converting the digital control data may be input to the frequency control circuit50. The logic circuit60is a control circuit and performs various types of control processing. For example, the logic circuit60controls the entire circuit device20or controls an operation sequence of the circuit device20. The logic circuit60performs various types of processing for controlling the oscillation circuit30, controls the temperature sensor48, the output circuit80, or the power supply circuit90, or controls reading and writing of information from and to the nonvolatile memory70. The logic circuit60can be implemented by, for example, an application specific integrated circuit (ASIC) using automatic placement and routing such as a gate array. The nonvolatile memory70is a memory that stores information even without power supply. For example, the nonvolatile memory70is a memory that can store information without power supply and in which information can be rewritten. The nonvolatile memory70stores various kinds of information necessary for operations of the circuit device20and the like. The nonvolatile memory70can be implemented by an electrically erasable programmable read-only memory (EEPROM) or the like that is implemented by a floating gate avalanche injection MOS memory (FAMOS memory) or a metal-oxide-nitride-oxide-silicon memory (MONOS memory). The nonvolatile memory70stores correction data such as linear correction data and high-order correction data used for temperature compensation of the temperature compensation circuit40. The output circuit80outputs the clock signal CK based on an oscillation signal from the oscillation circuit30. For example, the output circuit80buffers an oscillation signal, which is an oscillation clock signal from the oscillation circuit30, and outputs the buffered oscillation signal as the clock signal CK to the pad PCK. The clock signal CK is output to the outside via the clock output terminal TCK of the oscillator4. For example, the output circuit80outputs the clock signal CK in a single-ended CMOS signal format. The output circuit80may output the clock signal CK in a signal format other than the CMOS signal format. In addition, a clock signal generation circuit such as a PLL circuit that generates a clock signal CK having a frequency obtained by multiplying a frequency of an oscillation signal may be provided at a subsequent stage of the oscillation circuit30, and the output circuit80may buffer the clock signal CK generated by the clock signal generation circuit and output the buffered clock signal CK. The power supply circuit90is supplied with the power supply voltage VDD from the pad PVDD and the ground voltage GND from the pad PGND, and supplies various power supply voltages for an internal circuit in the circuit device20to the internal circuit. For example, the power supply circuit90supplies a regulated power supply voltage obtained by regulating the power supply voltage VDD to circuits in the circuit device20, such as the oscillation circuit30. FIG.3is a graph illustrating a positive voltage capacitance characteristic of the first variable capacitance circuit31.FIG.4is a graph illustrating a negative voltage capacitance characteristic of the second variable capacitance circuit32. The voltage capacitance characteristic is a characteristic of a capacitance C with respect to a capacitance control voltage VCC. Note thatFIGS.3and4schematically illustrate the voltage capacitance characteristic, and actually, the voltage capacitance characteristic is not a linear characteristic as illustrated inFIGS.3and4, but a characteristic having an inflection point at which an inclination is maximum in the vicinity of a center of a change range. As illustrated inFIG.3, the temperature compensation voltage VCP is input as the capacitance control voltage VCC to the first variable capacitance circuit31. According to the positive voltage capacitance characteristic of the first variable capacitance circuit31, when the temperature compensation voltage VCP increases, the capacitance C increases, and accordingly an oscillation frequency f of the oscillation circuit30decreases. On the other hand, as illustrated inFIG.4, the frequency control voltage VFC is input as the capacitance control voltage VCC to the second variable capacitance circuit32. According to the negative voltage capacitance characteristic of the second variable capacitance circuit32, when the frequency control voltage VFC increases, the capacitance C decreases, and accordingly the oscillation frequency f increases. FIG.5is a graph illustrating a frequency-temperature characteristic of the resonator10. Specifically, the frequency-temperature characteristic is a frequency-temperature characteristic of the AT-cut quartz crystal resonator10, for example. As illustrated inFIG.5, the resonator10has a frequency-temperature characteristic approximated by using a cubic curve. When the capacitance is to be adjusted with respect to the resonator10by using the second variable capacitance circuit32having a negative voltage capacitance characteristic, the temperature compensation circuit40needs to output the temperature compensation voltage VCP having a temperature characteristic as illustrated inFIG.6, for example. In this way, when the oscillation frequency of the resonator10increases in a high temperature range as indicated by A1inFIG.5, the temperature compensation voltage VCP output from the temperature compensation circuit40decreases as indicated by A2inFIG.6. Accordingly, the capacitance of the second variable capacitance circuit32having a negative characteristic increases, and the increase in the oscillation frequency of the resonator10is canceled out, whereby temperature compensation for maintaining the oscillation frequency constant can be implemented. However, in order to sufficiently decrease the temperature compensation voltage VCP in a high temperature range indicated by A2inFIG.6, it is necessary to widen a low-voltage-side operation range of an output amplifier of the temperature compensation circuit40. Therefore, in an amplifier circuit for a class A operation illustrated inFIG.17to be described later that serves as the output amplifier of the temperature compensation circuit40, it is difficult to widen the low-voltage-side operation range, and it is necessary to use an amplifier circuit for a class AB operation illustrated inFIG.18to be described later, which complicates the circuit and increases the circuit scale. On the other hand, when the capacitance is to be adjusted with respect to the resonator10by using the first variable capacitance circuit31having a positive voltage capacitance characteristic, the temperature compensation circuit40may output the temperature compensation voltage VCP having a temperature characteristic as illustrated inFIG.7, for example. In this way, when the oscillation frequency of the resonator10increases in the high temperature range as indicated by A1inFIG.5, the temperature compensation voltage VCP output from the temperature compensation circuit40increases as indicated by A3inFIG.7. Accordingly, the capacitance of the second variable capacitance circuit31having a positive characteristic increases, and the increase in the oscillation frequency of the resonator10is canceled out, whereby temperature compensation for maintaining the oscillation frequency constant can be implemented. In this case, in the high temperature range indicated by A3inFIG.7, in order to output a high temperature compensation voltage VCP from the temperature compensation circuit40, it is necessary to widen a high-voltage-side operation range of the output amplifier. In this regard, even in the amplifier circuit for a class A operation illustrated inFIG.17to be described later, since a P-type drive transistor constituting an output part is sufficiently turned on, the high temperature compensation voltage VCP as indicated by A3inFIG.7can also be appropriately output. Therefore, it is not essential to use the amplifier circuit for a class AB operation as illustrated inFIG.18as the output amplifier of the temperature compensation circuit40, and it is possible to achieve a reduction in scale and simplification of the circuit. As described above, the temperature compensation circuit40according to the present embodiment can include an amplifier circuit for a class A operation that outputs the temperature compensation voltage VCP. The amplifier circuit for a class A operation is an amplifier circuit including, for example, a differential part having a differential input and an output part coupled to the differential part, and the output part includes a P-type drive transistor and a transistor for current source that are coupled in series between a high potential side power supply node and a low potential side power supply node. In this way, appropriate temperature compensation for the oscillation frequency in a wide temperature range can be implemented by the temperature compensation circuit40including the amplifier circuit for a class A operation, which has a smaller circuit scale and a simpler configuration as compared with the amplifier circuit for a class AB operation, and it is possible to achieve both appropriate temperature compensation and a reduction in circuit scale. For example, the resonator10has a frequency-temperature characteristic approximated by using a cubic curve as illustrated inFIG.5. For example, the resonator10has a frequency-temperature characteristic approximated by using a polynomial such as a cubic polynomial. The resonator10is a resonator in which, in a state where temperature compensation is not performed as illustrated inFIG.5, an oscillation frequency fh at an upper limit of an operating temperature range indicated by A1is larger than a local maximum value fa of the oscillation frequency indicated by A4. The operating temperature range is a temperature range in which specification characteristics can be satisfied in the oscillator4and the circuit device20. For example, according to the cubic frequency-temperature characteristic of the resonator10illustrated inFIG.5, the oscillation frequency has the local maximum value fa at a temperature Ta as indicated by A4, and the oscillation frequency has a local minimum value fb at a temperature Tb as indicated by A5. The cubic frequency-temperature characteristic of the resonator10has an inflection point in the vicinity of, for example, 25° C. between the temperature Ta and the temperature Tb. Therefore, of the operating temperature range, a temperature range on the high temperature side is wider than a temperature range on the low temperature side. For example, when the operating temperature range is −40° C. to 125° C., the temperature range on the high temperature side is 25° C. to 125° C., which is wider than a range of −40° C. to 25° C. that is the temperature range on the low temperature side. The same applies to a case where the operating temperature range is −40° C. to 100° C. The oscillation frequency fh indicated by A1inFIG.5is an oscillation frequency at the upper limit of the operating temperature range, and is, for example, an oscillation frequency at the upper limit in a range of 100° C. to 125° C. Therefore, in FIG.5, a relationship that the oscillation frequency fh at the upper limit of the operating temperature range is larger than the local maximum value fa of the oscillation frequency is established. In this way, in the resonator10in which the oscillation frequency fh at the upper limit of the operating temperature range is larger than the local maximum value fa of the oscillation frequency, the temperature range on the high temperature side is wider than the temperature range on the low temperature side. Therefore, when the second variable capacitance circuit32having a negative characteristic is used as a variable capacitance circuit to which the temperature compensation voltage VCP is input, in order to sufficiently decrease the temperature compensation voltage VCP at the upper limit of the operating temperature range as indicated by A2inFIG.6, it is necessary to widen the low-voltage-side operation range of the output amplifier of the temperature compensation circuit40, and an amplifier circuit for a class AB operation is necessary. In contrast, in the present embodiment, the first variable capacitance circuit31having a positive characteristic is used as the variable capacitance circuit to which the temperature compensation voltage VCP is input. Therefore, even when an amplifier circuit for a class A operation is used as the output amplifier of the temperature compensation circuit40, the temperature compensation voltage VCP can be sufficiently increased at the upper limit of the operating temperature range as indicated by A3inFIG.7, and it is possible to achieve both appropriate temperature compensation and a reduction in circuit scale. 2. Oscillation Circuit Next, the oscillation circuit30will be described in detail.FIG.8is a diagram illustrating a first configuration example of the oscillation circuit30. The oscillation circuit30inFIG.8includes a drive circuit DV, a resistor RA, the first variable capacitance circuit31, and the second variable capacitance circuit32. The oscillation circuit30can include a capacitor CB5provided between a supply node of the temperature compensation voltage VCP and a low potential side power supply node, and a capacitor CB6provided between a supply node of the frequency control voltage VFC and a low potential side power supply node. The low potential side power supply node is, for example, a ground node. One end of the resonator10is coupled to a node N1, which is an input node of the drive circuit DV, via the pad PX1, and the other end of the resonator10is coupled to a node N2, which is an output node of the drive circuit DV, via the pad PX2. The resistor RA serving as a feedback element from the output to the input of the drive circuit DV has one end coupled to the node N1and the other end coupled to the node N2. InFIG.8, the first variable capacitance circuit31and the second variable capacitance circuit32are provided at both the node N1to which one end of the resonator10is coupled via the pad PX1and the node N2to which the other end of the resonator10is coupled via the pad PX2. However, the first variable capacitance circuit31and the second variable capacitance circuit32may be provided only at one of the nodes N1and N2. InFIG.8, the first variable capacitance circuit31is implemented by a transistor TR1, and the second variable capacitance circuit32is implemented by a transistor TR2. The transistors TR1and TR2are variable capacitance elements of a metal oxide semiconductor (MOS) type, and are also referred to as MOS varactors. InFIG.8, the first variable capacitance circuit31and the second variable capacitance circuit32are implemented by N-type transistors TR1and TR2, respectively. The MOS type variable capacitance element is a capacitance element, in which a source and a drain of a MOS transistor are short-circuited and an electrostatic capacitance generated between the short-circuited source and drain and a gate is variably controlled by using a capacitance control voltage. Although a case where each of the first variable capacitance circuit31and the second variable capacitance circuit32is implemented by one transistor that is a MOS type variable capacitance element will be mainly described as an example hereinafter, each of the first variable capacitance circuit31and the second variable capacitance circuit32may be implemented by two or more transistors provided in parallel. In addition, a modification in which a P-type transistor is used as a transistor constituting the first variable capacitance circuit31and the second variable capacitance circuit32can be made. Hereinafter, configurations of the first variable capacitance circuit31and the second variable capacitance circuit32provided at the node N1will be mainly described as an example. Since configurations of the first variable capacitance circuit31and the second variable capacitance circuit32coupled to the node N2are similar to those provided at the node N1, a detailed description thereof will be omitted. In the transistor TR1of the first variable capacitance circuit31on the node N1side, the gate thereof is supplied with the temperature compensation voltage VCP from the temperature compensation circuit40, and the source and the drain thereof are coupled to the node N1. Accordingly, a variable capacitance circuit having a positive voltage capacitance characteristic as illustrated inFIG.3is implemented. When the temperature compensation voltage VCP increases, a load capacitance of the node N1increases, and when the temperature compensation voltage VCP decreases, the load capacitance of the node N1decreases. Therefore, when the temperature compensation voltage VCP increases, the oscillation frequency decreases, and when the temperature compensation voltage VCP decreases, the oscillation frequency increases. As a result, with respect to the cubic frequency-temperature characteristic of the resonator10illustrated inFIG.5, the temperature compensation voltage VCP illustrated inFIG.7cancels the increase and decrease of the oscillation frequency, and thus the temperature compensation for making the oscillation frequency constant can be implemented. In the transistor TR2of the second variable capacitance circuit32on the node N1side, the source and the drain thereof are supplied with the frequency control voltage VFC from the frequency control circuit50, and the gate thereof is coupled to the node N1. Accordingly, a variable capacitance circuit having a negative voltage capacitance characteristic as illustrated inFIG.4is implemented. When the frequency control voltage VFC increases, the load capacitance of the node N1decreases, and when the frequency control voltage VFC decreases, the load capacitance of the node N1increases. Therefore, when the frequency control voltage VFC increases, the oscillation frequency increases, and when the frequency control voltage VFC decreases, the oscillation frequency decreases. Therefore, it is possible to implement VC-TCXO that is a voltage-controlled temperature compensation oscillator configured to control the oscillation frequency of the oscillation circuit30based on the control voltage VC from the outside. FIG.9is a diagram illustrating a second configuration example of the oscillation circuit30. The oscillation circuit30is not limited to the configurations inFIGS.8and9, and various modifications such as omitting some of the components, adding other components, and replacing some of the components with other components can be made. In the second configuration example inFIG.9, the first variable capacitance circuit31on the node N1side is implemented by the transistor TR1, and the second variable capacitance circuit32on the node N1side is implemented by the transistor TR2. The first variable capacitance circuit31is electrically coupled to the node N1via a capacitor CB1for DC cut, and the second variable capacitance circuit32is electrically coupled to the node N1via a capacitor CB2for DC cut. InFIG.9, a reference voltage generation circuit34that generates reference voltages VR1and VR2is provided. The first variable capacitance circuit31and the second variable capacitance circuit32on the node N2side are coupled to the node N2via capacitors CB3and CB4, respectively, and configurations thereof are similar to those of the first variable capacitance circuit31and the second variable capacitance circuit32on the node N1side, and thus a detailed description thereof will be omitted. The transistor TR1, which is a first variable capacitance element, and the capacitor CB1are provided in series between a supply node of the reference voltage VR1and the node N1. Specifically, one end of the capacitor CB1is coupled to the node N1, the other end of the capacitor CB1is coupled to the gate of the transistor TR1, and the reference voltage VR1is supplied to the source and the drain of the transistor TR1. The temperature compensation circuit40supplies the temperature compensation voltage VCP to a coupling node between the capacitor CB1and the transistor TR1via a resistor RB1. The transistor TR2, which is a second variable capacitance element, and the capacitor CB2are provided in series between a supply node of the reference voltage VR2and the node N1. Specifically, one end of the capacitor CB2is coupled to the node N1, the other end of the capacitor CB2is coupled to the source and the drain of the transistor TR2, and the reference voltage VR2is supplied to the gate of the transistor TR2. The frequency control circuit50supplies the frequency control voltage VFC to a coupling node between the capacitor CB2and the transistor TR2via a resistor RB2. InFIG.9, the first variable capacitance circuit31may be implemented by a plurality of transistors TR1provided in parallel, and the second variable capacitance circuit32may be implemented by a plurality of transistors TR2provided in parallel. In this case, a plurality of reference voltages VR1different from one another may be supplied to the sources and drains of the plurality of transistors TR1constituting the first variable capacitance circuit31. A plurality of reference voltages VR2different from one another may be supplied to the gates of the plurality of transistors TR2constituting the second variable capacitance circuit32. In this way, a linearity characteristic of a total capacitance of the first variable capacitance circuit31can be improved by overlapping different voltage capacitance characteristics of the plurality of transistors TR1. In addition, a linearity characteristic of a total capacitance of the second variable capacitance circuit32can be improved by overlapping different voltage capacitance characteristics of the plurality of transistors TR2. As illustrated inFIG.10, in the transistor TR1constituting the first variable capacitance circuit31, the temperature compensation voltage VCP is supplied to the gate, and the reference voltage VR1is supplied to the source and the drain. Accordingly, as illustrated inFIG.11, a voltage capacitance characteristic of the first variable capacitance circuit31is a positive voltage capacitance characteristic with respect to VGD=VCP−VR1, which is a gate-drain voltage. Therefore, as illustrated inFIG.12, the voltage capacitance characteristic of the first variable capacitance circuit31is also a positive voltage capacitance characteristic with respect to the temperature compensation voltage VCP. Specifically, inFIG.12, the positive voltage capacitance characteristic is shifted by the reference voltage VR1as indicated by B1with respect to that inFIG.11. InFIGS.11and12, Vth is a threshold voltage of the transistor TR1. As described above, the inclination of the voltage capacitance characteristic is maximum in the vicinity of the center of the change range, and inFIG.11, the inclination of the voltage capacitance characteristic is maximum when VGD=Vth. On the other hand, inFIG.12, the inclination of the voltage capacitance characteristic is maximum when VCP=Vth+VR1. As illustrated inFIG.13, in the transistor TR2constituting the second variable capacitance circuit32, the frequency control voltage VFC is supplied to the source and the drain, and the reference voltage VR2is supplied to the gate. Accordingly, as illustrated inFIG.14, a voltage capacitance characteristic of the second variable capacitance circuit32is a positive voltage capacitance characteristic with respect to VGD=VR2−VFC, which is a gate-drain voltage. InFIG.14, Vth is a threshold voltage of the transistor TR2. Therefore, as illustrated inFIG.15, the voltage capacitance characteristic of the second variable capacitance circuit32is a negative voltage capacitance characteristic with respect to the frequency control voltage VFC. Specifically, inFIG.15, the negative voltage capacitance characteristic has an inverted polarity and is shifted by the reference voltage VR2as indicated by B2with respect to that inFIG.14. 3. Temperature Compensation Circuit FIG.16is a diagram illustrating a configuration example of the temperature compensation circuit40. The temperature compensation circuit40is not limited to the configuration inFIG.16, and various modifications such as omitting some of the components, adding other components, and replacing some of the components with other components can be made. The temperature compensation circuit40is a circuit that outputs the temperature compensation voltage VCP according to polynomial approximation using a temperature as a variable. The temperature compensation circuit40includes a current generation circuit42and a current-voltage conversion circuit46. The current generation circuit42generates a function current based on a temperature detection result of the temperature sensor48. For example, the current generation circuit42generates a function current for temperature compensating the frequency-temperature characteristic of the resonator10as illustrated inFIG.5, based on a temperature detection voltage VTS that is a temperature detection result from the temperature sensor48. The current-voltage conversion circuit46converts the function current from the current generation circuit42into a voltage and outputs the temperature compensation voltage VCP. Specifically, the current-voltage conversion circuit46outputs the temperature compensation voltage VCP by an amplifier circuit AM for a class A operation. The current generation circuit42includes a linear correction circuit43and a high-order correction circuit44. The linear correction circuit43outputs, based on the temperature detection voltage VTS, a linear current approximating a linear function. For example, the linear correction circuit43outputs a linear function current based on linear correction data corresponding to a linear coefficient of a polynomial in polynomial approximation. The linear correction circuit43includes, for example, an operational amplifier, a first variable resistance circuit, a second variable resistance circuit, and a third variable resistance circuit. The operational amplifier, the first variable resistance circuit, and the second variable resistance circuit constitute a non-inverting amplifier circuit. The non-inverting amplifier circuit amplifies the temperature detection voltage VTS with reference to a reference voltage VRC, for example. The non-inverting amplifier circuit outputs the linear current to an input node of the current-voltage conversion circuit46via the third variable resistance circuit. The high-order correction circuit44outputs, based on the temperature detection voltage VTS, a high-order current approximating a high-order function to the current-voltage conversion circuit46. For example, the high-order correction circuit44outputs a high-order current based on high-order correction data corresponding to a high-order coefficient of a polynomial in polynomial approximation. As an example, the high-order correction circuit44outputs a cubic current that approximates a cubic function. In this case, the high-order correction circuit44includes a first differential circuit that performs a differential operation based on the temperature detection voltage VTS, and a second differential circuit that performs a differential operation based on an output voltage of the first differential circuit and the temperature detection voltage VTS to output a cubic current. InFIG.16, the temperature sensor48performs offset correction on the temperature detection voltage VTS based on zero-order correction data corresponding to a zero-order coefficient of a polynomial. That is, the temperature sensor48adjusts the offset of the temperature detection voltage VTS by offset indicated by the zero-order correction data. The offset correction on the temperature detection voltage VTS corresponds to zero-order correction in the temperature compensation for the oscillation frequency. The high-order correction circuit44may further include a correction circuit that performs fourth-order or higher-order correction. For example, the high-order correction circuit44may further include a quartic correction circuit that outputs a quartic current approximating a quartic function, a quintic correction circuit that outputs a quintic current approximating a quintic function, and the like. The current-voltage conversion circuit46adds the linear current and the high-order current, and performs current-voltage conversion on the added current to output the temperature compensation voltage VCP. Accordingly, the temperature compensation voltage VCP that approximates a polynomial function is generated. The current-voltage conversion circuit46includes an amplifier circuit AM, a resistor RC, and a capacitor CC. The amplifier circuit AM is implemented by an operational amplifier. The resistor RC and the capacitor CC are coupled in parallel between an output terminal and an inverting input terminal of the amplifier circuit AM. The reference voltage VRC is input to a non-inverting input terminal of the amplifier circuit AM. Accordingly, the current-voltage conversion circuit46outputs the temperature compensation voltage VCP by, for example, the amplifier circuit AM for a class A operation. According to the temperature compensation circuit40having such a configuration, the function current generated by the current generation circuit42based on the temperature detection result of the temperature sensor48can be converted into a voltage and output as the temperature compensation voltage VCP by the current-voltage conversion circuit46. Since the current-voltage conversion circuit46outputs the temperature compensation voltage by the amplifier circuit AM for a class A operation, the temperature compensation voltage VCP for appropriately temperature compensating the oscillation frequency in a wide temperature range can be output by the amplifier circuit AM for a class A operation that has a small circuit scale and a simple configuration. FIG.17is a diagram illustrating a configuration example of the amplifier circuit AM for a class A operation. The amplifier circuit AM includes a differential part DP and an output part QP. The differential part DP includes transistors TD1and TD2constituting a current mirror circuit, bipolar transistors BP1and BP2that are transistors as a differential pair, and a bipolar transistor BP3for a current source. The transistors TD1and TD2are P-type transistors whose gates are commonly coupled. A voltage VCP0from the current generation circuit42is input to an inverting input terminal that is a base of the bipolar transistor BP1, and the reference voltage VRC is input to a non-inverting input terminal that is a base of the bipolar transistor BP2. The output part QP includes a P-type drive transistor TD3and a bipolar transistor BP4for a current source, which are provided in series between a VDD node and a GND node. The amplifier circuit AM is provided with a resistor RD for phase compensation and a capacitor CD. Note that MOS transistors may be used instead of the bipolar transistors BP1to BP4. On the other hand,FIG.18is a diagram illustrating a configuration example of the amplifier circuit AM for a class AB operation.FIG.18is different fromFIG.17in that the output part QP includes a P-type drive transistor TD4and an N-type drive transistor TD5that are provided in series between a VDD node and a GND node. Further, inFIG.18, a switch circuit SW for controlling a gate of the drive transistor TD5and a bipolar transistor BP5are provided. As described above, when the second variable capacitance circuit32having a negative voltage capacitance characteristic is used in order to perform temperature compensation on the frequency-temperature characteristic of the resonator10illustrated inFIG.5, the temperature compensation circuit40needs to output the temperature compensation voltage VCP having the temperature characteristic illustrated inFIG.6. In this case, as indicated by A2inFIG.6, the amplifier circuit AM of the temperature compensation circuit40needs to output a sufficiently low voltage at the upper limit of the operating temperature range. However, when the amplifier circuit AM of a class A operation as illustrated inFIG.17is used as the output amplifier of the temperature compensation circuit40, it is difficult to output a sufficiently low temperature compensation voltage VCP indicated by A2inFIG.6by the bipolar transistor BP4that is a transistor for a current source of the output part QP. That is, since the bipolar transistor BP4for a current source is a transistor through which a constant current flows, the bipolar transistor BP4does not have a capability of drawing a large current to a low potential power supply side, and it is difficult to output a sufficiently low temperature compensation voltage VCP. In this case, when the constant current is set to a large current, it is easy to output a low voltage, but when the constant current is set to a large current, power consumption increases. Therefore, when the second variable capacitance circuit32having a negative characteristic is used for temperature compensating the frequency-temperature characteristic, it is essential to use the amplifier circuit AM for a class AB operation as illustrated inFIG.18or the like. This is because, according to the amplifier circuit AM for a class AB operation, a low temperature compensation voltage VCP can be output by sufficiently turning on the N-type drive transistor TD5of the output unit QP. However, when the amplifier circuit AM for a class AB operation is used, the circuit is complicated and the circuit scale is large. For example, in the amplifier circuit AM for a class AB operation, a filter circuit cannot be provided at the gate of the drive transistor of the output part QP. When a filter circuit is provided at an output node of the output part QP, the mirror effect is not obtained. Therefore, a capacitor having a large capacitance is required to set the cutoff frequency same as that of the amplifier circuit AM for a class A operation, and the scale of the circuit is increased. In this regard, in the present embodiment, the first variable capacitance circuit31having a positive voltage capacitance characteristic is used in order to perform temperature compensation on the frequency-temperature characteristic of the resonator10as illustrated inFIG.5. Therefore, the temperature compensation circuit40outputs the temperature compensation voltage VCP having the temperature characteristic as illustrated inFIG.7. In this case, the amplifier circuit AM of the temperature compensation circuit40needs to output a sufficiently high voltage at the upper limit of the operating temperature range as indicated by A3inFIG.7. Even when the amplifier circuit AM for a class A operation illustrated inFIG.17is used as the output amplifier of the temperature compensation circuit40, a high voltage as indicated by A3inFIG.7can be easily output by sufficiently turning on the P-type drive transistor TD3of the output part QP. Therefore, it is not essential to use the amplifier circuit AM for a class AB operation as illustrated inFIG.18, and thus a reduction in scale and simplification of the circuit can be achieved. 4. Frequency Control Circuit Next, the frequency control circuit50will be described in detail.FIG.19is a diagram illustrating a configuration example of the frequency control circuit50. The frequency control circuit50is not limited to the configuration inFIG.19, and various modifications such as omitting some of the components, adding other components, and replacing some of the components with other components can be made. The frequency control circuit50according to the present embodiment outputs, as the frequency control voltage VFC, a voltage generated by dividing the control voltage VC, which is input from the outside, using at least one variable resistor. For example, the frequency control circuit50generates the frequency control voltage VFC by dividing the control voltage VC using a variable resistor, without using an amplifier implemented by an operational amplifier or the like. For example, when an amplifier is used for gain adjustment of the frequency control circuit50, the circuit scale increases, the power consumption increases, and noise increases with an increase in the number of transistors. In this regard, the frequency control circuit50inFIG.19outputs, as the frequency control voltage VFC, a voltage generated by dividing the control voltage VC using a variable resistor. In this way, the oscillation frequency can be controlled by using the frequency control voltage VFC without using an amplifier that has a large circuit scale, a high power consumption and much noise, and thus a reduction in scale, a reduction in power consumption, a reduction in noise, and the like of the circuit device20can be achieved. Specifically, the frequency control circuit50inFIG.19includes a first variable resistor RA1, a second variable resistor RA2, a third variable resistor RA3, and a fourth variable resistor RA4. The first variable resistor RA1is provided between an input node NA1of the control voltage VC and an output node NA2of the frequency control voltage VFC. One end of the second variable resistor RA2is coupled to the output node NA2of the frequency control voltage VFC. The third variable resistor RA3is provided between an input node NA3of a reference voltage VREG and a coupling node NA4to which the other end of the second variable resistor RA2is coupled. The fourth variable resistor RA4is provided between the coupling node NA4and a low potential power supply node NA5. The low potential power supply node NA5is, for example, a GND node. In the frequency control circuit50inFIG.19, currents IA1, IA2, IA3, and IA4flowing through the first variable resistor RA1, the second variable resistor RA2, the third variable resistor RA3, and the fourth variable resistor RA4respectively are obtained using the following equations (1), (2), and (3). IA1=IA2=VC-VMRA1+RA2=VC-VFCRA1=VFC-VMRA2(1)IA3=VREG-VMRA3(2)IA4=IA2+IA3=VC-VMRA1+RA2+VREG-VMRA3(3) Accordingly, the voltage VM at the coupling node NA4, which is a voltage division node between the third variable resistor RA3and the fourth variable resistor RA4, is obtained by the following equations (4) and (5). VM=RA4×IA4=RA4×(VC-VMRA1+RA2+VREG-VMRA3)(4)VM=(RA1+RA2)×RA3×RA4(RA1+RA2)×(RA3+RA4)+RA3×RA4×(VCRA1+RA2+VREGRA3)(5) Therefore, the frequency control voltage VFC can be obtained using the following equations (6) and (7). VFC=RA2RA1+RA2×VC+RA1RA1+RA2×VM(6)VFC=(RA2(RA3+RA4)+RA3×RA4)VC+RA1×RA4×VREG(RA1+RA2)×(RA3+RA4)+RA3×RA4(7) Then, G=ΔVFC/ΔVC, which is gain in the frequency control circuit50, is expressed by the following equation (8). G=ΔVFCΔVC=RA2(RA3+RA4)+RA3×RA4(RA1+RA2)×(RA3+RA4)+RA3×RA4>0(8) As shown in the above equation (8), according to the frequency control circuit50of the present embodiment, it is possible to adjust the gain G of the frequency control voltage VFC with respect to the control voltage VC with a small-scale circuit configuration that does not use an amplifier. For example, the gain G in the frequency control circuit50can be adjusted by adjusting resistance values of the first variable resistor RA1to the fourth variable resistor RA4. Therefore, the gain G of the frequency control voltage VFC with respect to the control voltage VC can be adjusted to any value without using an amplifier that has a large circuit scale, a high power consumption and much noise. 5. Setting of Sensitivity In the present embodiment, an absolute value of frequency-voltage sensitivity of the first variable capacitance circuit31is set to be higher than an absolute value of frequency-voltage sensitivity of the second variable capacitance circuit32. For example, sensitivity KV=ΔF/ΔV, which is the frequency-voltage sensitivity, is expressed by the following equation (9). CL is an overall load capacitance of the oscillation circuit30, and CO and Cl are equivalent parallel capacitance and series capacitance of the resonator10. KV=ΔFΔV=-C12·(CO+CL)2×ΔCLΔV(9) As is clear from the above equation (9), ΔF, which indicates a frequency change, can be increased by increasing an absolute value of the sensitivity KV. Therefore, by setting the absolute value of the sensitivity KV of the first variable capacitance circuit31for temperature compensation to be higher than an absolute value of the sensitivity KV of the second variable capacitance circuit32for frequency control, a frequency fluctuation range of the first variable capacitance circuit31for temperature compensation can be made wider than a frequency fluctuation range of the second variable capacitance circuit32for frequency control. For example, according to specifications or the like, a frequency fluctuation range of about ±20 ppm to 30 ppm is required on the temperature compensation side, and a frequency fluctuation range of about ±5 ppm to 15 ppm is required on the frequency control side. Therefore, by setting the absolute value of the sensitivity KV of the first variable capacitance circuit31to be higher than the absolute value of the sensitivity KV of the second variable capacitance circuit32, it is possible to make the frequency fluctuation range on the temperature compensation side wider than the frequency fluctuation range on the frequency control side, and it is possible to meet the requirements according to the specifications or the like. In order to increase the absolute value of the sensitivity of a variable capacitance circuit, a size of a transistor constituting the variable capacitance circuit needs to be increased. Therefore, in order to set the absolute value of the sensitivity KV of the first variable capacitance circuit31to be higher than the absolute value of the sensitivity KV of the second variable capacitance circuit32, a size of the transistor TR1constituting the first variable capacitance circuit31may be made larger than a size of the transistor TR2constituting the second variable capacitance circuit32. In the following description, the absolute value of the sensitivity is simply referred to as sensitivity as appropriate. For example,FIG.20is a graph illustrating a voltage capacitance characteristic of a variable capacitance circuit. InFIG.20, an inclination of a voltage capacitance characteristic of D2is doubled as compared with that of D1. That is, capacitance voltage sensitivity of the variable capacitance circuit is doubled. However, even if the capacitance voltage sensitivity, which is the inclination of the voltage capacitance characteristic of the variable capacitance circuit, is doubled, the sensitivity KV=ΔF/ΔV is not doubled as illustrated inFIG.21. That is, even if the inclination of the voltage capacitance characteristic is doubled as indicated by D1and D2inFIG.20, the sensitivity KV expressed by the above equation (9) is not twice the sensitivity indicated by D3inFIG.21, but the sensitivity decreases by an amount corresponding to an increase in the load capacitance CL caused by an increase in a parasitic capacitance as indicated by D4, and is the sensitivity indicated by D5. For example, when the sensitivity KV of the second variable capacitance circuit32for frequency control is increased, the sensitivity KV of the first variable capacitance circuit31for temperature compensation decreases with an increase in the load capacitance CL caused by a parasitic capacitance of the second variable capacitance circuit32. Therefore, in the present embodiment, the sensitivity KV of the first variable capacitance circuit31for temperature compensation is preferentially made higher, and the sensitivity KV of the second variable capacitance circuit32for frequency control is made lower than that on the temperature compensation side. In this way, the transistor size of the second variable capacitance circuit32can be reduced, the circuit scale can be reduced, and a decrease in the sensitivity KV of the first variable capacitance circuit31caused by the parasitic capacitance of the second variable capacitance circuit32can also be prevented. 6. Oscillator FIG.22is a diagram illustrating a first structure example of the oscillator4according to the present embodiment. The oscillator4includes the resonator10, the circuit device20, and a package15that accommodates the resonator10and the circuit device20. The package15is made of, for example, ceramic, and has an accommodating space on an inner side. The resonator10and the circuit device20are accommodated in the accommodating space. The accommodating space is hermetically sealed, and is desirably in a depressurized state that is a state close to vacuum. With the package15, the resonator10and the circuit device20can be suitably protected from impact, dust, heat, moisture, and the like. The package15includes a base16and a lid17. Specifically, the package15includes the base16that supports the resonator10and the circuit device20, and the lid17that is joined to an upper surface of the base16so that the accommodating space is defined between the base16and the lid17. The resonator10is supported by a step portion, which is provided at an inner side of the base16, via a terminal electrode. The circuit device20is disposed on an inner bottom surface of the base16. Specifically, the circuit device20is disposed such that an active surface thereof faces the inner bottom surface of the base16. The active surface is a surface at which a circuit element of the circuit device20is formed. Bumps BMP are formed at terminals of the circuit device20. The circuit device20is supported by the inner bottom surface of the base16via the conductive bumps BMP. The conductive bumps BMP are, for example, metal bumps, and the resonator10is electrically coupled to the circuit device20via the bumps BMP, an internal wiring of the package15, a terminal electrode, and the like. The circuit device20is electrically coupled to external terminals18and19of the oscillator4via the bumps BMP and the internal wiring of the package15. The external terminals18and19are formed at an outer bottom surface of the package15. The external terminals18and19are coupled to an external device via an external wiring. The external wiring is, for example, a wiring formed at a circuit board mounted on the external device. Accordingly, a clock signal or the like can be output to the external device. Although the circuit device20is flip mounted such that the active surface of the circuit device20faces downward inFIG.22, the present embodiment is not limited to such mounting. For example, the circuit device20may be mounted such that the active surface of the circuit device20faces upward. That is, the circuit device20is mounted such that the active surface faces the resonator10. FIG.23is a diagram illustrating a second structure example of the oscillator4. The oscillator4includes the resonator10, the circuit device20, and the package15that accommodates the resonator10and the circuit device20. The package15includes the base16and the lid17. The base16includes a first substrate6that is an intermediate substrate, a second substrate7having a substantially rectangular frame shape that is laminated on an upper surface side of the first substrate6, and a third substrate8having a substantially rectangular frame shape that is laminated on a bottom surface side of the first substrate6. The lid17is joined to an upper surface of the second substrate7, and the resonator10is accommodated in an accommodating space S1that is defined by the first substrate6, the second substrate7, and the lid17. For example, the resonator10is hermetically sealed in the accommodating space S1, and the accommodating space S1is desirably in a depressurized state that is a state close to vacuum. Accordingly, the resonator10can be suitably protected from impact, dust, heat, moisture, and the like. The circuit device20that is a semiconductor chip is accommodated in an accommodating space S2defined by the first substrate6and the third substrate8. The external terminals18and19that are external coupling electrode terminals of the oscillator4are formed at a bottom surface of the third substrate8. In the accommodating space S1, the resonator10is coupled to, by conductive coupling portions CDC1and CDC2, a first electrode terminal and a second electrode terminal (not illustrated) formed at an upper surface of the first substrate6. For example, the conductive coupling portions CDC1and CDC2may be implemented by conductive bumps such as metal bumps, or may be implemented by conductive adhesives. Specifically, for example, a first electrode pad (not illustrated) formed at one end of the tuning-fork type resonator10is coupled to, via the conductive coupling portion CDC1, the first electrode terminal formed at the upper surface of the first substrate6. The first electrode terminal is electrically coupled to the pad PX1of the circuit device20. A second electrode pad (not illustrated) formed at the other end of the tuning-fork type resonator10is coupled to, via the conductive coupling portion CDC2, the second electrode terminal formed at the upper surface of the first substrate6. The second electrode terminal is electrically coupled to the pad PX2of the circuit device20. Accordingly, the one end and the other end of the resonator10can be electrically coupled to the pads PX1and PX2of the circuit device20via the conductive coupling portions CDC1and CDC2. The conductive bumps BMP are formed at a plurality of pads of the circuit device20that is a semiconductor chip, and these conductive bumps BMP are coupled to a plurality of electrode terminals formed at a bottom surface of the first substrate6. The electrode terminals coupled to the pads of the circuit device20are electrically coupled to the external terminals18and19of the oscillator4via an internal wiring or the like. The oscillator4may be an oscillator of a wafer level package (WLP). In this case, the oscillator4includes: a base that includes a semiconductor substrate and a penetration electrode penetrating between a first surface and a second surface of the semiconductor substrate; the resonator10that is fixed to the first surface of the semiconductor substrate via a conductive joining member such as a metal bump; and an external terminal that is provided at a second surface side of the semiconductor substrate via an insulating layer such as a re-wiring layer. An integrated circuit serving as the circuit device20is formed at the first surface or the second surface of the semiconductor substrate. In this case, by bonding a first semiconductor wafer disposed with a plurality of bases, each having the resonator10and the integrated circuit, to a second semiconductor wafer formed with a plurality of lids, the plurality of bases are joined to the plurality of lids, and then dicing of the oscillators4is performed using a dicing saw or the like. In this way, the oscillator4of the wafer level package can be implemented, and the oscillator4can be manufactured with high throughput and low cost. As described above, a circuit device according to the present embodiment includes: an oscillation circuit configured to oscillate a resonator; a temperature compensation circuit configured to output a temperature compensation voltage for temperature compensating an oscillation frequency of the oscillation circuit, based on a temperature detection result of a temperature sensor; and a frequency control circuit configured to output a frequency control voltage for the oscillation frequency. The oscillation circuit includes a first variable capacitance circuit whose capacitance change characteristic with respect to a capacitance control voltage is a positive characteristic, and a second variable capacitance circuit whose capacitance change characteristic with respect to the capacitance control voltage is a negative characteristic. The temperature compensation circuit supplies the temperature compensation voltage as the capacitance control voltage to the first variable capacitance circuit, and the frequency control circuit supplies the frequency control voltage as the capacitance control voltage to the second variable capacitance circuit. In this way, since the oscillation circuit is provided with the first variable capacitance circuit having a positive characteristic to which the temperature compensation voltage from the temperature compensation circuit is supplied as the capacitance control voltage, when the temperature rises, a capacitance of the first variable capacitance circuit increases and the oscillation frequency of the oscillation circuit decreases, and when the temperature falls, the capacitance of the first variable capacitance circuit decreases and the oscillation frequency of the oscillation circuit increases. Accordingly, it is possible to implement temperature compensation in which a change in the oscillation frequency due to a temperature change is canceled out and the oscillation frequency is made constant. In addition, since the oscillation circuit is provided with the second variable capacitance circuit having a negative characteristic to which the frequency control voltage from the frequency control circuit is supplied as the capacitance control voltage, when the frequency control voltage increases, a capacitance of the second variable capacitance circuit decreases and the oscillation frequency of the oscillation circuit increases, and when the frequency control voltage decreases, the capacitance of the second variable capacitance circuit increases and the oscillation frequency of the oscillation circuit decreases. Therefore, it is possible to implement control of the oscillation frequency based on the frequency control voltage while implementing appropriate temperature compensation for the oscillation frequency in a wide temperature range by the temperature compensation circuit. In the present embodiment, the temperature compensation circuit may include an amplifier circuit for a class A operation configured to output the temperature compensation voltage. In this way, appropriate temperature compensation for the oscillation frequency in a wide temperature range can be implemented by the temperature compensation circuit including an amplifier circuit for a class A operation that has a smaller circuit scale and a simpler configuration than an amplifier circuit for a class AB operation. In the present embodiment, the temperature compensation circuit may include a current generation circuit configured to generate a function current based on the temperature detection result of the temperature sensor, and a current-voltage conversion circuit configured to convert the function current into a voltage and output the temperature compensation voltage, and the current-voltage conversion circuit may output the temperature compensation voltage by the amplifier circuit for a class A operation. In this way, the function current generated by the current generation circuit based on the temperature detection result of the temperature sensor can be converted into a voltage and output as the temperature compensation voltage by the current-voltage conversion circuit. Since the current-voltage conversion circuit outputs the temperature compensation voltage by the amplifier circuit for a class A operation, the temperature compensation voltage for appropriately temperature compensating the oscillation frequency in a wide temperature range can be output by the amplifier circuit for a class A operation that has a small circuit scale and a simple configuration. In the present embodiment, the resonator may have a frequency-temperature characteristic approximated by using a cubic curve, and the resonator may be a resonator in which the oscillation frequency at an upper limit of an operating temperature range is larger than a local maximum value of the oscillation frequency in a state where temperature compensation is not performed. In this way, for the resonator in which the oscillation frequency at the upper limit of the operating temperature range is larger than the local maximum value of the oscillation frequency, a temperature range on the high temperature side is wider than a temperature range on the low temperature side, but in the present embodiment, the first variable capacitance circuit having a positive characteristic is used as the variable capacitance circuit to which the temperature compensation voltage is supplied. Therefore, the temperature compensation voltage can be sufficiently increased at the upper limit of the operating temperature range, and appropriate temperature compensation can be implemented. In the present embodiment, the frequency control circuit may output, as the frequency control voltage, a voltage generated by dividing a control voltage input from an outside by at least one variable resistor. In this way, the oscillation frequency can be controlled by using the frequency control voltage without using an amplifier that has a large circuit scale, a high power consumption and much noise, and a reduction in circuit scale, a reduction in power consumption and the like can be achieved. In the present embodiment, the frequency control circuit may include a first variable resistor provided between an input node of the control voltage and an output node of the frequency control voltage, a second variable resistor having one end coupled to the output node, a third variable resistor provided between an input node of a reference voltage and a coupling node to which the other end of the second variable resistor is coupled, and a fourth variable resistor provided between the coupling node and a low potential power supply node. In this way, by adjusting resistance values of the first variable resistor to the fourth variable resistor, gain of the frequency control voltage with respect to the control voltage can be adjusted without using an amplifier that has a large circuit scale, a high power consumption and much noise. In the present embodiment, an absolute value of frequency-voltage sensitivity of the first variable capacitance circuit may be higher than an absolute value of frequency-voltage sensitivity of the second variable capacitance circuit. In this way, a frequency fluctuation range on the temperature compensation side is made wider than a frequency fluctuation range on the frequency control side, and appropriate temperature compensation can be implemented. An oscillator according to the present embodiment includes a resonator and a circuit device. The circuit device includes: an oscillation circuit configured to oscillate the resonator; a temperature compensation circuit configured to output a temperature compensation voltage for temperature compensating an oscillation frequency of the oscillation circuit, based on a temperature detection result of a temperature sensor; and a frequency control circuit configured to output a frequency control voltage for the oscillation frequency. The oscillation circuit includes a first variable capacitance circuit whose capacitance change characteristic with respect to a capacitance control voltage is a positive characteristic, and a second variable capacitance circuit whose capacitance change characteristic with respect to the capacitance control voltage is a negative characteristic. The temperature compensation circuit supplies the temperature compensation voltage as the capacitance control voltage to the first variable capacitance circuit, and the frequency control circuit supplies the frequency control voltage as the capacitance control voltage to the second variable capacitance circuit. Although the present embodiment has been described in detail above, it will be easily understood by those skilled in the art that many modifications can be made without substantially departing from the novel matters and effects of the present disclosure. Therefore, all such modifications are intended to be included within the scope of the present disclosure. For example, a term cited with a different term having a broader meaning or the same meaning at least once in the specification or in the drawings can be replaced with the different term at any place in the specification or in the drawings. In addition, all combinations of the present embodiment and the modifications are also included in the scope of the present disclosure. The configurations, operations, and the like of the circuit device and the oscillator are not limited to those described in the present embodiment, and various modifications can be made. | 80,263 |
11863125 | MODE FOR CARRYING OUT THE INVENTION The present invention is a terahertz oscillator using a double barrier type resonant tunneling diode (RTD) and is a structure having no an MIM (Metal Insulator Metal) capacitor structure constituted by “metal (conductive material)/insulator/metal (conductive material)”. Since there is no provided the MIM capacitor structure, the structure is simple, and the steps of the producing process can be significantly reduced than the conventional producing steps. Further, since it is possible to obtain the oscillation in the terahertz frequency band due to the resonance of the RTD and the stabilizing resistors, it is especially desired to the application for an imaging field, a high speed communication and so on. Embodiments of the present invention will be described with reference to the accompanying drawings as follows. FIG.17is a perspective view of a terahertz oscillator100according to the present invention, andFIG.18is a three-dimensional perspective structure view. Further,FIG.19is a cross-sectional structure view taken along an X-X′ line inFIG.17,FIG.20is a cross-sectional structure view taken along a Y-Y′ line inFIG.17andFIG.22is a plane view showing the actually produced terahertz oscillator. An end surface of an electrode plate105facing to a slot portion surrounded by stabilizing resistors103and104is a slot antenna102(about 12 [μm]), and a rectangular conductive material member108is provided on a central portion of the slot antenna102, An RTD120is connected to a tip portion of the conductive material member108through a mesa121, and a portion below the conductive material member108is an air bridge structure to form the slot portion. The terahertz oscillator100according to the present invention does not have the MIM capacitor structure, and comprises the electrode plate105which is connected to a bias pad105A being grounded and a square shape electrode plate106which is connected to a bias pad106A to apply a DC bias Vb. The slot is formed between the end surface of the electrode plate105and an opposite end surface of the electrode plate106, and the end surface of the electrode plate106facing to the slot is the slot antenna102. Further, the electrode plates105and106are respectively connected by the stabilizing resistors103and104of both sides. A square recess in a plane view is disposed on a portion opposite to the slot antenna102of the electrode plate105, and a resonant tunneling diode (RTD)101is provided in the recess. The conductive material member108is suspended between the RTD101and the electrode plate106, and the slot is the air bridge structure as shown inFIGS.19and20. The RTD140is a double barrier of, for example, AlAs/InGaAs, which may be constituted by the layers: an “n+InGaAs” layer (5×1019[cm−3], 100 [nm])/spacer an “InGaAs” layer (undoped, 12 [nm])/barrier an “AlAs” layer (undoped, 0.9 [nm])/well an “InGaAs” layer (undoped, 3 [nm])/barrier an “AlAs” layer (undoped, 0.9 [nm])/spacer an “InAlGaAs” layer (undoped, 2 [nm])/an “n—InAlGaAs” layer (3×1018[cm−3], 25 [nm])/an “n+InGaAs” layer (5×1019[cm−3], 400 [nm]), from the top to the bottom. The equivalent circuit of the terahertz oscillator100is shown inFIG.21. The bias voltage Vb from the bias circuit107is applied to the RTD101through an inductance LWof the lines and an inductance LSof the orbital current of the slot antenna102, and the stabilizing resistors103and104(a total resistance value RS) are connected to the RTD101in parallel. In the low frequency band, since the inductances LWand LSmay be presumed as the short circuits and the resistance value RSof the stabilizing resistors103and104cancels the negative differential resistance (NDR) of the RTD101, the equivalent circuit becomesFIG.23A. On the contrary, in the high frequency band as the terahertz, the impedance of the inductance LWbecomes great and then the bias circuit is cut away, and further the equivalent circuit as shown inFIG.23Ais obtained by converting the series connected resistance R s and the inductance LSinto the parallel connection. However, since the loss G of the resistance RSof the stabilizing resistors103and104becomes small, the resonance of the inductance LSand the capacitance in the RTD101is occurred and oscillates. That is, assuming that the oscillation frequency of the oscillator is “f”, an angular frequency co is defined by “ω=2πf”. The loss G of the resistance RSis expressed by a following Expression 3. G=RsRs2+ω2LS2[Expression3] Since the resistance value RSis a small value (few Ω), “RS2” in the Expression 3 becomes almost zero. Accordingly, the Expression 3 can be approximated by a following Expression 4. G≈Rsω2LS2[Expression4] In the above Expression 4, since a square “ω2” of the angular frequency ω is a great value in the terahertz frequency, the Expression 4 becomes almost zero and the loss G of the resistance RScan be ignored. In this connection, the resonance of the inductance LSand the capacitance in the RTD101is occurred and oscillates. FIG.24shows an oscillation characteristic (frequency/intensity) of a prototype product (FIG.22) according to the present invention, and the oscillation of about 0.2 [THz] is obtained. FIGS.25A to25Fshow the producing processes of the terahertz oscillator100according to the present invention. First, the substrate structure110comprising four layers which are the resonant tunneling diode (RTD) layer114of the top, the etch stopper layer (n+-InP)113, the n+-InGaAs layer112and the Sl-InP substrate111of the bottom, is prepared as shown inFIG.25A. Then, the upper electrode121is formed on the RTD portion on the RTD layer114by the resist patterning with the exposure and the deposition, and the RTD mesa120is formed by removing a part of the RTD layer114with the wet etching as shown inFIG.25B. At a time of the RTD mesa formation with the wet etching, a side etch proceeds for the RTD mesa120existing under the conductive material member121as shown inFIG.26. That is, although the size of the RTD120immediately after the wet etching is started is a distance d1from an end surface of the mesa121as shown inFIG.26A, all of the RTD120existing under the portion108in which the conductive material member is thin is finally removed as shown inFIG.26Bwhen the wet etching further is continued. Thereby, the air bridge is formed, and the size of the RTD120becomes a distance d2(>d1) from the end surface of the mesa121. Since the size of the RTD120influences to the inner capacitance, the wet etching may be stopped when the air bridge is formed. On the contrary, by continuing the wet etching, the size of the RTD120may allow to be smaller. The width of the portion108in which the conductive material member is thin, is double of the distance d2or less. Next, the etch stopper of the etch stopper layer113is removed by the wet etching. Then, the photo resist130is layered on the whole surface as shown inFIG.25D, and the resist pattern to protect the portions becoming to the electrode portion and the stabilizing resistors104and105except for the air bridge is formed by the exposure with the photomask. Further, the n+-InGaAs layer112is removed by the wet etching as shown inFIG.25E. By removing the photo resist130, the terahertz oscillator to oscillate due to the resonance of the RTD120and the stabilizing resistors104and105is produced as shown inFIG.25F. INDUSTRIAL APPLICABILITY By using the minute devices according to the present invention, a compact chip, which measures absorption spectra of a material whose absorption spectra are existed in the terahertz frequency band and a light source chip for the terahertz imaging, can be easily produced. It is considered that the terahertz oscillator according to the present invention enables to facilitate a further development in the fields such as the chemistry, the medical regions and the security. Further, the terahertz oscillator according to the present invention has extensibility to large scale array as shown inFIGS.27A and27B, andFIG.27Ashows an example of a case having oscillating element spacing andFIG.27Bdoes an example of the high-density array. EXPLANATION OF REFERENCE NUMERALS 1resonant tunneling diode (RTD)2slot antenna3substrate4,41lower electrode40upper electrode7MIM10wafer20mesa30RTD100terahertz oscillator101resonant tunneling diode (RTD)102slot antenna103,104stabilizing resistor108conductive material member110substrate structure120RTD mesa121mesa130photoresist | 8,472 |
11863126 | DETAILED DESCRIPTION Throughout the specification, the same reference numerals refer to the same components. This specification does not describe all elements of the embodiments, and overlaps between general contents or embodiments in the technical field to which the present disclosure pertains are omitted. The term “unit, module, member, or block” used in the specification may be implemented by software or hardware, and according to embodiments, it is also possible that a plurality of “unit, module, member, or block” may be implemented as one component, or that one “part, module, member, or block” includes a plurality of components. Throughout the specification, when a part is “connected” to another part, this includes a case of being directly connected as well as being connected indirectly, and indirect connection includes connecting through a wireless communication network. Also, when a part is said to “comprise” a certain component, this means that other components may be further included instead of excluding other components unless specifically stated otherwise. Terms such as first and second are used to distinguish one component from other components, and the component is not limited by the above-described terms. In each of steps, an identification code is used for convenience of description, and the identification code does not describe the order of each of the steps, and each of the steps may be performed differently from the specified order, unless a specific order is explicitly stated in the context. Hereinafter, the principle and embodiments of the present disclosure will be described with reference to accompanying drawings. FIG.1is a block diagram of a phase shifter100according to an embodiment of the present disclosure.FIG.2illustrates a circuit diagram of the phase shifter100ofFIG.1. Referring toFIG.1, the phase shifter100may include a signal generator110, an operator120, and a signal converter130. The signal generator110may generate a first signal RF_I and a second signal RF_Q having a phase orthogonal to a phase of the first signal RF_I. InFIG.2, the first signal RF_I and the second signal RF_Q are denoted by RF_in as an example. For example, the signal generator110may include a resistor (not illustrated), an inductor (not illustrated), or a capacitor (not illustrated). The signal generator110may generate the first signal RF_I and the second signal RF_Q that have a phase orthogonal to each other by using the resistor (not illustrated), the inductor (not illustrated), or the capacitor (not illustrated). The operator120includes a cascode amplifier121and an output load122. The operator120receives the first signal RF_I and the second signal RF_Q from the signal generator110, and generates a first current corresponding to the first signal RF_I and a second current corresponding to the second signal RF_Q. The operator120may determine a path of the first current and a path of the second current, and may control phases of the first current and the second current. The operator120sums the first current and the second current and generates a summed first current and a summed second current. The operator120may generate a voltage signal based on the generated summed first and second currents. For example, the operator120may determine a phase of a sum of the first signal and the second signal by a vector sum method. The detailed configuration and operation of the operator120will be described in detail with reference toFIG.3. The signal converter130may convert a digital signal having information associated with the first current and the second current into an analog signal. The signal converter130mirrors the current to the operator120, based on the converted analog signal. The signal converter130receives the digital signal having a preset number of bits (N-bit), and mirrors the current to the operator120, based on the input digital signal. For example, the signal converter130may include two pairs of transistors connected in series, and may mirror the current to the operator120by using the two pairs of transistors. FIG.3illustrates a circuit diagram of the operator120illustrated inFIG.2. Referring toFIG.3, the operator120may include the cascode amplifier121and the output load122. The cascode amplifier121may include an input circuit121-1, a path selection circuit121-2, and a cascode amplification circuit121-3. The input circuit121-1generates the first current corresponding to the first signal RF_I and the second current corresponding to the second signal RF_Q, based on a current mirroring result of the signal converter130. For example, the input circuit121-1may include four transistors121-a,121-b,121-c, and121-d, and may include two resistors R1and R2. The transistors121-aand121-bmay generate the first current in response to the first signal RF_I input to their gate electrodes. In addition, the transistors121-cand121-dmay generate the second current in response to the second signal RF_Q input to their gate electrodes. The path selection circuit121-2may determine paths of the generated first and second currents. In this case, the determined paths of the first and second currents become a basis for phase control of the first and second currents. The path selection circuit121-2may include a transistor pair121-2aand a transistor pair121-2bthat are controlled by a first selection signal Quad_sel_I and an inverted first selection signal. In addition, the path selection circuit121-2may include a transistor pair121-2cand a transistor pair121-2dthat are controlled by a second selection signal Quad_sel_Q and an inverted second selection signal. For example, each of the four transistor pairs121-2a,121-2b,121-2c, and121-2dof the path selection circuit121-2may operate as a switch. As the transistors constituting the four transistor pairs121-2a,121-2b,121-2c, and121-2dare controlled by the first selection signal Quad_sel_I and the inverted first selection signal, and the second selection signal Quad_sel_Q and the inverted second selection signal, the operator120may control the paths of the first current and the second current. The cascode amplification circuit121-3buffers the first current and the second current of which the paths are determined, and transfers the buffered first current and the second current to the output load122. For example, the cascode amplification circuit121-3may include a pair of transistors121-3aand121-3bof which gate electrodes are connected to each other and which receive the first current of which the paths are determined through their source electrodes. In addition, the cascode amplification circuit121-3may include a pair of transistors121-3cand121-3dof which gate electrodes are connected to each other and which receive the second current of which the paths are determined through their source electrodes. As a result of the buffering, the vector sum of the first current and the second current may be facilitated. In other words, except for the transistors constituting the path selection circuit121-2, the transistors121-1a,121-1b,121-1c, and121-1dof the input circuit121-1are connected to the transistors121-3a,121-3b,121-3c, and121-3dof the cascode amplification circuit121-3in two stages. As a result, since only two stages of transistors are stacked, the operator120may operate even at a low voltage, and the isolation characteristics of the input terminal and the output terminal may be improved. The output load122may receive and add the first current and the second current, and may generate the summed first and second currents. As a result of the sum of the first current and the second current, the sum of a vector of the first current and a vector of the second current may be adjusted. In this case, the sum of the vectors outputs a corresponding intermediate phase vector value by summing the magnitudes of the first and second currents having an orthogonal relationship to each other, and the magnitude of the summed signal is uniform. The summed first current and second current flow through the output load to generate an output voltage signal. The output load122includes an inductor122a, a capacitor122b, and a signal controller123. For example, the output load122may match a resonant frequency of the inductor122aor the capacitor122bwith a frequency of the summed first and second currents. FIG.4illustrates a configuration of the signal controller123ofFIG.3. The signal controller123amay include the plurality of transistors123-a1to123-an, each of which is controlled by a control signal and connected in parallel to one another. Since each of the transistors123-a1to123-anhas an ON-resistance component when each of the transistors is turned on, the transistors123-a1to123-anare separately controlled, thereby adjusting the equivalent resistance of an entire RLC resonance circuit. As a result, the summed first current and second current flows to the load, such that a magnitude of the output voltage signal may be controlled. That is, a final input and output signal gain is controlled. FIG.5illustrates another configuration of the signal controller123ofFIG.3. The signal controller123bmay include a plurality of resistors R, and switches SW that are controlled by a control signal may be connected to both ends of each resistor. As the switches SW are turned on or off by the control signal, and then some or all of the resistors R are connected in parallel, a total resistance value may be adjusted. As a result, the magnitude of the summed first and second currents may be controlled. FIG.6illustrates a digital-to-analog converter131aand131baccording to an embodiment of the present disclosure. The digital-to-analog converter may include bias circuits131a-3and131a-4, and a plurality of transistors131a-1to131a-2that receive a plurality of digital input signals to generate corresponding currents. That is, like the transistors131a-1and131a-2, there are as many connected transistors as the number of digital input signals. In this case, as the plurality of transistors131a-2are turned on/off by the digital input signals, the bias current of the plurality of transistors131a-1corresponds to a current magnitude according to the digital input signals, which flows to a current mirror circuit below. The transistor131a-1and the transistor131a-2may be connected in series to each other. The source electrode of the transistor131a-1and the source electrode of the transistor131a-3may be connected to each other, and the gate electrode of the transistor131a-1, and the drain electrode and the gate electrode of the transistor131a-3may be connected to one another. For example, the digital-to-analog converter131amay receive the digital signal having the preset number of bits through the gate electrode of the transistor131a-2. In response to the digital signal having the preset number of bits, the digital-to-analog converter131aconverts a first digital signal into a first analog signal, and converts a second digital signal into a second analog signal. In addition, the signal converter130performs a current mirroring to the operator120, based on the first analog signal and the second analog signal. As a result of performing the current mirroring, the input circuit121-1may generate the first current and the second current. FIG.7is a graph illustrating a result of a phase control during a magnitude control for an antenna weight factor according to an embodiment of the present disclosure. In the graph, an x-axis represents a frequency band, and a y-axis represents the output magnitude. Referring toFIG.3together, the cascode amplifier121of the operator120buffers the first current and the second current. In addition, the signal controller123of the operator120controls the input and output gain by controlling the equivalent resistance value of the RLC load, and controls the antenna weight factor. The principle is that when the resonant frequency of the RLC load is the same as a center frequency of the transferred signal, the L and C disappear equivalently due to the resonance, and only the resistance value remains. Since only the resistance value changes equivalently, only the gain of the signal changes and the phase of the signal does not change. In this gain control method, the RLC load may be used for the output load of the phase shifter or may be used for the output load of an amplifier designed independently. This is necessary because when the output phase value of the phase shifter is set to a specific value and the gain of each cell in the phase array needs to be adjusted for the purpose of removing the side lobe of the beam, etc., a main beam may be continuously radiated in a desired direction only when the phase value of each cell that has already been set does not change. FIG.8illustrates how the phase shifter100according to an embodiment of the present disclosure controls a phase of an output signal. Referring toFIG.8, an x-axis of the graph represents a magnitude I of a final signal due to the first current, and a y-axis represents a magnitude Q of a final signal due to the second current. By summing the magnitude I and the magnitude Q of the two final signals, the final signal has an arbitrary phase value. In detail, as the operator120sums the magnitude I and the magnitude Q of the two final signals in the process of controlling the magnitudes of the first current and the second current, the final signal has an arbitrary phase value, and to further increase a phase resolution even with the input of the same N-bit digital-analog converter, the following method may be used. For example, in a signal in which the sum of the first current signal and the second current signal is 64, to increase the resolution, the operator120may allow the magnitude values32and32of the first current and the second current to be changed to the final magnitude values31and33through the intermediate values32and33. That is, the phase shifter100of the present disclosure may generate the intermediate values in the process of adjusting the vector sum. With a code sweep control described above, the operator120may generate an intermediate phase value, and may increase the resolution without increasing the number of bits of a phase control code value of the digital-analog converter. FIG.9illustrates how the phase shifter100according to an embodiment of the present disclosure synthesizes a phase of an output signal. Although a method of synthesizing the phase of the output signal by the phase shifter100inFIG.9is illustrated in steps, this is for ease of description. That is, it should be understood that the phase shifter100of the present disclosure operates organically to generate the output signal corresponding to a set value in response to the input signal under a predetermined setting. Hereinafter, reference will be made toFIG.3together to aid in understanding the description. The signal generator110generates the first signal and the second signal (S1001). In this case, the first signal and the second signal are signals having a phase orthogonal to each other. Also, the first signal and the second signal may be radio frequency (RF) voltage signals, but are not limited thereto. Further, the first signal includes first current information, and the second signal includes second current information. When the first signal and the second signal are generated, the signal converter130converts the input digital signal into the analog signal, based on a preset number of bits “N-bit” (S1002). For example, the signal converter130converts the first digital signal into the first analog signal and converts the second digital signal into the second analog signal. When the input digital signal is converted into the analog signal, the signal converter130generates the first current and the second current by performing the current mirroring to the operator120, based on the first and second current signals (S1003). The current mirroring may be performed by the plurality of transistors132a,132b,132c, and132dof the signal converter130. In detail, the plurality of transistors132a,132b,132c, and132dare connected in the cascode structure, and the signal converter130may perform the current mirroring depending on a symmetry of the circuit. As described above, the first current is generated based on the first signal RF_I, and the second current is generated based on the second signal RF_Q. Specifically, the first signal RF_I and the second signal RF_Q may be converted into the current signal by the input circuit121-1. When the first current and the second current are generated, the path selection circuit121-2selects a path of the first current and a path of the second current (S1004). For example, the path selection circuit121-2may select the paths of the first current and the second current, based on the switching operations of the transistors, and as a result of the path selections, the operator120may determine the phases of the first and second currents. Also, the transistors provided in the path selection circuit121-2are controlled by the digital signals, and are provided such that a phase range of the final output signal of the operator120is selected to be one of the quadrants of 360 degrees. In this case, the digital signal may be the first selection signal Quad_sel_I for determining the path of the first current and the second selection signal Quad_sel_Q for determining the path of the second current. When the paths of the first and second currents are determined, the cascode amplification circuit121-3buffers the first and second currents (S1005). When the first and second currents are buffered, the magnitudes of the currents of I and Q mirrored under the control of the signal converter130are changed, and the phase of the final summed signal changes based on the relative magnitudes of the currents of I and Q. The signal controller123includes a plurality of transistors or a plurality of resistors, and controls an equivalent resistance value of the RLC load. The principle is that when the equivalent resistance value of the RLC load is controlled and the resonant frequency of the RLC load is the same as the center frequency of the transferred signal, the L and C disappear equivalently due to the resonance, and only the resistance value remains. As only the resistance value changes equivalently, only the gain of the signal changes and the phase of the signal does not change. In this gain control method, the RLC load may be used for the output load of the phase shifter or may be used for the output load of an amplifier designed independently. The output load122sums the first current and the second current, and generates the final signal having the final phase value, by changing the signal to a voltage (S1006). In this case, the final signal refers to a signal having the intermediate phase vector depending on the relative magnitudes of the first current and the second current. FIG.10illustrates a phase array transceiver including the phase shifter100described inFIGS.1to9. A phase array transceiver200includes an antenna array210including two-dimensionally arranged antennas. The phase array transceiver200forms the beam based on energy radiated from each of the antennas. The phase array transceiver200may transmit or receive a wireless RF signal so as to overlap the formed beam. For example, the phase array transceiver200may be a radar apparatus. The beam formed by the phase array transceiver200may include one main lobe ML and side lobes SL1to SL4. The main lobe ML may be defined as a lobe in a direction in which the most energy is radiated in a beam formed by the phase array transceiver200. The side lobes SL1to SL4may be defined as lobes in a direction in which energy is radiated in the direction other than the main lobe ML. The side lobes SL1to SL4are formed based on less energy than the main lobe ML. The phase array transceiver200controls energy radiated from the antenna array210to form the main lobe ML in a reception direction of the RF signal. However, in the process of generating the main lobe ML by radiating energy through a plurality of antennas, the plurality of side lobes SL1to SL4may be generated in a direction other than the direction in which the main lobe ML is formed. The phase array transceiver200may control a size and a phase of signals for the plurality of antennas to form a beam for transmitting and receiving the RF signal. The phase array transceiver200may control each of the antennas to adjust the direction and the magnitude of the beam. The phase array transceiver200allows energy provided to the plurality of side lobes SL1to SL4to be minimize by using the phase shifter100described inFIGS.1to9. To this end, phase shifters corresponding to each of the antennas may be included in the phase array transceiver200. For example, when a specific antenna has the greatest influence on the formation of the first side lobe SL1, the phase shifter100corresponding to the specific antenna may control the magnitude of the beam to be low. As described above, in the phase shifter100, under the control of the signal converter130, the cascode amplifier121may buffer and transfer the current to the output load122, and the output load122may minimize a phase change according to the magnitude control. That is, since the phase array transceiver200according to an embodiment of the present disclosure controls a size of the phase shifter100corresponding to each of the antennas and suppresses the phase change according to the size control, the transmission and reception of RF signals by the side lobes SL1to SL4may be minimized. According to an embodiment of the present disclosure, a phase shifter may operate at a relatively low voltage, based on a vector sum circuit method using active elements. In addition, according to an embodiment of the present disclosure, the phase shifter may obtain a high phase resolution. In addition, according to an embodiment of the present disclosure, the phase shifter may minimize a phase change value according to gain control for array antenna weight control by controlling the equivalent resistance of the RLC output load, based on the output gain control by the output load control. While the present disclosure has been described with reference to embodiments thereof, it will be apparent to those of ordinary skill in the art that various changes and modifications may be made thereto without departing from the spirit and scope of the present disclosure as set forth in the following claims. | 22,589 |
11863127 | DETAILED DESCRIPTION The terms used in this specification and claims, unless otherwise stated, generally have their ordinary meanings in the art, within the context of the disclosure, and in the specific context where each term is used. Certain terms that are used to describe the disclosure are discussed below, or elsewhere in the specification, to provide additional guidance to the practitioner skilled in the art regarding the description of the disclosure. It is understood that, although the terms “first,” “second,” “third,” etc. may be used herein to describe various elements, signals and/or entitles, these elements, signals and/or entities should not be limited by these terms. These terms are only used to distinguish elements, signals and/or entities. Therefore, a first element, signal and/or entity in the description of the disclosure can be referred to a second element, signal and/or entity, and they are not intended to limit the scope of the present disclosure. The term “and/or” used in the description of the disclosure includes anyone and all combinations from one or more associated items. In the following description, the term “coupled” may be used to indicate that two or more elements are in direct physical or electrical contact with each other, or may also mean that two or more elements may not be in direct contact with each other. “Coupled” may still be used to indicate that two or more elements cooperate or interact with each other. Reference is made toFIG.1;FIG.1is a functional block diagram of an amplifier device100according to some embodiment of the present disclosure. As shown inFIG.1, the amplifier device100includes a regulator circuit110, a voltage converting circuit1201, a voltage converting circuit1202, a control circuit1301, a control circuit1302, an amplifier circuit140, a voltage dividing circuit150, an output unit160and an input unit170. In some embodiments, the regulator circuit110is configured to output a driving voltage VD1, the regulator circuit110may be any circuit capable of providing the driving voltage VD1, which is not limited in the present disclosure. In some embodiments, the voltage converting circuit1201is coupled between the regulator circuit110and the control circuit1301, and configured to receive and convert the driving voltage VD1 outputted by the regulator circuit110, to subsequently output an operating voltage VO1 to the control circuit1301. In some embodiments, the operating voltage VO1 outputted by the voltage converting circuit1201is substantially equal to the received the driving voltage VD1 (i.e., the voltage converting circuit1201may directly output the driving voltage VD1 without converting the driving voltage VD1). In some embodiments, the voltage converting circuit1201may utilize an included voltage divider unit (e.g., a voltage divider unit2211shown inFIG.2) and a switch set (e.g., a switch set2221shown inFIG.2), to convert the driving voltage VD1 into the operating voltage VO1, so as to output the operating voltage VO1 that is different from the driving voltage VD1. Detailed configurations and operations will be described in the following specification. In some embodiments, a node N1 is between the voltage converting circuit1201and the regulator circuit110, the voltage converting circuit1201is coupled to the regulator circuit110through the node N1, a capacitor C1 is disposed between the node N1 and a reference voltage terminal (i.e., the capacitor C1 is coupled between the node N1 and the reference voltage terminal). In some embodiments, the reference voltage terminal is a ground. In this specification, the reference voltage terminal being a ground is an example of the present disclosure, which is not limited. In some embodiments, a node N2 is between the voltage converting circuit1201and the control circuit1301, the voltage converting circuit1201is coupled to the control circuit1301through the node N2, a capacitor C2 is disposed between the node N2 and the reference voltage terminal (i.e., the capacitor C2 is coupled between the node N2 and the reference voltage terminal). In some embodiments, the control circuit1301is coupled between the voltage converting circuit1201and the amplifier circuit140. The control circuit1301is configured to receive an operating voltage VO1 transmitted from the voltage converting circuit1201, and generate an operating signal OS1 according to the operating voltage VO1 and a control signal Vcon1. Detailed operations regarding the control circuit1301will be described in the following specification. In some embodiments, the operating signal OS1 is substantially equal to the operating voltage VO1. In some embodiments, the amplifier circuit140is coupled to the control circuit1301and the regulator circuit110, and configured to receive the operating signal OS1 transmitted from the control circuit1301to generate an output voltage Vout. In some embodiments, the amplifier circuit140is coupled to the node N1 through an inductive element having a large impedance (e.g., inductor L1). In some embodiments, the amplifier circuit140is configured to receive an input voltage Vin, and amplify the input voltage Vin according to the operating signal OS1 to generate the output voltage Vout corresponding to the operating signal OS1. In other words, adjusting a voltage level of the operating signal OS1 by the voltage converting circuit1201and the control circuit1301, a linearity of the amplifier circuit140can be improved. In some embodiments, the voltage dividing circuit150is coupled between the regulator circuit110and the voltage converting circuit1202, and configured to output a driving voltage VD2 according to the driving voltage VD1. In some embodiments, a node N3 is between the voltage converting circuit1202and the control circuit1302, the voltage converting circuit1202is coupled to the control circuit1302through the node N3. The voltage converting circuit1202is configured to receive and convert the driving voltage VD2 to output an operating voltage VO2 to the control circuit1302. After the control circuit1302has received the operating voltage VO2, the control circuit1302is configured to generate an operating signal OS2 according to the operating voltage VO2 and a control signal Vcon2 to the amplifier circuit140. In this embodiment, the amplifier circuit140is configured to amplify the input voltage Vin according to the operating signal OS1 and the operating signal OS2 those are different from each other, to generate the output voltage Vout corresponding to the operating signal OS1 and the operating signal OS2. In some embodiments, a capacitor C3 is disposed between the node N3 and the reference voltage terminal, the capacitor C3 is coupled between the node N3 and the reference voltage terminal. In some embodiments, the control signal Vcon1 and the control signal Vcon2 may be generated by a digital control circuit (not shown inFIG.1). In some embodiments, operations and connections of the voltage converting circuit1202and the control circuit1302are similar to those of the voltage converting circuit1201and the control circuit1301, which will not be reiterated herein. In other embodiments, the voltage converting circuit1202may be connected to the regulator circuit110without the voltage dividing circuit150. That is, the voltage converting circuit1202may be directly connected to the regulator circuit110. In other embodiments, the amplifier device100may a combination including one or more voltage converting circuits and cooperative control circuits and voltage dividing circuits as above mentioned and the present disclosure can be provided without being limited to a number of circuits as above mentioned and the figures. For example, in the amplifier device100, at least one of the voltage dividing circuit150, the voltage converting circuit1202, the control circuit1302, the output unit160and the input unit170can be omitted. For another example, in addition to a first path including the voltage converting circuit1201and the control circuit1301and a second path including the voltage converting circuit1202and the control circuit1302, the amplifier device100may further include an additional path including a voltage converting circuit and a cooperative control circuit, e.g., a third path, a fourth path and the like. The additional path may be coupled between the regulator circuit110and the amplifier circuit140, operations and connection relationships of the additional path are similar to those of the voltage converting circuit1201and the control circuit1301, which will not be reiterated herein. In some embodiments, the amplifier device100further include an output unit160configured to receive and generate an output signal RFout according to the output voltage Vout. In some embodiments, the output signal RFout may be transmitted to a mixer (not shown inFIG.1) or any circuits and elements that can cooperate with the amplifier device100, which is not limited in the present disclosure. In some embodiments, the amplifier device100further include an input unit170configured to receive an input signal RFin to generate the input voltage Vin. In some embodiments, the input signal RFin may be transmitted from an antenna or a soldering pad, which is not limited in the present disclosure. Reference is made toFIG.2.FIG.2is a schematic circuit diagram of an amplifier device200inFIG.1according to some embodiment of the present disclosure. The circuit inFIG.2shows detailed circuit structures of the amplifier device100, which is not limited in the present disclosure. As shown inFIG.2, the voltage converting circuit2201is a feasible implementation of the voltage converting circuit1201or the voltage converting circuit1202inFIG.1. The voltage converting circuit2201may include a voltage divider unit2211and a switch set2221. As shown inFIG.2, a terminal of the switch set2221is coupled to the regulator circuit110through the node N1 and configured to receive a voltage V1, and another terminal of the switch set2221is coupled to the regulator circuit110through the voltage divider unit2211and configured to receive a voltage V2. Therefore, the voltage V1 received by the switch set2221is substantially equal to the driving voltage VD1, while the voltage V2 received by the switch set2221is different from the voltage V1. In some embodiments, the voltage divider unit2211includes a resistor R1 coupled between the regulator circuit110and the reference voltage terminal, and the resistor R1 is configured to receive the driving voltage VD1 and output the voltage V2 to the switch set2221according to the driving voltage VD1. In some embodiments, the switch set2221includes a switch SW1 and a switch SW2, and configured to respectively receive the voltage V1 and the voltage V2. When the switch SW1 is turned on, the switch set2221outputs the voltage V1 (which is the operating voltage VO1) to a control circuit2301. When the switch SW2 is turned on, the switch set2221outputs the voltage V2 (which is the operating voltage VO1) to the control circuit2301. In some embodiments, the switch SW1 is parallel to the switch SW2, a plurality of terminals of the switches SW1 and SW2 are coupled to the node N2, and a plurality of other terminals of the switches SW1 and SW2 are respectively coupled to different terminals of the resistor R1. In other words, the switch SW1 is coupled between the regulator circuit110(through the node N1) and the node N2, the switch SW2 is serially connected to the voltage divider unit2211(e.g., the resistor R1) and coupled between the regulator circuit110and the node N2, and the switch SW2 is also coupled between the reference voltage terminal and the node N2. In some embodiments, a voltage converting circuit2202is a feasible implementation of the voltage converting circuit1201or the voltage converting circuit1202inFIG.1, the voltage converting circuit2202may include a voltage divider unit2212and a switch set2222. In some embodiments, the voltage divider unit2212includes a resistor R2 and a resistor R3, a node N4 is between the resistor R2 and the resistor R3, the resistor R2 is coupled to the resistor R3 through the node N4, and the resistors R2 and R3 are serially connected between a voltage dividing circuit250and the reference voltage terminal. The voltage divider unit2212is configured to receive and output a voltage V4 and a voltage V5 to the switch set2222according to the driving voltage VD2. Likewise, the voltage V3 received by the switch set2222is substantially equal to the driving voltage VD2, while the voltage V4 and the voltage V5 received by the switch set2222are different from the voltage V3. In some embodiments, the switch set2222includes a switch SW3, a switch SW4 and a switch SW5. The switches SW3, SW4 and SW5 are configured to respectively receive the voltages V3, V4 and V5. As shown inFIG.2, a plurality of terminals of the switches SW3, SW4 and SW5 are coupled to the node N3, and a plurality of other terminals of the switches SW3, SW4 and SW5 are respectively coupled to a plurality of terminals of the resistors R2 or R3. For example, the switch SW3 is coupled between the voltage dividing circuit250and the node N3, the switch SW4 is coupled between the node N4 and the node N3, and the switch SW5 is coupled between the reference voltage terminal and the node N3. When the switch SW3 is turned on, the switch set2222outputs the voltage V3 (which is the operating voltage VO2) to a control circuit2302. When the switch SW4 is turned on, the switch set2222outputs the voltage V4 (which is the operating voltage VO2) to the control circuit2302. When the switch SW5 is turned on, the switch set2222outputs the voltage V5 (which is the operating voltage VO2) to the control circuit2302. In other embodiments, the voltage divider unit2211and the voltage divider unit2212may respectively include a plurality of serial-connected resistors and/or a plurality of parallel-connected resistors, the present disclosure can be provided without being limited to the embodiment as shown inFIG.2. In some embodiments, the control circuit2301is a feasible implementation of the control circuit1301or the control circuit1302inFIG.1. The control circuit2301may include an inverter INV1 and an inverter INV2. In some embodiments, the inverter INV2 is serially connected between the inverter INV1 and an amplifier circuit240, and the inverters INV1 and INV2 are coupled to the node N2 to receive the operating voltage VO1. In some embodiments, the inverter INV1 is configured to receive the control signal Vcon1, and cooperate with the inverter INV2 to generate the operating signal OS1 according to the control signal Vcon1 and the operating voltage VO1. In some embodiments, when the control signal Vcon1 is at a high level (e.g., logic 1), the operating signal OS1 received by the amplifier circuit240is also at the high level, then the amplifier circuit240in some embodiments is considered equivalent to be coupled to the node N2. In some embodiments, when the control signal Vcon1 is at a low level (e.g., logic 0), the operating signal OS1 received by the amplifier circuit240is also at the low level, then the amplifier circuit240in some embodiments is considered equivalent to be coupled to the reference voltage terminal, such that the amplifier circuit240is turned off. When the amplifier circuit240is turned off, the output voltage Vout received by an output unit260is associated with a driving voltage outputted by the regulator circuit110, or a voltage signal transmitted by an inductive element (e.g., inductor L1). In some embodiments, the control circuit2302is coupled between the node N3 and the amplifier circuit240, and includes an inverter INV3 and an inverter INV4, wherein the inverter INV3 is configured to receive the control signal Vcon2. Operations and connections of the inverters INV3 and INV4 are similar to those of the inverters INV1 and INV2 in the control circuit2301, which will not be reiterated herein for simplicity. In some embodiments, the amplifier circuit240is a feasible implementation of the amplifier circuit140inFIG.1. The amplifier circuit240includes transistors T1-T3 serially connected to each other. In some embodiments, a control terminal of the transistor T1 is coupled to the control circuit2301, and configured to receive the operating signal OS1, a first terminal of the transistor T1 is directly coupled or through an element having a large impedance (e.g., inductor L1) to the regulator circuit110; and a second terminal of the transistor T1 is coupled to a first terminal of the transistor T2. A control terminal of the transistor T2 is coupled to the control circuit2302, and configured to receive the operating signal OS2; a second terminal of the transistor T2 is coupled to a first terminal of the transistor T3. A control terminal of the transistor T3 is configured to receive the input voltage Vin, and a second terminal of the transistor T3 is coupled to the reference voltage terminal. In other embodiments, when the amplifier device100shown inFIG.1does not include the voltage converting circuit1202and the control circuit1302(i.e., the second path as mentioned inFIG.1), at least one transistor (e.g., the transistor T2 of the amplifier circuit240inFIG.2) can be omitted in the amplifier circuit140accordingly. In other words, a number of transistors configured and included in the amplifier circuit140may be increased or decreased depending on a number of voltage converting circuits, the present disclosure can be provided without being limited to the embodiment as shown inFIG.2. In some embodiments, the voltage dividing circuit250is a feasible implementation of the voltage dividing circuit150inFIG.1, the voltage dividing circuit250may include a resistor R4 and a resistor R5. As the embodiment shown inFIG.2, the resistors R4 and R5 are configured to perform voltage division to the driving voltage VD1 to output the driving voltage VD2. In some embodiments, a first terminal of the resistor R4 is coupled to the node N1 and the voltage converting circuit2201, a second terminal of the resistor R4 is coupled to a first terminal of the resistor R5 and the voltage converting circuit2202. The first terminal of the resistor R5 is coupled to the second terminal of the resistor R4 and the voltage converting circuit2202, a second terminal of the resistor R5 is coupled to the reference voltage terminal. In other embodiments, the voltage dividing circuit250may include a plurality of serial-connected resistors and/or a plurality of parallel-connected resistors, which is not limited in the present disclosure. In some embodiments, the output unit260is a feasible implementation of the output unit160inFIG.1. The output unit260includes an output matching circuit261and a capacitor C4, the capacitor C4 is coupled between the output matching circuit261and an output terminal of the amplifier circuit240(e.g., the first terminal of the transistor T1). In detail, the capacitor C4 is configured to block a direct signal of an output load from the amplifier device200. The output matching circuit261is configured to perform impedance matching between the amplifier device200and the output load, the output matching circuit261may include one or more serial-connected and/or parallel-connected elements such as inductors, resistors, and capacitors, which is not limited in the present disclosure. In some embodiments, an input unit270is a feasible implementation of the input unit170inFIG.1. The input unit270includes an input matching circuit271and a bias circuit272. In some embodiments, the input matching circuit271is configured to perform impedance matching between the amplifier device200and an input load, the input matching circuit271may include one or more serial-connected and/or parallel-connected elements such as inductors, resistors, and capacitors, which is not limited in the present disclosure. In some embodiments, the input matching circuit271is coupled to the amplifier circuit240(e.g., the control terminal of the transistor T3) through a capacitor C5, wherein the capacitor C5 is configured to block a direct signal of the input load from the amplifier device200. Moreover, as shown inFIG.2, the bias circuit272and the capacitor C5 are coupled to an input terminal of the amplifier circuit240, and the bias circuit272is configured to generate a fixed reference voltage, such that the fixed reference voltage and a signal transmitted through the capacitor C5 are superposed to be the input voltage Vin to be transmitted to the amplifier circuit240. In some embodiments, the bias circuit272may include a transistor T4 and a current source IS. The current source IS is coupled to a first terminal and a control terminal of the transistor T4, and a second terminal of the transistor T4 is coupled to the reference voltage terminal. In some embodiments, the bias circuit272is coupled to the amplifier circuit240(e.g., the control terminal of the transistor T3) through a resistor R6, wherein the resistor R6 has a large resistance (e.g., 10 kΩ). In some embodiments, the transistors T1-T4 are N-type metal oxide semiconductor (NMOS) transistors. In other embodiments, the transistors T1-T4 may be the same or different types of transistors (e.g., a bipolar transistor, a PMOS transistor, etc.), which is not limited in the present disclosure. In view of the foregoing, the amplifier device100and amplifier device200are provided in present disclosure, and the voltage divider unit and the switch set included in the voltage converting circuit output different operating voltages to the amplifier circuit according to the driving voltage, so as to adjust the linearity of the amplifier circuit. Although the present disclosure has been described in considerable detail with reference to certain embodiments thereof, other embodiments are possible. Therefore, the spirit and scope of the appended claims should not be limited to the description of the embodiments contained herein. It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present disclosure without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention cover modifications and variations of this invention provided they fall within the scope of the following claims. | 22,450 |
11863128 | DETAILED DESCRIPTION Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. The same elements will be denoted by the same reference numerals, and a duplicate description will be omitted as much as possible. A power amplifier circuit10according to a first embodiment will be described.FIG.1illustrates a circuit diagram of the power amplifier circuit10. The power amplifier circuit10includes transistors101to10n,111,112, and113, capacitors114and131, resistance elements115and121, an inductor141, and a matching circuit151. The transistors101to10n,111,112, and113are transistors, such as heterojunction bipolar transistors (HBTs), for example. The transistors101to10nare disposed in a disposition area A1on a semiconductor substrate (not illustrated). The transistors101to10n, each has a base connected to the capacitor131and the resistance element121, a collector connected to the matching circuit151, and an emitter connected to ground. The collectors of the transistors101to10nare supplied with a power supply voltage Vcc through the inductor141. Each of the transistors101to10n(first transistor) amplifies a signal RFin input from an input terminal161. Each of the transistors101to10namplifies the signal RFin input to the base thereof on the basis of a bias current Ib output from the transistor111(second transistor), which will be described below. The signal RFin amplified by each transistor is output as a signal RFout through the matching circuit151. The transistors111,112, and113, the capacitor114, and the resistance element115are disposed in a disposition area A3on the semiconductor substrate. The transistor111(second transistor) has a base connected to the resistance element115, a collector connected to a power supply, and an emitter connected to the resistance element121. The transistor111is switched between an ON-state and an OFF-state by a current I1supplied to the base thereof on the basis of a control current Ic supplied from a control input terminal181and flowing through the resistance element115. In an ON-state, the transistor111outputs the bias current Ib. The transistor112(current output element) is a diode-connected transistor and has a collector connected to an emitter of the transistor113and an emitter connected to ground. The collector of the transistor112is supplied with a current I2. The transistor112outputs a current I3from the emitter on the basis of the current I2. The transistor112is disposed in a disposition area A2on the semiconductor substrate. The transistor113(third transistor) is a diode-connected transistor and has a collector connected to the resistance element115and the base of the transistor111and an emitter connected to the collector of the transistor112. The transistor113outputs the current I2to the collector of the transistor112on the basis of a current I4flowing through the collector thereof on the basis of the control current Ic. The capacitor114has one end connected to the base of the transistor111and the other end connected to ground. The capacitor114functions to cause an alternating-current component of the control current Ic to flow to ground. The resistance element115is disposed to cause a predetermined voltage drop based on the control current Ic input from the control input terminal181. The transistors111,112, and113, the capacitor114, and the resistance element115constitute a bias circuit. The matching circuit151has one end connected to the collectors of the transistors101to10nand the other end connected to an output terminal171. The matching circuit151achieves impedance matching between the collectors of the transistors101to10nand the output terminal171. The layout of the power amplifier circuit10according to the first embodiment on the semiconductor substrate will be described with reference toFIG.2.FIG.2schematically illustrates the transistors101,102,111,112, and113. The transistors101and102are supplied with a bias current from the transistor111through a power supply wiring line201. The transistors101and102are also supplied with the signal RFin input from the input terminal161, which is not illustrated here, through the power supply wiring line201. The transistors101and102are arranged in a direction along an x-axis in the coordinate system inFIG.2. In the power amplifier circuit10, the plurality of transistors103to10n, which are not illustrated here, are also arranged in the x-axis direction. The transistor112is disposed along a direction in which the transistors101to10nare arranged. The transistor113and the transistor111are disposed at positions away from the transistor112in a y-axis direction. The disposition area A1is an internal area surrounded by the boundary of an isolation area that is disposed to electrically insulate, on the semiconductor substrate, the transistors101to10nfrom the other active devices disposed on the semiconductor substrate. In a case where isolation areas are disposed for the individual transistors101to10n, the disposition area A1is defined by the envelopes of the individual isolation areas so as to include the isolation areas of the individual transistors101to10n. The disposition area A3is based on an isolation area that is disposed to electrically insulate, on the semiconductor substrate, a plurality of active elements for supplying the bias current Ib to the transistors101to10nfrom the other active elements. The disposition area A3is an internal area enveloping, with straight lines parallel to the x-axis and the y-axis, the outermost sides in the x-axis direction and the y-axis direction of the boundaries between the individual active elements and the isolation area. The disposition area A2is, similarly to the disposition areas A1and A3, an internal area surrounded by the boundary of an isolation area that is disposed to insulate, on the semiconductor substrate, the transistor112from the other active elements. A second emitter wiring line202is disposed above the transistors101to10nand the transistor112so as to be connected to the transistors101to10nand the transistor112. The second emitter wiring line202extends so as to overlap an area extending from the disposition area A1to the disposition area A2. That is, the second emitter wiring line202extends in the x-axis direction. A bump203is disposed along the second emitter wiring line202so as to cover the transistors101to10nand the transistor112. The bump203extends so as to overlap an area extending from the disposition area A1to the disposition area A2. That is, the bump203extends in the x-axis direction. The bump203is a copper pillar bump, for example. In the layout illustrated inFIG.2, it is sufficient that the bump203be disposed so as to cover the transistors101to10nand the transistor112. The arrangement direction of the transistors101to10nand the positions of the transistors111,112, and113are not limited to those in the layout illustrated inFIG.2. Other layout examples are illustrated inFIG.4toFIG.7. The layout diagrams inFIG.4toFIG.7illustrate variations of the disposition areas based on the relationship between the individual transistors and the isolation areas of the individual transistors. InFIG.4toFIG.7, a semiconductor substrate301has an isolation area Bl, and the relationships between the isolation area Bl and the disposition areas A1to A3are illustrated. The cross-sectional structure of the power amplifier circuit10will be described with reference toFIG.3. The transistors101and102will be described. The transistors101and102are disposed on the semiconductor substrate301. A detailed structure will be described about the transistor102. A sub-collector layer302is disposed on the semiconductor substrate301. The semiconductor substrate301is made of, for example, semi-insulating GaAs. The sub-collector layer302is made of, for example, high-concentration n-type GaAs. The sub-collector layer302has a thickness of about 0.5 μm, for example. A collector layer303is disposed on the sub-collector layer302. The collector layer303is made of, for example, n-type GaAs. The collector layer303has a thickness of about 1 μm, for example. A base layer304is disposed on the collector layer303. The base layer304is made of, for example, p-type GaAs. The base layer304has a thickness of about 100 nm, for example. An emitter layer305is disposed on the base layer304. The emitter layer305includes an intrinsic emitter layer305A and an emitter mesa layer305B. The intrinsic emitter layer305A is disposed on the base layer304. The emitter mesa layer305B is disposed on the intrinsic emitter layer305A. The intrinsic emitter layer305A is made of, for example, n-type InGaP. The intrinsic emitter layer305A has a thickness of about 30 nm or more and about 40 nm or less, for example. The emitter mesa layer305B is made of, for example, high-concentration n-type GaAs and high-concentration n-type InGaAs. In the emitter mesa layer305B, for example, a high-concentration n-type InGaAs layer having a thickness of about 100 nm is disposed on a high-concentration n-type GaAs layer having a thickness of about 100 nm. A ledge layer306is disposed on an upper surface of the base layer304, in an area in which the emitter layer305is not disposed. The ledge layer306is deposited together with the intrinsic emitter layer305A and has the same composition as that of the intrinsic emitter layer305A. Because the emitter mesa layer305B is not disposed on the ledge layer306, the ledge layer306is depleted and thus does not function as an emitter of the transistor. Thus, the intrinsic emitter layer305A and the emitter mesa layer305B are referred to as the emitter layer305, and the ledge layer306is distinguished from the emitter layer305. The collector layer303, the base layer304, and the emitter layer305constitute the transistor102. The same applies to the transistor101. A base electrode331is disposed on the base layer304. The base electrode331is in ohmic contact with the base layer304through a cavity disposed in the ledge layer306. The base electrode331is disposed between portions of the emitter layer305in each of the transistors101and102. A collector electrode332is disposed on the sub-collector layer302. The collector electrode332is in ohmic contact with the sub-collector layer302. The collector electrode332is disposed between the transistor101and the transistor102in the x-axis direction. The collector electrode332is connected to the collector layer303via the sub-collector layer302. The collector electrode332is shared between the transistor101and the transistor102. An emitter electrode333is disposed on the emitter layer305. The emitter electrode333is in ohmic contact with the emitter layer305. The base electrode331is formed by stacking, for example, a Ti film, a Pt film, and a Au film in order on the base layer304. The collector electrode332is formed by stacking, for example, a AuGe film, a Ni film, and a Au film in order on the sub-collector layer302. The emitter electrode333is formed of, for example, a Ti film having a thickness of about 50 nm. In the sub-collector layer302, an isolation area3021for isolating elements from each other is disposed. The isolation area3021is formed by insulating a part of the sub-collector layer302by using, for example, an ion implantation technique. A part of the sub-collector layer302is a part in an x-y plane. The isolation area3021is disposed over the entire part in a z-axis direction of the sub-collector layer302. The isolation area3021may be disposed so as to include a part of the semiconductor substrate301in addition to the sub-collector layer302in the z-axis direction. The transistor112will be described. A sub-collector layer3022is disposed on the semiconductor substrate301. A collector layer3031is disposed on the sub-collector layer3022. A base layer3041is disposed on the collector layer3031. An emitter layer3051and a ledge layer3061are disposed on the base layer3041. The emitter layer3051includes an intrinsic emitter layer3051A and an emitter mesa layer3051B. The relationship among the intrinsic emitter layer3051A, the emitter mesa layer3051B, and the ledge layer3061is similar to the relationship among the intrinsic emitter layer305A, the emitter mesa layer305B, and the ledge layer306. The collector layer3031, the base layer3041, and the emitter layer3051constitute the transistor112. The transistor112is formed by the same process as the process of forming the transistors101and102. Thus, the transistor112has temperature characteristics similar to those of the transistors101and102. A base electrode3311is disposed on the base layer3041. The base electrode3311is in ohmic contact with the base layer3041through a cavity disposed in the ledge layer3061. A collector electrode3321is disposed on the sub-collector layer3022. The collector electrode3321is in ohmic contact with the sub-collector layer3022. The collector electrode3321is connected to the collector layer3031via the sub-collector layer3022. An emitter electrode3331is disposed on the emitter layer3051. The emitter electrode3331is in ohmic contact with the emitter layer3051. The base electrode3311, the collector electrode3321, and the emitter electrode3331are formed by the same processes as the processes of forming the base electrode331, the collector electrode332, and the emitter electrode333, respectively. A first insulating layer321is disposed so as to cover the transistors101,102, and112. The first insulating layer321has a multilayer structure formed of, for example, a SiN layer and a resin layer. Alternatively, the first insulating layer321may be formed of only a SiN layer. First-layer collector wiring lines341and3411are disposed on the first insulating layer321. The first-layer collector wiring line341is disposed through the first insulating layer321and connected to the collector electrode332. The first-layer collector wiring line3411is disposed through the first insulating layer321and connected to the collector electrode3321. First emitter wiring lines351and3511are disposed on the first insulating layer321. The first emitter wiring line351is disposed for each of the transistors101and102. The first emitter wiring line351connects the emitter electrodes333of each transistor. The first-layer collector wiring line341and the first emitter wiring line351, each has a multilayer structure formed of, for example, a Ti film having a thickness of about 10 nm or more and about 50 nm or less and a Au film having a thickness of about 1 μm or more and about 2 μm or less. The first emitter wiring line3511is disposed for the transistor112. The material and structure of the first emitter wiring line3511are similar to those of the first emitter wiring line351. A second insulating layer322is disposed on the first insulating layer321so as to cover the first-layer collector wiring lines341and3411and the first emitter wiring lines351and3511. The second insulating layer322has a multilayer structure formed of, for example, a SiN layer and a resin layer. Alternatively, the second insulating layer322may be formed of only a SiN layer. A second emitter wiring line202is disposed on the second insulating layer322. The second emitter wiring line202is connected to the first emitter wiring lines351through cavities disposed in the second insulating layer322. The first emitter wiring lines351of the transistors101and102arranged in the x-axis direction are connected to each other via the second emitter wiring line202. The second emitter wiring line202has a multilayer structure formed of, for example, a Ti film having a thickness of about 10 nm or more and about 50 nm or less and a Au film having a thickness of about 2 μm or more and about 4 μm or less. The second emitter wiring line202extends to a position above the transistor112. A third insulating layer323is disposed on the second emitter wiring line202so as to cover the second emitter wiring line202. The third insulating layer323has a multilayer structure formed of, for example, a SiN film and a resin film. Alternatively, the third insulating layer323may be formed of only a SiN film. The third insulating layer323functions as a protective film that protects the transistors101,102, and112. The bump203is disposed on the third insulating layer323. The bump203is connected to the second emitter wiring line202through a cavity of the third insulating layer323. The bump203has a multilayer structure in which an under bump metal layer3111, a metal post3112, and a solder layer3113are stacked in this order. The under bump metal layer3111is formed of, for example, a Ti film having a thickness of about 50 nm or more and about 100 nm or less. The under bump metal layer3111has a function of increasing adhesion of the bump203to the third insulating layer323. The metal post3112is formed of, for example, a Cu film having a thickness of about 30 μm or more and about 50 μm or less. The solder layer3113is formed of, for example, a film made of Sn or a SnAg alloy and having a thickness of about 10 μm or more and about 30 μm or less. Alternatively, a barrier metal layer made of Ni or the like and preventing mutual diffusion may be disposed between the metal post3112and the solder layer3113. The bump203functions as a metal member for dissipating heat generated by the transistors101and102to the outside. The second emitter wiring line202and the bump203, which are two metal layers, constitute a wiring portion W1. The second emitter wiring line202and the bump203are members having different shapes. The movement of heat generated by the transistors101and102and thermal coupling will be described. The transistors101and102generate heat when operating. The transistors101and102operate as a result of being supplied with the bias current Ib from the transistor111. In the transistors101and102, an operating current flows from the collector layer303through the base layer304to the emitter layer305. In the collector layer303, the base layer304, and the emitter layer305, the area through which the operating current substantially flows substantially matches the emitter layer305in plan view. The flow of the operating current in the collector layer303, the base layer304, and the emitter layer305generates Joule heat, which increases the temperatures of the transistors101and102. The heat generated in the transistors101and102is transmitted to the second emitter wiring line202through the emitter electrodes333. The heat transmitted to the second emitter wiring line202is transmitted in the second emitter wiring line202in the x-axis direction and the z-axis direction. The heat transmitted from the second emitter wiring line202in the z-axis direction reaches the bump203. The heat reached the bump203is transmitted in the x-axis direction and the z-axis direction. In a case where a semiconductor chip including the power amplifier circuit10is flip-chip connected, the heat transmitted in the z-axis direction in the bump203is transmitted to the substrate on which the semiconductor chip is installed. The heat transmitted in the x-axis direction in the second emitter wiring line202increases the temperature of the second emitter wiring line202above the transistor112. The heat transmitted in the x-axis direction in the bump203increases the temperature of the bump203above the transistor112. The increase in the temperatures of the second emitter wiring line202and the bump203near a position above the transistor112causes an increase in the temperature of the transistor112as a result of heat conduction in the z-axis direction. If the second emitter wiring line202and the bump203do not extend to the vicinity of a position above the transistor112, heat conduction from the second emitter wiring line202and the bump203to the transistor112does not sufficiently occur. In this case, the amount of increase in the temperature of a transistor of the bias circuit is smaller than the amount of increase in the temperature of the transistor112. Such a relationship between the transistor112and the transistors101and102, that is, a phenomenon in which a change in the temperature of a first element causes the temperature of a second element to change to approach the temperature of the first element, is referred to as thermal coupling. In the power amplifier circuit10, the temperature of the transistor112changes to become closer to the temperature of the transistors101and102compared to the case where the second emitter wiring line202and the bump203do not extend to the vicinity of a position above the transistor112. In other words, the strength of the thermal coupling between the transistor112and the transistors101and102increases. The operations of the circuit will be described also with reference toFIG.1. The operations of the transistors101to10ncause an increase in the temperatures of the transistors101to10n. The increase in the temperatures of the transistors101to10ncauses an increase in the collector currents of the transistors101to10n. In accordance with the increase in the collector currents, the gains of the transistors101to10nincrease. The heat generated by the transistors101to10ncauses an increase in the temperature of the transistor112via the second emitter wiring line202and the bump203. The increase in the temperature of the transistor112causes a decrease in the forward voltage of the transistor112serving as a diode. The decrease in the forward voltage causes in an increase in the current I2and the current I3. The increase in the current I2and the current I3causes a decrease in the current I1, which is the base current of the transistor111. The decrease in the current I1causes a decrease in the bias current Ib output from the transistor111. The decrease in the bias current Ib causes a decrease in the current supplied to the bases of the transistors101to10nand thus causes a decrease in the collector currents of the transistors101to10n. The power amplifier circuit10is capable of decreasing the collector currents more than in the case where thermal coupling is insufficient. As a result of suppressing more an increase in the collector currents, variations in gain, including an increase in gain, of the transistors101to10ncan be suppressed more. The transistor112serving as a current output element may be replaced with a diode in which the forward voltage decreases and the current I2increases in accordance with a rise in temperature. As the current output element, any element may be used as long as the current I2increases in accordance with a rise in temperature. An example will be given regarding suppression of variations in gain by the power amplifier circuit10.FIG.8illustrates chronological changes in the collector current of a transistor that performs an intermittent operation. The intermittent operation means an operation of causing, in addition to a signal amplified by the transistor, such as the one illustrated between time T2and time T3, an idle current flowing when the transistor is in an OFF-state, to rise and fall immediately before input of the signal. During the intermittent operation, the idle current changes in the following manner. At time T1, the idle current enters an ON-state. At time T2, the amplified signal enters an ON-state in addition to the idle current. At time T3, the amplified signal enters an OFF-state. At time T4, the idle current enters an OFF-state. In the transistor that performs such an intermittent operation, heat generation after time T2may cause variations in gain. In the power amplifier circuit10, the balance in temperature between the transistor112and the transistors101to10nis achieved by the second emitter wiring line202and the bump203, and thus an increase in gain can be suppressed. Additionally, in the transistor that performs the intermittent operation, the amount of heat generation in the emitter significantly changes also before time T1at which the idle current enters an ON-state and between time T1and time T2, and thus the temperature changes. The change in temperature may cause variations in gain. Also, in this case, in the power amplifier circuit10, the balance in temperature between the transistor112and the transistors101to10ncan be achieved and an increase in gain can be suppressed. Also, in a case where a current changes to cause an increase in temperature as a result of an operation other than the intermittent operation, an increase in gain can be suppressed by achieving the balance in temperature between the transistor112and the transistors101to10n. FIG.9illustrates a comparison in performance between a power amplifier circuit having the configuration of the power amplifier circuit10and a power amplifier circuit (comparative example) in which the wiring portion W1does not extend to the vicinity of a position above the transistor112. InFIG.9, the horizontal axis represents time t (ms) and the vertical axis represents output power Pout (dBm). An input signal has a frequency of about 3.75 GHz.FIG.9illustrates variations in the output power Pout until the output power Pout reaches 20 dBm, which is a target value. In the comparative example represented by a broken line inFIG.9, the amount of variation in the output power Pout until 20 dBm is reached is about 0.4 dBm. In the power amplifier circuit10represented by a solid line inFIG.9, the amount of variation in the output power Pout is about 0.2 dBm. In this way, in the power amplifier circuit10, variations in gain resulting from variations in temperature is suppressed, and thus the amount of variation in the output power Pout is smaller than in the comparative example. As illustrated in the layout diagram inFIG.10and the cross-sectional view inFIG.11, the second emitter wiring line202and the bump203do not necessarily need to completely cover the disposition area A2in which the transistor112is disposed. It is sufficient that the temperature state of the transistors disposed in the disposition area A1be reflected in the transistor disposed in the disposition area A2by the second emitter wiring line202and the bump203. In the layout diagram inFIG.10and the cross-sectional view inFIG.11, the second emitter wiring line202reaches a position above the transistor112that is disposed in the disposition area A2, whereas a bump203A, which includes an under bump metal layer3111A, a metal post3112A, and a solder layer3113A, does not reach a position above the transistor112. In this case, the second emitter wiring line202and the bump203A, which are two metal layers, constitute a wiring portion W2. In the layout diagram inFIG.12and the cross-sectional view inFIG.13, the bump203reaches a position above the transistor112that is disposed in the disposition area A2, whereas a second emitter wiring line202A does not reach a position above the transistor112. In this case, the second emitter wiring line202A and the bump203, which are two metal layers, constitute a wiring portion W3. The bump203may be disposed so as to overlap an area extending from the disposition area A1to an area between the disposition area A1and the disposition area A3including the disposition area A2. More specifically, the second emitter wiring line202and the bump203may cover the transistors111,112, and113disposed in the disposition area A3.FIG.14illustrates a layout diagram in a case where a second emitter wiring line202B and a bump203B extend to a position above the transistor113. FIG.15illustrates a cross-sectional view taken along line XV-XV inFIG.14. In the transistor112, the positions of the electrodes are inverted with respect toFIG.3. The transistor113will be described. The transistor113includes, like the transistor112, a collector layer15031, a base layer15041, an emitter layer15051, and a ledge layer15061that are disposed on a sub-collector layer15022. The emitter layer15051includes an intrinsic emitter layer15051A and an emitter mesa layer15051B. The collector layer15031, the base layer15041, and the emitter layer15051constitute the transistor113. In the transistor113, as in the transistor112, a base electrode15311, a collector electrode15321, and an emitter electrode15331are disposed. The transistor113is formed by the same process as the process of forming the transistors101and102. Thus, the transistor113has temperature characteristics similar to those of the transistors101and102. A first-layer collector wiring line15411is disposed on the first insulating layer321, similarly to the first-layer collector wiring line3411. A first emitter wiring line15511is disposed on the first insulating layer321, similarly to the first emitter wiring line3511. The first-layer collector wiring line3411and the first emitter wiring line15511are connected to each other by a wiring line1501disposed on the first insulating layer321. Above the transistor113, the second emitter wiring line202B and the bump203B extend from the transistor112side. In this case, the second emitter wiring line202B and the bump203B, which are two metal layers, constitute a wiring portion W4. With such extension of the bump203, variations in current based on changes in the temperature characteristics of the transistor112and the transistor113cancel each other out so as not to affect the bias current Ib. In other words, cancelling out occurs between a tendency in which the current I2of the transistor112increases and a tendency in which the collector current of the transistor113decreases as a result of an increase in the base current I1of the transistor111. Thus, the strength the thermal coupling between the transistor112and the transistors101to10ncan be increased as in the power amplifier circuit10. The second emitter wiring line202and the bump203may extend from a position above the transistors101and102to the vicinity of a position above only the transistor113. In this case, the transistor113functions as a current output element. Heat conduction occurs from the transistors101and102to the transistor113through the second emitter wiring line202or the bump203. Thus, the strength the thermal coupling between the transistor113and the transistors101to10ncan be increased as in the power amplifier circuit10. As a result of increasing the temperature of the transistor113by thermal coupling, the collector currents of the transistors101to10ncan be decreased and variations in gain during operation can be suppressed as in the case of thermal coupling with the transistor112. A second embodiment will be described. In the second embodiment and the following embodiments, a description of the same points as those of the first embodiment will be omitted, and only different points will be described. In particular, similar functions and effects obtained by similar configurations will not be described in each embodiment. In a power amplifier circuit according to the second embodiment, the individual transistors are disposed similarly to the first embodiment as illustrated inFIG.16.FIG.17illustrates a cross-sectional view taken along line XVII-XVII inFIG.16. The power amplifier circuit according to the second embodiment illustrated inFIG.17is different from the power amplifier circuit according to the first embodiment in that a second emitter wiring line202C has a protruding portion2021. The protruding portion2021extends toward the first emitter wiring line3511through a cavity of the second insulating layer322. The protruding portion2021is in ohmic contact with the first emitter wiring line3511. In the power amplifier circuit according to the second embodiment, the second emitter wiring line202C above the transistor112is at a position closer to the transistor112as illustrated inFIG.17. Accordingly, more heat is transmitted from the second emitter wiring line202C to the transistor112. Thus, the temperature of the transistor112increases more and becomes closer to the temperatures of the transistors101and102. In other words, the strength of thermal coupling is higher than in the power amplifier circuit according to the first embodiment. The higher strength of thermal coupling makes it possible to more appropriately achieve the balance in characteristic change between the transistors caused by variations in temperature, and an increase in gain can be further suppressed. In this case, the second emitter wiring line202C and the bump203, which are two metal layers, constitute a wiring portion W5. Alternatively, the bump203may have a protruding portion. In the cross-sectional view of another power amplifier circuit according to the second embodiment illustrated inFIG.18, a bump203C has a protruding portion2031which is a part of an under bump metal layer3111C and a metal post3112C. The protruding portion2031extends toward the second emitter wiring line202through a cavity of the third insulating layer323. The protruding portion2031is in ohmic contact with the second emitter wiring line202. Also, with this configuration, the strength of thermal coupling is higher than in the power amplifier circuit according to the first embodiment. Thus, an increase in gain can be further suppressed. In this case, the bump203C further includes a solder layer3113C, and the second emitter wiring line202and the bump203C, which are two metal layers, constitute a wiring portion W6. A third embodiment will be described.FIG.19is a layout diagram of a power amplifier circuit according to the third embodiment. In the power amplifier circuit according to the third embodiment, the second emitter wiring line202described in the foregoing embodiments includes second emitter wiring lines202aand202bisolated from each other. The second emitter wiring line202ais disposed above the transistors101to10nand the transistor112. The second emitter wiring line202bis disposed above the transistor113. In the power amplifier circuit according to the third embodiment, a wiring line1901is disposed between a bump203D and the second emitter wiring lines202aand202bin the z-axis direction. The wiring line1901is made of a metallic material. The wiring line1901is disposed along the second emitter wiring line202for the transistors101to10nin the disposition area A1so as to overlap the transistor113in the disposition area A3. In the layer in which the wiring line1901is disposed, the wiring line1901is isolated from the remaining portion for the purpose of insulation. The wiring line1901may be regarded as a rewiring line. FIG.20illustrates a cross-sectional view taken along line XX-XX inFIG.19. The wiring line1901includes a first metal portion2001and a second metal portion2002. The wiring line1901is isolated by an insulating layer2003as necessary. The second emitter wiring line202b, the wiring line1901, and the bump203D are located above the transistor113. The wiring line1901and the bump203D above the transistor113have a protruding portion20011aand the protruding portion2031, respectively. The second emitter wiring line202, the wiring line1901, and the bump203D, which are three metal layers, constitute a wiring portion W7. The bump203D includes an under bump metal layer3111D, a metal post3112D, and a solder layer3113D. FIG.21illustrates a cross-sectional view taken along line XXI-XXI inFIG.19. InFIG.21, the second emitter wiring line202aand the wiring line1901are located above the transistor112. The second emitter wiring line202aand the wiring line1901above the transistor112have a protruding portion2021A and protruding portions20011band20011c, respectively. In this case, the wiring portion W7is partially constituted by the second emitter wiring line202aand the wiring line1901, which are two metal layers. In such a case where the second emitter wiring line202includes the second emitter wiring lines202aand202bisolated from each other, heat can be transferred from the disposition area A1to the disposition areas A2and A3through the wiring line1901. Thus, variations in gain during operation can be suppressed more effectively. As a modification example of the third embodiment, the layout illustrated inFIG.22may be employed, in which heat transfer can be performed by providing a wiring line in a similar manner. In the layout illustrated inFIG.22, the second emitter wiring line202includes second emitter wiring lines202c,202d, and202eisolated from each other. The second emitter wiring line202cis disposed above the transistors101to10n. The second emitter wiring line202dis disposed above the transistor112. The second emitter wiring line202eis disposed above the transistor113. In the layout illustrated inFIG.22, a wiring line2201is disposed between a bump203E and the second emitter wiring lines202c,202d, and202ein the z-axis direction. The wiring line2201is made of a metallic material. The wiring line2201is disposed so as to overlap the individual transistors101to10n,112, and113in the disposition areas A1, A2, and A3. In the layer in which the wiring line2201is disposed, the wiring line2201is isolated from the remaining portion for the purpose of insulation. The wiring line2201may be regarded as a rewiring line. FIG.23illustrates a cross-sectional view taken along line XXIII-XXIII inFIG.22. The wiring line2201includes a first metal portion2301and a second metal portion2302. The wiring line2201is isolated by an insulating layer2303as necessary. The second emitter wiring line202eand the wiring line2201are located above the transistor113. The wiring line2201above the transistor113has a protruding portion23011a. The second emitter wiring line202, the wiring line2201, and the bump203E, which are three metal layers, constitute a wiring portion W8. The bump203E includes an under bump metal layer3111E, a metal post3112E, and a solder layer3113E. FIG.24illustrates a cross-sectional view taken along line XXIV-XXIV inFIG.22. InFIG.24, the second emitter wiring line202dand the wiring line2201are located above the transistor112. The second emitter wiring line202dabove the transistor112has the protruding portion2021A. The wiring line2201has protruding portions23011band23011c. As is clear from the modification example illustrated inFIG.22toFIG.24, the wiring line2201significantly increases the degree of freedom in designing thermal coupling in the disposition areas A1, A2, and A3. A fourth embodiment will be described.FIG.25illustrates a circuit diagram of a power amplifier circuit10A according to the fourth embodiment. The power amplifier circuit10A is different from the power amplifier circuits according to the foregoing embodiments in that the transistors101to10nare disposed both in two disposition areas A11and A12. The transistor111supplies the bias current Ib to the transistors101to10nthrough resistance elements121aand121b. FIG.26illustrates a layout diagram of the power amplifier circuit10A. In the power amplifier circuit10A, the second emitter wiring line202includes second emitter wiring lines202f,202g,202h, and202iisolated from each other. The second emitter wiring line202fis disposed above a subset of the transistors101to10n, that is, above the disposition area A11. The second emitter wiring line202gis disposed above the other subset of the transistors101to10n, that is, above the disposition area A12. The second emitter wiring line202his disposed above the transistor112. The second emitter wiring line202iis disposed above the transistor113. In the layout illustrated inFIG.26, a wiring line2601is disposed between bumps203aand203band the second emitter wiring lines202f,202g,202h, and202iin the z-axis direction. The wiring line2601is made of a metallic material. The wiring line2601is disposed so as to overlap the individual transistors101to10n,112, and113in the disposition areas A11, A12, A2, and A3. In the layer in which the wiring line2601is disposed, the wiring line2601is isolated from the remaining portion for the purpose of insulation. The wiring line2601may be regarded as a rewiring line. As illustrated inFIG.26, also in a case where the disposition area A1includes a plurality of areas, such as the disposition areas A11and A12divided from each other, variations in gain during operation can be suppressed by applying one embodiment of the present disclosure. In this case, the disposition area A1including the disposition areas A11and A12is connected to the disposition area A2via the wiring line2601. A fifth embodiment will be described.FIG.27illustrates a layout diagram of a power amplifier circuit according to the fifth embodiment. In the layout illustrated inFIG.27, the second emitter wiring line202includes second emitter wiring lines202j,202k, and202lisolated from each other. The second emitter wiring line202jis disposed on a first emitter wiring line351aof the transistor101. The second emitter wiring line202kis disposed on a first emitter wiring line351bof the transistor102. The second emitter wiring line202lis disposed above the transistor113. In addition, a wiring line2701is disposed along the collector wiring line341illustrated inFIG.27so as to connect grounds of the transistors101to10n. A bump2702is disposed above the wiring line2701. In the z-axis direction, a wiring line2703is disposed between a set of a bump203F and the bump2702, and a set of the second emitter wiring lines202j,202k, and202land the wiring line2701. The wiring line2703is made of a metallic material. The wiring line2703is disposed so as to overlap the individual transistors101to10n,112, and113in the disposition areas A1, A2, and A3. In the layer in which the wiring line2703is disposed, the wiring line2703is isolated from the remaining portion for the purpose of insulation. FIG.28illustrates a cross-sectional view taken along line XXVIII-XXVIII inFIG.27. The wiring line2703includes a first metal portion2801and a second metal portion2802. The wiring line2703is isolated by an insulating layer2803as necessary. The second emitter wiring line202land the wiring line2703are located above the transistor113. The wiring line2703above the transistor113has a protruding portion28011a. The second emitter wiring line202, the wiring line2703, and the bump203F, which are three metal layers, and a wiring line or the like described below constitute a wiring portion W9. The bump203F includes an under bump metal layer3111F, a metal post3112F, and a solder layer3113F. FIG.29illustrates a cross-sectional view taken along line XXIX-XXIX inFIG.27. InFIG.29, the wiring lines2701and2703are located above the transistor112. The wiring line2703above the transistor112has protruding portions28011band28011c. Also, with this configuration, the strength of the thermal coupling between the transistor112and the transistors101to10ncan be increased and variations in gain during operation can be suppressed. The wiring line2701, the wiring line2703, and the bump2702, which are three metal layers, constitute the wiring portion W9. Exemplary embodiments of the present disclosure have been described above. The power amplifier circuit10according to the first embodiment includes the transistor101disposed on the semiconductor substrate301; the transistor111disposed on the semiconductor substrate301and having a base configured to be supplied with the current I1which is a part of the control current Ic, the transistor111being configured to supply the bias current Ib based on the current I1to the transistor101; the transistor112disposed on the semiconductor substrate301and configured to be supplied with the current I2which is a part of the control current Ic and to output the current I3based on the current I2, the current I2increasing in accordance with a rise in temperature; and the wiring portion W1including the second emitter wiring line202and the bump203that are disposed so as to overlap at least a part of the disposition area A1in which the transistor101is disposed and an area between the disposition area A1and the disposition area A2in which the transistor112is disposed, that are electrically connected to the emitter of the transistor101, and that are stacked one on top of another so as to oppose the semiconductor substrate301. At least one of the second emitter wiring line202and the bump203extends so as to overlap an area extending from the at least the part of the disposition area A1to the disposition area A2in plan view of the semiconductor substrate301. This configuration enables heat transfer through the second emitter wiring line202and the bump203, and thus increases the strength of the thermal coupling between the transistor112and the transistors101and102that generate heat during operation. The increased strength of the thermal coupling makes it possible to adjust the bias current Ib in accordance with a change in characteristics resulting from a change in the temperature of the transistor101, and thus variations in gain during operation can be suppressed. In the power amplifier circuit10, both the second emitter wiring line202and the bump203of the wiring portion W1may extend so as to overlap the disposition area A1and the disposition area A2. This configuration makes it possible to transmit heat to a position closer to the transistor112. Thus, the strength of the thermal coupling between the transistor101and the transistor112is further increased. The uppermost metal layer of the wiring portion W1is the bump203. Thus, in a case where a semiconductor device including the power amplifier circuit10is flip-chip connected, the bump203serving as a heat path to the substrate extends to a position close to the transistor112. Thus, the strength of the thermal coupling between the transistor112and the transistors101and102can be increased. In the power amplifier circuit according to the second embodiment, the second emitter wiring line202C has the protruding portion2021extending toward the transistor112. Thus, the strength of the thermal coupling between the transistor101and the transistor112is further increased. In a power amplifier circuit according to another embodiment, a wiring portion may extend so as to overlap an area between the disposition area A1and the disposition area A3. Also, in this embodiment, the strength of the thermal coupling between the transistor101and the transistor112can be increased. Alternatively, the wiring portion may extend so as to overlap an area between the disposition area A1and the disposition area A2and to overlap an area which is between the disposition area A1and the disposition area A2and which is the disposition area A3except for the area. Also, in this embodiment, the strength of the thermal coupling between the transistor101and the transistor112can be increased. The power amplifier circuit according to the fifth embodiment includes the transistor101disposed on the semiconductor substrate301; the transistor111disposed on the semiconductor substrate301and having a base configured to be supplied with the current I1which is a part of the control current Ic, the transistor111being configured to supply the bias current Ib based on the current I1to the transistor101; the transistor112disposed on the semiconductor substrate301and configured to be supplied with the current I2which is a part of the control current Ic and to output the current I3based on the current I2, the current I2increasing in accordance with a rise in temperature; and the wiring portion W9including the wiring line2701, the wiring line2703, and the bump2702that are disposed so as to overlap the disposition area A1in which the transistor101is disposed and an area between the disposition area A1and the disposition area A2in which the transistor112is disposed, that are electrically connected to the collector of the transistor101, and that are stacked one on top of another so as to oppose the semiconductor substrate301. Any one of the wiring line2701, the wiring line2703, and the bump2702extends so as to overlap an area extending from the disposition area A1to the disposition area A2in plan view of the semiconductor substrate301. Also, in this embodiment, the strength of the thermal coupling between the transistor101and the transistor112can be increased. Each of the above-described embodiments can be applied to individual amplification stages of a power amplifier circuit. For example, in a power amplifier circuit having a three-stage configuration, the strength of the thermal coupling between a transistor and a bias circuit can be increased in at least one amplification stage. As an example, in a power amplifier circuit having a three-stage configuration, an increase in gain can be further suppressed by increasing the strength of the thermal coupling between a transistor and a bias circuit in the second stage and between a transistor and a bias circuit in the final stage. The individual embodiments described above are given for facilitating the understanding of the present disclosure and are not for interpreting the present disclosure in a limited manner. The present disclosure can be modified or improved without necessarily deviating from the gist thereof, and includes the equivalents thereof. That is, an embodiment obtained by appropriately changing the design of one of the embodiments by a person skilled in the art is included in the scope of the present disclosure as long as the embodiment has a feature of the present disclosure. For example, the individual elements of the individual embodiments, and the dispositions, materials, conditions, shapes, sizes, and so forth thereof are not limited to those described as an example and can be changed as appropriate. Each of the embodiments is merely an example. Obviously, configurations illustrated in different embodiments can be partially replaced or combined, and the replacement or combination is also included in the scope of the present disclosure as long as the replacement or combination has a feature of the present disclosure. The scope of the disclosure is to be determined solely by the following claims. | 49,899 |
11863129 | DETAILED DESCRIPTION Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. It should be noted that the embodiment described below is a comprehensive or specific instance. Specifics including numerical values, shapes, materials, constituent elements, arrangements of the constituent elements, and modes of connection given in the following embodiment are mere instances and are not intended to limit the present disclosure. Among the constituent elements in the following embodiment, constituent elements not recited in any of the independent claims are described as arbitrary constituent elements. Furthermore, the size or the size ratio of the constituent elements illustrated in the drawings is not necessarily presented in an exact manner. Like reference characters are used to denote substantially like configurations in the drawings, and redundant descriptions thereof may be omitted or simplified. Configurations less related to features of the present disclosure may be not assigned any reference character in the drawings including circuit diagrams or not described. Embodiment Firstly, a bias circuit according to a comparative example will be explained before a description of a bias circuit according to an embodiment. FIG.1is a circuit configuration diagram illustrating an example of a bias circuit100according to the comparative example. The bias circuit according to the embodiment partially includes the circuit configuration of the bias circuit100according to the comparative example, and constituent elements included in the bias circuit100are not described in detail here. The bias circuit100according to the comparative example uses a resistor R1instead of a current source using a P-ch transistor or depletion-mode field-effect transistor (FET) for the purpose of reducing costs of the bias circuit100. The resistor R1can be used instead of a current source because a current according to the supply voltage supplied from a terminal11flows through the resistor R1. However, for example, when a battery in a device, such as a portable terminal supplies the supply voltage, as the battery capacity decreases with the use of the device, the supply voltage also gradually decreases, which results in decrease in the current flowing through the resistor R1. As a result, the bias voltage outputted from a terminal12varies along with the supply voltage. Specifically, variations in the supply voltage vary the current flowing through a resistor R2and the current flowing through a resistor R3, and consequently, the bias voltage outputted from the terminal12varies. In consideration of such a problem, the present disclosure provides a bias circuit capable of supplying stable bias voltage regardless of supply voltage. The following will be a description of such a bias circuit. FIG.2is a circuit configuration diagram illustrating an example of a bias circuit10according to the embodiment.FIG.2also illustrates a load200. The bias circuit10is configured to produce a bias voltage and supply the produced bias voltage to the load200. For example, the bias circuit10has the terminals11and12. The terminal11is an example of a power supply terminal coupled to a power supply. The power supply supplies a supply voltage to the bias circuit10via the terminal11. The terminal12is an example of an output terminal coupled to the load200. The bias circuit10supplies a bias voltage to the load200via the terminal12. The bias circuit10includes transistors Tr1, Tr2, Tr3, Tr4, Tr5, and Tr6and resistors R1, R2, R3, R4, and R5. The transistor Tr1is an example of a first transistor including a first terminal functioning as a base or gate, a second terminal functioning as a collector or drain, and a third terminal functioning as an emitter or source. The transistor Tr2is an example of a second transistor including a fourth terminal functioning as a base or gate, a fifth terminal functioning as a collector or drain, and a sixth terminal functioning as an emitter or source. The transistor Tr3is an example of a third transistor including a seventh terminal functioning as a base or gate, an eighth terminal functioning as a collector or drain, and a ninth terminal functioning as an emitter or source. The transistor Tr4is an example of a fourth transistor including a tenth terminal functioning as a base or gate, an eleventh terminal functioning as a collector or drain, and a twelfth terminal functioning as an emitter or source. The transistor Tr5is an example of a fifth transistor including a thirteenth terminal functioning as a base or gate, a fourteenth terminal functioning as a collector or drain, and a fifteenth terminal functioning as an emitter or source. The transistor Tr6is an example of a sixth transistor including a sixteenth terminal functioning as a base or gate, a seventeenth terminal functioning as a collector or drain, and an eighteenth terminal functioning as an emitter or source. The resistor R1is an example of a first resistor. The resistor R2is an example of a second resistor. The resistor R3is an example of a third resistor. The resistor R4is an example of a fourth resistor. The resistor R5is an example of a fifth resistor. The following description includes expressions, such as one end of a resistor and the other end of the resistor. For example, one end denotes a terminal on the upper side of a corresponding resistor in the drawings, and the other end denotes a terminal on the lower side of a corresponding resistor in the drawings. The transistors Tr5and Tr6together form a current mirror circuit having a function of equalizing the amount of current flowing from a node N1(refer toFIG.2) into the fourteenth terminal of the transistor Tr5and the amount of current flowing into the resistor R5. For example, the transistors Tr1, Tr2, Tr3, Tr4, Tr5, and Tr6are bipolar transistors, such as heterojunction bipolar transistors. In this case, the first terminal, the fourth terminal, the seventh terminal, the tenth terminal, the thirteenth terminal, and the sixteenth terminal are bases; the second terminal, the fifth terminal, the eighth terminal, the eleventh terminal, the fourteenth terminal, and the seventeenth terminal are collectors; the third terminal, the sixth terminal, the ninth terminal, the twelfth terminal, the fifteenth terminal, and the eighteenth terminal are emitters. The collector of the transistor Tr1, one end of the resistor R1, and one end of the resistor R5are coupled to each other and further coupled to the terminal11. The base of the transistor Tr1, the collector of the transistor Tr2, and the other end of the resistor R1are coupled to each other. The emitter of the transistor Tr1, one end of the resistor R2, and the one end of the resistor R3are coupled to each other and further coupled to the terminal12. The base of the transistor Tr2, the collector of the transistor Tr3, and the other end of the resistor R2are coupled to each other. The base of the transistor Tr3, the base of the transistor Tr4, the collector of the transistor Tr4, the collector of the transistor Tr5, and the other end of the resistor R3are coupled to each other. The node N1is a node in a path connecting the collector of the transistor Tr4and the other end of the resistor R3. The base of the transistor Tr5, the base of the transistor Tr6, the collector of the transistor Tr6, and the other end of the resistor R5are coupled to each other. The emitter of the transistor Tr3and the one end of the resistor R4are coupled to each other. The emitter of the transistor Tr2, the emitter of the transistor Tr4, the emitter of the transistor Tr5, the emitter of the transistor Tr6, and the other end of the resistor R4are coupled to each other and further coupled to the ground. The transistors Tr2, Tr3, and Tr4and the resistors R2, R3, and R4together form a Widlar bandgap reference circuit. When Vbgris a bias voltage outputted from the terminal12, VBE2is a voltage between the base and emitter of the transistor Tr2, and VR2is a voltage applied to the resistor R2(that is, a voltage between the emitter of the transistor Tr1and the base of the transistor Tr2), Vbgris given by Equation 1. Vbgr=VBE2+VR2(Equation 1) When T is an absolute temperature of the transistor Tr2, for example, Equation 2 is satisfied. ∂VBE2∂T=-1.8(mV/°C.)(Equation2) According to this equation, the temperature coefficient of the voltage between the base and emitter of the transistor Tr2is −1.8 mV/° C. The temperature coefficient −1.8 mV/° C. is a mere example. The temperature coefficient of the voltage between the base and emitter of the transistor Tr2is dependent on the transistor structure of the transistor Tr2. When R2is a resistance of the resistor R2, R3is a resistance of the resistor R3, R4is a resistance of the resistor R4, k is the Boltzmann constant, q is the electric charge carried by an electron, Equation 3 is satisfied. ∂VR2∂T=R2R4·kq·lnR2R3(Equation3) According to Equation 3, controlling the ratio of the resistance of the resistor R2to the resistance of the resistor R3and the ratio of the resistance of the resistor R2to the resistance of the resistor R4can control the temperature coefficient of the voltage applied to the resistor R2. For example, by setting the temperature coefficient of the voltage applied to the resistor R2to 1.8 mV/° C., the negative temperature coefficient (−1.8 mV/° C.) of the voltage between the base and emitter of the transistor Tr2is canceled, and as a result, a fixed bias voltage can be outputted from the terminal12regardless of variations in temperature. Although the bias circuit10uses the resistor R1instead of a current source, a stable bias voltage can be supplied regardless of supply voltage because the bias circuit10includes the transistors Tr5and Tr6and the resistor R5. The following describes how stable bias voltage can be supplied regardless of supply voltage. When the supply voltage from a power supply coupled to the terminal11varies, the current flowing in the resistor R5varies along with the variations. When the supply voltage from the power supply coupled to the terminal11varies, the current flowing into the resistor R3also varies. Because the transistors Tr5and Tr6(that is, a current mirror circuit) has a function of equalizing the current flowing from the node N1into the collector of the transistor Tr5and the current flowing into the resistor R5, when the current flowing into the resistor R5varies, the current flowing from the node N1into the collector of the transistor Tr5also varies in the same manner. The amount of variation in the current flowing into the resistor R3flows from the node N1into the collector of the transistor Tr5, whereas the current flowing from the node N1into the collector of the transistor Tr4is maintained at a fixed amount regardless of the variations in the supply voltage. Because the transistors Tr3and Tr4form a current mirror circuit, a fixed current flows through the resistor R2, the transistor Tr3, and the resistor R4, and as a result, the bias voltage is maintained at a fixed level regardless of variations in the supply voltage. FIG.3is a graph illustrating the relationship between supply voltage and bias voltage with respect to the bias circuit10according to the embodiment and the bias circuit100according to the comparative example. As for the bias circuit100according to the comparative example, it can be seen that the bias voltage varies with variations in the supply voltage. By contrast, as for the bias circuit10according to the embodiment, the bias voltage is maintained at a substantially fixed level regardless of variations in the supply voltage. For example, it can be seen that, in the range of 3 to 3.3 V, which is a range of supply voltage actually used in devices, such as portable terminals, the bias voltage is maintained at approximately 2.4 V. Next, a specific example of the load200will be described. FIG.4is a circuit configuration diagram illustrating an example of the load200to which the bias circuit10according to the embodiment supplies bias voltage.FIG.4also illustrates the bias circuit10that supplies bias voltage to the load200. The terminal11is provided on the upper side of the bias circuit10inFIG.4, and a power supply (indicated by “VCC” inFIG.4) is coupled to the terminal11; the terminal12is provided on the lower side of the bias circuit10inFIG.4, and the load200(sensing circuit) is coupled to the terminal12. The terminals11and12are not illustrated inFIG.4. Although detailed descriptions of constituent elements included in the load200illustrated inFIG.4are omitted, the load200has a function of sensing output power of a power amplifier. An output signal from a power amplifier is partially inputted to “PAin” indicated inFIG.4. The load200outputs from “Vout” a voltage according to the signal inputted to “PAin”. This means that the voltage outputted from “Vout” corresponds to output power of the power amplifier, and thus, the output power can be measured by measuring the voltage outputted from “Vout”. An enable signal is inputted to “PAen”. While the enable signal is inputted to “PAen”, the function of sensing output power of a power amplifier is active in the load200. The transistors in the load200are, for example, bipolar transistors (for example, gallium arsenide-based bipolar transistors), which contributes to implementation of the load200with reduced size and high performance. The transistors Tr2, Tr3, Tr4, Tr5, and Tr6in the bias circuit10may be bipolar transistors (for example, gallium arsenide-based bipolar transistors), which contributes to implementation of the bias circuit10with reduced size and high performance. The transistors Tr1, Tr2, Tr3, Tr4, Tr5, and Tr6may be FETs. In this case, the first terminal, the fourth terminal, the seventh terminal, the tenth terminal, the thirteenth terminal, and the sixteenth terminal are gates; the second terminal, the fifth terminal, the eighth terminal, the eleventh terminal, the fourteenth terminal, and the seventeenth terminal are drains; the third terminal, the sixth terminal, the ninth terminal, the twelfth terminal, the fifteenth terminal, and the eighteenth terminal are sources. In the above description, base may be replaced with gate, collector may be replaced with drain, and emitter may be replaced with source. As described above, the bias circuit10includes the transistor Tr1including the first terminal functioning as a base or gate, the second terminal functioning as a collector or drain, and the third terminal functioning as an emitter or source, the transistor Tr2including the fourth terminal functioning as a base or gate, the fifth terminal functioning as a collector or drain, and the sixth terminal functioning as an emitter or source, the transistor Tr3including the seventh terminal functioning as a base or gate, the eighth terminal functioning as a collector or drain, and the ninth terminal functioning as an emitter or source, the transistor Tr4including the tenth terminal functioning as a base or gate, the eleventh terminal functioning as a collector or drain, and the twelfth terminal functioning as an emitter or source, the transistor Tr5including the thirteenth terminal functioning as a base or gate, the fourteenth terminal functioning as a collector or drain, and the fifteenth terminal functioning as an emitter or source, the transistor Tr6including the sixteenth terminal functioning as a base or gate, the seventeenth terminal functioning as a collector or drain, and the eighteenth terminal functioning as an emitter or source, the resistor R1, the resistor R2, the resistor R3, the resistor R4, and the resistor R5. The second terminal, the one end of the resistor R1, and the one end of the resistor R5are coupled to each other and further coupled to the terminal11. The first terminal, the fifth terminal, and the other end of the resistor R1are coupled to each other. The third terminal, the one end of the resistor R2, and the one end of the resistor R3are coupled to each other and further coupled to the terminal12. The fourth terminal, the eighth terminal, and the other end of the resistor R2are coupled to each other. The seventh terminal, the tenth terminal, the eleventh terminal, the fourteenth terminal, and the other end of the resistor R3are coupled to each other. The thirteenth terminal, the sixteenth terminal, the seventeenth terminal, and the other end of the resistor R5are coupled to each other. The ninth terminal and the one end of the resistor R4are coupled to each other. The sixth terminal, the twelfth terminal, the fifteenth terminal, the eighteenth terminal, and the other end of the resistor R4are coupled to each other and further coupled to the ground. With this configuration, a current according to variations in supply voltage flows into the resistor R5; the current mirror circuit constituted by the transistors Tr5and Tr6passes the current according to the variations from the node N1into the fourteenth terminal of the transistor Tr5; as a result, the current flowing from the node N1into the eleventh terminal of the transistor Tr4is maintained at a fixed amount regardless of the variations in the supply voltage. Because the transistors Tr3and Tr4form a current mirror circuit, a fixed current flows through the resistor R2, the transistor Tr3, and the resistor R4, and as a result, the bias voltage is maintained at a fixed level regardless of variations in the supply voltage. As such, the behavior of bias voltage dependent on supply voltage is eliminated, and thus, a stable bias voltage can be supplied regardless of supply voltage. Additionally, because the bias circuit10uses the resistor R1instead of a current source, cost reduction can be achieved. For example, the transistors Tr2, Tr3, and Tr4may be bipolar transistors. This can contribute to implementation of the bias circuit10with reduced size and high performance. For example, when the load200coupled to the terminal12is a small load, the bias circuit10also needs to be miniaturized to downsize the entire system. In this case, because the transistors in the bias circuit10are bipolar transistors, the entire system can be downsized. For example, a sensing circuit for sensing output power of a power amplifier may be coupled to the terminal12. In this case, regardless of supply voltage, a stable bias voltage can be supplied to the sensing circuit for sensing output power of a power amplifier. First Modification Next, a bias circuit according to a first modification will be described. FIG.5is a circuit configuration diagram illustrating an example of a bias circuit10aaccording to the first modification. The bias circuit10aaccording to the first modification differs from the bias circuit10according to the embodiment in that the transistor Tr1is a FET (specifically, an enhancement-mode FET). Other points of the bias circuit10aaccording to the first modification are identical to those of the bias circuit10according to the embodiment, and descriptions thereof are not repeated. Because the transistor Tr1is a FET, the bias circuit10acan supply more stable bias voltage than if the transistor Tr1is a bipolar transistor. This is because voltage drop between the gate and source of a FET is less likely to be dependent on temperature than a bipolar transistor. Moreover, the amount of voltage drop between the gate and source of the transistor Tr1is less than that of a bipolar transistor, and thus, a low-power supply voltage can be used as a power supply. Alternatively, bias voltage may be set at a level close to supply voltage. Second Modification Next, a bias circuit according to a second modification will be described. FIG.6is a circuit configuration diagram illustrating an example of a bias circuit10baccording to the second modification. The bias circuit10baccording to the second modification differs from the bias circuit10aaccording to the first modification in the following points: a resistor R6is further included; the first terminal of the transistor Tr1, the fifth terminal of the transistor Tr2, and the other end of the resistor R1are coupled to each other and further coupled to the terminal12(specifically, the first terminal of the transistor Tr1and the other end of the resistor R1are coupled to the terminal12via the resistor R6); the third terminal of the transistor Tr1, the one end of the resistor R2, and the one end of the resistor R3are coupled to each other. Other points of the bias circuit10baccording to the second modification are identical to those of the bias circuit10aaccording to the first modification, and descriptions thereof are not repeated. It is assumed that a fixed load current flows through the load200. The description of the bias circuit10aaccording to the first modification has used an example in which the terminal12is coupled to the third terminal of the transistor Tr1, whereas in the second modification the terminal12is coupled to the first terminal of the transistor Tr1(specifically, the terminal12is coupled to the first terminal of the transistor Tr1via the resistor R6). When the terminal12is coupled to the third terminal of the transistor Tr1, the bias voltage outputted from the terminal12is lowered by the amount of voltage drop across the resistor R1and the amount of voltage drop between the gate and source of the transistor Tr1in comparison to supply voltage. Thus, a power supply with a supply voltage higher than these voltage drops needs to be prepared. By contrast, when the terminal12is coupled to the first terminal of the transistor Tr1, the bias voltage outputted from the terminal12is not lowered by the amount of voltage drop between the gate and source of the transistor Tr1. Thus, a power supply with a supply voltage at least higher than voltage drop across the resistor R1needs to be prepared. The resistor R6is an example of a sixth resistor for controlling the level of bias voltage. The bias voltage outputted from the terminal12is lowered by the amount of voltage drop across the resistor R6in comparison to supply voltage. For example, the resistance of the resistor R6is set at a small value that enables the voltage drop caused by current flowing through the resistor R6to be smaller than the voltage drop between the gate and source of the transistor Tr1. As described above, the bias circuit10bincludes the transistor Tr1including the first terminal functioning as a base or gate, the second terminal functioning as a collector or drain, and the third terminal functioning as an emitter or source, the transistor Tr2including the fourth terminal functioning as a base or gate, the fifth terminal functioning as a collector or drain, and the sixth terminal functioning as an emitter or source, the transistor Tr3including the seventh terminal functioning as a base or gate, the eighth terminal functioning as a collector or drain, and the ninth terminal functioning as an emitter or source, the transistor Tr4including the tenth terminal functioning as a base or gate, the eleventh terminal functioning as a collector or drain, and the twelfth terminal functioning as an emitter or source, the transistor Tr5including the thirteenth terminal functioning as a base or gate, the fourteenth terminal functioning as a collector or drain, and the fifteenth terminal functioning as an emitter or source, the transistor Tr6including the sixteenth terminal functioning as a base or gate, the seventeenth terminal functioning as a collector or drain, and the eighteenth terminal functioning as an emitter or source, the resistor R1, the resistor R2, the resistor R3, the resistor R4, and the resistor R5. The second terminal, the one end of the resistor R1, and the one end of the resistor R5are coupled to each other and further coupled to the terminal11. The first terminal, the fifth terminal, and the other end of the resistor R1are coupled to each other and further coupled to the terminal12. The third terminal, the one end of the resistor R2, and the one end of the resistor R3are coupled to each other. The fourth terminal, the eighth terminal, and the other end of the resistor R2are coupled to each other. The seventh terminal, the tenth terminal, the eleventh terminal, the fourteenth terminal, and the other end of the resistor R3are coupled to each other. The thirteenth terminal, the sixteenth terminal, the seventeenth terminal, and the other end of the resistor R5are coupled to each other. The ninth terminal and the one end of the resistor R4are coupled to each other. The sixth terminal, the twelfth terminal, the fifteenth terminal, the eighteenth terminal, and the other end of the resistor R4are coupled to each other and further coupled to the ground. This yields the bias circuit10bcapable of supplying stable bias voltage regardless of supply voltage. Furthermore, because the terminal12is coupled to the first terminal of the transistor Tr1, the bias voltage outputted from the terminal12is not lowered by the amount of voltage drop between the gate and source of the transistor Tr1. As a result, a power supply with low supply voltage can be used. Alternatively, bias voltage may be set at a level close to supply voltage. For example, the bias circuit10bmay further include the resistor R6. Via the resistor R6, the first terminal and the other end of the resistor R1may be coupled to the terminal12. With this configuration, controlling the resistance of the resistor R6can control the level of bias voltage. OTHER EMBODIMENTS The bias circuit according to the present disclosure has been described by using the embodiment, but the present disclosure is not limited to the embodiment described above. The present disclosure also embraces other embodiments implemented as any combination of the constituent elements of the embodiment, other modified examples obtained by making various modifications to the embodiment that occur to those skilled in the art without necessarily departing from the scope of the present disclosure, and various hardware devices including the amplifier circuit according to the present disclosure. For example, the second modification uses the example in which the transistor Tr1is a FET, but the transistor Tr1may be a bipolar transistor in the second modification similarly to the embodiment. For example, the second modification uses the example in which the bias circuit10bincludes the resistor R6, but the bias circuit10bdoes not necessarily include the resistor R6. For example, the embodiment uses the example in which the load200is a sensing circuit for sensing output power of a power amplifier, but the load200is not necessarily such a sensing circuit. The present disclosure can be used for a wide variety of communication devices, such as mobile phones, as a bias circuit for supplying bias voltage to, for example, a sensing circuit for sensing output power of a power amplifier. While embodiments of the disclosure have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without necessarily departing from the scope and spirit of the disclosure. The scope of the disclosure, therefore, is to be determined solely by the following claims. | 27,369 |
11863130 | DETAILED DESCRIPTION Conventional Group III nitride-based RF transistor amplifiers, such as the RF transistor amplifier100ofFIGS.1A-1B, may use bond wires to connect the RF transistor amplifier die110to gate and drain leads172,174. These bond wires have inherent inductance that may be used to implement some of the inductors in the impedance matching and/or harmonic termination circuits of the RF transistor amplifiers. The amount of inductance provided may be varied by changing the length and/or the cross-sectional area (e.g., the diameter) of the bond wires so that the bond wires provide a desired amount of inductance. Unfortunately, as applications move to higher frequencies, the inductance of the bond wires may exceed a desired amount of inductance for the impedance matching and/or harmonic termination circuits. When this occurs, bond wires that are very short and/or that have large cross-sectional areas may be used in an effort to decrease the inductance thereof to suitable levels. Very short bond wires, however, may be difficult to solder in place, which may increase manufacturing costs, and/or may result in higher device failure rates. Bond wires having large cross-sectional areas may require larger gate and drain bond pads on the RF transistor amplifier die, which require an increase in the overall size of the RF transistor amplifier die, which is also undesirable. Moreover, in some higher frequency applications, even very short bond wires having large cross-sectional areas may have too much inductance such that the matching networks cannot, for example, properly terminate the second or third order harmonics. While the RF transistor amplifiers may be implemented as MMIC devices in order to avoid the problem of too much inductance in the bond wires, MMIC RF amplifiers are more expensive to fabricate and can only be used in the frequency range of the matching circuits, reducing flexibility. Pursuant to embodiments of the present invention, Group III nitride-based RF transistor amplifiers are provided that include RF transistor amplifier dies that have source terminals and at least one of their drain terminals and/or their gate terminals all located on the back side of the RF transistor amplifier die. The gate, drain and source terminals may all be connected to corresponding gate, drain and source pads on an interconnection structure using conductive contacts such as, for example, conductive bump technology (e.g., solder bumps), die attach material, conductive epoxies, or other low inductance electrical connections. In some embodiments, the RF transistor amplifiers may not include any bond wires. The RF amplifier die may include one or more conductive gate vias and/or one or more conductive drain vias that are used to connect a gate bus and/or a drain bus that are on the top side of the RF transistor amplifier die to the respective gate and drain terminals that are on the back side of the RF transistor amplifier die. The length of the conductive vias may be a small fraction (e.g., 10-30%) of the length of conventional bond wires, and hence the inductance of the connections between the gate and drain buses and the interconnection structure may be reduced significantly. As a result, the impedance matching and/or harmonic termination circuits may be configured to have a desired amount of inductance without the need for implementing the RF transistor amplifier as a MMIC device. Thus, the size of the RF transistor amplifier dies may be reduced without compromising the performance thereof, and the RF transistor amplifier dies can be used for applications in a variety of different frequency bands, as the frequency-specific portions of the device (e.g., the matching circuits) may be implemented as separate chips or circuits. Moreover, the wire bonding equipment that is typically used for high volume manufacturing may have a tolerance of +/−1 mil, meaning that the length of any particular wire bond may vary by as much a 4 mils (i.e., +/−1 mil on each end of the bond wire). For high frequency applications, the variation in inductance associated with 4 mils of wire bond may be significant, and hence the performance of the matching circuits may be degraded if the bond wires are 1-2 mils too short or long from a desired nominal length. Forming the gate and drain terminals on the back side of the device and using contacts to connect these terminals to corresponding pads on the interconnection structure may largely eliminate this process variation, resulting in improved performance. Pursuant to some embodiments of the present invention, RF transistor amplifiers are provided that include an interconnection structure and a Group III nitride-based RF transistor amplifier die that is mounted on top of interconnection structure. The Group III nitride-based RF transistor amplifier die includes a semiconductor layer structure. A plurality of unit cell transistors are provided in an upper portion of the semiconductor layer structure, and a gate terminal, a drain terminal and a source terminal are provided on a lower surface of the semiconductor layer structure that is adjacent the interconnection structure. The gate terminal is electrically connected to the unit cell transistors through one or more conductive gate vias, the drain terminal is electrically connected to the unit cell transistors through one or more conductive drain vias, and the source terminal is electrically connected to the unit cell transistors through one or more conductive source vias. The gate, drain and source vias may extend completely through the semiconductor layer structure. In some embodiments, the RF transistor amplifiers may comprise a Group III nitride-based RF transistor amplifier die that has a semiconductor layer structure having a source region therein, a conductive source via and an additional conductive via that each extend through the semiconductor layer structure. A first end of the additional conductive via is connected to a first external circuit and a second, opposed end of the additional conductive via is connected to a first matching circuit. The additional conductive via can be a conductive gate via that is connected to a gate electrode or a conductive drain via that is connected to a drain electrode of the RF transistor amplifier die. In other embodiments, the RF transistor amplifier may comprise a Group III nitride-based RF transistor amplifier die that includes a semiconductor layer structure and a conductive via that extends through the semiconductor layer structure. A first impedance matching circuit is coupled between a first end of the conductive via and a first external electrical connection and a first harmonic termination circuit is coupled between a second opposed end of the additional conductive via and a second external electrical connection. In still other embodiments, the RF transistor amplifier comprises (1) an interconnection structure such as, for example, a redistribution layer (“RDL”) laminate substrate, a printed circuit board, an interposer or a substrate having a dielectric layer or pattern on a surface thereof with conductive traces on the dielectric pattern/layer opposite the substrate and (2) a Group III nitride-based RF transistor amplifier die on a top surface of the interconnection structure. The Group III nitride-based RF transistor amplifier die includes a semiconductor layer structure that has a plurality of unit cell transistors in an upper portion thereof, a conductive source via, a conductive gate via, and a conductive drain via, each of which extends through the semiconductor layer structure, and a plurality of contacts on a bottom surface of the RDL laminate substrate. Embodiments of the present invention will now be discussed in further detail with reference to the accompanying figures. FIGS.2A-2Gdepict a Group III nitride-based RF transistor amplifier200according to certain embodiments of the present invention. In particular,FIG.2Ais a schematic side view of the Group III nitride-based RF transistor amplifier200.FIG.2Bis a schematic cross-sectional view of an RF transistor amplifier die210that is part of the Group III nitride-based RF transistor amplifier200ofFIG.2Athat is taken along line2B-2B ofFIG.2A.FIGS.2C through2Fare schematic cross-sectional views of the RF transistor amplifier die210that are taken along lines2C-2C through2F-2F ofFIG.2B, respectively. Finally,FIG.2Gis a schematic bottom view of the RF transistor amplifier die210. As shown inFIG.2A, the Group III nitride-based RF transistor amplifier200includes an RF transistor amplifier die210that is mounted on the upper surface of an interconnection structure270. The RF transistor amplifier die210has a top side212and a bottom side214. The RF transistor amplifier die210includes a bottom side metallization structure220, a semiconductor layer structure230, and a top side metallization structure240that are sequentially stacked. The bottom side metallization structure220comprises a gate terminal222, a drain terminal224, and a source terminal226. The RF transistor amplifier200may be a HEMT-based RF transistor amplifier, in which case the semiconductor layer structure230may include at least a channel layer and a barrier layer, as will be discussed in greater detail with reference toFIGS.2C and2D. The top side metallization structure240will be discussed in greater detail with reference toFIG.2B. The interconnection structure270may comprise any structure that is electrically connected to the RF transistor amplifier die210that provides a suitable mounting surface for the RF transistor amplifier die210. In some cases, the interconnection structure270may comprise an RDL laminate structure. An RDL laminate structure refers to a substrate that has conductive layer patterns and/or conductive vias for electrical and/or thermal interconnection. RDL laminate structures may be fabricated using semiconductor processing techniques by depositing conductive and insulating layers and/or patterns on a base material and by forming vias and copper routing patterns within the structure for transmitting signals through the RDL laminate structure. Other interconnection structures270may alternatively be used such as, for example, a printed circuit board (e.g., a multi-layer printed circuit board), a metal core printed circuit board, or a ceramic substrate that includes conductive vias and/or pads. In still other embodiments, the interconnection structure270may comprise a metal flange that has an insulating pattern on a top surface thereof, and conductive traces on the insulating layer that, for example, provide electrical connections to the gate terminal222and the drain terminal224. The source terminal226may be electrically connected to the metal flange via, for example, electrically conductive die attach material such as solder. In some embodiments, the insulating pattern formed on the top surface of the metal flange may be a solder mask layer. In any event, it will be appreciated that the interconnection structure270may be any suitable mounting surface for the RF transistor amplifier die210that can make electrical connections to the back side214of the RF transistor amplifier die210. More than one interconnection structure270may be provided in a stacked manner. The RF transistor amplifier die210may be mounted on the interconnection structure270(e.g., on an RDL laminate structure) by the die manufacturer. In other cases, the RF transistor amplifier die210may be directly mounted in a package on a package submount, such as a metal flange, where dielectric and traces are formed on the metal flange so that the metal flange can act as interconnection structure270. A gate pad272, a drain pad274and a source pad276are provided on the top surface of the interconnection structure270. Each of these pads272,274,276may comprise, for example, an exposed copper pad. The gate terminal222may overlap the gate pad272along a first vertical axis that extends perpendicular to the top surface of the semiconductor layer structure230, the drain terminal224may overlap the drain pad274along a second vertical axis that extends perpendicular to the top surface of the semiconductor layer structure230, and the source terminal226may overlap the source pad276along a third vertical axis that extends perpendicular to the top surface of the semiconductor layer structure230. By “overlap” it is meant that the axis extends through both the terminal and its corresponding pad, and “vertical” refers to a direction that is perpendicular to a major surface of the semiconductor layer structure230. Each overlapping terminal and pad (e.g., gate terminal222and gate pad272) may be physically and electrically connected to each other by any suitable contacts including, for example, a conductive bump (e.g., a solder bump or a conductive epoxy), a die attach material, or the like (not shown). It will be appreciated that any type of bump grid array technology may be used to connect the gate, drain and source terminals222,224,226to the respective gate, drain and source pads272,274,276while facilitating dissipation of heat from the RF amplifier die210. The interconnection structure270may further includes a plurality of heat dissipation structures290. In the depicted embodiment, the heat dissipation structures290comprise metal-filled (or partly metal-filled) vias that extend through the interconnection structure270. Heat that is generated in the RF transistor amplifier die210may be dissipated through the metal-filled vias290. The RF transistor amplifier die210may comprise a Group III nitride-based HEMT RF transistor amplifier that includes a plurality of unit cell transistors216that are electrically connected to each other in parallel. This can best be seen inFIG.2B, which schematically depicts a cut through the top side metallization structure240of RF transistor amplifier die210. As shown inFIG.2B, the top side metallization structure240includes a gate bus242and a drain bus244, a plurality of gate fingers252, a plurality of drain fingers254and a plurality of source fingers256, all of which may be formed on an upper surface of the semiconductor layer structure230. The gate bus242and gate fingers252are part of a gate electrode of the RF transistor amplifier die210. The gate bus242and the gate fingers252may be implemented as a first monolithic metal pattern. The drain bus244and drain fingers254are part of a drain electrode of the RF transistor amplifier die210, and may be implemented as a second monolithic metal pattern. The gate fingers252may be formed of materials that are capable of making a Schottky contact to a Group III nitride-based semiconductor material, such as Ni, Pt, Cu, Pd, Cr, W and/or WSiN. The drain fingers254and source fingers256may include a metal, such as TiAlN, that can form an ohmic contact to Group III nitride-based materials. A dielectric layer (or a series of dielectric layers) that help isolate the gate metallization242,252, the drain metallization244,254and the source metallization256from each other is not shown inFIG.2Bto better illustrate the elements of the top side metallization structure240. A conductive gate bond pad243and/or a conductive drain bond pad253may optionally be provided on the upper surface of the RF transistor amplifier die210. The gate bond pad243may be electrically connected to the gate terminal222, and the drain bond pad253may be electrically connected to the drain terminal224. One of the unit cell transistors216is also shown inFIG.2B. As shown, the unit cell transistor216includes a gate finger252, a drain finger254and a source finger256along with the underlying portion of the semiconductor layer structure230. Since all of the gate fingers252are electrically connected to a common gate bus242, all of the drain fingers254are electrically connected to a common drain bus244, and all of the source fingers256are electrically connected together via the conductive source vias266(discussed below) and the source terminal226, it can be seen that the unit cell transistors216are all electrically connected together in parallel. The unit cell transistors216may by HEMT devices. Suitable structures for Group III-nitride-based HEMT devices that may utilize embodiments of the present invention are described, for example, in commonly assigned U.S. Patent Publication No. 2002/0066908A1 published Jun. 6, 2002, for “Aluminum Gallium Nitride/Gallium Nitride High Electron Mobility Transistors Having A Gate Contact On A Gallium Nitride Based Cap Segment And Methods Of Fabricating Same,” U.S. Patent Publication No. 2002/0167023A1 for “Group-III Nitride Based High Electron Mobility Transistor (HEMT) With Barrier/Spacer Layer,” published Nov. 14, 2002, U.S. Patent Publication No. 2004/0061129 for “Nitride-Based Transistors And Methods Of Fabrication Thereof Using Non-Etched Contact Recesses,” published on Apr. 1, 2004, U.S. Pat. No. 7,906,799 for “Nitride-Based Transistors With A Protective Layer And A Low-Damage Recess” issued Mar. 15, 2011, and U.S. Pat. No. 6,316,793 entitled “Nitride Based Transistors On Semi-Insulating Silicon Carbide Substrates,” issued Nov. 13, 2001, the disclosures of which are hereby incorporated herein by reference in their entirety. As is further shown inFIG.2B, a plurality of metal-plated vias are provided that extend from the top metallization structure240through the semiconductor layer structure230. The metal-plated vias include metal-plated gate vias262, metal-plated drain vias264, and metal-plated source vias266. The metal-plated gate vias262physically and electrically connect the gate bus242to the gate terminal222, the metal-plated drain vias264physically and electrically connect the drain bus244to the drain terminal224, and metal-plated source vias262physically and electrically connect the source fingers256to the source terminal226. As is further shown inFIG.2B, the conductive gate vias262and/or the conductive drain vias264may be offset (in the Y-direction ofFIG.2B) from the conductive source vias266. In particular, two or more conductive source vias266may be formed in each source finger256, and the conductive source vias266that are formed in a particular source finger256may extend (at least generally) along a horizontal (X-direction) axis. Thus, the conductive source vias266included in each source finger256may define respective horizontal axes in the view ofFIG.2B, with line2C-2C inFIG.2Billustrating one such horizontal axis. As shown inFIG.2B, the conductive gate vias262and/or the conductive drain vias264may be positioned between these horizontal axes (as opposed to, for example, being aligned along these horizontal axes). In some cases, the conductive gate vias262and/or the conductive drain vias264may be positioned along the longitudinal axes defined by the respective drain fingers254. Offsetting the conductive gate vias262and the conductive drain vias264from the conductive source vias266may increase the distance between conductive vias,262,264,266, which can reduce the possibility that the wafer or die cracks due to mechanical weaknesses. This arrangement also reduces parasitic gate-to-source and/or parasitic source-to-drain coupling that may occur between the various vias262,264,266. Such parasitic coupling may lead to gain loss and/or instability. Referring toFIGS.2C and2D, the semiconductor layer structure230includes a plurality of semiconductor layers. In the depicted embodiment, a total of two semiconductor layers are shown, namely a channel layer234and a barrier layer236that is on a top side of the channel layer234. The semiconductor layer structure230may include additional semiconductor and/or non-semiconductor layers. For example, the semiconductor layer structure230may include a growth substrate232on which the other semiconductor layers are grown. The growth substrate232may comprise, for example, a 4H-SiC or 6H-SiC substrate. In other embodiments, the growth substrate may be comprise a different semiconductor material (e.g., silicon or a Group III nitride-based material, GaAs, ZnO, InP) or a non-semiconductor material (e.g., sapphire). SiC has a much closer crystal lattice match to Group III nitrides than does sapphire (Al2O3), which is a very common substrate material for Group III nitride devices. The closer lattice match of SiC may result in Group III nitride films of higher quality than those generally available on sapphire. SiC also has a very high thermal conductivity so that the total output power of Group III nitride devices on silicon carbide is, typically, not as limited by thermal dissipation of the substrate as in the case of the same devices formed on sapphire. Also, the availability of semi-insulating SiC substrates may provide for device isolation and reduced parasitic capacitance. Optional buffer, nucleation and/or transition layers (not shown) may be provided on the growth substrate232beneath the channel layer234. For example, an AlN buffer layer may be included to provide an appropriate crystal structure transition between a SiC growth substrate232and the remainder of the semiconductor layer structure230. Additionally, strain balancing transition layer(s) may also be provided as described, for example, in commonly assigned U.S. Patent Publication 2003/0102482A1, published Jun. 5, 2003, and entitled “Strain Balanced Nitride Heterojunction Transistors And Methods Of Fabricating Strain Balanced Nitride Heterojunction Transistors,” the disclosure of which is incorporated herein by reference as if set forth fully herein. In some embodiments, the channel layer234is a Group III nitride material, such as AlxGa1-xN where 0≤x<1, provided that the energy of the conduction band edge of the channel layer234is less than the energy of the conduction band edge of the barrier layer236at the interface between the channel and barrier layers234,236. In certain embodiments of the present invention, x=0, indicating that the channel layer234is gallium nitride (“GaN”). The channel layer234may also be other Group III nitrides such as InGaN, AlInGaN or the like. The channel layer234may be undoped or unintentionally doped and may be grown to a thickness of, for example, greater than about 20 Å. The channel layer234may also be a multi-layer structure, such as a superlattice or combinations of GaN, AlGaN or the like. The channel layer234may have a bandgap that is less than the bandgap of at least a portion of the barrier layer236, and the channel layer234may also have a larger electron affinity than the barrier layer236. In certain embodiments, the barrier layer236is AlN, AlInN, AlGaN or AlInGaN with a thickness of between about 0.1 nm and about 10 nm or more. In particular embodiments, the barrier layer236is thick enough and has a high enough Al composition and doping to induce a significant carrier concentration at the interface between the channel layer234and the barrier layer236. The barrier layer236may be a Group III nitride and may have a bandgap larger than that of the channel layer234and a smaller electron affinity than the channel layer234. In certain embodiments, the barrier layer236is undoped or doped with an n-type dopant to a concentration less than about 1019cm−3. In some embodiments of the present invention, the barrier layer236is AlxGa1-xN where 0<x<1. In particular embodiments, the aluminum concentration is about 25%. However, in other embodiments of the present invention, the barrier layer236comprises AlGaN with an aluminum concentration of between about 5% and about 100%. In specific embodiments of the present invention, the aluminum concentration is greater than about 10%. Due to the difference in bandgap between the barrier layer236and the channel layer234and piezoelectric effects at the interface between the barrier layer236and the channel layer234, a two dimensional electron gas (2DEG) is induced in the channel layer234at a junction between the channel layer234and the barrier layer236. The 2DEG acts as a highly conductive layer that allows conduction between the source region of each unit cell transistor216and its associated drain region, where the source region is the portion of the semiconductor layer structure230that is directly underneath the source finger256and the drain region is the portion of the semiconductor layer structure230that is directly underneath the corresponding drain finger254. An interlayer insulating layer238is formed over the gate fingers252, the drain fingers254, and the source fingers256. The interlayer insulating layer238may include a dielectric material, such as SiN, SiO2, etc. FIGS.2C-2Gillustrate the metal-plated gate vias262, metal-plated drain vias264, and metal-plated source vias266in more detail. As shown inFIGS.2C through2F, the metal-plated gate vias262, metal-plated drain vias264, and metal-plated source vias266may extend the entire way through the semiconductor layer structure230in order to physically and electrically connect the gate bus242to the gate terminal222, the drain bus244to the drain terminal224, and the source fingers256to the source terminal226. In some embodiments, the metal-plated gate vias262, metal-plated drain vias264, and metal-plated source vias266may all have the same shape and horizontal cross-section (i.e., a cross-section taken through the vias in a plane that is parallel to a major surface of the semiconductor layer structure230). For example, all of the vias262,264,266may be substantially cylindrical or oval vias having the same diameter, or may all be truncated fustoconical vias that have the same diameter when measured at the same height above the bottom surface214of the RF amplifier die210. Such an arrangement may allow all of the vias262,264,266to be readily formed in a single manufacturing step. In other embodiments, the metal-plated gate vias262and/or the metal-plated drain vias264may have a larger cross-sectional area as compared to the metal-plated source vias266. This technique may be used to further reduce the inherent inductance of the metal-plated gate vias262and/or the metal-plated drain vias264if necessary for certain applications. The metal-plated gate vias262, metal-plated drain vias264, and metal-plated source vias266may each be implemented by forming openings though the semiconductor layer structure (e.g., by anisotropic etching) and by then depositing metal-plating that coats the sidewalls of the openings. In some applications, the metal may completely fill the openings so that the metal-plated vias are metal-filled vias. However, in many applications, the RF transistor amplifier die210may operate over a wide temperature range (due to outdoor applications and/or the high levels of heat that may be generated within the RF transistor amplifier die during device operation), which may lead to high stress levels in the device due to the metal and semiconductor materials having significantly different coefficients of thermal expansion. In such cases, the center of the metal-plated vias262,264,266may be left open (i.e., air-filled) in order to reduce the amount of stress that occurs due to thermal cycling. The cross-sectional areas of the vias262,264,266may be selected, for example, based on heat dissipation considerations and/or a desired amount of series inductance. Whether a metal-plated via will dissipate more or less heat than the semiconductor material that the metal-plated via penetrates will depend upon a variety of considerations, including the thermal dissipation qualities of the semiconductor material and the metal used, the thickness of the metal plating, the cross-sectional area(s) of the vias, etc. Generally speaking, metals such as copper dissipate heat more efficiently than Group III nitride-based and silicon carbide semiconductor materials, but any central air-filled opening in the vias will dissipate heat less efficiently than the semiconductor materials. As shown inFIG.2G, the gate terminal222, the drain terminal224and the source terminal226may each comprise a metallization pattern on a lower surface of the semiconductor layer structure230. Gaps may be provided between the gate terminal222and the drain terminal224and between the drain terminal224and the source terminal226in order to electrically insulate the gate, drain and source terminals222,224,226from each other. These gaps may expose the growth substrate232when the RF transistor amplifier die210is viewed from the back side. In some embodiments, an insulating pattern (not shown) may be deposited in the gaps. The gate vias262, the drain vias264and the source vias266each physically and electrically connect to the respective gate terminal222, drain terminal224and source terminal226. FIG.3is a schematic top view of an example embodiment of the interconnection structure270included in the RF amplifier200ofFIGS.2A-2G. As discussed above, the interconnection structure270may comprise, for example, an RDL laminate structure or a multilayer printed circuit board. The gate pad272, drain pad274and source pad276are implemented on an upper surface of the interconnection structure270. Each of these pads272,274,276may comprise a respective metal pattern (e.g., a copper pattern). The gate pad272, drain pad274and source pad276may have the same or similar sizes and shapes as the respective gate terminal222, drain terminal224and source terminal226on the RF amplifier die210. A plurality of metal filled vias290(or alternatively, a solid conductive slug) may be provided underneath the source pad276that extend through the interconnection structure270. The metal-filled vias (or conductive slug)290may act as a heat sink that carries heat that is generated in the RF amplifier die210and passed to the interconnection structure270to the bottom side of the interconnection structure270where it is vented into the ambient environment or passed to a heat sink in an underlying structure such as a printed circuit board. As is also shown inFIG.3, in some embodiments, additional metal-filled vias290may be provided under the gate pad272and/or under the source pad276to provide additional thermal dissipation. As is further shown inFIG.3, a plurality of additional components281may be mounted on the interconnection structure270. These components281may include, for example, passive RF components such as integrated passive devices or printed circuit boards that include resistors, capacitors and/or inductors. These passive components may form input and/or output matching circuits that are used to (1) match the impedance of the input and/or output of the RF transistor amplifier die210to the impedance at the fundamental frequency of the respective input and output RF transmission lines or (2) terminate harmonics of the fundamental frequency that may be present at either the input or output of the RF transistor amplifier die210. Some of the matching circuitry may also be implemented in the interconnection structure270. For example, the interconnection structure270may include meandered or spiral trace patterns (not shown) that implement inductors that are included in the input and/or output matching circuits. Other RF circuitry may also be mounted on the interconnection structure270such as transmit/receive switches, circulators, filters or the like. One advantage of the having the gate terminal222, the drain terminal224and the source terminal226all on the same side of the RF amplifier die210is that it may enable more wafer level processing, which may lead to more efficient manufacturing. As shown inFIG.4A, in many applications, a plurality of RF transistor amplifier dies210are fabricated from a single semiconductor wafer201. The semiconductor wafer201may comprise, for example, a silicon carbide wafer, and a plurality of gallium nitride based epitaxial layers may be grown on the silicon carbide wafer201using semiconductor epitaxial growth techniques. Then, conventional semiconductor processing techniques such as metal and insulating material deposition, photolithography, masking and/or etching may be performed to form the bottom side and top side metallization structures220,240and the conductive gate vias262, the conductive drain vias264and the conductive source vias266in order to form a plurality of RF transistor amplifier dies210in the silicon carbide wafer210(with a portion of the silicon carbide wafer201forming the growth substrate232of each individual RF transistor amplifier die210). Ultimately, the wafer201is cut along horizontal and vertical “scribe” lines (not shown) to singulate the individual RF transistor amplifier dies210. It should be noted thatFIG.4Ais an diagram that is provided for illustrative purposes and that typically a much larger number of RF transistor die210are formed on a wafer and the RF transistor die210are typically located in more dense fashion. FIG.4Bis a schematic cross-sectional view of one of the RF amplifier dies210that is included in the wafer201ofFIG.4A. As shown inFIG.4B, contacts280(e.g., solder bumps) are affixed to each of the gate terminal222, the drain terminal224and the source terminal226. These contacts280may be used to mechanically and electrically attach the RF transistor amplifier die210to an interconnection structure (not shown) such as interconnection structure270. While not shown inFIGS.4A and4B, the contacts280may be applied as part of the wafer level processing steps, (i.e., before the semiconductor wafer201is diced into a plurality of individual RF transistor amplifier dies210. Such wafer level processing is faster and more efficient than applying contacts280to each individual RF transistor amplifier die210. Additionally, because the gate terminal222and the drain terminal224may be electrically connected to corresponding gate and drain pads272,274on the interconnection structure270(seeFIGS.2A and3) in the same processing step that is used to connect the source terminal226to the source pad276on the interconnection structure270, all of the electrical connections to the RF transistor amplifier die210can potentially be established in a single processing step. In contrast, when conventional RF transistor amplifier die are employed (e.g., RF transistor amplifier die110ofFIGS.1A-1B), additional, time consuming wire bonding processes are employed to make the electrical connections to the gate and drain terminals142,144. Eliminating these processing steps may significantly simplify the manufacturing process. As described above, provision of the conductive gate vias262and the conductive drain vias264results in all three of the gate, drain and source terminals222,224,226for the RF transistor amplifier die210being on the same surface of the die, and hence in the same plane. This makes it possible to employ a variety of different types of wafer level packaging techniques such as, for example, various fan-in, fan-out and interposer topologies. The RF transistor amplifier dies according to embodiments of the present invention may be mounted directly on interconnection structures or on intervening structures such as RDL laminate structures or interposers (which may be a custom RDL laminate structure) using contacts such as, for example, conductive bumps or conductive die attach materials. When the RF transistor amplifier dies according to embodiments of the present invention are mounted on, for example, RDL laminate structures or interposers, contacts may be pre-mounted on the bottom surfaces of the RDL laminate structures/interposers which may allow end users to readily mount the RF amplifier dies on or other structures. Moreover, as noted above, the provision of the conductive gate vias262and the conductive drain vias264reduces the variation in the electrical path lengths, which improves performance, and may reduce or eliminate the need for costly and time-consuming wire bonding processes. The reduced or eliminated need for wire bonds may also allow for reduced die size in some applications (where the sizes of the wire bond pads drive die size), and hence the RF transistor amplifier dies according to embodiments of the present invention may also exhibit increased integration density. Thus, the RF amplifier die according to embodiments of the present invention may exhibit improved product assembly consistency, higher yields, increased product integration, reduced cost and improved RF performance, especially for products operating at high frequencies such as millimeter wave frequencies. The techniques disclosed herein may be particularly beneficial in higher frequency applications as the inductance required in the matching circuits may be much lower in such applications, and hence the use of traditional bond wires may inject too much inductance. Additionally, the tolerances in the bond wire lengths may have a larger impact at higher frequencies, and in high frequency applications (particularly if lower power) the size of the bond pads may drive the size of the die. In some embodiments, any of the RF transistor amplifier dies disclosed herein may be configured to operate at frequencies greater than 1 GHz. In other embodiments, these RF transistor amplifier dies may be configured to operate at frequencies greater than 2.5 GHz. In still other embodiments, these RF transistor amplifier dies may be configured to operate at frequencies greater than 3.1 GHz. In yet additional embodiments, these RF transistor amplifier dies may be configured to operate at frequencies greater than 5 GHz. In some embodiments, these RF transistor amplifier dies may be configured to operate in at least one of the 2.5-2.7 GHz, 3.4-4.2 GHz or 5.1-5.8 GHz frequency bands or sub-portions thereof. FIGS.4C through4Eillustrate example packaged RF transistor amplifiers that each include RF transistor amplifier dies according to embodiments of the present invention.FIGS.5A-5Cthen illustrate how the planarized terminal configuration of the RF transistor amplifier dies according to embodiments of the present invention also allow the RF transistor amplifier dies to be used in a variety of different wafer level packaging topologies. FIG.4Cis a schematic cross-sectional view of a packaged RF transistor amplifier300that includes the RF transistor amplifier die210ofFIG.4Bin an open cavity package. As shown inFIG.4C, the open-cavity package310includes a base320, such as a metal flange, and an upper housing330which may include, for example, sidewalls332and a lid334. In an example embodiment, the base320may be a multilayer copper/molybdenum/copper metal flange that comprises a core molybdenum layer with copper cladding layers on either major surface thereof. The ceramic sidewalls332and lid334may be formed of, for example, Al2O3. The ceramic lid334may be glued to the ceramic sidewalls332using an epoxy glue. The ceramic sidewalls332may be attached to the metal base320via braising. The RF transistor amplifier die210may, for example, be mounted on an interconnection structure270using, for example, the conductive contacts, such as bumps280, shown inFIG.4B, and the interconnection structure270is mounted on the base320using, for example, a conductive die attach material. The base320may dissipate heat carried through the heat dissipation structures290in interconnection structure270outside of ceramic package310. Additional components350,360are mounted on the interconnection structure270. These additional components may include, for example, input matching components350and output matching components360that are used to impedance match at the fundamental frequency and/or to terminate intermodulation products to ground. As discussed above, these matching components350,360may be passive RF components that include resistors, capacitors and/or inductors that are implemented (at least partially) in integrated passive devices or printed circuit boards, for example. Conductive leads340extend through the housing310to allow the RF transistor amplifier300to be connected to external devices/circuits/power sources. In the depicted embodiment, wire bonds370are used to connect the conductive leads340to passive RF components350,360on the interconnection structure270. It will be appreciated, however, that the wire bonds370may be omitted in other embodiments and different electrical connections ay be used. An RF signal input to the RF transistor amplifier300on a first lead340-1may be passed through the wire bond370-1to input matching circuits350and from there to a gate terminal222(seeFIG.4B) of the RF transistor amplifier die210, and the amplified output RF signal may be passed from the drain terminal224of the RF transistor amplifier die210to the output matching circuits360and from there to the bond wire370-2where the RF signal is output through lead340-2. FIG.4Dis a schematic cross-sectional view of a packaged RF transistor amplifier400that includes the RF transistor amplifier die210ofFIG.4Bin an overmold plastic package. As shown inFIG.4D, the packaged RF transistor amplifier400includes a base420, such as a metal heat sink that is part of a lead frame or metal slug, that is at least partially surrounded by a plastic overmold410. The RF transistor amplifier die210is mounted on an interconnection structure270using, for example, the conductive bumps280shown inFIG.4B, and the interconnection structure270is mounted on the base420. The base420may comprise, for example, a metal base that may dissipate heat carried through the heat dissipation structures290in interconnection structure270. Additional components450,460are mounted on the interconnection structure270. These additional components may include, for example, input matching components450and output matching components460that are used to impedance match at the fundamental frequency and/or to terminate intermodulation products to ground. As discussed above, these matching components may be passive RF components that include resistors, capacitors and/or inductors that are implemented (at least partially) in integrated passive devices or printed circuit boards, for example. Conductive leads440extend through the plastic overmold410to allow the RF transistor amplifier400to be connected to external devices/circuits/power sources. In the depicted embodiment, wire bonds470are used to connect the conductive leads440to the passive RF components450,460on the interconnection structure270, although the wire bonds470may be omitted in other embodiments. FIG.4Eis a schematic cross-sectional view of a packaged RF transistor amplifier300A that includes the RF transistor amplifier die ofFIG.4Bin a printed circuit board based package. The packaged RF transistor amplifier300A is very similar to the packaged RF transistor amplifier300discussed above with reference toFIG.4C, except that the leads340-1,340-2of packaged RF transistor amplifier300are replaced with a printed circuit board322that includes traces342-1,342-2that act as the input and output leads. The printed circuit board322may be attached to the metal base320via, for example, a conductive glue. The printed circuit board322includes a central opening and the interconnection structure270is mounted within this opening on the base (e.g., metal flange)320. The RF transistor die210and the matching networks350-1,350-2,360-1,360-2are mounted on the interconnection structure270. It will be appreciated that any of the RF transistor amplifiers according to embodiments of the present invention that are discussed herein may be mounted in packages such as the open cavity and overmold packages shown inFIGS.4C through4E. Thus, the RF transistor die210and interconnection structures270shown inFIGS.4C-4Emay be replaced with the RF transistor die and interconnection structures according to any of the embodiments of the present invention that are discussed herein to provide many further embodiments of packaged RF transistor amplifiers. Depending on the embodiment, the packaged RF transistor amplifier can include a monolithic microwave integrated circuit (MMIC) as the RF transistor amplifier die where the RF transistor amplifier die incorporates multiple discrete circuits in a single integrated die. Additionally and/or alternatively, the package can comprise multiple RF transistor amplifier die in a path that are connected in series to form a multiple stage RF transistor amplifier and/or multiple RF transistor amplifier die that are disposed in multiple paths (e.g., in parallel) to form an RF transistor amplifier with multiple transistor amplifier die and multiple paths, such as in a Doherty amplifier configuration. In some embodiments, the packaged RF transistor amplifier may include RF transistor amplifier die according to embodiments of the present invention that have conducive gate vias and/or conductive drain vias that provide electrical connections to a back side interconnection structure as well as traditional RF transistor amplifier die such as the RF transistor die110ofFIG.1Athat have gate and drain terminals that are connected to other structures via wire bonds. FIG.5Ais a schematic cross-sectional view of an RF transistor amplifier500according to embodiments of the present invention that includes the RF transistor amplifier die210mounted on a RDL laminate structure510in a fan-in topology. As is known in the art, integrated circuit chips may be mounted on and electrically connected to various underlying substrates using contacts such as conductive bumps or other conductive attachment mechanisms. The contacts may provide electrical connections between terminals on the integrated circuit chip and corresponding electrical connection points (e.g., conductive pads) on the substrate. The substrate may be used to rearrange the configuration of the gate, drain and source terminals to, for example, align with terminals on another substrate. As shown inFIG.5A, the RF transistor amplifier die210may be mounted on an RDL laminate structure510. The RDL laminae structure510may include an upper gate terminal522, an upper drain terminal524and an upper source terminal526that may be aligned with the respective gate terminal222, drain terminal224and source terminal226on RF transistor amplifier die210so that the gate terminal222, drain terminal224and source terminal226may be physically and electrically connected to the respective upper gate terminal522, upper drain terminal524and upper source terminal526using, for example, conductive epoxies or bumps (not shown). The RDL laminate structure510further includes a lower gate terminal532a lower drain terminal534and a lower source terminal536. As shown inFIG.5A, one or more conductive gate vias542, conductive drain vias544and conductive source vias546are provided that electrically connect the upper gate terminal522to the lower gate terminal532, the upper drain terminal524to the lower drain terminal534, and the upper source terminal526to the lower source terminal536. The conductive gate vias542and the conductive drain vias544are located inwardly on the bottom surface of the RF transistor amplifier die210of the respective gate terminal222and drain terminal224. Conductive bumps280are attached to the lower gate terminal532the lower drain terminal534and the lower source terminal536for attaching the RF transistor amplifier500to another substrate, such as a customer printed circuit board. The RF transistor amplifier500has a fan-in topology where the RDL laminate structure510relocates the electrical connections for the gate, drain and source (here conductive bumps280) generally inwardly towards the center of the bottom surface of the RF transistor amplifier die210. Since the conductive bumps280are all within the “footprint” of the RF transistor amplifier die210, the conductive bumps280may be applied during wafer level processing to the bottom side of the wafer201shown inFIG.4A, and the wafer201may then be diced after the conductive bumps280have been applied into individual RF transistor amplifier die210. Typically, the individual RF transistor amplifier die210are mounted on a large RDL laminate structure (or other interconnection structure), which is later diced to provide a plurality of the RF transistor amplifiers500ofFIG.5A. It will be appreciated, however, that in other embodiments an RDL laminate structure could be bonded to the wafer201and the wafer201could thereafter be diced to provide a plurality of the RF transistor amplifiers500ofFIG.5A. FIG.5Bis a schematic cross-sectional view of an RF transistor amplifier500′ according to embodiments of the present invention that includes the RF transistor amplifier die210mounted on a redistribution layer substrate510′ in a fan-out topology. The RF transistor amplifier500′ is very similar to the RF transistor amplifier500discussed above, except that the RDL laminate structure510′ included therein has a fan-out topology in which the lower gate terminal532and the lower drain terminal534are located outwardly (when the RF transistor amplifier die210is viewed from below) of the respective gate terminal222and drain terminal224. Conductive bumps (or other contacts)280are attached to the lower gate terminal532the lower drain terminal534and the lower source terminal536for attaching the RF transistor amplifier500′ to another substrate, such as a customer printed circuit board. FIG.5Cis a schematic cross-sectional view of an RF transistor amplifier500″ according to embodiments of the present invention that includes an RF transistor amplifier die210that is mounted on a custom interposer510″ in a fan-out topology. Interposers may be custom RDL laminate structure designs that allow increased flexibility with respect to the location of the contacts280. Additionally, in some cases passive circuits such as capacitors or inductors (not shown) may be implemented within the interposer510″, reducing the need for additional components280(seeFIG.3). As discussed above, Group III nitride-based RF transistor amplifiers often include one or more of an input impedance matching network, an input harmonic termination circuit, an output harmonic termination circuit, and an output impedance matching network. Each of these matching circuits may include one or more capacitors and/or inductors. In conventional RF transistor amplifiers, the inductances are often at least partly implemented using bond wires that form connections between the RF transistor amplifier die, various passive RF components and input/output leads of the amplifier. As applications move to higher frequencies, the amount of inductance needed to properly impedance match at the fundamental frequency and/or to terminate certain harmonics such as the second and/or third order harmonics typically decreases. In some applications, even if very short, thick bond wires are used, the inductance of the bond wires may exceed the optimum amount of inductance required by one or more of the matching circuits. If the inductance is larger than the optimum amount of inductance for an impedance matching circuit, then the return loss of the RF transistor amplifier may be increased, and/or the operating bandwidth may be reduced. If the inductance is larger than the optimum amount of inductance for a harmonic termination circuit, then less reduction in the harmonic at issue may be achieved, which may degrade the efficiency, power and/or gain performance of the RF transistor amplifier, and result in increased levels of passive intermodulation distortion that may degrade other aspects of a communication system in which the RF transistor amplifier is used. The Group III nitride-based RF transistor amplifiers according to embodiments of the present invention may avoid the above-discussed problem of having more series inductance than the amount of series inductance that provides for optimum impedance matching. In particular, the conductive gate and drain vias that are used in the RF transistor amplifiers according to embodiments of the present invention may have lengths of less than 8 mils, and often less than 5 mils, less than 4 mils or even less than 3 mils in example embodiments. In contrast, the gate and drain bond wires that are used in the conventional RF transistor amplifiers typically are at least 20 mils in length, with lengths of mils or more being common. As such, the inductance injected by the gate and drain vias may be a small fraction of the inductance injected by comparable gate and drain bond wires (e.g., perhaps on the order of 15-20% the inductance), which may ensure that the inductance is less than or equal to the optimum amount of inductance required by the various matching circuits of the Group III nitride-based RF transistor amplifier. Any additional inductance required to obtain the optimum amount of inductances for the matching networks may be added using inductor chips and/or inductive traces (or other structures) that are mounted on or implemented in the interconnection structure, in RF passive components or the like. Mounting the gate and drain terminals on the bottom side of the device may also reduce process variation during high volume manufacturing, as the ball bonders that are used to solder the bond wires to the gate and drain terminals on RF transistor amplifier die typically have a tolerance of +/−1 mil, resulting in potentially as much as 4 mils of variation in the length of each bond wire. The amount of inductance associated with such variation in the lengths of the bond wires can be significant, particularly at higher frequencies, and can degrade the performance of the impedance matching circuits, and hence the performance of the RF transistor amplifier. Additionally, connecting the gate and drain terminals to corresponding gate and drain pads on the interconnection structure through a surface mount process using conductive bumps, die attach material or the like may allow for the use of smaller gate and drain terminals than could be used when bond wire connections are required, and hence the RF transistor amplifier dies according to embodiments of the present invention may be smaller in applications where the gate and drain terminal sizes impacted the size of the die. Additionally, using ball bonding techniques as opposed to wire bonds may reduce manufacturing costs. Another advantage provided by the conductive gate and drain vias that are included in the RF transistor amplifiers according to embodiments of the present invention is that more flexibility is provided for implementing the matching networks, since connections may be made to both the tops and bottoms of the conductive gate and drain vias. This feature of the RF transistor amplifiers according to embodiments of the present invention is schematically shown in the circuit diagram ofFIG.6. As shown inFIG.6, the RF transistor amplifier200has a pair of RF inputs, namely a first “top” RF input that connects directly to the gate (i.e., an upper gate terminal that connects directly to the gate bus) of the RF transistor amplifier die210, and a “bottom” RF input that connects to the bottom of the conductive gate vias262. These RF inputs are electrically connected to each other through an inductance Lgate-via which represents the inherent inductance of the conductive gate vias262. Likewise, the RF transistor amplifier200has a pair of RF outputs, namely a first “top” RF output that connects directly to the drain (i.e., an upper drain terminal that connects directly to the drain bus) of the RF transistor amplifier die210, and a “bottom” RF output that connects to the bottom of the conductive drain vias264. These RF outputs are electrically connected to each other through an inductance Ldrain-viawhich represents the inherent inductance of the conductive drain vias264. This arrangement may provide increased flexibility to implement certain matching topologies. For example,FIG.7Ais a circuit diagram of a conventional RF transistor amplifier600that has an input series impedance matching circuit, an input harmonic termination circuit for termination of the harmonic frequencies (e.g., second harmonics or “2f0”), and an output shunt-L impedance matching circuit. The input and output series transmission lines610-1,610-2may be selected to provide appropriate impedance transformation between the RF transistor amplifier die110and the RF input (e.g., a gate lead) and output (e.g., a drain lead). These series transmission lines610-1,610-2can be treated as an extension of the transmission line matching network on an underlying substrate (not shown) such as, for example, a customer printed circuit board, and electrical widths can be selected or configured to achieve the desired characteristic impedance for the impedance matching. In conventional designs, these matching circuits are implemented through bond wires (for the inductances) and RF passive components (for the capacitances). This arrangement may result in parasitic coupling between the input and output side bond wires that can compromise RF performance and, as described above, at higher frequencies the bond wires may inject too much inductance which can compromise the impedance match and/or the harmonic termination. FIGS.7B and7Cillustrate two possible implementations of the matching topology shown inFIG.7Ausing RF transistor amplifier die according to embodiments of the present invention. As shown inFIG.7B, the RF input and the input impedance matching network are connected to the upper gate terminal, while the input harmonic termination circuit may be coupled to the lower gate terminal. On the output side, the output impedance matching network is connected to the lower drain terminal as a shunt circuit, while the RF output is connected to the upper drain terminal. As shown inFIG.7C, in an alternative embodiment, the RF input and the input impedance matching network are connected to the lower gate terminal, while the input harmonic termination circuit may be coupled to the upper gate terminal as a shunt circuit. On the output side, the output impedance matching network is connected to the upper drain terminal as a shunt circuit, while the RF output is connected to the lower drain terminal. FIGS.8A and8Billustrate an RF transistor amplifier700according to further embodiments of the present invention. In particular,FIG.8Ais a schematic cross-sectional view of the RF transistor amplifier700that illustrates the circuits components included therein and the electrical interconnections therebetween, whileFIG.8Bis a circuit diagram of the RF transistor amplifier700. As shown inFIG.8A, the RF transistor amplifier700includes an RF transistor die, which may be implemented, for example, using the RF transistor amplifier die210that is described above, or any of the other RF transistor amplifier die according to embodiments of the present invention. The RF transistor amplifier die210is mounted on an RDL laminate structure710, although other mounting structures may be used in other embodiments such as multi-layer printed circuit boards or integrated passive devices or “IPDs” that comprise capacitors (and perhaps other passive devices such as inductors) formed on thin film substrates such as silicon, alumina, or glass using semiconductor processing techniques. The RDL laminate structure710includes conductive regions712and dielectric regions714. A plurality of RF passive components720-1through720-4are mounted on the RDL laminate structure710. Interconnections amongst and between the RF transistor amplifier die210and the RF passive components720-1through720-4are made using bond wires730and through electrical connections in the RDL laminate structure710. In particular, the RF input740and the RF output742are formed as conductive structures in the RDL laminate structure710. The RF input740may be connected to a first external circuit and the RF output742may be connected to a second external circuit. Focusing first on the input (left) side ofFIG.8A, a first bond wire730-1connects the RF input740to an upper terminal of an RF passive component720-1that comprises a shunt capacitor to ground. The RF passive component720-1may be implemented as, for example, a capacitor IPD or a surface mount capacitor chip. A lower terminal of RF passive component720-1is connected to a grounded region in the RDL laminate structure710. The first bond wire730-1implements the inductance “Input_L2” shown inFIG.8B, and the RF passive component720-1implements the shunt capacitance “Input_C1” shown inFIG.8B. A second bond wire730-2connects the upper terminal of RF passive component720-1to the upper gate terminal243of RF transistor amplifier die210. The second bond wire730-2implements the series inductance “Input_L1” shown inFIG.8B. The inherent inductance of the conductive gate vias262in RF transistor amplifier die210is shown inFIG.8Bas the inductance “Lvia_G.” The lower gate terminal222of RF transistor amplifier die210, which is connected to the lower ends of the conductive gate vias262, is connected via a contact280to a conductive trace716-1on RDL laminate structure710. Conductive trace716-1is connected to RF passive component720-2, which may include capacitors and/or inductors. RF passive component720-2may be implemented, for example, as an IPD or as a surface mount chip. The combination of conductive trace716-1and RF passive component720-2may implement the series C-L circuit “Input_2f” shown inFIG.8B. Focusing next on the output (right) side ofFIG.8A, a third bond wire730-3connects the upper drain terminal253of RF transistor amplifier die210to an upper terminal of RF passive component720-4, which forms a shunt capacitor to ground. RF passive component720-4may be implemented as, for example, a capacitor IPD or as a surface mount capacitor chip. A lower terminal of RF passive component720-4is connected to a grounded region in the RDL laminate structure710via contacts280. The third bond wire730-3implements the series inductance “Output_L1” shown inFIG.8B, and the RF passive component720-4implements the shunt capacitance “Output_C1” shown inFIG.8B. A fourth bond wire730-4connects the upper terminal of RF passive component720-4to the RF output742in the RDL laminate structure710. The fourth bond wire730-4implements the series inductance “Output_L2” shown inFIG.8B. The inherent inductance of the conductive drain vias264in RF transistor amplifier die210is shown inFIG.8Bas the inductance “Lvia_D.” The lower drain terminal224of RF transistor amplifier die210, which is connected to the lower ends of the conductive drain vias264, is connected via a contact280to a conductive trace716-2on RDL laminate structure710. Conductive trace716-2is connected to RF passive component720-3, which may include capacitors and/or inductors. RF passive component720-3may be implemented, for example, as IPD or a surface mount chip. The combination of conductive trace716-2and RF passive component720-3may implement the series C-L circuit “Output_f0” shown inFIG.8B. As can be seen fromFIGS.8A-8B, in the RF transistor amplifier700, the RF input and RF output are routed through the upper gate and drain terminals, respectively, and the input harmonic termination circuit and the output impedance matching circuit are routed through the lower gate and drain terminals, respectively.FIGS.9A and9Bare a schematic cross-sectional view and a circuit diagram, respectively, of an RF transistor amplifier800in which the RF input and RF output are routed through the lower gate and drain terminals, respectively, and the input harmonic termination circuit and the output impedance matching circuit are routed through the respective upper gate and drain terminals. As shown inFIG.9A, the RF transistor amplifier800includes an RF transistor amplifier die210(which may be implemented as any of the other RF transistor amplifier die according to embodiments of the present invention) that is mounted on an RDL laminate structure810(which may alternatively be another mounting structure such as a multi-layer printed circuit boards or IPD). The RDL laminate structure810includes conductive regions812and dielectric regions814. A pair of RF passive components820-1,820-2are mounted on the RDL laminate structure810. An RF input840is implemented as a conductive structure in the RDL laminate structure810. This RF input840may be connected to a first external circuit. The RF input840is connected by a contact280to the lower gate terminal222of RF transistor amplifier die210, and is connected to the gate of RF transistor amplifier die210through the conductive gate vias262. The inherent inductance of the conductive gate vias262in RF transistor amplifier die210is shown inFIG.9Bas the inductance “Lvia_G.” A first bond wire830-1connects the upper gate terminal243of RF transistor amplifier die210to an upper terminal of RF passive component820-1. RF passive component820-1may comprise a lump capacitance and may be implemented as, for example, a capacitor IPD or a surface mount capacitor chip. The lower terminal of RF passive component820-1is connected to a grounded region in the RDL laminate structure810. The first bond wire830-1implements the inductance included in circuit “Input_2f” shown inFIG.9B, and RF passive component820-1implements the capacitance included in circuit “Input_2f.” A second bond wire830-2connects the upper drain terminal253of RF transistor amplifier die210to an upper terminal of RF passive component820-2, which forms a shunt capacitor to ground. RF passive component820-2may be implemented as, for example, a capacitor IPD or as a surface mount capacitor chip. A lower terminal of RF passive component820-2is connected to a grounded region in the RDL laminate structure810via contacts280. The second bond wire830-2and a lump capacitance implemented in RF passive component820-2together implement the series L-C circuit labelled “Output_f0” inFIG.9B. The RF output842is implemented as a conductive structure in the RDL laminate structure810, and may be connected to a second external circuit. The RF output842is connected by a contact280to the lower drain terminal224of RF transistor amplifier die210, and is connected to the drain of RF transistor amplifier die210through the conductive drain vias264. The inherent inductance of the conductive drain vias264in RF transistor amplifier die210is shown inFIG.9Bas the inductance “Lvia_D.” FIGS.10A and10Bare schematic cross-sectional views that illustrate the structure of the top metallization of two RF transistor amplifier dies according to further embodiments of the present invention. As shown inFIG.10A, an RF transistor amplifier die210′ according to embodiments of the present invention is very similar to RF transistor amplifier die210, except that RF transistor amplifier die310does not include drain vias264, and the drain terminal in RF transistor amplifier die210′ may be implemented on the top side of the semiconductor layer structure230and connected to, for example, a drain lead via bond wire(s), in the manner discussed above with reference to the RF transistor amplifier100ofFIGS.1A-1B. The RF transistor amplifier die210′ may be used, for example, when bond wires do not provide too much inductance for any of the output matching networks. The remainder of RF transistor amplifier die210′ may be identical to RF transistor amplifier210, and hence further description thereof will be omitted. As shown inFIG.10B, an RF transistor amplifier die210″ according to embodiments of the present invention is also very similar to RF transistor amplifier die210, except that RF transistor amplifier die210″ does not include gate vias262, and the gate terminal in RF transistor amplifier die210″ may be implemented on the top side of the semiconductor layer structure230and connected to, for example, a gate lead via bond wire(s), in the manner discussed above with reference to the RF transistor amplifier100ofFIGS.1A-1B. The RF transistor amplifier die210″ may be used, for example, when bond wires do not provide too much inductance for any of the input matching networks. The remainder of RF transistor amplifier die210″ may be identical to RF transistor amplifier210, and hence further description thereof will be omitted. RF transistor amplifier dies210′,210″ may be used in place of RF transistor amplifier die210in any of the above-described embodiments of the present invention. As described above with reference toFIGS.8A-9B, the RF transistor amplifiers according to embodiments of the present invention may include RF passive components in the form of IPDs that are mounted on an RDL laminate structure or other substrate. In the embodiments ofFIGS.8A-9B, the ground connections to the RF passive components720,820are formed using contacts280, while the other connections to the IPDs are formed using bond wires. Pursuant to further embodiments of the present invention, all of the electrical connections to the RF passive components may be formed using conductive bumps or other electrical connections other than wire bonding connections. This may further simplify the manufacturing operations, allow for smaller device footprints (since large wire bond pads are no longer required), and remove some of the RF performance issues that may arise when wire bond connections are used such as variation in the inductance (due to variation in the lengths of the bond wires), parasitic inductances, and the problem of too much inductance, particularly in high frequency applications. FIG.11Ais a schematic cross-sectional view of one such RF transistor amplifier900according to embodiments of the present invention. As shown inFIG.11A, the RF transistor amplifier900includes an RF transistor amplifier die210(or any of the other RF transistor amplifier dies according to embodiments of the present invention) and an RF passive component920-1, which are both mounted on an RDL laminate structure910. In the embodiment ofFIG.11A, the RF transistor amplifier900includes an input impedance matching circuit and an input harmonic termination circuit, but does not include any output matching circuits. The input matching circuits are primarily implemented in RF passive component920-1. The RDL laminate structure910includes a plurality of conductive traces912and conductive vias914that are formed within a dielectric base916. The conductive traces912and conductive vias914are used to electrically connect various terminals on the RF transistor amplifier die210and the RF passive component920-1. The RDL laminate structure910further includes electrical connections to external circuits including connections to a gate lead940, a drain lead942and a source connection944. The source connection944may be connected to electrical ground in some embodiments. The RDL laminate structure910further includes a metal slug946(or, alternatively, a dense array of metal-filled or mostly filled, e.g., at least 75% filled or at least 85% filled vias such as copper-filled vias) that dissipates heat generated in the RF transistor amplifier die210to outside a package (not shown) of the RF transistor amplifier900. The electrical connections to the RF transistor amplifier die210are made to the back side of the die210at the lower ends of the conductive gate, drain and source vias262,264,266. The RF transistor amplifier die210may be directly attached to the RDL laminate structure910using typical die attach techniques such as eutectic materials, precoats (e.g., a gold-tin precoat), solder pre-forms, sintering (e.g., Ag-sintering) and the like. The RF passive component920-1, which may be, for example, an IPD, is flip-chip attached to the RDL laminate structure910. The RF passive component920-1may have a plurality of terminals on an “upper” side thereof, and a plurality of contacts such as conductive bumps280may be pre-attached to there terminals. The RF passive component920-1may then be turned upside-down and the conductive bumps280may be mounted on corresponding conductive pads on the RDL laminate structure910to physically and electrically attach the RF passive component920-1to the RDL laminate structure. The RF passive component920-1may include one or more capacitors and/or one or more inductors that may be used to implement at least a potion of the input matching networks. In the embodiment shown inFIG.11A, the RF passive comment920-1includes a pair of capacitors922-1,922-2and a pair of inductors924-1,924-2, which are illustrated schematically inFIG.11A. The inductors924may be implemented, for example, as conductive traces which may be narrowed, elongated, spiraled or the like so that they will generate a desired amount of inductance. The amount of inductance generated by each such conductive trace may be carefully controlled, unlike the inductance generated by the aforementioned wire bonding processes which may have variation in wire lengths as great as 4 mils. As shown inFIG.11A, the gate lead940on RDL laminate structure910is connected by a conductive via to a conductive pad912on the upper side of RDL laminate structure910. A first contact280electrically connects the conductive pad912to a first terminal926-1of RF passive component920-1. The first terminal926-1is electrically connected to a first electrode of the first capacitor922-1. The second electrode of the first capacitor922-1may be connected a second terminal926-2of RF passive component920-1, which in turn is connected to a corresponding pad on the RDL laminate structure910by a second contact280. The second contact280may be electrically connected to a source connection on the RDL laminate structure910, which may be connected to electrical ground. The first electrode of the first capacitor922-1is connected to a first electrode of the second capacitor922-2by a first inductive trace segment924-1. The second electrode of the second capacitor922-2may be connected a third terminal926-3of RF passive component920-1, which in turn is connected to a corresponding pad on the RDL laminate structure910by a third contact280. The third contact280may be electrically connected to the source connection on the RDL laminate structure910. The first electrode of the second capacitor922-2is connected by a second inductive trace segment924-2to a fourth terminal926-4of RF passive component920-1, which in turn is connected to a corresponding pad on the RDL laminate structure910by a fourth contact280. The gate terminal222on the RF transistor amplifier die210is connected via a fifth contact280to the same pad912on RDL laminate structure910, so that RF signals input at the gate lead940on RDL laminate structure910may be passed through RF passive component920-1to the gate of RF transistor amplifier die210. The drain terminal224of RF transistor amplifier die210is connected to the drain lead942on RDL laminate structure910by contact280and a conductive via in RDL laminate structure910. FIG.11Bis a schematic cross-sectional view of an RF transistor amplifier900′ according to further embodiments of the present invention. RF transistor amplifier900′ is similar to RF transistor amplifier900ofFIG.11A, but further includes an output matching network. The description of RF transistor amplifier900′ will focus on the output matching network as the remainder of RF transistor amplifier900′ is identical to RF transistor amplifier900. As shown inFIG.11B, RF transistor amplifier900′ includes a second RF passive component920-2that is electrically interposed between the drain terminal224of RF transistor amplifier die210and drain lead942on RDL laminate structure910. RF passive component920-2may also be, for example, an IPD that is flip-chip attached to the RDL laminate structure910. RF passive component920-2may have a plurality of terminals on an “upper” side thereof, and a plurality of contacts280may be pre-attached to the terminals. In the depicted embodiment, RF passive component920-2includes a shunt L-C network including capacitor922-3and inductor924-3for impedance matching and a series transmission line connects the drain terminal224of RF transistor amplifier die210to the drain lead942of RDL laminate structure910. The impedance of the series transmission line can be adjusted by, for example, adjusting the width (or thickness) of the conductive trace to further enhance the impedance match at the output of RF transistor amplifier900′. FIG.11Cis a schematic circuit diagram of RF transistor amplifier900′ ofFIG.11B. InFIG.11B, the circuit elements included in RF transistor amplifier die210, RF passive components920-1,920-2and RDL laminate structure910are shown. FIG.11Dis a schematic top view of an RF transistor amplifier900″ that is similar to RF transistor amplifier900′ ofFIGS.11B-11C. As shown inFIG.11D, the RF transistor amplifier900″ includes the RDL laminate structure910, the RF transistor amplifier die210and the RF passive components920-1,920-2of RF transistor amplifier900′. As these components and the electrical connections therebetween have already been described above, further description thereof will be omitted. RF transistor amplifier900″ further includes two additional RF passive components920-3,920-4in the form of high density (i.e., high capacitance) capacitor chips. RF passive component920-3is inductively connected to the capacitors922in RF passive component920-1, and RF passive component920-4is inductively connected to the capacitors922in RF passive component920-2. The capacitances in RF passive components920-3,920-4may improve the video bandwidth performance of RF transistor amplifier900″ as compared to RF transistor amplifier900′. A minimum amount of inductance may be required to in the connection between RF passive components920-1and920-3and in the connection between RF passive components920-2and920-4in order to isolate the RF signal path from the resistive losses in the high density capacitor chips920-3,920-4. In some embodiments, the necessary inductances may be implemented in RF passive devices920-1,920-2. FIG.11Dalso illustrates the gate lead940, drain lead942and source lead944that are implemented in RDL laminate structure910. The gate and drain leads940,942may be implemented on opposed sides of RDL laminate structure910, with the source lead944therebetween. The source lead944may be implemented as a dense array of conductive vias and/or as a large conductive pad that may sit on a corresponding conductive pad/slug on an interconnection structure such as a printed circuit board of an end device that includes RF transistor amplifier900″ in order to electrically connect the source lead944to electrical ground and to provide a thermal dissipation path through the interconnection structure. The gate and drain leads940,942may similarly be electrically connected to corresponding pads on the interconnection structure. FIGS.12A-12Billustrate an RF transistor amplifier1000according to further embodiments of the present invention. RF transistor amplifier1000is similar to RF transistor amplifier900″, but has a different output matching circuit. In particular, as shown inFIG.12A, the drain lead942connects directly to the drain terminal224on RF transistor amplifier die210, and the shunt L-C circuit922-3,924-3is likewise coupled to the drain terminal224.FIG.12Bis an equivalent circuit diagram of the RF transistor amplifier ofFIG.12A. FIG.13is a schematic cross-sectional view of an RF transistor amplifier1300according to still further embodiments of the present invention. The RF transistor amplifier1300ofFIG.13is similar to the RF transistor amplifier900ofFIG.11A, with the primary difference being that RF transistor amplifier1300is a multi-stage amplifier that includes two RF transistor amplifier dies210-1,210-2mounted on an RDL laminate structure1110, and RF transistor amplifier1300also includes an inter-stage impedance matching circuit that is implemented in RF passive component1120-2. The input impedance matching RF passive component1120-1also has a slightly different design than the corresponding RF passive component920-1inFIG.11A. It will be appreciated that while in some embodiments, each of the two RF transistor amplifier dies210-1,210-2may be identical, this need not be the case. For example, in other embodiments, one of the RF transistor amplifier dies210-1,210-2may be smaller than the other or may have a different configuration. It will also be appreciated that one of the two RF transistor amplifier dies210-1,210-2may comprise a Group III nitride based RF transistor amplifier while the other may be implemented in a different technology such as, for example, a silicon LDMOS RF transistor amplifier. Moreover, while the RF transistor amplifier dies210-1,210-2shown inFIG.13each have both conductive gate vias and conductive drain vias, it will be appreciated that one or both may only have conductive gate vias or only have conductive drain vias, or have neither conductive gate vias or conductive drain vias in further embodiments. The RF transistor amplifiers according to embodiments of the present invention that include RF transistor amplifier dies that are mounted on a RDL laminate structures may be particularly well-suited to overmold packaging.FIG.14is a schematic cross-sectional view of a packaged RF transistor amplifier1200according to embodiments of the present invention that includes such overmold packaging. As shown inFIG.14, a plastic overmold1210may be formed on the top surface of the RF amplifier die700ofFIG.8A. The plastic overmold1210may be formed as part of a wafer level packaging process where a plurality of RF amplifier die700are mounted on a large RDL laminate structure (not shown). After the plastic overmold is formed on the large RDL laminate structure and on the individual RF transistor amplifier die700, the large RDL laminate structure with the RF transistor amplifier die700mounted thereon may be diced to provide a plurality of the packaged RF transistor amplifiers1200shown inFIG.14. In other embodiments, the plastic overmold1210may be applied directly to the RF transistor amplifier structure shown inFIG.8A. In that case, the plastic overmold may also be formed to cover the sidewalls of the RDL laminate structure710. This technique may be applied, for example, when fan-out configurations are used since the RF transistor amplifier die may be applied to the interconnection structure after the wafer is diced and the plastic overmold may be applied after the RF transistor amplifier die is mounted on the interconnection structure. It will be appreciated that any of the RF transistor amplifiers according to embodiments of the present invention that include RF transistor amplifier dies that are mounted on a RDL laminate structure that are disclosed herein may be packaged in either of the above-described plastic overmold packaging configurations. Note that herein the term “overmold” is used broadly to encompass protective plastic coatings and the like that are deposited on top of a wafer before the wafer is diced into individual die, as is shown, for example, inFIG.14. It will be appreciated that any of the RF transistor amplifiers according to embodiments of the present invention that are discussed above may be mounted in packages such as the open cavity and overmold packages shown inFIGS.4C and4D, respectively to provide packaged RF transistor amplifiers that can be readily shipped to customers. Thus, the RF transistor die210and interconnection structures270shown inFIGS.4C-4Dmay be replaced with the RF transistor die and interconnection structures according to any of the embodiments of the present invention that are discussed herein to provide many further embodiments of packaged RF transistor amplifiers. It will also be appreciated that protective plastic packaging may be applied to any of the RF transistor amplifiers according to embodiments of the present invention that are disclosed herein.FIGS.15A through15Cillustrate additional examples of RF transistor amplifiers according to embodiments of the present invention that include protective plastic packaging. As shown inFIG.15A, the wafer201ofFIG.4A(only a portion of which is visible inFIG.15A) has a plurality of RF transistor die210formed thereon. The wafer201is mounted on a composite RDL laminate substrate1310. The composite RDL laminate substrate1310includes a plurality of individual RDL laminate substrates1312that are positioned underneath the respective RF transistor amplifier die210. Each individual RDL laminate substrate1312includes a metal gate slug1342, a metal drain slug1344and a metal source slug1346that are provided within a dielectric of the composite RDL laminate substrate1310. The wafer201may be mounted on the composite RDL laminate substrate1310by any appropriate means, such as using contacts (e.g., conductive solders or die attach material). The metal gate slug1342in each individual RDL laminate substrate1312is electrically connected to the conductive gate via262of its associated RF transistor amplifier die210, the metal drain slug1344in each individual RDL laminate substrate1312is electrically connected to the conductive drain via264of its associated RF transistor amplifier die210, and the metal source slug1346in each individual RDL laminate substrate1312is electrically connected to the conductive source vias266of its associated RF transistor amplifier die210. A protective plastic coating1301may be applied to the top surface of the wafer201(either before or after the wafer201is mounted on the composite RDL substrate1310). The wafer201with the protective plastic coating1301thereon may then be diced along the vertical dashed scribe lines shown inFIG.15Ato form individual RF transistor amplifiers1300that each comprise an RF transistor die210mounted on an individual RDL laminate substrate1312. One such individual RF transistor amplifier1300is schematically depicted inFIG.15B. The example ofFIGS.15A-15Billustrates one example way in which wafer level processing may be used to form the individual RF transistor amplifiers1300. The wafer level processing technique described above with reference toFIGS.15A-15Bmay be particularly suitable for use with individual RDL laminate substrates1312that have a fan-in design, since each individual RDL laminate substrate1312will have about the same “footprint” (i.e., area when viewed from above) as its associated RF transistor amplifier die210. When the individual RDL laminate substrates1312have a fan-out topology, and hence have a footprint that is larger than the individual RDL laminate substrates1312, it may not always be practical to attach the above discussed composite RDL laminate structure1310to the wafer201and then dice both together, as this would require spacing the individual RF transistor die210farther apart on the wafer201so that an RF transistor amplifier die210would be positioned above each individual RDL laminate structure1312. Thus, for such fan-out RDL laminate substrates1312, the plastic coating1301may be applied as a wafer level processing step and the wafer201may then be diced into individual RF amplifier dies210(each with a plastic coating on a top surface thereof). Thereafter, the individual RF transistor amplifier dies210may be mounted on the composite RDL laminate substrate1310, which may then be diced. Alternatively, each RF transistor amplifier die210may be mounted on a respective individual RDL laminate substrate1312. Referring next toFIG.15C, in still other embodiments, the protective plastic packaging may be applied after the wafer201ofFIG.4Ais diced into individual RF transistor amplifier die210. When the protective plastic packaging is applied after dicing it may be applied as a plastic overmold1402that covers the top surface and sidewalls of each RF amplifier die210. In the embodiment ofFIG.15C, the RF amplifier die210with the plastic overmold1402thereon is mounted on an individual RDL laminate substrate1410using any appropriate contacts such as, for example, die attach material to provide an RF transistor amplifier1400. As yet another example, protective plastic coatings may be applied to RF transistor amplifier die according to embodiments of the present invention that are mounted on custom interposers.FIG.16Aillustrates one such example embodiment in which the RF transistor die210is coated with a protective plastic coating1501as part of a wafer level processing step. The coated RF transistor amplifier die210is then mounted on a custom interposer1510to provide RF transistor amplifier1500, as shown inFIG.16A. The interposer1510included in RF transistor amplifier1500includes conductive gate, drain and source connections1542,1544,1546which are implemented as large metal slugs. In other embodiments, a protective plastic overmold1502may be applied after the wafer201of FIG.4A is diced into individual RF transistor amplifier die210. When the protective plastic overmold1502is applied after dicing it may cover both the upper surface and sidewalls of the RF transistor amplifier die210to provide an RF transistor amplifier1500′, as shown inFIG.16B. It will be appreciated that in the embodiments ofFIGS.16A and16Bthe interposers1510,1510′ may have either fan-in or fan-out topologies. To illustrate this, the interposer1510is shown having a fan-in topology while the interposer1510′ is shown having a fan-out topology. Alternatively, the RF transistor amplifier die may be provided as stand-alone parts that may be mounted by a customer on an interconnection structure such as a customer printed circuit board. The stand-alone RF transistor amplifier die may include a protective plastic package. In some embodiments, the protective plastic package may be applied as part of waver level packaging. Using the RF transistor die210ofFIGS.4A and4Bas an example, a protective plastic coating211may be applied to the top surface of the wafer201ofFIG.4Aas part of a wafer level packaging step. The wafer201with the protective plastic coating211thereon may be diced to singulate the individual RF transistor die210. As shown inFIG.17A, this will provide RF transistor amplifier die210A that each have a protective plastic coating211on an upper surface thereof. A customer may then mount the RF transistor amplifier die210A on, for example, a customer printed circuit board or other interconnection structure. In other situations, it may be advantageous to apply the protective plastic packaging as a die level process. Once again, using the wafer201ofFIG.4Aas an example, the wafer201may be singulated into individual RF transistor amplifier die210as discussed above and as shown inFIG.4B. Thereafter, a protective plastic overmold213may be applied to the individual RF transistor amplifier die210. When the protective plastic overmold213is applied after singulation, the protective plastic overmold213may cover both the upper and side surfaces of the RF transistor amplifier die210to provide an RF transistor amplifier die210B. This is schematically shown inFIG.17B. A customer may then mount the RF transistor amplifier die210B on, for example, a customer printed circuit board or other interconnection structure. Depending on the embodiment, the packaged RF transistor amplifier can include a monolithic microwave integrated circuit (MMIC) as the RF transistor amplifier die where the RF transistor amplifier die incorporates multiple discrete devices in a single integrated die. Additionally or alternatively, the package can comprise multiple RF transistor amplifier die in a path that are connected in series to form a multiple stage RF transistor amplifier and/or multiple RF transistor amplifier die that are disposed in multiple paths (e.g., in parallel) to form an RF transistor amplifier with multiple RF transistor amplifier die and multiple paths, such as in a Doherty amplifier configuration. In any of these multiple RF transistor amplifier die embodiments one or more, including all of the RF transistor amplifier die may be RF transistor amplifier die according to any of the embodiments described above. While the example embodiments discussed above include a single RF amplifier die having a single stage amplifier, it will be appreciated that embodiments of the present invention are not limited thereto. In other embodiments, the amplifiers may include multiple stages, may have a Doherty configuration, etc. The RF transistor amplifiers according to embodiments of the present invention may have a number of advantages as compared to conventional RF transistor amplifiers. The provision of conductive gate and drain vias in the RF transistor amplifier die may reduce or eliminate the need for bond wires. The elimination of bond wire connections may reduce costs and simplify manufacturing, and may improve the RF performance of the device since the amount of inductance in the impedance matching networks may be tightly controlled, and the problem of too much inductance in the matching networks can be avoided. Additionally, the elimination of bod wires may reduce the size of the device. Moreover, increased wafer level packaging becomes possible with the RF transistor amplifiers according to embodiments of the present invention, which may further simplify manufacturing and/or reduce production costs. Embodiments of the present disclosure can be used, for example, in RF power products for 5G and base-station and/or handset applications, as well as in radar applications. Embodiments of the present inventive concepts have been described above with reference to the accompanying drawings, in which embodiments of the invention are shown. This inventive concepts may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the inventive concepts to those skilled in the art. Like numbers refer to like elements throughout. It will be understood that, although the terms first, second, etc. may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another. For example, a first element could be termed a second element, and, similarly, a second element could be termed a first element, without departing from the scope of the present invention. As used herein, the term “and/or” includes any and all combinations of one or more of the associated listed items. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention. As used herein, the terms “comprises” “comprising,” “includes” and/or “including” specify the presence of stated features, integers, steps, operations, elements, and/or components, but do not preclude the presence or addition of one or more other features, integers, steps, operations, elements, components, and/or groups thereof. It will be understood that when an element such as a layer, region or substrate is referred to as being “on” or extending “onto” another element, it can be directly on or extend directly onto the other element or intervening elements may also be present. In contrast, when an element is referred to as being “directly on” or extending “directly onto” another element, there are no intervening elements present. It will also be understood that when an element is referred to as being “connected” or “coupled” to another element, it can be directly connected or coupled to the other element or intervening elements may be present. In contrast, when an element is referred to as being “directly connected” or “directly coupled” to another element, there are no intervening elements present. Relative terms such as “below” or “above” or “upper” or “lower” or “horizontal” or “lateral” or “vertical” may be used herein to describe a relationship of one element, layer or region to another element, layer or region as illustrated in the figures. It will be understood that these terms are intended to encompass different orientations of the device in addition to the orientation depicted in the figures. In the drawings and specification, there have been disclosed typical embodiments of the invention and, although specific terms are employed, they are used in a generic and descriptive sense only and not for purposes of limitation, the scope of the invention being set forth in the following claims. | 96,441 |
11863131 | DETAILED DESCRIPTION FIG.1shows a schematic of a very narrowband amplifier according to the present invention. As used herein, a “very narrow bandwidth” may comprise a bandwidth less than 0.1%. The amplifier consists of a network20providing a negative resistance between its output terminals22and24. Terminals22and24are connected to the input terminals26and28of reactance network30. Output terminal32of reactance network30is connected to a resistor36which is connected to port2of circulator40and a resistor38which is connected to ground. Resistor36or resistor38or both may have a value of zero and can then be deleted. Circulator40has at least three ports. For a three port circulator, a substantially unidirectional signal path is provided between a first (or input) port1and a second (or intermediate) port2and between the second port2and a third (or output) port3. When a signal with power P0 is applied to port1of the circulator and port2is terminated in an impedance Z2, the power at port3of the circulator is equal to P0 times the magnitude squared of the reflection coefficient at port2, the reflection coefficient being equal to |(Z0−Z2)/(Z0+Z2)|2, where Z0 is the characteristic impedance of the circulator. Port1of circulator40is the input terminal of the amplifier and port3is the output port of the amplifier. The negative resistance network20may be any known circuit element characterized by a negative ratio of the voltage as between its output terminals22and24to the current flowing through the element over a frequency range. The amplifier shown inFIG.1operates as follows. When a signal with power Pin is delivered to port1of a circulator with characteristic impedance Z0 and terminal2terminated in an impedance ZL, the output power Pout at terminal3is equal to: Pin|ρ|2, where p the reflection coefficient is given by Eq. 1. ρ=(Z0−ZL)/(Z0+ZL) Equation 1 If ZL is a resistance of value Z0, ρ is equal to zero and the output power Pout is equal to zero. If ZL is a reactance, ρ is equal to 1.0 and the output power Pout is equal to Pin. If ZL is equal to a negative resistance −RL then ρ given by Eq. 2 is greater than one and the input power Pin will be amplified. ρ=(Z0+RL)/(Z0−RL) Equation 2 If for example ZL=−44.68 and Z0 equals 50 ohms then |ρ|2=316.4 and Pout/Pin=25 dB where Pin is the input power introduced to port1of circulator40and Pout is the output power at port3of circulator40. FIG.2shows a lossless two port network with an input port and an output port. It is characterized by jX11, jX12, jX21 and jX22, where jX11, jX12, jX21 and jX22 are defined by Equations 3a and 3b. V1=(I1)JX11+I2(JX12) Equation 3a V2=(I1)JX21+I2)(JX22) Equation 3b The input voltage V1, the output voltage V2, the input current I1 and the output current I2 are as show inFIG.2. For a reciprocal network X21=X12. If the network ofFIG.2is terminated in a resistance Rout then V2=−(I2)Rout and solving Eqs. 3a and 3b one finds that the input impedance Rin+ is given by Eq. 4. Rin+=jX11+(Rout(X12)2−jX122(X22))/(Rout2+X222)=R2+jX2 Equation 4 Where R2 and X2 are the real and imaginary parts of the input impedance Rin+ when the two port is terminated in Rout. If the network ofFIG.2is terminated in a resistance −Rout then the input impedance Rin− is given by Equation 5. Rin−=jX11+(−Rout(X12)2−jX122(X22))/(Rout2+X222) Equation 5 It can be seen from Equations 4 and 5 that Rin−=−R2+jX2. Substituting Equation 4 into Equation 2 one finds that the reflection coefficient ρ+ when the network ofFIG.2is terminated in Rout is given in Eq. 6. ρ+=((Z0−R2)−jX2)/((Z0+R2)+jX2) Equation 6 Substituting Equation 5 into Equation 2 one finds that the reflection coefficient ρ-when the network ofFIG.2is terminated in −Rout is given in Equation 7. ρ−=((Z0+R2)−jX2)/((Z0−R2)+jX2) Equation 7 and that |ρ−|2=1/|ρ+|2Equation 8 If the two port network ofFIG.2has a monotonic attenuation characteristic, is resonant at frequency F0 and is designed for an impedance level of 44.68 ohms then when it is terminated in a negative resistance of −44.68, Eq. 2 yields |ρ−|2=316.2. If the amplifier inFIG.1utilizes the two port shown inFIG.2, for reactance network30, the value of the negative resistance network is −44.68 and resistor36and resistor38are deleted the amplifier will have a gain of 25 dB at F0. If the two port network when terminated in 44.68 ohms has an attenuation of −3db (|ρ+|2=½) at F1 and F2, then at F1 and F2|ρ−|2=2. The gain of the amplifier will then be 3 dB at F1 and F2. The attenuation of the negative resistance amplifier at F1 and F2 is therefore 22 dB below the gain at F0. If at frequencies Fb and Fa the gain of the negative resistance amplifier is 22 dB; 3 dB below the gain of the amplifier at F0, then Fa and Fb define the edges of the 3 dB bandwidth of the negative resistance amplifier. The bandwidth of the amplifier is then equal to Fb-Fa. The gain of the amplifier at Fb and Fa equals 22 dB and hence |ρ−|2=158.5 for this case and Equation 8 yields |ρ+|2=1/158.5=0.0063 at these frequencies. The bandwidth edges of the amplifier Fa and Fb are therefore equal to the frequencies where |ρ+|2=0.0063. If the reactance network30is a filter such as the Butterworth filter this result is independent of the number of sections of the filter. Since more sections yield a flatter filter response the amplifiers bandwidth will increase with the increasing number of sections since |ρ+|2=0.0063 will occur at frequencies which are further from F0. One or two sections should therefore be used for narrow bandwidths. Multi-section filters can be used for wider bandwidths. FIG.3shows an embodiment of the invention of the amplifier inFIG.1which can be used to realize narrow or wideband negative resistance amplifiers. For this embodiment negative resistance network20has a value of −94.68 ohms and is connected to a series resonator230which is resonant at 81.87 MHz, consisting of a 50.4 pf capacitor232and a 75 nh inductor221. Series resonator230is connected to a shunt resonator220, which is resonant at 81.87 MHz, consisting of a 1.7 mfd capacitor222and a 2.22 nh inductor221. A 50 ohm resistor236is connected between shunt resonator220and port2of circulator40. Negative resistance network20has a value equal to −94.68 ohms. The series connection of the negative resistance of −96.68 ohms and the 50 ohm resistor236yields a negative impedance of −44.68 ohms and therefore the negative resistance amplifier has a gain of 25 dB at frequency F0. This combination yields a narrower bandwidth for the amplifier, since at frequencies differing from F0 shunt resonator220approximates a short circuit; terminating port2of the circulator40in approximately 50 ohms and hence greater attenuation. This circuit was built and tested where the negative resistance20was realized using an operational amplifier with appropriate feedback. Measured results are shown inFIG.4. The bandwidth of the amplifier is 410 KHz a 0.5% bandwidth. FIG.5shows an alternative embodiment of the invention of the amplifier inFIG.1which can be used to realize narrow or wideband negative resistance amplifiers. For this embodiment, negative resistance network20with a value of −23.6 is connected to reactance network30which is realized by a shunt resonator420, which is resonant at 800 MHz, consisting of a 96.5 pf capacitor422and a 0.41 nh inductor421. Shunt resonator420, is connected to a series resonator430which is resonant at 800 MHz, consisting of a 0.066 pf capacitor432and a 600 nh inductor431. Series resonator430is connected to a 50 ohm resistor438which is connected to ground and to port2of circulator40. The 50 ohm resistor438in parallel with −23.6 ohms yields a negative resistance of −44.68 ohms at F0. Setting the value of the negative resistance network20equal to −23.6 and the value of resistor438equal to 50 ohms results in a narrower bandwidth for the amplifier since at frequencies differing from F0 the series resonator approximates an open circuit; terminating port2of the circulator40in approximately 50 ohms. This circuit was computer analyzed using “Microwave Office”. The results are plotted inFIG.6. The bandwidth was found to be equal to 914 KHz or a 0.114% bandwidth. The gain falls to zero at a deviation from the center frequency of 8 MHz or 1%. FIG.7shows an alternative embodiment of the invention of the amplifier inFIG.1which can be used to realize narrow or wideband negative resistance amplifiers. This embodiment utilizes transmission lines resonators. Port2of circulator40is connected to the input port538of a series connected transmission line540which has a characteristic impedance Z01 and length L1 and is terminated in an open circuited transmission line542which has a characteristic impedance Z02 and length L2, such that series connected transmission line540terminated in transmission line542is resonant at F0. If Z01 equals Z02 then transmission lines540and542degenerate into a single transmission line of length L1+L2. The output port544of transmission line540is connected to the input port545of a shunt connected transmission line546which has a characteristic impedance Z03 and length L3, and is terminated in a short circuited transmission line548which has a characteristic impedance Z04 and length L4, such that shunt connected transmission line546terminated in transmission line548is resonant at F0. The input port545of transmission line546is connected to negative resistance network20. The resultant amplifier has a center frequency F0. FIG.8shows an alternative embodiment of the invention of the amplifier inFIG.1which can be used to realize narrow or wideband negative resistance amplifiers. This embodiment utilizes transmission lines resonators and a balun to effect the series connection of the series transmission line. Port2of circulator40is connected to a balanced port652of balun650, balun650having balanced ports652and654and unbalanced port656. Unbalanced port656is connected to the input port657of transmission line658which has a characteristic impedance Z01 and length L1, and which terminated in on open circuited transmission line660, which has a characteristic impedance Z02 and length L2, such that series connected transmission line658terminated in on open circuited transmission line660is resonant at F0. Balanced port654is connected to the input port662of a shunt connected transmission line664which has a characteristic impedance Z03 and length L3 and is terminated in a short circuited transmission line666which has a characteristic impedance Z04 and length L4, such that shunt connected transmission line664terminated in short circuited transmission line666is resonant at F0. Input terminal662of transmission line664is connected to negative resistance network20. The resultant amplifier has a center frequency F0. FIG.9shows an embodiment of the invention of the amplifier which can be used to realize tunable narrow or wideband negative resistance amplifiers. Reactance network30is realized by the series connection of a varactor732and an inductor734. Inductor734is connected between port2of circulator40and varactor732. A variable DC voltage supply742is connected through an RF choke740to varactor732. An RF choke is connected between the junction of varactor732and inductor734and ground. Negative resistance network20is connected to the junction of varactor732and RF choke740. A 50 ohm resistor is connected between port2of circulator40and ground. Varying the dc voltage of DC voltage supply742varies the center frequency of the amplifier. Computer analysis of three cases was performed using “Microwave Office”. In all cases varactor732was assumed to be model GVD1203 manufactured by Sprague Goodman which has a tuning range from 0.6 pf to 2.5 pf. In the first case negative resistance network had a value of −23.6 ohms and the inductor734was set to 20 nh. The result is shown inFIG.10, where varactor732was tuned over its range. The center frequency of the amplifier varied from 698 MHz to 1.45 GHz, while the gain remained constant at 25 dB and the bandwidth remained approximately constant at 11 MHz. This corresponds to a 1.1% bandwidth at 1.0 GHz. In the second case the negative resistance network had a value of −23.6 ohms and inductor734was set to 2.64 nh. The result is shown inFIG.11where varactor732was tuned over its tuning range. The center frequency of the amplifier varied from 2 GHz to 4.0 GHz. The gain remained constant at 25 dB and the bandwidth remained constant at 8 MHz. This corresponds to a 0.27% bandwidth at 3.0 GHz. In the third case the negative resistance network had a value of −44.68 ohms, inductor734was set to 2.64 nh and resistor738was deleted. The result is shown inFIG.12where varactor732was tuned over its tuning range. The center frequency of the amplifier varied from 2 GHz to 4.0 GHz. The gain remained constant at 25 dB and the bandwidth remained constant at 320 MHz. This corresponds to a 10.67% bandwidth at 3.0 GHz. Thus for wide bandwidths resistors738should be deleted, as should resistor38inFIG.1. The present invention may be embodied in other specific forms without departing from its spirit or characteristics. The described embodiments are to be considered in all respects only as illustrative and not restrictive. The scope of the invention is, therefore, indicated by the appended claims rather than by the foregoing description. All changes which come within the meaning and range of equivalency of the claims are to be embraced within their scope. | 13,495 |
11863132 | DETAILED DESCRIPTION OF THE INVENTION Embodiments of the present invention will be described below with reference to the drawings. Throughout the description in each of the embodiments and the accompanying drawings below, components substantially identical or equivalent are designated by identical reference symbols. First Embodiment FIG.1is a block diagram illustrating a configuration of an infrared sensor device mounted with an amplifier circuit according to the present embodiment. The infrared sensor device100has a thermopile sensor110, an amplifier circuit120, and an A/D conversion circuit130. The thermopile sensor110includes a thermopile constituted of a plurality of thermocouples connected in series or in parallel. The thermopile sensor110converts thermal energy radiated from a measurement target into electrical energy, and outputs the electrical energy as a measurement voltage corresponding to surface temperature of the measurement target. The thermopile sensor110supplies the measurement voltage to the amplifier circuit120as an input voltage Vin. The thermopile sensor110has an impedance of the order of a few hundred kΩ, for example. Upon reception of the input voltage Vin from the thermopile sensor110, the amplifier circuit120amplifies the input voltage Vin, and outputs the amplified voltage as an output voltage Vout. The amplifier circuit120performs a first operation and a second operation, while periodically switching the first operation and the second operation in response to a clock signal. The first operation is configured to retain an electric charge corresponding to the input voltage Vin. The second operation is configured to output the output voltage Vout. The A/D conversion circuit130performs analog to digital conversion of the output voltage Vout, and outputs the converted voltage as a sensor output Sout. FIG.2is a circuit diagram illustrating a configuration of the amplifier circuit120. The amplifier circuit120includes an operational amplifier10, a capacitor C1, a capacitor C2, a switching circuit S1, a switching circuit S2, and an impedance converter circuit11. The operational amplifier10has a positive input terminal (non-inverting input terminal) connected to a ground potential. The operational amplifier10has a negative input terminal (inverting input terminal) connected to one end of the capacitor C1 and to one end of the capacitor C2 through a node n1. The operational amplifier10has an output terminal that outputs the output voltage Vout. The impedance converter circuit11is a circuit that converts an output impedance of the signal source (thermopile circuit110) into a specified impedance. The impedance converter circuit11is configured as a voltage amplifier circuit having an operational amplifier12, a resistance R1, and a resistance R2. In the following description, the impedance converter circuit11is also called a voltage amplifier circuit11. The operational amplifier12has an output terminal connected to the other end of the capacitor C1 and to one end of the resistance R2 through a node n2. The operational amplifier12has a negative input terminal connected to a node n3between the resistance R1 and the resistance R2. The resistances R1 and R2 are connected in series between the node n2and a ground potential. The resistance R1 has one end connected to the ground potential. The resistance R1 has the other end connected to the node n3. The resistance R2 has one end connected to the node n2. The resistance R2 has the other end connected to the node n3. The resistance R1 has a resistance value of 1 kΩ, for example. The resistance R2 has a resistance value of 10 kΩ, for example. The switching circuit S1is a switching circuit that switches connection upon reception of a first switching signal ϕ1and a second switching signal ϕ2. The switching circuit S1is constituted of a switch M1and a switch M2. The first switching signal ϕ1and the second switching signal ϕ2have a signal level that complementarily changes to a logic level 0 (L level) or a logic level 1 (H level). The operating state of the amplifier circuit120switches to the first operation and the second operation in response to the change in the signal level of the first switching signal ϕ1and the second switching signal ϕ2. For example, during the first operation, the first switching signal ϕ1has the logic level 1 (H level), while the second switching signal ϕ2has the logic level 0 (L level). During the second operation, the first switching signal ϕ1has the logic level 0 (L level), while the second switching signal ϕ2has the logic level 1 (H level). The switch M1has one end connected to an output terminal of the thermopile sensor110that is a signal source of the input voltage Vin (hereinafter also simply called a signal source). The switch M1has the other end connected to a positive input terminal of the operational amplifier12. The switch M1is a switch element turned on or off in response to the first switching signal ϕ1. The switch M1is constituted of an N-channel MOS transistor, for example. The switch M1is controlled to be turned on during the first operation and to be turned off during the second operation. When the switch M1is turned on, the positive input terminal of the operational amplifier12and the signal source are connected, so that the input voltage Vin is supplied to the positive input terminal of the operational amplifier12. The switch M2has one end connected to the positive input terminal of the operational amplifier12. The switch M2has the other end connected to a ground potential. The switch M2is a switch element turned on or off in response to the second switching signal ϕ2. The switch M2is constituted of an N-channel MOS transistor, for example. The switch M2is controlled to be turned off during the first operation and to be turned on during the second operation. When the switch M2is turned on, the ground potential is supplied to the positive input terminal of the operational amplifier12. Namely, it can be said that the positive input terminal of the operational amplifier12is connected to the ground potential. The switching circuit S2is a switching circuit that switches connection upon reception of the first switching signal ϕ1and the second switching signal ϕ2. The switching circuit S2is constituted of a switch M3, a switch M4, and a switch M5. The switch M3has one end connected to the other end of the capacitor C2. The switch M3has the other end connected to the output terminal of the operational amplifier10. The switch M3is a switch element turned on or off in response to the second switching signal ϕ2. The switch M3is constituted of an N-channel MOS transistor, for example. The switch M3is controlled to be turned off during the first operation and to be turned on during the second operation. When the switch M3is turned on, the other end of the capacitor C2 and the output terminal of the operational amplifier10are connected to each other. Accordingly, feedback connection between the output terminal and the negative input terminal of the operational amplifier10is established through the capacitor C2. The switch M4has one end connected to the other end of the capacitor C2 and to the one end of the switch M3through a node n4. The switch M4has the other end connected to a ground potential. The switch M4is a switch element turned on or off in response to the first switching signal ϕ1. The switch M4is constituted of an N-channel MOS transistor, for example. The switch M4is controlled to be turned on during the first operation and to be turned off during the second operation. When the switch M4is turned on, the ground potential is supplied to the other end of the capacitor C2. Namely, it can be said that the other end of the capacitor C2 is connected to the ground potential. The switch M5has one end connected to the one end of the capacitor C1 and to the negative input terminal of the operational amplifier10through the node n1. The switch M5has the other end connected to the output terminal of the operational amplifier10. The switch M5is a switch element tuned on or off in response to the first switching signal ϕ1. The switch M5is constituted of an N-channel MOS transistor, for example. The switch M5is controlled to be turned on during the first operation and to be turned off during the second operation. When the switch M5is turned on, the output terminal and the negative input terminal of the operational amplifier10are short-circuited. During the first operation and the second operation of the amplifier circuit120, the capacitors C1 and C2 serve as capacitative elements (capacitors) that store (retain) an electric charge. The capacitor C1 is interposed between the output terminal of the operational amplifier12and the negative input terminal of the operational amplifier10. The capacitor C2 is connected between the negative input terminal of the operational amplifier10and the one end of the switch M3. When the switch M3is turned on, the capacitor C2 serves as a feedback capacitance inserted to a connection line that connects the output terminal and the negative input terminal of the operational amplifier10. Next, the operation of the amplifier circuit120will be described with reference toFIGS.3,4and5. FIG.3illustrates the state of the amplifier circuit120during the first operation. In the first operation, the switches M1, M4, and M5are turned on, while the switches M2and M3are turned off. As a consequence, the input voltage Vin is supplied to the positive input terminal of the operational amplifier12, and the negative input terminal and the output terminal of the operational amplifier10are short-circuited. Since the positive input terminal of the operational amplifier10is grounded, and the negative input terminal and the output terminal of the operational amplifier10are short-circuited, an offset voltage (−Vos2) of the operational amplifier10is output as the output voltage Vout. In this case, an electric charge Q1 stored in the capacitor C1 (capacitance C1) is expressed by the following Expression (1). An electric charge Q2 stored in the capacitor C2 (capacitance C2) is expressed by the following Expression (2). [Expression 1] Q1=C1×V1=C1×((1+R2/R1)×(Vin−Vout)+Vos2) (1) [Expression 2] Q2=C2×V2=C2×Vos2(2) FIG.4illustrates the state of the amplifier circuit120during the second operation. In the second operation, the switches M2and M3are turned on, while the switches M1, M4, and M5are turned off. As a consequence, the positive input terminal of the operational amplifier12is grounded, and the output terminal of the operational amplifier10is connected to the negative input terminal through the capacitor C2. In this case, an electric charge Q1′ stored in the capacitor C1 is expressed by the following Expression (3). An electric charge Q2′ stored in the capacitor C2 is expressed by the following Expression (4). [Expression 3] Q′1=C1×V′1=C1×((1+R2/R1)(−Vos1)+Vos2) (3) [Expression 4] Q′2=Q2+Q1−Q′1=C2×Vos2+C1×((1+R2/R1)×Vin(4) Based on Expressions (3) and (4), a voltage V2′ applied across both the ends of the capacitor C2 is calculated as in Expression (5). The output voltage Vout of the operational amplifier10is calculated as in Expression (6). [Expression 5] V′2=Q′2/C2=Vos2+C1/C2×(1+R2/R1)×Vin(5) [Expression 6] Vout=V′2−Vos2=Vos2+C1/C2×(1+R2/R1)×Vin−Vos2=C1/C2×(1+R2/R1)×Vin(6) As is seen from Expression (6), the output voltage Vout is obtained by multiplying a ratio between the capacitance C1 and the capacitance C2 by a value obtained by adding one to a ratio between the resistance R1 and the resistance R2. As a result, the offset voltages Vos1and Vos2are canceled. When R1=1 kΩ, R2=10 kΩ, C1=10 pF, and C2=1 pF, a voltage gain of 110 is obtained. In accordance with time change in the signal level of the first switching signal ϕ1and the second switching signal ϕ2, the amplifier circuit120repeatedly performs the first operation and the second operation. FIG.5is a time chart illustrating the time change in the first switching signal ϕ1, the second switching signal ϕ2, the input voltage Vin, and the output voltage Vout. During the period from the time when the first switching signal ϕ1becomes H level to the time when the second switching signal ϕ2becomes H level, the offset voltage (−Vos2) of the operational amplifier10is output as the output voltage Vout. During the period from the time when the second switching signal ϕ2becomes H level to the time when the first switching signal ϕ1becomes H level, input voltages Vin (illustrated as V(t1), V(t2), and V(t3) in the drawing) at the times (illustrated as t1, t2, and t3 in the drawing) when each of the preceding first switching signals ϕ1becomes L level are amplified, and the amplified input voltages are output as the output voltages Vout. For example, when the voltage amplifier circuit11has an amplification factor expressed by G, the output voltages Vout are equal to G×C1/C2×V(t1), G×C1/C2×V(t2), and G×C1/C2×V(t3), respectively. As described in the foregoing, in the amplifier circuit120of the present embodiment, the impedance converter circuit11is interposed between the input terminal that receives supply of the input voltage Vin from the thermopile sensor110and the capacitor C1. Therefore, the capacitor C1 is charged and discharged not directly by the thermopile sensor110but by the impedance converter circuit11serving as a voltage amplifier circuit. Meanwhile, the thermopile sensor110charges and discharges an input stray capacitance of the impedance converter circuit11, which is extremely lower in capacitance value than the capacitor C1. As described before, the thermopile sensor110has a relatively high impedance. In contrast, the impedance converter circuit11is constituted of the operational amplifier12, the resistance R1, and the resistance R2, and therefore the impedance converter circuit11is lower in impedance than the thermopile sensor110. The input stray capacitance of the impedance converter circuit11is extremely lower than the capacitance value of the capacitor C1. Therefore, the amplifier circuit120of the present embodiment can charge and discharge the capacitor C1 in a short time. Hence, it becomes possible to obtain a desired voltage gain by cancelling an offset voltage, while switching the charging operation of an electric charge and the output operation of an amplified voltage. Second Embodiment FIG.6is a circuit diagram illustrating a configuration of an amplifier circuit220according to a second embodiment. The amplifier circuit220is different from the amplifier circuit120according to the first embodiment in the point that the amplifier circuit220includes a clock feedthrough cancellation circuit21constituted of a switch M6and a capacitor C3. The capacitor C3 has one end connected to the positive input terminal of the operational amplifier10. The capacitor C3 has the other end connected to a ground potential. The capacitor C3 has a capacitance value that is identical to a sum of the capacitance values of the capacitor C1 and the capacitor C2 (i.e., C3=C1+C2). The switch M6has one end connected to the positive input terminal of the operational amplifier10. The switch M6has the other end connected to a ground potential. The switch M6is a switch element turned on or off in response to the first switching signal ϕ1. The switch M6is controlled to be turned on during the first operation, and to be turned off during the second operation. When the switch M6is turned on, the positive input terminal of the operational amplifier10is connected to the ground potential. The switch M6is an N-channel MOS transistor, for example. The switch M6is produced to have a dimension identical to that of a transistor constituting the switch M5. The clock feedthrough cancellation circuit21is provided in order to cancel clock feedthrough generated in the amplifier circuit220. The clock feedthrough is a phenomenon in which the electric charge of a capacitance that constitutes the switched capacitor amplifier circuit is discharged or charged due to a stray capacitance between the gate and drain or between the gate and source, when a MOS transistor as a switch element transitions from ON to OFF. Hereinafter, the clock feedthrough phenomenon will be described with reference toFIGS.7to9. The following description relates to an example in which the switches M1to M6are each constituted of an N-channel MOS transistor. The switches M1to M6are also called transistors M1to M6. FIG.7illustrates a normal switched capacitor amplifier circuit as a comparative example, the circuit having neither the impedance converter circuit11according to the first embodiment nor the clock feedthrough cancellation circuit21according to the present embodiment. A stray capacitance Cst is generated between the gate and drain of the transistor M5. However, a junction point between the transistor M1and the transistor M2, and a junction point between the transistor M3and the transistor M4are connected to the fixed potential (sensor output, ground potential, and operational amplifier output) during both the first operation and the second operation. Even with the stray capacitance present in the transistors M1to M4, the capacitance is charged and discharged to keep the specified potential. Accordingly, the stray capacitance does not affect the operation of the switched capacitor amplifier circuit. FIG.8illustrates the state where the transistors M1, M4, and M5that receive supply of the first switching signal ϕ1are turned on, while the transistors M2and M3that receive supply of the second switching signal ϕ2are turned off, with each transistor being replaced with a switch. InFIG.8, Vth represents a threshold voltage of the transistor M5, and V3represents a voltage across both the ends of the stray capacitance Cst. An electric charge Q1 stored in the capacitor C1 (capacitance C1), an electric charge Q2 stored in the capacitor C2 (capacitance C2), and an electric charge Qst stored in the stray capacitance Cst in this state are expressed by the following Expressions (7) to (9), respectively. [Expression 7] Q1=C1×V1=C1(V1in+Vos) (7) [Expression 8] Q2=C2×V2=C2×Vos(8) [Expression 9] Qst=Cst×V3=Cst(Vth+Vos) (9) FIG.9illustrates the state where the transistors M1, M4, and M5that receive supply of the first switching signal ϕ1are turned off, while the transistors M2and M3that receive supply of the second switching signal ϕ2are turned on, with each transistor being replaced with a switch. The stray capacitance Cst is directly connected to a ground potential, without the threshold voltage of the transistor M5interposed therebetween. An electric charge Q1′ stored in the capacitor C1, an electric charge Q2′ stored in the capacitor C2, and an electric charge Qst′ stored in the stray capacitance Cst in this case are expressed by the following Expressions (10) to (12), respectively. [Expression 10] Q′1=C1×V′1=C1×Vos(10) [Expression 11] Q′2=Q2+Q1−Q′2+Qst−Q′st(11) [Expression 12] Q′st−Cst×V′2=Cst×Vos(12) According to Expressions (10) to (12), the electric charge Q2′ stored in the capacitor C2 is calculated as in the following Expression (13). The output voltage Vout of the operational amplifier10is calculated as in the following Expression (14). [Expression13]Q2′=C2×Vos+C1(Vin+Vos)-C1×Vos+Cst(Vth+Vos)-Cst×Vos=C2×Vos+C1×Vin+Cst×Vth(13)[Expression14]Vout=Q2′/C2-Vos=Vos+C1/C2·Vin+Cst/C2·Vth-Vos=C1/C2·Vin+Cst/C2·Vth(14) Thus, an error of the output voltage caused by the clock feedthrough is generated in proportion to (Cst/C2)Vth. FIG.10illustrates a circuit formed by adding a clock feedthrough cancellation circuit, constituted of a transistor M6and a capacitor C3, to the switched capacitor amplifier circuit ofFIG.7. A stray capacitance Cst is generated between the gate and drain of the transistor M6. FIG.11illustrates the state where the transistors M1, M4, M5, and M6that receive supply of the first switching signal ϕ1are turned on, while the transistors M2and M3that receive supply of the second switching signal ϕ2are turned off, with each transistor being replaced with a switch. An electric charge Q1 stored in the capacitor C1, an electric charge Q2 stored in the capacitor C2, and an electric charge Qst stored in the stray capacitance Cst in this state are expressed by the following Expressions (15) to (17). A voltage V4across both the ends of the capacitor C3 is expressed by the following Expression (18). [Expression 15] Q1=C1×V1=C1(Vin+Vos) (15) [Expression 16] Q2=C2×V2=C2×Vos(16) [Expression 17] Qst=Cst×V3=Cst(Vth+Vos) (17) [Expression 18] V4=0 (18) FIG.12illustrates the state where the transistors M1, M4, M5, and M6that receive supply of the first switching signal ϕ1are turned off, while the transistors M2and M3that receive supply of the second switching signal ϕ2are turned on, with each transistor being replaced with a switch. An electric charge Q1′ stored in the capacitor C1, an electric charge Q2′ stored in the capacitor C2, and an electric charge Qst′ stored in the stray capacitance Cst in this case are expressed by the following Expressions (19) to (21). A voltage V4′ across both the ends of the capacitor C3 is expressed by the following Expression (22). [Expression 19] Q′1=C1×V′1=C1(Vos+V′4) (19) [Expression 20] Q′2=Q2+Q1−Q′1+Qst−Q′st(20) [Expression 21] Q′st=Cst×V′3=Cst(Vos+V′4) (21) [Expression 22] V′4=Cst/(C1+C2+Cst)·Vth(22) According to Expressions (19) to (22), the electric charge Q2′ stored in the capacitor C2 is calculated as in Expression (23). The output voltage Vout of the operational amplifier10is calculated as in Expression (24). [Expression23]Q2′=C2×Vos+C1(Vin+Vos)-C1(Vos+Cst/(C1+C2+Cst)·Vth)+Cst×(Vth+Vos)-Cst(Vos+Cst/(C1+C2+Cst)·Vth)=C2×Vos+C1×Vin-C1/(C1+C2+Cst)·Cst×Vth+Cst×Vth-Cst/(C1+C2+Cst)·Cst×Vth)=C2×Vos+C1×Vin+(1-(C1+Cst)/(C1+C2+Cst))·Cst×Vth(23)[Expression24]Vout=(Q2′/C2)-(Vos+V4′)=Vos+(C1/C2)·Vin+(1-(C1+Cst)/(C1+C2+Cst)·(Cst/C2)×Vth-(Vos+Cst/(C1+C2+Cst)·Vth)=(C1/C2)·Vin+(1-(C1+Cst)/(C1+C2+Cst)·(Cst/C2)×Vth-Cst/(C1+C2+Cst)·Vth)=(C1/C2)·Vin+(1-(C1+Cst)/(C1+C2+Cst)-C2/(C1+C2+Cst))(Cst/C2)·Vth=(C1/C2)·Vin(24) According to a comparison between Expression (24) and Expression (14), an error ((Cst/C2) (Vth) of the output voltage caused by the clock feedthrough is canceled and eliminated in Expression (24). With reference toFIG.6again, the amplifier circuit220according to the present embodiment has the clock feedthrough cancellation circuit21in addition to the impedance converter circuit11. Therefore, the amplifier circuit220according to the present embodiment can switch the charging operation of an electric charge and the output operation of an amplified voltage at high speed, while suppressing the influence of the clock feedthrough. Third Embodiment FIG.13is a circuit diagram illustrating a configuration of an amplifier circuit320according to a third embodiment. The amplifier circuit320is different from the amplifier circuit220according to the second embodiment in the point that the switching signals supplied to the switches M1and M2are exchanged. More specifically, the switch M1is turned on or off upon reception of the second switching signal ϕ2. The switch M2is turned on or off upon reception of the first switching signal ϕ1. FIG.14is a time chart illustrating time change in the first switching signal ϕ1, the second switching signal ϕ2, the input voltage Vin, and the output voltage Vout. During the period when the first switching signal ϕ1is at H level, i.e., during the period when the switches M2, M4, M5, and M6are turned on, an offset voltage (−Vos2) of the operational amplifier10is output as the output voltage Vout. During the period when the second switching signal ϕ2is at H level, i.e., during the period when the switches M1and M3are turned on, a voltage obtained by amplifying the input voltage Vin with an amplification factor −1×G×(C1/C2) is output as the output voltage Vout. More specifically, in the subject period, an output voltage Vout=−1×G×(C1/C2)×Vin that is opposite in phase to the input voltage Vin is output. Here, G is an amplification factor of the voltage amplifier circuit11. The amplifier circuit320according to the present embodiment can provide an output voltage Vout having a signal waveform corresponding to change in the input voltage Vin during the second operation. Fourth Embodiment FIG.15is a circuit diagram illustrating a configuration of an amplifier circuit420according to a fourth embodiment. The amplifier circuit420is different from the amplifier circuit220according to the second embodiment in the point that the output terminal of the impedance converter circuit11is connected to the positive input terminal of the operational amplifier10through a capacitor C1b. The negative input terminal of the operational amplifier10is connected to a ground potential through a capacitor C1a. The capacitors C1a and C1b have the same capacitance value (capacitance value C1). The negative input terminal of the operational amplifier10is connected to one end of the switch M3through a capacitor C2a. The positive input terminal of the operational amplifier10is connected to a ground potential through a clock feedthrough cancellation circuit constituted of a switch M6and a capacitor C2b. The capacitors C2a and C2b have the same capacitance value (capacitance value C2). FIG.16is a time chart illustrating time change in the first switching signal ϕ1, the second switching signal ϕ2, the input voltage Vin, and the output voltage Vout. During the period from the time when the first switching signal ϕ1becomes H level to the time when the second switching signal ϕ2becomes H level, an offset voltage (−Vos2) of the operational amplifier10is output as the output voltage Vout. Meanwhile, during the period from the time when the second switching signal ϕ2becomes H level to the time when the first switching signal ϕ1becomes H level, input voltages Vin (V(t1), V(t2), (t3)) at the times (t1, t2, t3) when each of the preceding first switching signals ϕ1becomes L level are amplified with an amplification factor −1×G×(C1/C2), the amplified input voltages are output as the output voltages Vout. More specifically, the output voltages Vout are −G×C1/C2×V(t1), −G×C1/C2×V(t2), and −G×C1/C2×V(t3). Here, G is an amplification factor of the voltage amplifier circuit11. Thus, in the amplifier circuit420according to the present embodiment, the output voltage Vout that is opposite in phase to the output voltage Vout of the amplifier circuit120according to the first embodiment is output. Therefore, it is possible to provide the output voltage Vout of an opposite phase by switching the charging operation of an electric charge and the output operation of an amplified voltage at high speed, while suppressing the influence of the clock feedthrough. Embodiment 5 FIG.17is a circuit diagram illustrating a configuration of an amplifier circuit520according to a fifth embodiment. The amplifier circuit520is different from the amplifier circuit420according to the fourth embodiment in the point that the switching signals supplied to the switches M1and M2are exchanged. More specifically, the switch M1is turned on or off upon reception of the second switching signal ϕ2. The switch M2is turned on or off upon reception of the first switching signal ϕ1. FIG.18is a time chart illustrating time change in the first switching signal ϕ1, the second switching signal ϕ2, the input voltage Vin, and the output voltage Vout. During the period when the first switching signal ϕ1is at H level, i.e., during the period when the switches M2, M4, M5, and M6are turned on, an offset voltage (−Vos2) of the operational amplifier10is output as the output voltage Vout. During the period when the second switching signal ϕ2is at H level, i.e., during the period when the switches M1and M3are turned on, a voltage obtained by amplifying the input voltage Vin with an amplification factor G×(C1/C2) is output as the output voltage Vout. More specifically, in the pertinent period, an output voltage Vout=G×(C1/C2)×Vin that is identical in phase to the input voltage Vin is output. Here, G is an amplification factor of the voltage amplifier circuit11. The amplifier circuit520according to the present embodiment can provide an output voltage Vout having a signal waveform corresponding to change in the input voltage Vin during the second operation. Sixth Embodiment FIG.19is a circuit diagram illustrating a configuration of an amplifier circuit620according to a sixth embodiment. The amplifier circuit620has an impedance converter circuit61constituted of voltage amplifier circuits G1and G2. The amplifier circuit620is different from the amplifier circuit220according to the second embodiment in the point that the amplifier circuit620receives input voltages Vin(+) and Vin(−) that are differential input signals supplied from first and second signal sources not illustrated. The switching circuit S1receives a positive input voltage Vin(+) supplied from the first signal source. When the switch M1is turned on and the switch M2is turned off, an input terminal of the voltage amplifier circuit G1and the signal source are connected to each other. As a result, the input voltage Vin(+) is supplied to the voltage amplifier circuit G1. When the switch M1is turned off and the switch M2is turned on, the input terminal of the voltage amplifier circuit G1is connected to a ground potential. The input terminal of the voltage amplifier circuit G1is connected to the switching circuit S1. An output terminal of the voltage amplifier circuit G1is connected to the negative input terminal of the operational amplifier10through a capacitor C1a (capacitance C1). The voltage amplifier circuit G1has an amplification factor G. A switching circuit S3is a switching circuit that switches connection upon reception of the first switching signal ϕ1and the second switching signal ϕ2. The switching circuit S3is constituted of a switch M7and a switch M8. The switch M7is a switch element tuned on or off in response to the first switching signal ϕ1. The switch M7is controlled to be turned on during the first operation and to be turned off during the second operation. The switch M7is constituted of an N-channel MOS transistor, for example. The switch M8has one end connected to an input terminal of the voltage amplifier circuit G2. The switch M8has the other end connected to a ground potential. The switch M8is a switch element turned on or off in response to the second switching signal ϕ2. The switch M8is controlled to be turned off during the first operation and to be turned on during the second operation. The switch M8is constituted of an N-channel MOS transistor, for example. The switching circuit S3receives a negative input voltage Vin(−) supplied from the second signal source. When the switch M7is turned on and the switch M8is turned off, the input terminal of the voltage amplifier circuit G2and the signal source are connected to each other. As a result, an input voltage Vin(−) is supplied to the voltage amplifier circuit G2. When the switch M7is turned off and the switch M82is turned on, the input terminal of the voltage amplifier circuit G2is connected to a ground potential. The input terminal of the voltage amplifier circuit G2is connected to the switching circuit S3. An output terminal of the voltage amplifier circuit G2is connected to the positive input terminal of the operational amplifier10through a capacitor C1b (capacitance C1). The voltage amplifier circuit G2has the same amplification factor (amplification factor G) as the voltage amplifier circuit G1. The capacitor C1b has one end connected to the output terminal of the voltage amplifier circuit G2. The capacitor C1b has the other end connected to the positive input terminal of the operational amplifier10. The capacitor C1b has the same capacitance value as the capacitor C1a. The switch M6has one end connected to the positive input terminal of the operational amplifier10and to the other end of the capacitor C1b. The switch M6is a switch element tuned on or off in response to the first switching signal ϕ1. The switch M6is constituted of an N-channel MOS transistor, for example. The capacitor C2b is a capacitative element having the same capacitance value (capacitance value C2) as the capacitor C2a. The capacitor C2b has one end connected to the positive input terminal of the operational amplifier10and to the other end of the capacitor C1b. The capacitor C2b has the other end connected to a ground potential. The amplifier circuit620according to the present embodiment is a circuit that receives a positive input voltage Vin(+) and a negative input voltage Vin(−) as differential signals, amplifies a voltage difference Vin(+)−Vin(−) that is a difference between the input voltages, and outputs the amplified voltage as the output voltage Vout. FIG.20is a time chart illustrating time change in the first switching signal ϕ1, the second switching signal ϕ2, the voltage difference Vin(+)−Vin(−), and the output voltage Vout. During the period from the time when the first switching signal ϕ1becomes H level to the time when the second switching signal ϕ2becomes H level, an offset voltage (−Vos2) of the operational amplifier10is output as the output voltage Vout. During the period from the time when the second switching signal ϕ2becomes H level to the time when the first switching signal ϕ1becomes H level, values (V(t1), V(t2), V(t3)) of the voltage difference Vin(+)−Vin(−) at the times (t1, t2, t3) when each of the preceding first switching signals ϕ becomes L level are amplified with an amplification factor G×(C1/C2), and the amplified voltages are output as the output voltages Vout. More specifically, the output voltages Vout are G×C1/C2×V(t1), G×C1/C2×V(t2), and G×C1/C2×V(t3), respectively. Thus, the amplifier circuit620according to the present embodiment can provide the output voltage Vout obtained by amplifying the differential input signals (Vin(+), Vin(−)) by switching the charging operation of an electric charge and the output operation of an amplified voltage at high speed, while suppressing the influence of the clock feedthrough. As in the case of the third embodiment and the fifth embodiment, the switching signals supplied to the switches M1and M2may be exchanged, and the signals supplied to the switches M7and M8may also be exchanged. More specifically, the switch M1is controlled to be turned on or off upon reception of the second switching signal ϕ2, and the switch M2is controlled to be turned on or off upon reception of the first switching signal ϕ1. The switch M7is controlled to be turned on or off upon reception of the second switching signal ϕ2, and the switch M8is controlled to be turned on or off upon reception of the first switching signal ϕ1. Accordingly, as in the case of the third embodiment and the fifth embodiment, the output voltage Vout having a signal waveform corresponding to change in the differential input signals (Vin(+), Vin(−)) can be provided during the second operation. Seventh Embodiment FIG.21is a circuit diagram illustrating a configuration of an amplifier circuit720according to a seventh embodiment. The amplifier circuit720is different from the amplifier circuit620of the sixth embodiment in the point that an impedance converter circuit71is configured as a programmable gain amplifier. The impedance converter circuit71has operational amplifiers71aand71b. The operational amplifier71ahas an output terminal connected to the negative input terminal of the operational amplifier10through a capacitor C1a. The operational amplifier71bhas an output terminal connected to the positive input terminal of the operational amplifier10through a capacitor C1b. Between the output terminal and a negative input terminal of the operational amplifier71a, switches M9to M12are connected in parallel with each other. The switches M9to M12are each constituted of an N-channel MOS transistor, for example. Each of the switches M9to M12has one end connected to the negative input terminal of the operational amplifier71a. Each of the switches M13to M16has one end connected to a negative input terminal of the operational amplifier71b. The switch M9has the other end connected to the output terminal of the operational amplifier71athrough a node n5and a resistance R5a. The switch M16has the other end connected to the output terminal of the operational amplifier71bthrough a node n6and a resistance R5b. Between the node n5and the node n6, resistances R4a, R3a, R2a, R1, R2b, R3b, and R4b are connected in series. The resistance R2b has a resistance value R2 same as the resistance R2a. The resistance R3b has a resistance value R3 same as the resistance R3a. The resistance R4b has a resistance value R4 same as the resistance R4a. The resistance R5b has a resistance value R5 same as the resistance R5a. The switch M9has one end connected to the negative input terminal of the operational amplifier71a. The switch M9has the other end connected to the output terminal of the operational amplifier71athrough the resistance R5a. The switch M9is controlled to be turned on or off in response to a selecting signal S3. When the switch M9is turned on, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistance R5a. The switch M10has one end connected to the negative input terminal of the operational amplifier71a. The switch M10has the other end connected to the output terminal of the operational amplifier71athrough the resistances R4a and R5a. The switch M10is controlled to be turned on or off in response to a selecting signal S2. When the switch M10is turned on, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistances R4a and R5a. The switch M11has one end connected to the negative input terminal of the operational amplifier71a. The switch M11has the other end connected to the output terminal of the operational amplifier71athrough the resistances R3a, R4a, and R5a. The switch M11is controlled to be turned on or off in response to a selecting signal S1. When the switch M11is turned on, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistances R3a, R4a, and R5a. The switch M12has one end connected to the negative input terminal of the operational amplifier71a. The switch M12has the other end connected to the output terminal of the operational amplifier71athrough the resistances R2a, R3a, R4a, and R5a. The switch M12is controlled to be turned on or off in response to a selecting signal S0. When the switch M12is turned on, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistances R2a, R3a, R4a, and R5a. The switch M13has one end connected to the negative input terminal of the operational amplifier71b. The switch M13has the other end connected to the output terminal of the operational amplifier71bthrough the resistances R2b, R3b, R4b, and R5b. The switch M13is controlled to be turned on or off in response to the selecting signal S0. When the switch M13is turned on, the negative input terminal and the output terminal of the operational amplifier71bare connected to each other through the resistances R2b, R3b, R4b, and R5b. The switch M14has one end connected to the negative input terminal of the operational amplifier71b. The switch M14has the other end connected to the output terminal of the operational amplifier71bthrough the resistances R3b, R4b, and R5b. The switch M14is controlled to be turned on or off in response to the selecting signal S1. When the switch M14is turned on, the negative input terminal and the output terminal of the operational amplifier71bare connected to each other through the resistances R3b, R4b, and R5b. The switch M15has one end connected to the negative input terminal of the operational amplifier71b. The switch M15has the other end connected to the output terminal of the operational amplifier71bthrough the resistances R4b and R5b. The switch M15is controlled to be turned on or off in response to the selecting signal S2. When the switch M15is turned on, the negative input terminal and the output terminal of the operational amplifier71bare connected to each other through the resistances R4b and R5b. The switch M16has one end connected to the negative input terminal of the operational amplifier71b. The switch M16has the other end connected to the output terminal of the operational amplifier71bthrough the resistance R5b. The switch M16is controlled to be turned on or off in response to the selecting signal S3. When the switch M16is turned on, the negative input terminal and the output terminal of the operational amplifier71bare connected to each other through the resistance R5b. The selecting signals S0to S3are controlled such that any one signal is set to signal level H (i.e., turned on) and the remaining three signals are set to signal level L (i.e., turned off). Hence, among a pair of the switches M9and M16, a pair of the switches M10and M15, a pair of the switches M11and M14, and a pair of the switches M12and M13, any one pair is controlled to be turned on, while the remaining three pairs are controlled to be turned off. The state (ON or OFF) of the switches M9to M16changes in response to ON/OFF of the selecting signals S0to S3, and the resistance values of the resistances connected to the output terminals and the negative input terminals of the operational amplifiers71aand71bare switched. Accordingly, the voltage gain G of the impedance converter circuit71as a voltage amplifier circuit changes. FIG.22is a table illustrating a relation between the signal level of the selecting signals S0to S3and the voltage gain G. When the selecting signal S0is turned on, and the selecting signals S1to S3are turned off, the switches M12and M13are turned on, and the other switches are turned off. Consequently, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistances R2a, R3a, R4a, and R5a. The negative input terminal and the output terminal of the operational amplifier71bare also connected to each other through the resistances R2b, R3b, R4b, and R5b. The operational amplifiers71aand71bare connected to each other through the resistance R1. As a consequence, the voltage gain is defined by G=2×(R5+R4+R3+R2)/R1. When the selecting signal S1is turned on, the selecting signals S0, S2, and S3are turned off, the switches M11and M14are turned on, and the other switches are turned off. As a consequence, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistances R3a, R4a, and R5a. The negative input terminal and the output terminal of the operational amplifier71bare also connected to each other through the resistances R3b, R4b, and R5b. The operational amplifiers71aand71bare connected to each other through the resistances R1, R2a, and R2b. As a consequence, the voltage gain is defined by G=2×(R5+R4+R3)/(2×R2+R1). When the selecting signal S2is turned on, and the selecting signals S0, S1, and S3are turned off, the switches M10and M15are turned on, and the other switches are turned off. As a consequence, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistances R4a and R5a. The negative input terminal and the output terminal of the operational amplifier71bare also connected to each other through the resistances R4b and R5b. The operational amplifiers71aand71bare connected to each other through the resistances R1, R2a, R3a, R2b, and R3b. As a result, the voltage gain is defined by G=2×(R5+R4)/{2×(R3+R2)+R1}. When the selecting signal S3is turned on and the selecting signals S0, S1, and S2are turned off, the switches M9and M16are turned on, and the other switches are turned off. As a consequence, the negative input terminal and the output terminal of the operational amplifier71aare connected to each other through the resistance R5a. The negative input terminal and the output terminal of the operational amplifier71bare also connected to each other through the resistance R5b. The operational amplifiers71aand71bare connected to each other through the resistances R1, R2a, R3a, R4a, R2b, R3b, and R4b. As a result, the voltage gain is defined by G=2×R5/{2×(R4+R3+R2)+R1} As described in the foregoing, the amplifier circuit720according to the present embodiment can selectively change the amplification gain in the amplifier circuit configured to amplify an input voltage that is a differential input signal and to generate an output voltage. As in the second embodiment, the amplifier circuit according to the present embodiment can switch the charging operation of an electric charge and the output operation of an amplified voltage at high speed, while suppressing the influence of the clock feedthrough. As in the case of the third embodiment and the fifth embodiment, the switching signals supplied to the switches M1and M2may be exchanged, and the signals supplied to the switches M7and M8may also be exchanged. More specifically, the switch M1is controlled to be turned on or off upon reception of the second switching signal ϕ2, and the switch M2is controlled to be turned on or off upon reception of the first switching signal ϕ1. The switch M7is controlled to be turned on or off upon reception of the second switching signal42, and the switch M8is controlled to be turned on or off upon reception of the first switching signal ϕ1. Accordingly, as in the case of the third embodiment and the fifth embodiment, the output voltage Vout having a signal waveform corresponding to change in the differential input signal (Vin(+), Vin(−)) can be provided during the second operation. The present invention is not limited to the embodiments described above. For example, although the impedance converter circuit is a voltage amplifier circuit in the aforementioned embodiments, the impedance converter circuit may be a circuit with amplification gain of unity. In short, the impedance converter circuit may be any circuit as long as the output impedance of a signal source is converted to a smaller impedance. Although the switches are each an N-channel MOS transistor in the aforementioned embodiments, the switches may be constituted of a P-channel MOS transistor having an opposite conductivity type. Although the amplifier circuit is used for the infrared sensor device in the aforementioned embodiments, the amplifier circuit according to the present invention may be applied to other devices. When the amplifier circuit according to the present invention is applied to the devices having a relatively high output impedance of a signal source, high-speed switching operation can be performed by converting the output impedance. In the seventh embodiment described above, the gain of the amplifier circuit changes in four stages as the switches M9to M16are turned on or off in response to the selecting signals S0to S3. However, a value of the amplification factor usable as the amplification gain and the number thereof are not limited thereto. For example, when the number of the selecting signals is defined as n (n being natural numbers), and the number of the switches and resistances are set in accordance with n, the amplification gain may be changed in n stages. It is understood that the foregoing description and accompanying drawings set forth the preferred embodiments of the present invention at the present time. Various modifications, additions and alternative designs will, of course, become apparent to those skilled in the art in light of the foregoing teachings without departing from the spirit and scope of the disclosed invention. Thus, it should be appreciated that the present invention is not limited to the disclosed Examples but may be practiced within the full scope of the appended claims. This application is based upon and claims the benefit of priority from the prior Japanese Patent Application No. 2017-093049 filed on May 9, 2017, the entire contents of which are incorporated herein by reference. | 48,716 |
11863133 | DETAILED DESCRIPTION Aspects of the present application relate to amplification circuitry for an ultrasound device. An ultrasound device may include one or more ultrasonic transducers configured to receive ultrasound signals and produce electrical output signals. Thus, the ultrasonic transducers may be operated as ultrasound sensors. The ultrasound device may include one or more amplifiers for amplifying the electrical output signals. The power consumed by, the noise generated by, and the linear signal amplification quality provided by, the amplifier may depend on an amount of current consumed by the amplifier. In some embodiments, the amplifier has a variable current source. The variable current source is adjusted during acquisition of an ultrasound signal to maintain the noise level of the amplifier below the signal level and to maintain linear amplification, while at the same time reducing the amount of power consumed by the amplifier. In some embodiments, the amplifier is a TIA. The aspects and embodiments described above, as well as additional aspects and embodiments, are described further below. These aspects and/or embodiments may be used individually, all together, or in any combination of two or more, as the application is not limited in this respect. FIG.1illustrates a circuit for processing received ultrasound signals, according to a non-limiting embodiment of the present application. The circuit100includes N ultrasonic transducers102a. . .102n, wherein N is an integer. The ultrasonic transducers are sensors in some embodiments, producing electrical signals representing received ultrasound signals. The ultrasonic transducers may also transmit ultrasound signals in some embodiments. The ultrasonic transducers may be capacitive micromachined ultrasonic transducers (CMUTs) in some embodiments. The ultrasonic transducers may be piezoelectric micromachined ultrasonic transducers (PMUTs) in some embodiments. Further alternative types of ultrasonic transducers may be used in other embodiments. The circuit100further comprises N circuitry channels104a. . .104n. The circuitry channels may correspond to a respective ultrasonic transducer102a. . .102n. For example, there may be eight ultrasonic transducers102a. . .102nand eight corresponding circuitry channels104a. . .104n. In some embodiments, the number of ultrasonic transducers102a. . .102nmay be greater than the number of circuitry channels. The circuitry channels104a. . .104nmay include transmit circuitry, receive circuitry, or both. The transmit circuitry may include transmit decoders106a. . .106ncoupled to respective pulsers108a. . .108n. The pulsers108a. . .108nmay control the respective ultrasonic transducers102a. . .102nto emit ultrasound signals. The receive circuitry of the circuitry channels104a. . .104nmay receive the electrical signals output from respective ultrasonic transducers102a. . .102n. In the illustrated example, each circuitry channel104a. . .104nincludes a respective receive switch110a. . .110nand an amplifier112a. . .112n. The receive switches110a. . .110nmay be controlled to activate/deactivate readout of an electrical signal from a given ultrasonic transducer102a. . .102n. More generally, the receive switches110a. . .110nmay be receive circuits, since alternatives to a switch may be employed to perform the same function. The amplifiers112a. . .112n, as well as amplifier300ofFIG.3(described below), may be TIAs in some embodiments. One or more of the amplifiers112a. . .112nmay be variable current amplifiers. As will be described further below, the current of the amplifiers may be varied during an acquisition period, thus adjusting the power consumption, noise level, and linearity of the amplifiers. The amplifiers112a. . .112nmay output analog signals. The circuit100further comprises an averaging circuit114, which is also referred to herein as a summer or a summing amplifier. In some embodiments, the averaging circuit114is a buffer or an amplifier. The averaging circuit114may receive output signals from one or more of the amplifiers112a. . .112nand may provide an averaged output signal. The averaged output signal may be formed in part by adding or subtracting the signals from the various amplifiers112a. . .112n. The averaging circuit114may include a variable feedback resistance. The value of the variable feedback resistance may be adjusted dynamically based upon the number of amplifiers112a. . .112nfrom which the averaging circuit receives signals. In some embodiments, the variable resistance may include N resistance settings. That is, the variable resistance may have a number of resistance settings corresponding to the number of circuitry channels104a. . .104n. Thus, the average output signal may also be formed in part by application of the selected resistance to the combined signal received at the inputs of the averaging circuit114. The averaging circuit114is coupled to an auto-zero block116. The auto-zero block116is coupled to a programmable gain amplifier118which includes an attenuator120and a fixed gain amplifier122. The programmable gain amplifier118is coupled to an ADC126via ADC drivers124. In the illustrated example, the ADC drivers124include a first ADC driver125aand a second ADC driver125b. The ADC126digitizes the signal(s) from the averaging circuit114. WhileFIG.1illustrates a number of components as part of a circuit of an ultrasound device, it should be appreciated that the various aspects described herein are not limited to the exact components or configuration of components illustrated. For example, aspects of the present application relate to the amplifiers112a. . .112n, and the components illustrated downstream of those amplifiers in circuit100are optional in some embodiments. The components ofFIG.1may be located on a single substrate or on different substrates. For example, as illustrated, the ultrasonic transducers102a. . .102nmay be on a first substrate128aand the remaining illustrated components may be on a second substrate128b. The first and/or second substrates may be semiconductor substrates, such as silicon substrates. In an alternative embodiment, the components ofFIG.1may be on a single substrate. For example, the ultrasonic transducers102a. . .102nand the illustrated circuitry may be monolithically integrated on the same semiconductor die. Such integration may be facilitated by using CMUTs as the ultrasonic transducers. According to an embodiment, the components ofFIG.1form part of an ultrasound probe. The ultrasound probe may be handheld. In some embodiments, the components ofFIG.1form part of an ultrasound patch configured to be worn by a patient. FIG.2illustrates a non-limiting example of the amplifier112aofFIG.1in greater detail. The same configuration may be used for the other amplifiers112nofFIG.1. For context, the ultrasonic transducer102aand averaging circuit114are also illustrated, while for simplicity the receive switch110ais omitted. In this non-limiting embodiment, the amplifier112ais implemented as a two-stage operational amplifier (“op-amp” for short). The first stage202is coupled to the ultrasonic transducer102a. The second stage204is coupled between the first stage202and the averaging circuit114. The second stage204provides the output signal of the amplifier112a, in this non-limiting example. The first stage202and second stage204each have a variable current source. The variable current source203is provided for the first stage202and sinks a current I1. The variable current source205is provided for the second stage204and sinks a current I2. Although the variable current sources203and205are illustrated as distinct from the respective stages202and204, they may be considered part of the respective stages. With a two-stage amplifier construction as shown inFIG.2, the noise and linearity of the amplified signal may be controlled independently. The noise of the amplifier112ais impacted primarily by the first stage202. The linearity of the amplifier112ais impacted primarily by the second stage204. More generally, the same may be true for a multi-stage amplifier having two or more stages, such that the noise of the amplifier is impacted primarily by the first stage and the linearity of the amplifier is impacted primarily by the last stage. Applicant has appreciated that during acquisition of an ultrasound signal, referred to herein as an acquisition period, the noise and linearity of the amplified signal may vary in importance. When the ultrasound signal is initially received, early in the acquisition period and corresponding to shallow depths when the ultrasound signal is a reflected signal, the associated noise will be relatively low compared to the received signal amplitude, but the linearity of the amplified signal may be of relatively high importance. However, later during the acquisition period, corresponding to greater depths when the ultrasound signal is a reflected signal, the ultrasound signal is likely to become smaller, and thus the noise of the signal is of increased importance. Thus, the amplifier112aofFIG.2is designed to allow for independent and variable control of noise and linearity. The control may be provided via the variable current sources203and205. Early during an acquisition period, the variable current source203may be controlled to sink a relatively small amount of current, while the current source205may be controlled to sink a relatively large amount of current. In such a scenario, the second stage204may operate to control the linearity of the amplified signal produced by the amplifier112a, while the first stage202may control the noise of the amplified signal202to a lesser extent than that to which it is capable. Later in the acquisition period, the current sunk by the variable current source203may be increased while the current sunk by the variable current source205may be decreased. As the current sunk by the variable current source203is increased, the first stage202may operate to control the noise of the amplifier112ato a greater extent. As the current sunk by the variable current source205is decreased, the second stage204may operate to control the linearity of the amplifier112ato a lesser extent. Thus, dynamic current biasing of the amplifier112a, and first stage202and second stage204more specifically, may be implemented to control the power, noise, and linearity characteristics of the amplifier during an acquisition period. The dynamic control of current sources203and205may be achieved using a digital controller, an example being shown inFIG.3A. The variable current sources203and205may each include two or more programmable current settings. The greater the number of settings, the greater the control over the current sunk by the current sources203and205. The amplifier112aalso includes a variable feedback impedance206. In some embodiments, the variable feedback impedance is a variable RC feedback circuit. An example of the variable RC feedback circuit is illustrated inFIG.3Aand described in connection with that figure. The feedback impedance determines the transimpedance gain of the transimpedance amplifier, such that the input current signal may be converted into an output voltage of varying amplitude. It should be appreciated fromFIG.2and the foregoing description that an embodiment of the application provides a multi-stage TIA having two or more independently controllable variable current sources, with a variable feedback impedance. The variable current sources may allow for dynamic current biasing of the TIA, for example during an acquisition period. Thus, the power consumption, noise, and linearity of the amplifier may be adjusted during the acquisition period. FIG.3Ais a circuit diagram illustrating an implementation of the amplifier112aofFIG.2, according to a non-limiting embodiment of the present application. The amplifier300has an input302and an output304. The input302may be coupled to an ultrasonic transducer or a receive switch, as described previously in connection withFIGS.1and2, and may receive an electrical signal representing an ultrasound signal received by the ultrasonic transducer. The output304may provide an amplified output signal of the amplifier112a, and may be coupled to an averaging circuit or other component to which it is desired to provide the amplified output signal. The amplifier300includes a first stage306and a second stage308, which may be implementations of the first stage202and second stage204ofFIG.2, respectively. The first stage306includes an NMOS transistor310having a gate configured to receive the signal at input302. PMOS transistor312and PMOS transistor314have their gates coupled, with the drain of PMOS transistor312coupled to the drain of NMOS transistor310. The gate of transistor312is coupled to its drain. Transistors312and314are also configured to receive a power supply voltage VDDA. The first stage306further comprises NMOS transistor316having a gate configured to receive a bias voltage provided by an RC circuit. The RC circuit includes two resistors, of value R, with a capacitor Cbcoupled in parallel with one of the resistors. The other resistor receives the power supply voltage VDDA. The drain of PMOS transistor314is coupled to the drain of NMOS transistor316. An example value for R is50kOhm and an example value for Cbis10pF, although alternatives for both are possible, such as +/−20% of those values listed, or any value or range of values within such ranges. The second stage308includes a PMOS transistor318configured to receive the output of the first stage306. In particular, the gate of PMOS transistor318is coupled to a node between transistors314and316of the first stage306. The source of PMOS transistor318receives VDDA. A variable impedance circuit320is also provided in the second stage308. The variable impedance circuit320includes a variable capacitor CCin series with a variable resistor RZ, and thus is a variable RC circuit in this embodiment. Variable impedance circuit320may provide stable operation of the amplifier300when the gain of the amplifier, or the currents of the currents sources, are varied. Thus, the variable impedance circuit may be provided to maintain stable operation of the amplifier300for all the current magnitudes sunk by the variable current sources321and325. That is, the values of CCand RZmay be adjusted during operation of the amplifier300to account for the different current settings programmed by the digital controller330 A variable current source is provided for each of the stages306and308. The variable current source321for the first stage306includes three parallel connected current sources322a,322b, and322c. Current source322asinks a current IA, current source322bsinks a current2IA, and current source322csinks a current4IA. The current sources322a-322care coupled to the first stage306by respective switches324a,324b, and324c, which effectively provides 3 bits (8 states) of control of the current. The current IAmay equal 100 microAmps or +/−20% of that value, or any value or range of values within such ranges, as examples. The variable current source325for the second stage308includes three parallel connected current sources326a,326b, and326c. Current source326asinks a current IB, current source326bsinks a current2IB, and current source326csinks a current4IB. The current sources326a-326care coupled to the second stage308by respective switches328a,328b, and328c, which effectively provides 3 bits (8 states) of control of the current. The current IBmay equal 50 microAmps or +/−20% of that value, or any value or range of values within such ranges, as examples. WhileFIG.3Aillustrates variable current sources each include three parallel-coupled current sources, it should be appreciated that not all aspects of the present application are limited in this manner. That is, variable current sources may be implemented in various manners, including alternative manners to those illustrated. For example, more or fewer than three current sources may be coupled in parallel to create a variable current source. Also, the magnitudes of the current sources may be different than those illustrated inFIG.3A. Any suitable magnitudes may be provided to allow for operation over a desired range of currents. A digital controller330is provided to control operation of the variable current sources321and325. The digital controller provides control signals to (digitally) program the currents of the variable current sources. In the illustrated example, the digital controller330provides one or more switching signals S1to control operation of the switches324a-324c, and one or more switching signals S2to control operation of the switches328a-328c. In this manner, the amount of current sunk by the variable current sources may be varied independently during operation of the amplifier300, for example during an acquisition period. According to a non-limiting example, the digital controller330decreases the current sunk by variable current source325during the acquisition period and increases the current sunk by variable current source321during the acquisition period through suitable operation of the switching signals S1and S2. The digital controller330may be any suitable type of controller. The digital controller may include integrated circuitry. In some embodiments, the digital controller330may include or be part of an application specific integrated circuit (ASIC). In some embodiments, the digital controller330may not be specific to the amplifier300. For example, a digital controller may be provided to control more than one component of the circuit ofFIG.1, one of which may be the amplifiers112a. . .112n. The amplifier300further includes a variable feedback impedance332formed by variable capacitor Cfand variable resistor Rf. The capacitor Cfand resistor Rfmay be coupled between the output304and the input302, and may be in parallel with each other. The variable feedback impedance332may control the gain of the amplifier300. Thus, the values of Cfand Rfmay be adjusted to vary the amplifier's gain. The variable feedback impedance332and the variable impedance circuit320may be controlled in any suitable manner. In one embodiment, the digital controller330may set the values of the feedback impedances. However, alternatives manners of control may be used. It should be appreciated that the described groupings of components in connection withFIG.3Aare not limiting. For example, while certain components illustrated in that figure are described as being part of a first stage or a second stage, the identification of the first and second stages is not limiting. The first and second stages may include more, fewer, or different components than those illustrated. FIG.3Bis a circuit diagram of an implementation of the variable impedance circuit320ofFIG.3A, according to a non-limiting embodiment of the present application. The variable impedance circuit320includes a number of switches340a. . .340nconfigured in parallel and configured to receive respective control signals SWa . . . SWn. In some embodiments, the digital controller330may provide the control signals SWa . . . SWn, although alternatives may be used. Each switch is coupled in series with a respective capacitor Ccand resistor RZ. The impedance of the variable impedance circuit320may be adjusted during an acquisition period through suitable provision of the control signals SWa . . . SWn. Any suitable number of parallel signal paths may be provided, so that the illustrated example of two parallel signal paths is non-limiting. The number of parallel signal paths and the capacitance and resistive values provided may be selected to provide sufficient control of the feedback impedance to account for the variable operation of the amplifier across the range of operating scenarios resulting from the variation of the variable current sources. For example, for a given amplifier gain dictated by variable feedback impedance332, appropriate settings of variable impedance circuit320may be selected. In some embodiments, a lookup table may be utilized to determine the appropriate settings of variable impedance circuit320based on a given gain set by variable feedback impedance332. In bothFIGS.3A and3B, the values of CC and RZ may be selected to provide desired operating characteristics. As examples, Rzmay be equal to 3 kOhms in some embodiments, and CCmay be equal to 300 fF. Alternatives for both are possible. For example, they may assume values within +/−20% of those values listed, or any value or range of values within such ranges. FIG.3Cis a circuit diagram of an implementation of the variable impedance circuit332ofFIG.3A, according to a non-limiting embodiment of the present application. The variable impedance circuit332includes a number of complementary switches350a,350b. . .350n. Each switch receives respective control signals SLa, SLb . . . SLn and SHa, SHb . . . SHn. The control signals may be provided by the digital controller330in some embodiments, although alternatives may be used. The complementary switches are coupled to respective parallel-connected RC circuits Cf, Rf. While three complementary switches are shown inFIG.3C, any suitable number may be provided to allow for sufficient control of the gain of the amplifier300. In bothFIGS.3A and3C, the values of Cfand Rfmay be selected to provide desired operating characteristics. As examples, Rfmay be equal to 180 kOhms in some embodiments, and Cfmay be equal to 84 fF. Alternatives for both are possible. For example, they may assume values within +/−20% of those values listed, or any value or range of values within such ranges. FIG.4is a graph illustrating the behavior of two variable current sources of a variable current amplifier during an acquisition period, as may be implemented by the amplifier ofFIGS.2and3A, which again may be a TIA. For example, the illustrated behavior may be implemented by the variable current sources203and205ofFIG.2. The x-axis represents time during an acquisition period, ranging from t0to t8. The y-axis represents the current of the current source, having values ranging from I0to I8. The values of t0-t8and I0-I8may be any suitable values for operation of a given ultrasound system, as the various aspects described herein are not limited to implementation of any specific time or current values. Also, the number of time intervals during an acquisition period is non-limiting, as more or fewer may be implemented. The number of current values which may be implemented is non-limiting, as more or fewer may be implemented. Curve402represents the current of a variable current source of a second stage of a variable current amplifier. Thus, curve402may represent the current of current source205of FIG.2. Curve404represents the current of a variable current source of a first stage of the variable current amplifier. Thus, curve404may represent the current of current source203ofFIG.2. FIG.4illustrates that the currents of the first and second stages of the variable current amplifier move in opposing directions during the acquisition period. That is, curve402decreases moving from time t0to time t8, while curve404increases during the same time. As previously described in connection withFIG.2, the first and second stages of the variable current amplifier may impact different characteristics of the variable current amplifier behavior, such as noise and linearity. Thus, when operating in the manner illustrated inFIG.4, the impact of the two stages of the variable current amplifier may vary during the acquisition period. That is, the impact of the second stage may be greater initially, up to time t4, while the impact of the first stage may be greater thereafter, from time t4to time t8. As previously described in connection withFIG.3A, the currents of the two stages of a two-stage op-amp being used to implement a variable current amplifier may be controlled by digital codes. Thus, the current values I0-I7ofFIG.4may correspond to different digital codes set by a digital controller, such as digital controller330ofFIG.3A. WhileFIG.4illustrates that the currents in the first and second stages of the amplifier switch at the same times, not all embodiments are limited in this respect. For example, the current in the second stage could be adjusted at times offset from those at which the current in the first stage is adjusted. Likewise, the currents of the two stages need not be adjusted the same number of times during an acquisition period. As described previously, an aspect of the present application provides an amplifier with a variable current source which is controlled to adjust the noise of the amplifier during an acquisition period.FIG.5illustrates an example of such operation. InFIG.5, the voltage of an electrical signal502output by an ultrasonic transducer, and thus representing a detected ultrasound signal, is illustrated as a function of time. Dashed line504represents the noise floor of an amplifier used to amplify the electrical signal502, and may correspond to the noise floor of an amplifier of the types described herein, such as amplifier112a. It can be seen that during the acquisition period, the magnitude of the electrical signal decreases. Likewise, the noise floor of the amplifier is decreased. Such a decrease in the noise floor may be achieved by controlling the current sunk by a variable current source of the amplifier in the manner described previously herein. For example, referring toFIG.2, the variable current source203may be increased during the acquisition period to decrease the noise floor of the amplifier112a. The noise floor may be adjusted to a level which provides an acceptable signal-to-noise ratio (SNR). FIG.5also illustrates a constant noise floor506. It can be seen that while the constant noise floor506is at the same level as dashed line504toward the end of the acquisition period, the constant noise floor506is lower than the value of the dashed line504up to that point. As has been described herein, the noise level of an amplifier may be dependent on the current consumed by the amplifier, and in such situations it should be appreciated that operating with a constant noise floor506requires significantly more current (and therefore power) than operating according to dashed line504. Thus, aspects of the present application providing for a variable current amplifier to amplify ultrasound signals may provide substantial power savings compared to amplifiers operating with a constant noise level. The amount of power savings may be significant. For example, in the circuit100, the amplifiers112a. . .112nmay consume a significant amount of power. In some embodiments, the amplifiers112a. . .112nmay consume more power than any other components of the circuit100. Accordingly, reducing the power consumption of the amplifiers112a. . .112nmay provide a significant reduction in power of the circuit100. In some embodiments, utilizing variable current amplifiers of the types described herein may provide up to a 25% power reduction, up to a 40% power reduction, up to a 50% power reduction, or any range or value within such ranges, in terms of the operation of the amplifier. The resulting power reduction for the circuit100may be up to 10%, up to 20%, up to 25%, or any range or value within such ranges. Having thus described several aspects and embodiments of the technology of this application, it is to be appreciated that various alterations, modifications, and improvements will readily occur to those of ordinary skill in the art. Such alterations, modifications, and improvements are intended to be within the spirit and scope of the technology described in the application. It is, therefore, to be understood that the foregoing embodiments are presented by way of example only and that, within the scope of the appended claims and equivalents thereto, inventive embodiments may be practiced otherwise than as specifically described. As an example, certain embodiments described herein have focused on two-stage amplifiers. However, the techniques described herein may apply to multi-stage amplifiers having two or more stages. When more than two stages are used, the first stage may predominantly control the noise of the amplifier, while the last stage may predominantly control the linearity of the amplifier. As described, some aspects may be embodied as one or more methods. The acts performed as part of the method(s) may be ordered in any suitable way. Accordingly, embodiments may be constructed in which acts are performed in an order different than illustrated, which may include performing some acts simultaneously, even though shown as sequential acts in illustrative embodiments. All definitions, as defined and used herein, should be understood to control over dictionary definitions, definitions in documents incorporated by reference, and/or ordinary meanings of the defined terms. The phrase “and/or,” as used herein in the specification and in the claims, should be understood to mean “either or both” of the elements so conjoined, i.e., elements that are conjunctively present in some cases and disjunctively present in other cases. As used herein in the specification and in the claims, the phrase “at least one,” in reference to a list of one or more elements, should be understood to mean at least one element selected from any one or more of the elements in the list of elements, but not necessarily including at least one of each and every element specifically listed within the list of elements and not excluding any combinations of elements in the list of elements. As used herein, the term “between” used in a numerical context is to be inclusive unless indicated otherwise. For example, “between A and B” includes A and B unless indicated otherwise. In the claims, as well as in the specification above, all transitional phrases such as “comprising,” “including,” “carrying,” “having,” “containing,” “involving,” “holding,” “composed of,” and the like are to be understood to be open-ended, i.e., to mean including but not limited to. Only the transitional phrases “consisting of” and “consisting essentially of” shall be closed or semi-closed transitional phrases, respectively. | 30,352 |
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